LINER LTC3890 High voltage surge stopper Datasheet

LTC4366
High Voltage Surge Stopper
FEATURES
DESCRIPTION
Rugged Floating Topology
nn Wide Operating Voltage Range: 9V to >500V
nn Adjustable Output Clamp Voltage
nn Controls N-Channel MOSFET
nn Adjustable Protection Timer
nn Internal 9-Second Cool-Down Timer
nn Shutdown I < 14µA
Q
nn 8-Lead TSOT and 3mm × 2mm DFN Packages
The LTC®4366 surge stopper protects loads from high
voltage transients. By controlling the gate of an external
N-channel MOSFET, the LTC4366 regulates the output
during an overvoltage transient. The load may remain
operational while the overvoltage is dropped across the
MOSFET. Placing a resistor in the return line isolates the
LTC4366 and allows it to float up with the supply; therefore,
the upper limit on the output voltage depends only on the
availability of high valued resistors and MOSFET ratings.
nn
APPLICATIONS
Industrial, Automotive and Avionic Surge Protection
High Voltage DC Distribution
nn 28V Vehicle Systems
nn
nn
An adjustable overvoltage timer prevents MOSFET damage during the surge while an additional 9-second timer
provides for MOSFET cool down. A shutdown pin reduces
the quiescent current to less than 14µA during shutdown.
After a fault the LTC4366-1 latches off while the LTC4366‑2
will auto-retry.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property
of their respective owners. Patents pending.
TYPICAL APPLICATION
Overvoltage Protected 1.5A, 28V Supply
IXTK90N25L2
VIN
28V
VOUT
1.5A
2nF
10Ω
0.47µF
324k
VDD
GATE
SD
LTC4366-2
TIMER
VSS
Overvoltage Protector Regulates Output at 43V During Transient
OUT
VIN
100V/DIV
28V
12.4k
FB
BASE
250V INPUT SURGE
422k
VOUT
20V/DIV
43V CLAMP
28V
1µF
436612 TA01a
100ms/DIV
46.4k
436612 TA01b
436612fe
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1
LTC4366
ABSOLUTE MAXIMUM RATINGS
(Notes 1, 2) All voltages relative to VSS, unless otherwise noted.
Supply Voltage (VDD) ................................. –0.3V to 10V
Supply Voltage (OUT) .................................. –0.3V to 5V
Input Voltages
FB............................................... –0.3V to OUT + 0.3V
TIMER.................................................... –0.3V to 3.5V
SD........................................................... –0.3V to 10V
Output Voltages
BASE..........................................................–1.5V to 4V
OUT – BASE .......................................... –0.3V to 5.5V
GATE (Note 3)......................................... –0.3V to 15V
GATE – OUT (Note 3).............................. –0.3V to 10V
Currents
IVDD....................................................................10mA
IOUT ....................................................................10mA
BASE.................................................. –300µA to 10µA
SD........................................................–10mA to 10µA
Operating Ambient Temperature Range (Note 4)
LTC4366C................................................. 0°C to 70°C
LTC4366I..............................................–40°C to 85°C
LTC4366H........................................... –40°C to 125°C
LTC4366MP........................................ –55°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
TSOT-23 Package Only..................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
VDD 1
SD 2
TIMER 3
VSS 4
8 GATE
7 OUT
6 FB
5 BASE
TS8 PACKAGE
8-LEAD PLASTIC TSOT-23
TJMAX = 150°C, θJA = 195°C/W
2
VSS 1
8
BASE
TIMER 2
7
FB
6
OUT
5
GATE
SD 3
VDD 4
9
DDB PACKAGE
8-LEAD (3mm × 2mm) PLASTIC DFN
TJMAX = 150°C,
θJA = 75°C/W IF VSS IS SOLDERED TO PCB, θJA = 135°C/W IF VSS IS NOT SOLDERED TO PCB
EXPOSED PAD (PIN 9), PCB VSS CONNECTION OPTIONAL
436612fe
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LTC4366
ORDER INFORMATION
Lead Free Finish
TAPE AND REEL (MINI)
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4366CTS8-1#TRMPBF
LTC4366CTS8-1#TRPBF
LTFMC
8-Lead Plastic TSOT-23
0°C to 70°C
LTC4366ITS8-1#TRMPBF
LTC4366ITS8-1#TRPBF
LTFMC
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC4366HTS8-1#TRMPBF
LTC4366HTS8-1#TRPBF
LTFMC
8-Lead Plastic TSOT-23
–40°C to 125°C
LTC4366CDDB-1#TRMPBF
LTC4366CDDB-1#TRPBF
LFMD
8-Lead (3mm × 2mm) Plastic DFN
0°C to 70°C
LTC4366IDDB-1#TRMPBF
LTC4366IDDB-1#TRPBF
LFMD
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC4366HDDB-1#TRMPBF
LTC4366HDDB-1#TRPBF
LFMD
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 125°C
LTC4366CTS8-2#TRMPBF
LTC4366CTS8-2#TRPBF
LTFMF
8-Lead Plastic TSOT-23
0°C to 70°C
LTC4366ITS8-2#TRMPBF
LTC4366ITS8-2#TRPBF
LTFMF
8-Lead Plastic TSOT-23
–40°C to 85°C
LTC4366HTS8-2#TRMPBF
LTC4366HTS8-2#TRPBF
LTFMF
8-Lead Plastic TSOT-23
–40°C to 125°C
LTC4366CDDB-2#TRMPBF
LTC4366CDDB-2#TRPBF
LFMG
8-Lead (3mm × 2mm) Plastic DFN
0°C to 70°C
LTC4366IDDB-2#TRMPBF
LTC4366IDDB-2#TRPBF
LFMG
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC4366HDDB-2#TRMPBF
LTC4366HDDB-2#TRPBF
LFMG
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 125°C
LTC4366MPTS8-1#TRMPBF
LTC4366MPTS8-1#TRPBF
LTFMC
8-Lead Plastic TSOT-23
–55°C to 125°C
LTC4366MPTS8-2#TRMPBF
LTC4366MPTS8-2#TRPBF
LTFMF
8-Lead Plastic TSOT-23
–55°C to 125°C
LTC4366MPDDB-1#TRMPBF LTC4366MPDDB-1#TRPBF
LFMD
8-Lead (3mm × 2mm) Plastic DFN
–55°C to 125°C
LTC4366MPDDB-2#TRMPBF LTC4366MPDDB-2#TRPBF
LFMG
8-Lead (3mm × 2mm) Plastic DFN
–55°C to 125°C
TRM = 500 pieces. *Temperature grades are identified by a label on the shipping container.
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
436612fe
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3
LTC4366
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. All voltages relative to VSS, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VZ(VDD)
VDD Shunt Regulator Voltage
I = 1mA
∆VZ(VDD)
VDD Shunt Regulator Load Regulation
I = 1mA to 5mA
VDD
VDD Supply Voltage (Note 3)
MIN
TYP
MAX
UNITS
11.5
12
12.5
V
30
30
90
130
mV
mV
VDD Regulator
l
LTC4366C/I/H
LTC4366MP
l
l
l
4.5
VZ(VDD)
V
IVDD(STLO)
VDD Pin Current – Start-Up, Gate Low
GATE = 0V, VDD = 7V, OUT = 0V
l
15
23
µA
IVDD(STHI)
VDD Pin Current – Start-Up, Gate High
GATE Open, VDD = 7V, OUT = 0V
l
9
13
µA
IVDD(SD)
VDD Pin Current – Shutdown
VDD = 7V, OUT = 0V
l
5
8
µA
OUT Shunt Regulator Voltage
I = 1mA, BASE = 0V
l
5.7
6.0
V
∆VZ(OUT)
OUT Shunt Regulator Load Regulation
I = 1mA to 5mA
l
70
mV
OUT
OUT Supply Voltage (Note 3)
VUVLO1
OUT Undervoltage Lockout 1
∆VUVH1
OUT Undervoltage Lockout 1 Hysteresis
VUVLO2
OUT Undervoltage Lockout 2
∆VUVH2
OUT Regulator
VZ(OUT)
5.0
30
l
3.0
l
l
2.42
2.42
l
l
OUT Undervoltage Lockout 2 Hysteresis
l
IOUT(AMP)
OUT Pin Current – Regulation Amplifier On
l
IOUT(CP)
OUT Pin Current – Charge Pump On
l
IOUT(SD)
OUT Pin Current – Shutdown
l
Rising
LTC4366C/I/H
LTC4366MP
Rising
VZ(OUT)
V
2.55
2.55
2.75
2.80
V
V
0.2
0.28
0.4
V
4.5
4.75
4.9
V
0.3
0.4
0.5
V
37
54
µA
150
220
µA
3
6
µA
6.2
6.6
V
125
200
mV
BASE, VSS
VZ(BASE)
BASE Shunt Regulator Voltage (OUT – BASE)
I = –10µA, OUT = 4.5V
l
∆VZ(BASE)
BASE Shunt Regulator Load Regulation
I = –10µA to –80µA, OUT = 4.5V
l
OUT = 4.5V, BASE = –0.5V
l
5.5
IBASE
BASE Pin Leakage Current
–0.1
–0.8
–5.5
µA
IVSS(AMP)
VSS Pin Current – Regulation Amplifier On
l
–30
–45
–72
µA
IVSS(CP)
VSS Pin Current – Charge Pump On
l
–108
–160
–230
µA
IVSS(SD)
VSS Pin Current – Shutdown
l
–7
–12
µA
GATE Drive
∆VGATE
External N-Channel Gate Drive (GATE – OUT)
OUT = 4.9V, I = 0, –1µA
IGATE(ST)
GATE Pin Current – Start-Up
GATE = OUT = 0V
IGATE(CP)
GATE Pin Current – Charge Pump On
IGATE(FD)
IGATE(FLT)
4
l
11.2
12
12.5
V
l
l
–4.5
–3.2
–7.5
–7.5
–11
–11
µA
µA
GATE = 5V, OUT = 4.9V
l
–14
–20
–28
µA
GATE Pin Current – Fast Discharge
GATE = 10V, OUT = 4.9V
l
122
200
300
mA
GATE Pin Current – Fault
GATE = 10V, OUT = 4.9V
l
0.3
0.7
1.2
mA
LTC4366C/I/H
LTC4366MP
436612fe
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LTC4366
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. All voltages relative to VSS, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
1.193
1.23
1.267
V
FB, SD, TIMER
VFB(REG)
3% FB pin Regulation Threshold (OUT – FB)
IFB
FB Pin Leakage Current
OUT – FB = 1.2V
l
0
±1
µA
VSD(TH)
SD Pin Threshold Voltage (VDD – SD)
Falling
l
1.0
1.5
2.3
V
VSD(HYST)
SD Pin Hysteresis
LTC4366C/I/H
LTC4366MP
l
l
147
129
280
280
530
530
mV
mV
ISD
SD Pin Input Pull-Up Current
VDD – SD = 0.7V
LTC4366C/I/H
LTC4366MP
l
l
–0.7
–0.5
–1.6
–1.6
–3.5
–3.5
µA
µA
VTIMER(H)
TIMER Pin Threshold
TIMER Rising, VDD = 7V, OUT = VZ(OUT)
l
2.6
2.8
3.1
V
ITIMER(UP)
TIMER Pin Pull-Up Current
TIMER = 1V
LTC4366C/I/H
LTC4366MP
l
l
–5.1
–4
–9
–9
–13
–13
µA
µA
ITIMER(DN)
TIMER Pin Pull-Down Current
TIMER = 1V
LTC4366C/I/H
LTC4366MP
l
l
0.9
0.7
1.8
1.8
2.8
2.8
µA
µA
l
15
20
25
%
420
700
1200
µs
l
ITIMER(RATIO) TIMER Pin Current Ratio ITIMER(DN)/ITIMER(UP)
AC Characteristics
tDLY – SD
SD Low to Gate Low Filter Time
Step VDD – SD from 0V to 3V
l
tDLY – FAST
FB Low to Gate Low Delay Time
Step OUT – FB from 0V to 1.3V
l
60
150
300
ns
tD(COOL)
Cool-Down Timer (Internal)
VDD = VZ(VDD)
l
l
5.9
5.9
9
9
16
19
Seconds
Seconds
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: All currents into pins are positive.
LTC4366C/I/H
LTC4366MP
Note 3: Limits on the maximum rating is defined as whichever limit occurs
first. An internal clamp limits the GATE pin to a maximum of 12V above
source. Driving this pin to voltages beyond the clamp may damage the
device.
Note 4: TJ is calculated from the ambient temperature, TA, and power
dissipation, PD, according to the formula:
TJ = TA + (PD • θJA)
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5
LTC4366
TYPICAL PERFORMANCE CHARACTERISTICS
13.0
VDD Shunt Regulator vs
VDD Current
VDD Shunt Regulator vs
Temperature
VDD Start-Up Current vs
Temperature (Gate High)
13.0
15
12.5
12
IVDD(STHI) (µA)
VZ(VDD) (V)
VZ(VDD) (V)
12.5
12.0
12.0
6
11.5
11.5
5
0
10
IVDD (mA)
15
20
9
11.0
–50
–25
0
25
50
75
TEMPERATURE (°C)
436612 G01
100
3
–50
125
–25
0
25
50
75
TEMPERATURE (°C)
436612 G02
OUT Shunt Regulator vs
OUT Current
125
436612 G03
OUT Shunt Regulator vs
Temperature
5.9
100
VSS Current (Regulation AMP On)
vs Temperature
7
75
6
50
IVSS(AMP) (µA)
VZ(OUT) (V)
VZ(OUT) (V)
5.8
5.7
5
25
5.6
5.5
0
5
10
IOUT (mA)
15
4
–50
20
0
25
50
75
TEMPERATURE (°C)
436612 G04
100
16
–40
12
–30
IGATE(CP) (µA)
∆VGATE (V)
8
–100
100
125
436612 G11
6
0
100
125
–20
–10
4
0
25
50
75
TEMPERATURE (°C)
0
25
50
75
TEMPERATURE (°C)
Gate Current (Charge Pump On)
vs Temperature
–200
–25
–25
436612 G10
Gate Drive vs Gate Pull-Up
Current
–300
0
–50
0
–50
125
436612 G05
VSS Current (Charge Pump On)
vs Temperature
IVSS(CP) (µA)
–25
0
–10
–20
IGATE (µA)
–30
436612 G12
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
436612 G14
436612fe
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LTC4366
TYPICAL PERFORMANCE CHARACTERISTICS
Base Shunt Regulator vs
Base Current
Gate Start-Up Current vs
Temperature
–12
Timer Pull-Up Current vs
Temperature
7.0
–12
–10
ITIMER(UP) (µA)
6.5
VZ(BASE) (V)
IGATE(ST) (µA)
–10
–8
6.0
–6
–8
–6
–4
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
5.5
125
0
–100
436612 G15
–200
–300
IBASE (µA)
–400
–4
–50
–500
100
125
436612 G18
Cool-Down Time vs Temperature
1.24
–3
0
25
50
75
TEMPERATURE (°C)
436612 G16
FB Regulation Threshold vs
Temperature
SD Pull-Up Current vs
Temperature
–25
16
14
1.23
tD(COOL) (s)
ISD (µA)
VFB(REG) (V)
–2
12
10
1.22
–1
8
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
436612 G21
1.21
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
436612 G22
6
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
436612 G23
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7
LTC4366
PIN FUNCTIONS
BASE: Base Driver Output for External PNP Shunt Regulator. This pin is connected to the anode of an internal 6.2V
Zener with the cathode tied to OUT. In cases where lower
Zener (Z3) clamp current is desired but a large VSS resistor is prohibited, connect an external PNP base to this pin
(PNP collector is grounded, emitter is tied to VSS). Tie this
pin to VSS if unused.
Exposed Pad: The exposed pad may be left open or connected to VSS.
FB: Overvoltage Regulation Amplifier Feedback Input.
Connect this pin to an external resistive divider from OUT
to ground. The overvoltage regulation amplifier controls
the gate of the external N-channel MOSFET to regulate
the FB pin voltage at 1.23V below OUT. The overvoltage
amplifier will activate a 200mA pull-down on the GATE pin
during a fast overvoltage event.
GATE: Gate Drive for External N-Channel MOSFET. During start-up an internal 7.5µA current source charges the
gate of the external N-channel MOSFET from the VDD pin.
Once the OUT voltage is above VSS by 4.75V, the charge
pump will finish charging the GATE to 12V above OUT.
During a fast overvoltage event, a 200mA pull-down current source between GATE and OUT is activated, followed
by regulation of the GATE pin voltage by the overvoltage
regulation amplifier.
OUT: Charge Pump and Overvoltage Regulation Amplifier
Supply Voltage. Supply input for floating circuitry powered
from the MOSFET source. Once the OUT voltage is 4.75V
(UVLO2) above VSS, the charge pump will turn on and draw
power from this pin. When OUT exceeds 2.55V (UVLO1)
it is used as a power supply and reference input for overvoltage regulation amplifier. This pin is clamped at 5.7V
and requires a 0.22µF or greater bypass to the VSS pin.
8
SD: Shutdown Comparator Input. Tie to VDD if unused.
Connect pin to a limited current pull down created by adding
a resistor in series with an open-drain or open-collector
pull-down transistor. Activating the external pull down
overcomes the internal 1.6µA pull-up current source and
allows the SD pin to cross the shutdown threshold. This
threshold is defined as 1.5V below VDD with a 280mV
hysteresis. To prevent false triggers this pin must stay
below the threshold for 700µs to activate the shutdown
state. The shutdown state lowers the total quiescent current (IVDD plus IOUT) below 20µA. This quiescent current
does not include shunt current in the VDD, OUT and BASE
regulators. After a fault on the LTC4366, putting the part
in shutdown will clear the fault and allow operation to
resume. Clearing the fault during the 9-second cool-down
period will shorten the timeout for the LTC4366-2 (autoretry) version.
TIMER: Timer Input. Leave this pin open for a 1µs overvoltage regulation period before fault off. Connect a capacitor
between this pin and VSS to set a 311ms/µF duration for
overvoltage regulation before the switch is turned off.
The LTC4366-2 version will restart after a nine second
cool-down period.
VDD: Start-Up Supply. Supply input for 7.5µA start-up current source that charges the gate of the external N-channel
MOSFET. Also provides supply for timer and logic circuits
active when the external MOSFET is off. This pin is clamped
at 12V above VSS. Do not bypass this pin with a capacitor.
VSS: Device Return and Substrate. The capacitors on the
TIMER and OUT pins should be returned to this pin.
436612fe
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LTC4366
SIMPLIFIED DIAGRAM
Start
Run
Regulate
RIN
7.5µA
1.23V
20µA
CP
Z1
12V
Z3
5.7V
+
–
–
+
Z3
5.7V
RFB1
RFB2
RSS
RSS
RSS
436612 SD
FUNCTIONAL DIAGRAM
M1
VIN
VOUT
RG
RIN
CG
VDD
Z4
12V
GATE
7.5µA
OUT
Z1
12V
LOGIC
SUPPLY
D1
20µA
VCC
OUT
CHARGE
PUMP
f = 2MHz
UVLO2
4.75V
UVLO1
2.55V
VDD
VCC
1.6µA
+
–
SHUTDOWN
1.5V COMPARATOR
+
–
SD
1.23V
–
+
LOGIC
AND
TIMER
VCC
+
–
C1
FB
RFB1
RFB2
OVERVOLTAGE
AMPLIFIER
9µA
TIMER
COMPARATOR
TIMER
CT
1.8µA
+
–
+
–
2.8V
OUT
Z2
6.2V
Z3
5.7V
BASE
VSS
436612 FD
RSS
436612fe
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9
LTC4366
OPERATION
The Simplified Diagram shows three states of operation:
the start, run and regulate mode. Previous surge stopper
parts are powered off the input supply, therefore the surge
voltage is limited to the breakdown voltage of the input pins
of the part. As demonstrated in run and regulate modes,
the majority of this part is powered off the output, so the
MOSFET isolates the surge from the power pins of the
part. This allows surge voltages up to the breakdown of
the external MOSFET.
Once the OUT to VSS voltage exceeds the 2.55V UVLO1
threshold, the overvoltage amplifier is enabled. Next, the
UVLO2 threshold of 4.75V is crossed and the charge pump
turns on. The charge pump charges the GATE pin with
20µA to its final value 12V above OUT (clamped by Z4).
This allows the capacitor between OUT and VSS to charge
until clamped by Z3 to 5.7V. In this run mode the MOSFET
is configured as a low resistance pass transistor with little
voltage drop and power dissipation in the MOSFET.
In the start mode a 15µA trickle current flows through RIN,
half is used to charge the gate with the other half used as
bias current. As the GATE pin charges, the external MOSFET
brings up the OUT pin. This leads to the run mode where
the output is high enough to become a supply voltage for
the charge pump. The charge pump is then used to fully
charge the gate 12V above the source.
The powered up LTC4366 is now ready to protect the load
against an overvoltage transient. The overvoltage regulation amplifier monitors the load voltage between OUT and
ground by sensing the voltage on the FB pin with respect
to the OUT pin (drop across RFB1). In an overvoltage
condition the OUT rises until the amplifier drives the M1
gate to regulate and limit the output voltage. This is the
regulate mode.
With the output voltage equal to the input voltage, it is
necessary to protect the load from an input supply overvoltage. In the regulate mode, the overvoltage regulation
amplifier is referenced to the output through a 1.23V
reference. If the voltage drop across the upper feedback
resistor, RFB1, exceeds 1.23V the regulation amplifier pulls
the gate down to regulate the RFB1 voltage back to 1.23V.
Therefore, the output voltage is clamped by setting the
proper ratio between RFB1 and RFB2.
For example, if the output voltage is regulated at 100V then
the voltage drop across the RFB2 is 98.77V. If the Zener Z3
is 5.7V then the voltage drop across RSS is 94.3V. Therefore, when the output is at a high voltage, the majority of
the voltage is dropped across the two resistors RFB2 and
RSS. This demonstrates how the LTC4366 floats up with
the supply. The adjustable 3-terminal regulators, such
as the LT®1085 and LM117, are also based on this idea.
During regulation the excess voltage is dropped across the
MOSFET. To prevent overheating the MOSFET, the LTC4366
limits the overvoltage regulation time using the TIMER pin.
The TIMER pin is charged with 9µA until the pin exceeds
2.8V. At that point an overvoltage fault is set, the MOSFET
is turned off, and the part enters a cool-down period of
9 seconds. The logic and timer block are active during
cool down while the GATE pin is pulled to OUT.
The latched-off version, LTC4366-1, will remain in fault
until the SD pin is toggled low and then high. Once the fault
is cleared, the GATE is permitted to turn the MOSFET on
again. The auto-retry version, LTC4366-2, waits 9 seconds
then clears the fault and restarts.
The Functional Diagram shows the actual circuits. An
external RIN resistor on the VDD pin powers up the 12V
shunt regulator which then powers up logic supply, VCC.
After verifying that the shutdown input is not active, the
GATE pin is charged with a 7.5µA current from VDD. This
is the start mode.
10
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The typical LTC4366 application is a protected system
that distributes power to loads safe from overvoltage
transients. External component selection is discussed in
the following sections.
Dual Shunt Regulators
The LTC4366 uses two shunt regulators coupled with
the external voltage dropping resistors, RSS and RIN, to
generate internal supply rails at the VDD and OUT pins.
These shunt-regulated rails allow overvoltage protection
from unlimited high voltage transients irrespective of the
voltage rating of the LTC4366’s internal circuitry.
At the beginning of start-up, during shutdown, or after an
overvoltage fault, the GATE pin is clamped to the OUT pin
thereby shutting off the MOSFET. This allows the VSS and
OUT pins to be pulled to ground by output load and RSS.
Under this condition the VDD pin is clamped with a 12V
shunt regulator to VSS. The full supply voltage minus 12V
is then impressed on the RIN resistor which sets the shunt
current. The shunt current can be as high as 10mA which
is several orders of magnitude higher than the typical 9µA
VDD pin quiescent current.
In normal operation the OUT voltage is equal to the input
supply. With C1 fully charged IC1 is zero at this point. Under
this condition the voltage between the OUT and VSS pins
are clamped with a 5.7V shunt regulator. The input supply
M1
FQA62N25C
VIN
28V
(18V DC TO 250V DC)
RIN
324k
R1
470k
SD
R2
100k
VDD
SD
Q1
MMBT3904
C1
0.47µF
GATE
LTC4366-2
TIMER
VSS
Turn-On Sequence
The voltage between the VDD and VSS pins is shunt regulated to 12V after ramping up the input supply. Next, the
internally generated supply, VCC, produces a 30µs poweron-reset pulse which clears the fault latch and initializes
internal latches. Next, the shutdown comparator determines
if the SD pin is externally pulled low, thereby requesting a
low bias current shutdown state. Otherwise the external
MOSFET, M1, is allowed to turn on.
Turning on the 7.5µA GATE pull-up current source from the
VDD pin begins what can be described as a “bootstrapped”
method for powering up the MOSFET gate. Once the GATE
reaches the VDD pin voltage (minus a Schottky diode), the
7.5µA source loses voltage headroom and stops charging
the GATE (middle of waveforms in Figure 2.). The bootstrap
method relies on charging C1 to a sufficient voltage after
GATE stops increasing. The voltage on C1 is then used
as a supply for a charge pump that charges the gate to
its final value 12V above OUT. C1 will discharge if the
charge pump current exceeds the C1 charging current.
If the voltage drops below 4.35V, the charge pump will
pause allowing C1 to recharge.
VOUT
1.5A
(43V CLAMP)
CG
10nF
RG
10Ω
voltage minus 5.7V is impressed on RSS. The RSS current
is divided into three areas: the 5.7V shunt current, bias
current between OUT and VSS and finally the RIN current.
The 5.7V shunt current can be as high as 10mA which
greatly exceeds the typical OUT (160µA) bias current.
OUT
FB
BASE
RFB1
12.4k
RFB2
422k
VGATE
10V/DIV
CHARGE
PUMP PAUSE
VOUT
10V/DIV
C1 RECHARGING
CT
8.2nF
RSS
46.4k
436612 F01
VC1
5V/DIV
CHARGE
PUMP STARTS
C1 CHARGING
20ms/DIV
Figure 1. Typical Application
436612 TA01b
Figure 2. Turn-On Waveforms
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LTC4366
APPLICATIONS INFORMATION
Starting up with a supply voltage insufficient to charge
C1 with large load current may result in overheating the
MOSFET and subsequent damage. While the gate and
output are ramping the drop across the MOSFET is the
input supply minus the output. If the supply is lower than
necessary to charge C1, then the output fails to ramp higher
than the supply minus the threshold of the MOSFET. This
3V to 5V MOSFET drop with high load current will result in
power dissipation without any protection or timeout limit.
Overvoltage Fault
The LTC4366 prevents an overvoltage on the input supply
from reaching the load. Normally, the pass transistor is
fully on, powering the load with very little voltage drop. As
the input voltage increases the OUT voltage increases until
it reaches the regulation point (VREG). From that point any
further voltage increase is dropped across the MOSFET.
Note the MOSFET is still on so the LTC4366 allows uninterrupted operation during a short overvoltage event.
The VREG point is configured with the two FB resistors, RFB1
and RFB2. The regulation amplifier compares the FB pin to
a threshold 1.23V below the OUT pin. During regulation
the drop across RFB1 is 1.23V, while the remainder of the
VREG voltage is dropped across RFB2.
When the output is at the regulation point a timer is started
to prevent excessive power dissipation in the MOSFET.
Normally the TIMER pin is held low with a 1.8µA pulldown current. During regulation the TIMER pin charges
with 9µA. If the regulation point is held long enough for
the TIMER pin to reach 2.8V then an overvoltage fault is
latched. The equation for setting the timer capacitor is:
CT = 3.2 • t [nF / ms]
12
Depending on which version, the part will cool down and
self start (LTC4366-2), or remain latched off until the SD
pin activates a shutdown followed by a start-up command
(LTC4366-1). The cool-down time is typically nine seconds
which provides a very low pulsed power duty cycle.
Starting up with an input supply overvoltage and full
load current does increase the power dissipation in the
MOSFET well beyond the case for an overvoltage surge.
During the gate and output ramp up, the partial supply
voltage (at full current) is dropped across the MOSFET.
After start-up the normal overvoltage surge (with timeout)
occurs before the shutting off the MOSFET. The Design
Example section only considers the normal overvoltage
surge for safe operating area (SOA) calculations for the
MOSFET. Start-up into overvoltage will require additional
SOA considerations.
Shutdown
The LTC4366 has a low current (<20µA) shutdown state
that turns off the pass FET by tying the GATE and OUT pins
together with a switched resistor. In the normal operating
condition, the SD pin is pulled up to the VDD pin voltage
with a 1.6µA current source. Tie the SD pin to VDD when
the shutdown state is not used.
Bringing the SD pin more than 1.5V below VDD pin voltage for greater than the 700µs filter time activates the
shutdown state. This filter time prevents unwanted activation of shutdown during transients. The SD pin is diode
clamped 0.7V below VSS which requires current limiting
(maximum 10mA) on the pull-down device. One way to
limit the current is to connect an external 470k resistor in
series with the open-collector pull-down device. Activating the external pull-down overcomes the internal 1.6µA
pull-up current source and allows the SD pin to cross the
shutdown threshold.
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Following an overvoltage fault, putting the part in shutdown
will clear the fault, allowing operation to resume once the
LTC4366 leaves shutdown.
Output Short
A sudden short on the output can result in excessive current into the LTC4366 GATE pin supplied from the gate
capacitor, CG. The GATE pin is internally clamped to OUT
with a 10V to 12V clamp. If the OUT pin is pulled low
while the GATE pin is held up with CG, then the clamp will
be damaged trying to discharge CG when clamp voltage
is exceeded. One solution is to add a 1k RS resistor in
series with CG with a bypass diode as shown in Figure 3.
The diode allows the capacitor to function as a bypass for
energy coming from the MOSFET drain to gate capacitor
during an supply overvoltage.
The full supply voltage minus 12V can appear across
RIN during the overvoltage cool-down period. Normally
the value for RIN is several times larger than RSS which
lowers the power and size requirements for this resistor.
External PNP
In some cases the power resistor for RSS may be physically
large. A large value RSS (with lower power and size) may
be used in conjunction with a PNP as shown in Figure 4.
In addition to the 0.8µA sourced from the BASE pin, the
base current from the PNP must flow through RSS which
will limit the maximum RSS value. In some cases the
minimum PNP Beta is as low as 35. The base current
becomes 10µA when the VSS current is 350µA. One can
see this allows a 35 (Beta) times larger RSS than the application without the PNP.
M1
RG
LTC4366
DBYPASS
LTC4366
10V TO
12V
GATE
BASE
RS
1k
CG
VSS
RSS
OUT
436612 F04
436612 F03
Figure 4. External PNP Option
Figure 3. Output Short Protection
Resistor Power Ratings
Minimum Supply Start-Up
The proper rating for the RSS resistor in Figure 1 must be
considered. During an overvoltage event the OUT pin is
at regulation voltage (VREG), so the voltage across RSS is
VREG minus 5.7V. A small minimum supply voltage reduces
the value of RSS. Therefore, large differences between
minimum supply voltage and the regulation voltage may
require a large power resistor for RSS.
When designing for the minimum supply condition, it is
important that RSS and RIN are chosen to provide enough
current to sufficiently charge C1 to 4.75V. The parameters
that determine the minimum supply voltage include: C1
voltage, MOSFET threshold voltage, a series Schottky diode
voltage drop, resistance of RSS and RIN, current in the VDD
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pin, and finally the current from the VSS pin (see Figure 5).
VIN(MIN) = (IVDD • RIN) + VD + VTH + VC1 + (IVSS • RSS)
Using the Electrical Characteristics table for above
parameters:
VC1 = VUVLO2 = 4.75V (UVLO2 threshold)
IVDD = IVDD(STHI) = 9µA (IVDD start-up, gate high)
IVSS = IVSS(AMP) = 45µA (IVSS w/regulation amp)
VD = 0.58V
VIN(MIN) = (9µA • RIN) + 0.58V + VTH + 4.75V + (45µA • RSS)
When the MOSFET gate is fully enhanced, the OUT pin
voltage is equal to the supply voltage. This places another
constraint on the minimum supply voltage because the
charge pump increases the VSS current to 160µA. The C1
voltage is assumed to be clamped at 5.7V. These values
are specified as VZ(OUT) and IVSS(CP) (charge pump on)
in the table of Electrical Characteristics:
VIN(MIN) = VZ(OUT) + (IVSS(CP) • RSS)
VIN(MIN) = 5.7V + (160µA • RSS)
VIN
RIN
7.5µA
VDD
D1
GATE
M1
CG
VOUT
OUT
RLOAD
Z1
12V
LOGIC
TIMER
ISHUNT1
9µA
Z3
5.7V CIRCUITS
ISHUNT2
+
C1
–
VC1
IC1
IBIAS
VSS
RSS
IRSS
436612 F05
Figure 5. Simplified Block Diagram
14
RSS(MAX) =
VIN(MIN) – 5.7V
160µA
RIN(MAX) =
VIN(MIN) – 4.75V – 0.58V – VTH – ( 45µA • RSS )
9µA
These two equations maximize the values of RSS and
RIN (reducing power dissipation) while still providing
the necessary VC1 voltage to turn the charge pump on.
Increasing the supply voltage beyond the minimum supply voltage increases the current and power in RSS while
reducing the time required to charge C1. Conditions that
may require an even smaller RSS(MAX) will be discussed
in the Maximum Supply Start-Up section.
Maximum Supply Start-Up
The maximum overvoltage supply may also exist during
start-up. The overvoltage protection circuitry has to wake
up before high voltage is passed to the load. Dynamically
the GATE is ramping up while C1 is charging. Capacitor
C1 must charge to the 2.55V UVLO1 threshold to turn on
the regulation amplifier and reference before the OUT pin
voltage exceeds the overvoltage regulation point, VREG.
These conditions may reduce the value of RSS below the
maximum value dictated by the minimum supply start-up
discussed above.
or
IRIN
The last VIN(MIN) equation sets the maximum value for
RSS. After choosing RSS the maximum value for RIN (for
that particular RSS) is calculated from the first VIN(MIN)
equation:
When current in RSS exceeds the current sourced from
the VSS pin (essentially IRIN), the capacitor C1 begins to
charge. The voltage at the VSS pin when IRIN = IRSS is now
labeled VSS(MATCH). The VSS pin voltage is the center of a
voltage divider between RIN and RSS after the Zener clamp
voltage from VDD to VSS is subtracted from the supply.
VSS(MATCH) =
RSS
• ( VIN(MAX) – VZ(VDD) )
RSS +RIN
As VIN increases the VSS(MATCH) voltage increases. If the
match voltage exceeds the overvoltage regulation point
(VREG), then load is unprotected. This is true because
C1 will still need to charge to 2.55V while VSS already
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has exceeded VREG. Since the OUT pin voltage is at least
2.55V larger than VSS it exceeds the specified maximum.
Choosing the match point (with supply at the maximum)
sufficiently below VREG (by at least 2.55V), allows C1 to
charge up in time to protect the load from overvoltage.
In reality having VSS pin voltage 7V below VREG provides
required margin for charging C1.
VSS(MATCH)(MAX) = VREG – 7V
Increasing RSS increases the match point, so determining the maximum RSS value while still protecting from
overvoltage is useful. Using IRIN = IRSS:
RSS = RIN •
VRSS
VRIN
GATE Capacitor, CG
The gate capacitor is used for three functions. First, CG
absorbs charge from the gate-to-drain capacitance of
the MOSFET during overvoltage transients. Second, the
capacitor also acts as a compensation element for the
overvoltage regulation amplifier. The minimum value for
CG to guarantee stability is 2nF. Finally, CG sets the slew
rate of the GATE and OUT pins. The voltage at the GATE pin
rises at a slope equal to 20µA/CG. This slope determines
the charging current into the load capacitor.
IINRUSH =
CLOAD
•IG
CG
The voltage rating for CG must be greater than the regulation voltage (VREG).
Using:
VRSS = VSS(MATCH)(MAX) = VREG – 7V
MOSFET Selection
VRIN = VIN – VZ(VDD) – VRSS
The LTC4366 drives an N-channel MOSFET to conduct
the load current. The important features of the MOSFET
are on-resistance, RDS(ON), the maximum drain-source
voltage, V(BR)DSS, the threshold voltage, and the SOA.
Substituting:
RSS(MAX) =
RIN • ( VREG – 7V )
VIN(MAX) – 12V – ( VREG – 7V )
RSS(MAX) =
RIN • ( VREG – 7V )
VIN(MAX) – 5V – VREG
If we guarantee that RSS < RSS(MAX) then the following
is true:
VSS(MATCH) < VSS(MATCH)(MAX)
C1 bypasses the charge pump, and requires at least a
0.22µF. The size of C1 needs limits also. The gate capacitor (CG) dictates the maximum output capacitor C1(MAX)
that will charge to the 2.55V UVLO1 threshold (VUVLO1)
before the OUT voltage exceeds the overvoltage threshold.
C1(MAX ) =
–C G • (RSS +RIN ) ( VREG – VSS(MATCH ) )


2 • VUVLO1
IG • RSS • RIN •In1–

 VREG – VSS(MATCH ) 
The maximum allowable drain-source voltage must be
higher than the supply voltage. If the output is shorted
to ground or during an overvoltage event, the full supply
voltage will appear across the MOSFET.
The threshold voltage of the MOSFET is used in the minimum supply start-up calculation. For applications with
supplies less than 12V, a logic-level MOSFET is required.
Above 12V a standard threshold N-channel MOSFET is
sufficient.
The SOA of the MOSFET must encompass all fault conditions. In normal operation the pass transistor is fully on,
dissipating very little power. But during overvoltage faults,
the GATE pin is servoed to regulate the output voltage
through the MOSFET. Large current and high voltage drop
across the MOSFET can coexist in these cases. The SOA
curves of the MOSFET must be considered carefully along
with the selection of the fault timer capacitor.
In most cases:
C1(MAX) = 10 • CG to 100 • CG
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Layout Considerations
DESIGN EXAMPLE
Due to the high impedances on the SD, VDD, and GATE
pins, these pins are susceptible to leakages to ground.
For example, a leakage to ground on SD will activate the
shutdown state if greater than 1.6µA. Providing adequate
spacing away from grounded traces and adding conformal
coating on exposed pins lowers the risk that leakage current will interrupt system operation.
Overview
The design process starts with minimum input voltage
start-up equations to calculate values for RSS and RIN.
These values need further refinement to meet two other
conditions: the maximum input voltage start-up conditions and proper current for the charging of C1. The the
remaining element values are calculated based on the
input parameters.
It is important to put the bypass capacitor, C1, as close as
possible to the OUT and VSS pins. Place the 10Ω resistor
as close as possible to the MOSFET gate pin. This will
limit the parasitic trace capacitance that leads to MOSFET
self-oscillation.
Following are the input parameters for this example:
VSUPPLY(MIN) = 18V, VREG = 43V, VIN(MAX) = 250V, ILOAD
= 1.5A at start-up, ILOAD = 3A after start-up, VTH = 5V
The FB pin is sensitive to parasitic capacitance when the
regulation loop is closed. One result from this capacitive
loading is output oscillations during overvoltage regulation. It is suggested that the resistors RFB1 and RFB2 be
placed close to the pin and that the FB trace itself be
minimized in size.
Important Electrical Characteristics table parameters used
in this example are summarized in Table 1.
Step 1: Maximum RSS
In this design example (Figure 6.) the component sizing
first considers the start-up phase after the charge pump
is active. The goal is to maximize the resistance of RSS
which still allows operation when the input voltage is at
the minimum value.
M1
FQA62N25C
VIN
28V
(18V DC TO 250V DC)
VOUT
1.5A
(43V CLAMP)
CG
10nF
RG
10Ω
C1
0.47µF
RIN
324k
R1
470k
SD
R2
100k
Q1
MMBT3904
VDD
GATE
SD
LTC4366-2
TIMER
VSS
OUT
FB
BASE
RFB1
12.4k
RFB2
422k
CT
8.2nF
RSS
46.4k
436612 F06
Figure 6. Overvoltage Protected 28V, 1.5A Supply
16
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Table 1. Electrical Parameters Used in Design Example
SYMBOL
PARAMETER
CONDITIONS
TYP
MAX
VZ(OUT)
OUT Shunt Reg. Voltage
I = 1mA, BASE = 0V
5.7V
6.0V
VUVLO2
OUT Undervoltage Lockout 2
Rising
4.75V
4.9V
IVSS(CP)
VSS Pin Current – Charge Pump On
–160µA
–230µA
IVSS(AMP)
VSS Pin Current – Regulation Amplifier On
–45µA
–72µA
IVDD(STHI)
VDD Pin Current – Start-Up, Gate High
GATE Open, VDD = 7V, OUT = 0V
9µA
13µA
IGATE(ST)
GATE Pin Current – Start-Up
GATE = OUT = 0V
–7.5µA
–11µA
VUVLO1
OUT Undervoltage Lockout 1
Rising
2.55V
2.75V
After the charge pump is active the VSS current increases
to 160µA (worst-case 230µA, see Table 1) current while
the final value OUT voltage is equal to the minimum supply
voltage. The C1 voltage is clamped at 5.7V (worst-case
6.0V):
RSS(MAX) =
VIN(MIN) – VZ(OUT)
IVSS(CP)
RSS(MAX) =
18V – 6V
= 52.3k
230µA
Step 2: Determine RIN
The value for resistor RIN is calculated using the calculated
RSS value. RIN is chosen to provide enough headroom to
sufficiently charge C1 to 4.9V the maximum undervoltage
lockout 2 threshold (VUVLO2) which starts the charge pump.
The parameters that determine RIN include: minimum
supply voltage, the final C1 voltage, MOSFET threshold
voltage, RSS, 72µA maximum VSS pin current (regulation
amplifier on, IVSS(AMP)), and finally the 13µA maximum
start-up current in the VDD pin (IVDD(STHI)):
RIN(MAX) =
RIN(MAX) =
VIN(MIN) – VUVLO2 − VD − VTH − (ISS(AMP) • RSS )
IVDD(STHI)
18V − 4.9V − 0.58V − 5V − (72µA • 52.3k )
13µA
RIN(MAX) = 287k
Step 3: Find RSS(MAX)
In some cases this value for RSS is too large to charge C1
and power the overvoltage amplifier before the maximum
input voltage passes to the output. The voltage at the VSS
pin when IRIN = IRSS is called the match point (VSS(MATCH)).
Choosing the match point (with supply at the maximum)
sufficiently below VREG (by at least 7V), allows C1 to charge
up in time to protect the load from overvoltage:
RSS(MAX) =
RIN • ( VREG – 7V )
VIN(MAX) − 5V − VREG
RSS(MAX) =
287k • ( 43V – 7V )
= 51.1k
250V – 5V – 43V
In this case the RSS value of 52.3k calculated in Step 1
is too large.
Step 4: Iterate Smaller RSS
Using 51.1k (RSS(MAX)) as the next guess for RSS, we can
now calculate RIN and RSS(MAX):
RIN =
18V – 4.9V − 0.58V − 5V − (72µA • 51.1k)
13µA
RIN = 294k
RSS(MAX)
294k • ( 43V – 7V )
= 52.3k
250V – 5V − 43V
In this case the RSS value of 51.1k is less than RSS(MAX)
and the solution is acceptable.
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Step 5: Determine CG, C1(MAX), Check RSS
The gate capacitor (CG) determines the gate slew rate and
therefore the slew rate of the OUT pin since the output
voltage follows the GATE pin. The voltage at the GATE pin
rises with a slope equal to 7.5µA/CG at startup and 20µA/CG
when the charge pump is on. Limiting this slope will limit
the inrush current charging the load capacitance where:
IINRUSH =
CLOAD
•IG
CG
C1 is used as a bypass capacitor for the circuitry between
the OUT and VSS pins. C1 also stabilizes the shunt regulator that clamps the voltage between these pins where the
minimum value for regulator stability is 0.22µF. An even
greater 0.47µF value is desired for C1 to protect the OUT
to VSS circuitry from transients on the OUT pin.
The startup into an overvoltage creates an upper boundary on the value of C1. The value of CG, RSS and RVIN
determines a maximum C1 that will reach UVLO1 and
power the regulation amplifier before the OUT pin voltage
exceeds the overvoltage threshold. If our desired value
for C1 (0.47µF) exceeds the maximum allowed C1 then
a smaller RSS must be used to iterate a new solution for
C1(MAX). We start with calculating VSS(MATCH):
RSS
• ( VIN – VZ(VDD) )
RSS +R VIN
If we use the worst-case 1% maximum value for RSS
(51.6k) and minimum value for RVIN (291k):
VSS(MATCH) = 35.8V
C1(MAX) =
–CG • (RSS +RIN ) ( VREG – VSS(MATCH) )


2 • VUVLO1
IG • RSS • RIN •In1–

 VREG – VSS(MATCH) 
Use the worst-case maximum gate current of 11µA instead
of the typical 7.5µA and the worst-case minimum UVLO1
18
C1(MAX ) =
–10nF • (51.6k + 291k ) ( 43V – 35.8V )

2 • 2.75V 
11µA • 51.6k • 291k •In1–
 43V − 35.8V 
or
C1(MAX) = 0.1µF
This limit on C1 does not meet the shunt regulator stability
requirements (C1 > 0.22µF).
In this example we choose CG to be 10nF which limits the
inrush current to be 660mA for a 330µF CLOAD.
VSS(MATCH) =
threshold, 2.75V:
If we desire a larger value of C1 then a lower size of RSS
is required. A lower value for RSS is 48.7k, which calls
out an RIN value of 309k and a max C1 value of 0.27µF.
The next lower value of 46.4k with RVIN of 324k, results
in the worst-case maximum C1 value of 0.49µF. A larger
C1 increases circuit immunity to transients in exchange
for slightly higher current. Therefore, a selection of components that allow a 0.47µF C1 is recommended.
The lowered RSS value of 46.4k now considers the tolerances of all the components that set the C1 ramp rate to
guarantee it charges to the 2.55V UVLO1 threshold before
the OUT voltage exceeds the overvoltage threshold.
Step 6: Determine RFB1, RFB2
The feedback resistors, RFB1 and RFB2, are chosen to
regulate the overvoltage at 43V. One way to quickly choose
these resistors is to assign 100µA or 1.2V across a 12.4k
RFB1. RFB2 would need to drop the remainder of the regulated voltage. Dividing this remainder by 100µA yields the
value for RFB2. In this example RFB2 drops 41.8V. When
divided by 100µA it results in a 422k value.
Step 7: Determine CT, R1
During an overvoltage the power dissipated in the MOSFET
is dependent on the load current and the difference between the supply and regulated voltages. It is necessary
to keep the device power in a safe range. In the power
MOSFET data sheets there is a maximum safe operating
curve displaying current versus drain to source voltage
for a fixed pulsed time. Other pulsed time data from DC
to 10µs are plotted on the one graph. The different lines
of operation generally follow a constant power squared
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LTC4366
APPLICATIONS INFORMATION
times time or P2t. Knowing the power we then adjust
the time using the timer capacitor to limit the P2t during
overvoltage. In this example the MOSFET data sheet has
a 6400W2s P2t for a 10ms single pulse.
In order to limit the SD pin current (10mA max) a collector
resistor, R1, in series with Q1 is required. The maximum
value for this resistor is around 5M. This requirement occurs when the pull-down is required to sink 1.6µA from
SD and VDD is clamped at 12V. High valued resistors are
susceptible to leakage currents so we chose a 470k resistor
for R1. Resistor R2 provides ESD protection for Q1’s base.
In this application 250V minus 43V is applied across the
MOSFET at 3A. If the power is applied for less than 16.5ms
then MOSFET P2t limit is not exceeded:
The gate resistor RG limits the parasitic trace capacitance
on M1’s gate node that could lead to parasitic MOSFET
self-oscillation. The recommended value for RG is 10Ω.
P = (250V – 43V) • 3A = 621W
P2t = (621W)2 • 16.5ms = 6363W2s
Prior to the moment when the output is regulated at 43V,
the output is ramping from 28V to 43V. This ramp time is
based on the 20µA gate current charging the 10nF capacitor. Using the equation for ramp time:
∆t =
High Voltage Application
In Figure 7 the circuit accepts 110V AC (rectified to 160V)
and protects the load from accidental connection to 220V
AC by limiting the output to less than 200V. The circuit
has a 100V to 800V VIN operating range where the FET
breakdown voltage limits the maximum input voltage. The
C1 is set to 0.47µF to provide a bypass for the charge pump
that is large enough to provide good noise immunity from
outside voltage transients. The timer capacitor is sized to
give a 1ms overvoltage regulation time that keeps the P2t
below the 640W2s specified for this MOSFET.
CG • ∆V 10nF • 15V
=
= 7.5ms
IG
20µA
To be safe we set the overvoltage time to 10ms. We set the
regulation time to be 2.5ms (the remainder of the 10ms
overvoltage time minus the ramp time). In this example
it is assumed the 250V overvoltage is a constant DC voltage for 10ms. This duration exceeds Mil-Std-1275 which
specifies a 70µs surge to 250V that decays in 1.6ms. Using the following equation (based on charging with 9µA)
to set the CT:
CT = IT •
∆t
2.5ms
= 9µA •
≈ 8.2nF
∆V
2.8V
M1
IXTH12N100L
VIN
160V (RECTIFIED 110V AC)
100V TO 800V
VOUT
0.5A
(200V CLAMP)
CG
2nF
RG
10Ω
RIN
4.64M
R1
470k
SD
R2
100k
Q1
BF722
VDD
C1
0.47µF
GATE
LTC4366-2
SD
TIMER
VSS
OUT
FB
BASE
RFB1
12.4k
RFB2
2M
CT
3.3nF
DANGER! Lethal Voltages Present
RSS
412k
436612 F07
Figure 7. Rectified 110V AC Supply Protected from 220V AC
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19
LTC4366
APPLICATIONS INFORMATION
28V Vehicle Application
The circuit in Figure 8 adds reverse voltage protection to
the standard 28V application shown in Figure 6. There are
three modes to this circuit: pass FET On when the input
is 18V to 41V, clamping the output to 43V when more
than 43V appears at the input and finally reverse voltage
protection when up to –250V DC is present at the input.
The reverse voltage protection consists of the circuitry
inside the dotted box in Figure 8. When a positive voltage
is first applied to the input, D3 and the forward biased
base-collector junction of Q2 allow the gate of M2 to
follow the input voltage minus a two diode drop. During
this condition the body diode of M2 is used to transmit
power to the LTC4366. Once the LTC4366 is powered up
it fully enhances the gate of M1 and M2 (via D1). The M1
and M2 pass FETs then provide a low impedance path to
the load. In an overvoltage condition, D1 blocks excessive
positive voltage from the input supply passing to the GATE
pin of the LTC4366. D4 eliminates current flow through
R6 when the input is positive while D3 prevents emitter
base breakdown of Q2 when the input is powering up.
VIN
18V TO 41V
(±250V DC)
REVERSE VOLTAGE
PROTECTION
During negative input voltages Q2 turns on when current
from R6 (via D4) develops a forward diode drop on R5.
Q2 then holds the gate of M2 at the input voltage which
turns M2 off. This blocks negative input voltages from
reaching M1 and the load. D2 prevents damage to the
LTC4366’s GATE pin by clamping it at ground when the
M2’s gate is negative.
Low Voltage Application
The circuit on the last page (Surge Protected Automotive
Supply) starts up with minimum input voltage of 9V. In
order to successfully start up at 9V and clamp the output
voltage at 18V for input voltages up to 100V the value of
RSS has to be small (1.91k). The FET used in this case has
a 3V threshold to ease the start-up requirements. The timer
capacitor is sized to give a 2.5ms overvoltage regulation
time that keeps the P2t below the 420W2s specified for
this MOSFET.
M2
FDB33N25
M1
FQA62N25C
Q2
MMBT3904
D3
BAV3004W
D4
BAV3004W
R5
470k
R4
270k
D1
BAV3004W
CG
10nF
RG
10Ω
RIN
324k
VOUT
1.5A
(43V CLAMP)
C1
0.47µF
D2
BAV3004W
R1
470k
R6
270k
SD
R2
100k
Q1
MMBT3904
VDD
GATE
SD
LTC4366-2
TIMER
VSS
OUT
FB
BASE
RFB1
12.4k
RFB2
422k
CT
8.2nF
RSS
46.4k
436612 F08
Figure 8. 28V Vehicle Application with Reverse Voltage Protection
20
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LTC4366
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
TS8 Package
8-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1637 Rev A)
0.40
MAX
2.90 BSC
(NOTE 4)
0.65
REF
1.22 REF
1.4 MIN
3.85 MAX 2.62 REF
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.22 – 0.36
8 PLCS (NOTE 3)
0.65 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.95 BSC
TS8 TSOT-23 0710 REV A
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
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21
LTC4366
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DDB Package
8-Lead Plastic DFN (3mm × 2mm)
(Reference LTC DWG # 05-08-1702 Rev B)
0.61 ±0.05
(2 SIDES)
0.70 ±0.05
2.55 ±0.05
1.15 ±0.05
PACKAGE
OUTLINE
0.25 ±0.05
0.50 BSC
2.20 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
5
R = 0.05
TYP
0.40 ±0.10
8
2.00 ±0.10
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.56 ±0.05
(2 SIDES)
0.200 REF
0.75 ±0.05
0 – 0.05
4
0.25 ±0.05
1
PIN 1
R = 0.20 OR
0.25 × 45°
CHAMFER
(DDB8) DFN 0905 REV B
0.50 BSC
2.15 ±0.05
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
22
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LTC4366
REVISION HISTORY
REV
DATE
DESCRIPTION
A
1/12
Added Patents Pending statement
PAGE NUMBER
1
Revised Figure 4 in Applications Information section
11
B
2/12
Removed reference to overcurrent faults under MOSFET Selection
13
Fixed orientation of M2 in Figure 8
18
C
8/12
Updated Shutdown current from <20µA to <14µA
1
Changed MOSFET part number and Gate Capacitor value used in the Typical Application
Added MP-grade order information and specifications
Added negative sign to graphs G12 x-axis and G18, G21 y-axis
Changed MOSFET part number in Figure 1 and Figure 6
D
E
8/13
8/15
1
2, 3, 4, 5
6, 7
11, 16
Added section GATE Capacitor, CG
15
Changed ILOAD current from 5A to 3A in Design Example
16
Updated C1(MAX) values in Step 5 calculations to 0.27µF and worst case 0.49µF
18
Updated calculated values in Step 7, added supporting text
19
Changed MOSFET part number and GATE capacitor used in Figure 7
19
Simplified Diagram: Corrected amplifier’s input polarity in Regulate Diagram
9
Functional Diagram: Added switch in series with TIMER pull-down current
9
Clarified “Ambient” on Operating Temperature Range; raised TJMAX to 150°C
2
TIMER Pin Function: Changed 278ms/µF to 311ms/µF
Figures 1, 6, 8: Changed CT to 8.2nF from 10nF
In CT equation, changed constant to 3.2 from 3.5; updated CT calculation
8
11, 16, 20
12, 19
436612fe
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LTC4366
23
LTC4366
TYPICAL APPLICATION
Surge Protected Automotive 12V Supply
M1
HUF76639S3S
VIN
12V
(9V TO 100V)
VOUT
4A
(18V CLAMP)
CG
2nF
RG
10Ω
RIN
29.4k
R1
470k
SD
R2
100k
Q1
MMBT3904
C1
0.47µF
VDD
GATE
SD
OUT
FB
LTC4366-2
TIMER
VSS
BASE
RFB1
12.4k
RFB2
169k
CT
3.3nF
RSS
1.91k
436612 TA02
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1696
Overvoltage Protection Controller
ThinSOT™ Package, 2.7V to 28V
LTC2909
Triple/Dual Inputs UV/OV Negative Monitor
Pin Selectable Input Polarity Allows Negative and OV Monitoring
LTC2912/LTC2913
Single/Dual UV/OV Voltage Monitor
Ads UV and OV Trip Values, ±1.5% Threshold Accuracy
LTC2914
Quad UV/OV Monitor
For Positive and Negative Supplies
LTC3827/LTC3827-1 Low IQ, Dual, Synchronous Controller
4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, 80µA Quiescent Current
LTC3835/LTC3835-1 Low IQ, Synchronous Step-Down Controller
LT3845
Low IQ, Synchronous Step-Down Controller
Single Channel LTC3827/LTC3827-1
4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, 120µA Quiescent Current
LTC3850
Dual, 550kHz, 2-Phase Synchronous Step-Down Dual 180° Phased Controllers, VIN 4V to 24V, 97% Duty Cycle, 4mm × 4mm
Controller
QFN-28, SSOP-28 Packages
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Low IQ, Dual 2-Phase, Synchronous Step-Down
Controller
4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V, 50µA Quiescent Current
LT4256
Positive 48V Hot Swap Controller with
Open-Circuit Detect
Foldback Current Limiting, Open-Circuit and Overcurrent Fault Output, Up to 80V
Supply
LTC4260
Positive High Voltage Hot Swap Controller with
8-Bit ADC and I2C
Wide Operating Range 8.5V to 80V
LTC4352
Ideal MOSFET ORing Diode
External N-Channel MOSFETs Replace ORing Diodes, 0V to 18V
LTC4354
Negative Voltage Diode-OR Controller
Controls Two N-Channel MOSFETs, 1.2µs Turn-Off, –80V Operation
LTC4355
Positive Voltage Diode-OR Controller
Controls Two N-Channel MOSFETs, 0.4µs Turn-Off, 80V Operation
LT4363
High Voltage Surge Stopper
100V Overvoltage and Overcurrent Protection, Latch-Off and Auto-Retry Options
LTC4365
Window Passer – OV, UV and Reverse Supply
Protection Controller
2.5V to 34V Operation, Protects 60V to –40V
24 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LTC4366
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LTC4366
436612fe
LT 0815 REV E • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2011