TI OPA861ID Wide bandwidth operational transconductance amplifier (ota) Datasheet

 OPA861
SBOS338 – AUGUST 2005
Wide Bandwidth
OPERATIONAL TRANSCONDUCTANCE
AMPLIFIER (OTA)
FEATURES
DESCRIPTION
•
•
•
•
•
The OPA861 is a versatile monolithic component
designed for wide-bandwidth systems, including high
performance video, RF and IF circuitry. The OPA861
is a wideband, bipolar operational transconductance
amplifier (OTA).
Wide Bandwidth (80MHz, Open-Loop, G = +5)
High Slew Rate (900V/µs)
High Transconductance (95mA/V)
External IQ-Control
Low Quiescent Current (5.4mA)
APPLICATIONS
•
•
•
•
•
•
•
Video/Broadcast Equipment
Communications Equipment
High-Speed Data Acquisition
Wideband LED Drivers
Control Loop Amplifiers
Wideband Active Filters
Line Drivers
The OTA or voltage-controlled current source can be
viewed as an ideal transistor. Like a transistor, it has
three terminals—a high impedance input (base), a
low-impedance input/output (emitter), and the current
output (collector). The OPA861, however, is
self-biased and bipolar. The output collector current is
zero for a zero base-emitter voltage. AC inputs
centered about zero produce an output current, which
is bipolar and centered about zero. The
transconductance of the OPA861 can be adjusted
with an external resistor, allowing bandwidth, quiescent current, and gain trade-offs to be optimized.
Used as a basic building block, the OPA861 simplifies the design of AGC amplifiers, LED driver
circuits for fiber optic transmission, integrators for fast
pulses, fast control loop amplifiers and control amplifiers for capacitive sensors and active filters. The
OPA861 is available in SO-8 and SOT23-6 surface-mount packages.
0
−10
R
C1
R
V IN
V OUT
C2
Gain (dB)
−20
−30
10MHz
Low−Pass Filter
−40
20kHz
Low−Pass Filter
−50
−60
−70
−80
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Low−Pass Negative Impedance Converter (NIC) Filter
Frequency Response of 20kHz and 10MHz
Low−Pass NIC Filters
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005, Texas Instruments Incorporated
OPA861
www.ti.com
SBOS338 – AUGUST 2005
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated
circuits be handled with appropriate precautions. Failure to observe proper handling and installation
procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision
integrated circuits may be more susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
PRODUCT
PACKAGE
PACKAGE
DESIGNATOR
OPA861
SO-8
D
–45°C to +85°C
OPA861
OPA861
SOT23-6
DBV
–45°C to +85°C
N5R
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
OPA861ID
Rails, 75
OPA861IDR
Tape and Reel, 2500
OPA861IDBVT
Tape and Reel, 250
OPA861IDBVR
Tape and Reel, 3000
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
±6.5VDC
Power Supply
Internal Power Dissipation
See Thermal Information
±1.2V
Differential Input Voltage
±VS
Input Common-Mode Voltage Range
Storage Temperature Range: D
–40°C to +125°C
Lead Temperature (soldering, 10s)
+260°C
Junction Temperature (TJ)
+150°C
ESD Rating:
(1)
(2)
Human Body Model (HBM) (2)
1500V
Charge Device Model (CDM)
1000V
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operations of the device at these and any other conditions
beyond those specified is not supported.
Pin 2 for the SO-8 package > 500V HBM. Pin 4 for the SOT23-6 package > 500V HBM.
PIN CONFIGURATION
Top View
I Q Adjust
1
8
C
E
2
7
V+ = +5V
B
3
6
NC
V− = −5V
4
5
NC
SO−8
2
I Q Adjust
1
6
+VS
−VS
2
5
C
B
3
4
E
SOT23−6
OPA861
www.ti.com
SBOS338 – AUGUST 2005
ELECTRICAL CHARACTERISTICS: VS = ±5V
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
G = +5, VO = 200mVPP,
RL = 500Ω
80
77
75
74
MHz
min
B
G = +5, VO = 1VPP
80
MHz
typ
C
G = +5, VO = 5VPP
80
MHz
typ
C
G = +5, VO = 5V Step
900
V/µs
min
B
VO = 1V Step
4.4
ns
typ
C
OTA — Open-Loop (see Figure 30)
AC PERFORMANCE
Bandwidth
Slew Rate
Rise Time and Fall Time
Harmonic Distortion
860
850
840
G = +5, VO = 2VPP, 5MHz
2nd-Harmonic
RL = 500Ω
–68
–55
–54
–53
dB
max
B
3rd-Harmonic
RL = 500Ω
–57
–52
–51
–49
dB
max
B
Base Input Voltage Noise
f > 100kHz
2.4
3.0
3.3
3.4
nV/√Hz
max
B
Base Input Current Noise
f > 100kHz
1.7
2.4
2.45
2.5
pA/√Hz
max
B
Emitter Input Current Noise
f > 100kHz
5.2
15.3
16.6
17.5
pA/√Hz
max
B
Minimum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
95
80
77
75
mA/V
min
A
Maximum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
95
150
155
160
mA/V
max
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±12
±15
±20
mV
max
A
±67
±120
µV/°C
max
B
±6
±6.6
µA
max
A
±20
±25
nA/°C
max
B
±125
±140
µA
max
A
±500
±600
nA/°C
max
B
±30
±38
µA
max
A
±250
±300
nA/°C
max
B
±3.6
±3.6
OTA DC PERFORMANCE (4) (see Figure 30)
B-Input Offset Voltage
Average B-Input Offset Voltage Drift
B-Input Bias Current
Average B-Input Bias Current Drift
E-Input Bias Current
Average E-Input Bias Current Drift
C-Output Bias Current
Average C-Output Bias Current Drift
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
±1
±5
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
±30
±100
VB = 0V, VC = 0V
VB = 0V, VC = 0V
±5
±18
VB = 0V, VC = 0V
OTA INPUT (see Figure 30)
±4.2
B-Input Voltage Range
B-Input Impedance
±3.7
455 || 2.1
V
min
B
kΩ || pF
typ
C
Min E-Input Resistance
10.5
12.5
13.0
13.3
Ω
max
B
Max E-Input Resistance
10.5
6.7
6.5
6.3
Ω
min
B
IE = ±1mA
±4.2
±3.7
±3.6
±3.6
V
min
A
VE = 0
±15
±10
±9
±9
mA
min
A
IC = ±1mA
±4.7
±4.0
±3.9
±3.9
V
min
A
VC = 0
±15
±10
±9
±9
mA
min
A
kΩ || pF
typ
C
OTA OUTPUT
E-Output Voltage Compliance
E-Output Current, Sinking/Sourcing
C-Output Voltage Compliance
C-Output Current, Sinking/Sourcing
C-Output Impedance
(1)
(2)
(3)
(4)
54 || 2
Test levels: (A) 100% tested at 25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for 25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient + 7°C at high temperature limit for over
temperature specifications.
Current is considered positive out of node.
3
OPA861
www.ti.com
SBOS338 – AUGUST 2005
ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
Maximum Operating Voltage
±6.3
±6.3
±6.3
Minimum Operating Voltage
±2.0
±2.0
PARAMETER
CONDITIONS
+25°C
MIN/
MAX
TEST
LEVEL (1)
V
typ
C
V
max
A
±2.0
V
min
B
UNITS
POWER SUPPLY
±5
Specified Operating Voltage
Maximum Quiescent Current
RADJ = 250Ω
5.4
5.9
7.0
7.4
mA
max
A
Minimum Quiescent Current
RADJ = 250Ω
5.4
4.9
4.3
3.4
mA
min
A
∆IC/∆VS
±20
±50
±60
±65
µA/V
max
A
–40 to +85
°C
typ
C
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
Thermal Resistance θJA
D
SO-8
Junction-to-Ambient
125
°C/W
typ
C
DBV
SOT23-6
Junction-to-Ambient
150
°C/W
typ
C
4
OPA861
www.ti.com
SBOS338 – AUGUST 2005
ELECTRICAL CHARACTERISTICS: VS = +5V
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
Bandwidth
G = +5, VO = 200mVPP,
RL = 500Ω
73
72
72
70
MHz
min
B
G = +5, VO = 1VPP
73
MHz
typ
C
Slew Rate
G = +5, VO = 2.5V Step
410
395
390
390
V/µs
min
B
VO = 1V Step
4.4
ns
typ
C
PARAMETER
OTA—Open-Loop (see Figure 30)
AC PERFORMANCE
Rise Time and Fall Time
Harmonic Distortion
G = +5, VO = 2VPP, 5MHz
2nd-Harmonic
RL = 500Ω
–67
–55
–54
–54
dB
max
B
3rd-Harmonic
RL = 500Ω
–57
–50
–49
–48
dB
max
B
Base Input Voltage Noise
f > 100kHz
2.4
3.0
3.3
3.4
nV/√Hz
max
B
Base Input Current Noise
f > 100kHz
1.7
2.4
2.45
2.5
pA/√Hz
max
B
Emitter Input Current Noise
f > 100kHz
5.2
15.3
16.6
17.5
pA/√Hz
max
B
Minimum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
85
70
67
65
mA/V
min
A
Maximum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
85
140
145
150
mA/V
max
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±12
±15
±20
mV
max
A
±67
±120
µV/°C
max
B
±6
±6.6
µA
max
A
±20
±25
nA/°C
max
B
±125
±140
µA
max
A
±500
±600
nA/°C
max
B
µA
typ
C
B
OTA DC PERFORMANCE (4) (see Figure 30)
B-Input Offset Voltage
Average B-Input Offset Voltage Drift
B-Input Bias Current
Average B-Input Bias Current Drift
E-Input Bias Current
Average E-Input Bias Current Drift
C-Output Bias Current
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
±1
±5
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
±30
±100
VB = 0V, VC = 0V
VB = 0V, VC = 0V
±15
OTA INPUT (see Figure 30)
Most Positive B-Input Voltage
4.2
3.7
3.6
3.6
V
min
Least Positive B-Input Voltage
0.8
1.3
1.4
1.4
V
max
B
kΩ || pF
typ
C
B-Input Impedance
455 || 2.1
Min E-Input Resistance
11.8
14.4
14.9
15.4
Ω
max
B
Max E-Input Resistance
11.8
7.1
6.9
6.7
Ω
min
B
OTA OUTPUT
Maximum E-Output Voltage Compliance
IE = ±1mA
4.2
3.7
3.6
3.6
V
min
A
Minimum E-Output Voltage Compliance
IE = ±1mA
0.8
1.3
1.4
1.4
V
max
A
VE = 0
±8
±7
±6.5
±6.5
mA
min
A
Maximum C-Output Voltage Compliance
IC = ±1mA
4.7
4.0
3.9
3.9
V
min
A
Minimum C-Output Voltage Compliance
IC = ±1mA
0.3
1.0
1.1
1.1
V
max
A
VC = 0
±8
±7
±6.5
±6.5
mA
min
A
kΩ || pF
typ
C
E-Output Current, Sinking/Sourcing
C-Output Current, Sinking/Sourcing
C-Output Impedance
(1)
(2)
(3)
(4)
54 || 2
Test levels: (A) 100% tested at 25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for 25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient + 3°C at high temperature limit for over
temperature specifications.
Current is considered positive out of node.
5
OPA861
www.ti.com
SBOS338 – AUGUST 2005
ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
Maximum Operating Voltage
12.6
12.6
12.6
Minimum Operating Voltage
4
4
PARAMETER
CONDITIONS
+25°C
MIN/
MAX
TEST
LEVEL (1)
V
typ
C
V
max
A
4
V
min
B
UNITS
POWER SUPPLY
Specified Operating Voltage
5
Maximum Quiescent Current
RADJ = 250Ω
4.7
5.2
6.0
6.4
mA
max
A
Minimum Quiescent Current
RADJ = 250Ω
4.7
4.2
3.4
3.0
mA
min
A
∆IC/∆VS
±20
±50
±60
±65
µA/V
max
A
–40 to +85
°C
typ
C
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
Thermal Resistance θJA
D
SO-8
Junction-to-Ambient
125
°C/W
typ
C
DBV
SOT23-6
Junction-to-Ambient
150
°C/W
typ
C
6
OPA861
www.ti.com
SBOS338 – AUGUST 2005
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT
1000
150
IO UT
VIN = 100mVPP
RL = 50Ω
VIN = 10mVPP
50Ω
Transconductance (mA/V)
Transconductance (mA/V)
VI N
50Ω
I Q = 5.4mA (102mA/V)
IQ = 6.5mA (117mA/V)
100
IQ = 1.9mA (51mA/V)
120
90
I OUT
60
VIN
50Ω
30
50Ω
IQ = 3.4mA (79mA/V)
10
0
1M
10M
100M
1G
6
140
6
IQ = 6.5mA
IQ = 5.4mA
100
IQ = 3.4mA
80
60
IQ = 1.9mA
40
−30
−20
−10
0
IQ = 5.4mA
4
2
IQ = 3.4mA
0
IQ = 1.9mA
−2
IOUT
VIN
−4
50Ω
50Ω
20
30
−70 −60 −50 −40 −30 −20 −10
40
20
30
Figure 3.
Figure 4.
40
50
60
70
OTA LARGE-SIGNAL PULSE RESPONSE
3
0.2
0
G = +5V/V
RL = 500Ω
VIN = 0.25VPP
f IN = 20MHz
See Figure 48
Time (10ns/div)
Figure 5.
Output Voltage (V)
2
0.4
−0.8
10
OTA Input Voltage (mV)
0.6
−0.6
0
Input Voltage (mV)
OTA SMALL-SIGNAL PULSE RESPONSE
−0.4
13
−8
10
0.8
−0.2
12
IQ = 6.5mA
−6
Small signal around input voltage.
−40
11
OTA TRANSFER CHARACTERISTICS
8
OTA Output Current (mA)
Transconductance (mA/V)
10
Figure 2.
20
Output Voltage (V)
9
Figure 1.
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
0
8
Quiescent Current (mA)
160
120
7
Frequency (Hz)
1
0
−1
−2
G = +5V/V
RL = 500Ω
VIN = 1VPP
fIN = 20MHz
See Figure 48
−3
Time (10ns/div)
Figure 6.
7
OPA861
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SBOS338 – AUGUST 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
490
110
OTA C−Output Resistance (kΩ )
OTA B−Input Resistance (kΩ )
B-INPUT RESISTANCE vs QUIESCENT CURRENT
500
480
470
460
450
440
100
430
80
70
60
50
40
7
8
9
10
11
12
13
7
10
11
Figure 7.
Figure 8.
12
13
INPUT VOLTAGE AND CURRENT NOISE DENSITY
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
100
50
40
30
20
10
0
E−Input Current Noise (5.2pA/√Hz)
10
B−Input Voltage Noise (2.4nV/√Hz)
B−Input Current Noise (1.65pA/√Hz)
1
7
8
9
10
11
12
100
13
Quiescent Current (mA)
1k
10k
100k
1M
10M
Frequency (Hz)
Figure 9.
Figure 10.
QUIESCENT CURRENT vs RADJ
1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY
vs QUIESCENT CURRENT ADJUST RESISTOR
16
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
8
7
Quiescent Current (mA)
9
Quiescent Current (mA)
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
6
5
4
3
2
1
E−Input Current Noise (pA/√Hz)
14
12
10
8
B−Input Voltage Noise (nV/√Hz)
6
B−Input Current Noise (pA/√Hz)
4
2
0
0
0.1
1
10
100
1k
10k
Quiescent Current Adjust Resistor (Ω)
Figure 11.
8
8
Quiescent Current (mA)
60
OTA E−Output Resistance (Ω)
90
100k
0
200 400 600 800 1000 1200 1400 1600 1800 2000
Quiescent Current Adjust Resistor (Ω )
Figure 12.
OPA861
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SBOS338 – AUGUST 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
QUIESCENT CURRENT vs TEMPERATURE
3
9
4
2
8
1
B−Input Offset Voltage
0
0
−2
−4
−6
−40
−1
B−Input Bias Current
−20
0
20
40
60
80
100
Input Bias Current (µA)
2
Quiescent Current (mA)
6
7
6
5
−2
4
−3
3
−40
120
−20
0
Ambient Temperature ( C)
20
40
60
80
100
120
Ambient Temperature ( C)
Figure 13.
Figure 14.
C-OUTPUT BIAS CURRENT vs TEMPERATURE
40
OTA C−Output Bias Current (µA)
Offset Voltage (mV)
B-INPUT OFFSET VOLTAGE AND BIAS CURRENT
vs TEMPERATURE
Five Representative Units
30
20
10
0
−10
−20
−30
−40
−40
−20
0
20
40
60
80
100
120
Ambient Temperature ( C)
Figure 15.
9
OPA861
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SBOS338 – AUGUST 2005
TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
100
OTA TRANSCONDUCTANCE vs IQ
150
OTA Transconductance (mA/V)
IQ = 5.8mA
(93mA/V)
Transconductance (mA/V)
IQ = 4.7mA (80mA/V)
IQ = 3.1mA (60mA/V)
IQ = 1.65mA (37mA/V)
I OU T
VIN
50Ω
50Ω
90
60
30
VIN = 100mVPP
10
100
1k
0
1
2
3
4
5
Frequency (Hz)
Quiescent Current (mA)
Figure 16.
Figure 17.
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
6
7
OTA TRANSFER CHARACTERISTICS
120
6
IQ = 5.8mA
IQ = 5.8mA
IQ = 4.7mA
100
80
OTA Output Current (mA)
Transconductance (mA/V)
50Ω
50Ω
0
1
IQ = 3.1mA
60
IQ = 1.65mA
40
20
Small−signal around input voltage.
0
−30
−20
−10
0
10
20
4
IQ = 3.1mA
2
IQ = 4.7mA
I OUT
−2
V IN
−4
−6
−50 −40 −30 −20 −10
30
20
Figure 18.
Figure 19.
1.0
−0.05
G = +5V/V
R L = 500Ω
VIN = 0.07VPP
f IN = 20MHz
Time (10ns/div)
Figure 20.
Output Voltage (V)
1.5
0.10
0
30
40
OTA LARGE-SIGNAL PULSE RESPONSE
0.15
−0.20
10
OTA Input Voltage (mV)
2.0
−0.15
0
Input Voltage (mV)
0.05
50 Ω
50 Ω
0.20
−0.10
IQ = 1.65mA
0
OTA SMALL-SIGNAL PULSE RESPONSE
Output Voltage (V)
VIN
120
RL = 50Ω
VIN = 10mVPP
10
10
IOUT
0.5
0
−0.5
−1.0
−1.5
−2.0
G = +5V/V
R L = 500Ω
VIN = 0.7VPP
fIN = 20MHz
Time (10ns/div)
Figure 21.
50
OPA861
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SBOS338 – AUGUST 2005
TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
490
110
OTA C−Output Resistance (kΩ )
OTA B−Input Resistance (kΩ )
B-INPUT RESISTANCE vs QUIESCENT CURRENT
500
480
470
460
450
440
430
100
90
80
70
60
50
40
420
0
1
2
3
4
5
6
0
7
1
3
4
5
Quiescent Current (mA)
Figure 22.
Figure 23.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
7
50
6
40
30
20
10
6
7
QUIESCENT CURRENT vs RADJ
60
Quiescent Current (mA)
OTA E−Output Resistance (Ω )
2
Quiescent Current (mA)
5
4
3
2
1
0
0
0
1
2
3
4
Quiescent Current (mA)
Figure 24.
5
6
7
0.1
1
10
100
1k
10k
100k
Quiescent Current Adjust Resistor (Ω)
Figure 25.
11
OPA861
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APPLICATION INFORMATION
The
OPA861
is
a
versatile
monolithic
transconductance
amplifier
designed
for
wide-bandwidth systems, including high-performance
video, RF, and IF circuitry. The operation of the
OPA861 is discussed in the OTA (Operational
Transconductance Amplifier) section of this data
sheet. Over the years and depending on the writer,
the OTA section of an op amp has been referred to
as a Diamond Transistor, Voltage-Controlled Current
source, Transconductor, Macro Transistor, or positive
second-generation current conveyor (CCII+). Corresponding symbols for these terms are shown in
Figure 26.
C
3
VIN1
B
1
IOUT
2
VIN2
E
Diamond
Transistor
Transconductor
(used here)
Voltage−Controlled
Current Source
C
VIN1
VIN2
Z
CCII+
IOUT
B
The symbol for the OTA section is similar to a
transistor (see Figure 26). Applications circuits for the
OTA look and operate much like transistor circuits—the transistor is also a voltage-controlled current source. Not only does this characteristic simplify
the understanding of application circuits, it aids the
circuit optimization process as well. Many of the
same intuitive techniques used with transistor designs
apply to OTA circuits. The three terminals of the OTA
are labeled B, E, and C. This labeling calls attention
to its similarity to a transistor, yet draws distinction for
clarity. While the OTA is similar to a transistor, one
essential difference is the sense of the C-output
current: it flows out the C terminal for positive B-to-E
input voltage and in the C terminal for negative B-to-E
input voltage. The OTA offers many advantages over
a discrete transistor. The OTA is self-biased, simplifying the design process and reducing component
count. In addition, the OTA is far more linear than a
transistor. Transconductance of the OTA is constant
over a wide range of collector currents—this feature
implies a fundamental improvement of linearity.
BASIC CONNECTIONS
E
Current Conveyor II+
TRANSCONDUCTANCE (OTA) SECTION—AN
OVERVIEW
Macro Transistor
Figure 26. Symbols and Terms
Regardless of its depiction, the OTA section has a
high-input impedance (B-input), a low-input/output
impedance (E-input), and a high-impedance current
source output (C-output).
Figure 27 shows basic connections required for
operation. These connections are not shown in subsequent circuit diagrams. Power-supply bypass capacitors should be located as close as possible to the
device pins. Solid tantalum capacitors are generally
best.
RQ = 250Ω, roughly sets I Q = 5.4mA.
1
RS
(25Ωto 200Ω)
RADJ
250Ω
2
7
+5V(1)
0.1µF
+VS
+
VIN
−5V(1)
RC
8
−VS
3
6
4
5
2.2µF
Solid Tantalum
0.1µF
+
2.2µF
Solid
Tantalum
NOTE: (1) VS = ±6.5V absolute maximum.
Figure 27. Basic Connections
12
OPA861
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QUIESCENT CURRENT CONTROL PIN
The quiescent current of the transconductance
portion of the OPA861 is set with a resistor, RADJ,
connected from pin 1 to –VS. The maximum quiescent current is 6mA. RADJ should be set between 50Ω
and 1kΩ for optimal performance of the OTA section.
This range corresponds to the 5mA quiescent current
for RADJ = 50Ω, and 1mA for RADJ = 1kΩ. If the IQ
adjust pin is connected to the negative supply, the
quiescent current will be set by the 250Ω internal
resistor.
Reducing or increasing the quiescent current for the
OTA section controls the bandwidth and AC behavior
as well as the transconductance. With RADJ = 250Ω,
this sets approximately 5.4mA total quiescent current
at 25°C. It may be appropriate in some applications to
trim this resistor to achieve the desired quiescent
current or AC performance.
Applications circuits generally do not show the
resistor RQ, but it is required for proper operation.
With a fixed RADJ resistor, quiescent current increases with temperature (see Figure 11 in the
Typical Characteristics section). This variation of
current with temperature holds the transconductance,
gm, of the OTA relatively constant with temperature
(another advantage over a transistor).
It is also possible to vary the quiescent current with a
control signal. The control loop in Figure 28 shows
1/2 of a REF200 current source used to develop
100mV on R1. The loop forces 125mV to appear on
R2. Total quiescent current of the OPA861 is approximately 37 × I1, where I1 is the current made to flow
out of pin 1.
V+
OPA861
1/2 REF200
100µA
R1
1.25kΩ
With this control loop, quiescent current will be nearly
constant with temperature. Since this method differs
from the temperature-dependent behavior of the
internal current source, other temperature-dependent
behavior may differ from that shown in the Typical
Characteristics. The circuit of Figure 28 will control
the IQ of the OPA861 somewhat more accurately than
with a fixed external resistor, RQ. Otherwise, there is
no fundamental advantage to using this more complex biasing circuitry. It does, however, demonstrate
the possibility of signal-controlled quiescent current.
This capability may suggest other possibilities such
as AGC, dynamic control of AC behavior, or VCO.
BASIC APPLICATIONS CIRCUITS
Most applications circuits for the OTA section consist
of a few basic types, which are best understood by
analogy to a transistor. Used in voltage-mode, the
OTA section can operate in three basic operating
states—common emitter, common base, and common collector. In the current-mode, the OTA can be
useful for analog computation such as current amplifier, current differentiator, current integrator, and current summer.
Common-E Amplifier or Forward Amplifier
Figure 29 compares the common-emitter configuration for a BJT with the common-E amplifier for the
OTA section. There are several advantages in using
the OTA section in place of a BJT in this configuration. Notably, the OTA does not require any biasing,
and the transconductance gain remains constant over
temperature. The output offset voltage is close to 0,
compared with several volts for the common-emitter
amplifier.
The gain is set in a similar manner as for the BJT
equivalent with Equation 1:
R
G 1 L
RE
gm
(1)
IQ Adjust
1 I1
R2
425Ω
TLV2262
Just as transistor circuits often use emitter degeneration, OTA circuits may also use degeneration. This
option can be used to reduce the effects that offset
voltage and offset current might otherwise have on
the DC operating point of the OTA. The
E-degeneration resistor may be bypassed with a
large capacitor to maintain high AC gain. Other
circumstances may suggest a smaller value capacitor
used to extend or optimize high-frequency performance.
Figure 28. Optional Control Loop for Setting
Quiescent Current
13
OPA861
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The forward amplifier shown in Figure 30 and Figure 31 corresponds to one of the basic circuits used
to
characterize
the
OPA861.
Extended
characterization of this topology appears in the Typical Characteristics section of this datasheet.
V+
RS
RL
VO
VO
VI
Inverting Gain
VOS = Several Volts
RS
R1
160Ω
RE
VI
8
C
3 B
OPA861
E
2
V−
VI
8
C
3 B
VO
OPA861
Figure 30. Forward Amplifier Configuration and
Test Circuit
RL
E
2
RE
G = 5V/V
IQ = 5.4mA
RE
78Ω
(a) Transistor Common−Emitter Amplifier
Transconductance varies over temperature.
100Ω
RC
500Ω
RL1
Noninverting Gain
VOS = 0V
VO
Network
Analyzer
8
3
(b) OTA Common−E Amplifier
Transconductance remains constant over temperature.
Figure 29. Common-Emitter vs Common-E
Amplifier
(2)
A positive voltage at the B-input, pin 3, causes a
positive current to flow out of the C-input, pin 8. This
gives a noninverting gain where the circuit of Figure 29a is inverting. Figure 29b shows an amplifier
connection of the OPA861, the equivalent of a
common-emitter transistor amplifier. Input and output
can be ground-referenced without any biasing. The
amplifier is non-inverting because of the sense of the
output current.
14
RL2
rE
2
VI
The transconductance of the OTA with degeneration
can be calculated by Equation 2:
g m_deg 1 1
gm R E
RIN
50Ω
OPA861
R1
100Ω
RL = RL1 + RL2 || RIN
RE
G
RL
r E g1
m
RE rE
At I Q 5.4mA
G
rE 1
10.5
95mAV
RL
at I Q 5.4mA
R E 10.5
Figure 31. Forward Amplifier Design Equations
OPA861
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Common-C Amplifier
Current-Mode Analog Computations
Figure 32b shows the OPA861 connected as an
E-follower—a voltage buffer. It is interesting to notice
that the larger the RE resistor, the closer to unity gain
the buffer will be. If the OPA861 is to be used as a
buffer, use RE ≥ 500Ω for best results. For the
OPA861 used as a buffer, the gain is given by
Equation 3:
1
G
1
1
1 g R
m
As mentioned earlier, the OPA861 can be used
advantageously for analog computation. Among the
application possibilities are functionality as a current
amplifier, current differentiator, current integrator, current summer, and weighted current summer. Table 1
lists these different uses with the associated transfer
functions.
(3)
E
These functions can easily be combined to form
active filters. Some examples using these current-mode functions are shown later in this document.
V+
V+
G=1
VOS = 0.7V
VO
VI
RL
VO
RE
Noninverting Gain
VOS = Several Volts
V−
(a) Transistor Common−Collector Amplifier
(Emitter Follower)
G
100Ω
VI
RE
OPA861
1g
1
1
V−
mR E
R O g1 R E
m
8
C
3 B
RE
1
(a) Transistor Common−Base Amplifier
G
G=1
VOS = 0V
E
2
VO
100Ω
(b) OTA Common−C Amplifier
(Buffer)
RL
R
L
1
R
R E gm
E
8
C
3 B
OPA861
E
2
VO
Inverting Gain
VOS = 0V
RL
RE
Figure 32. Common-Collector vs Common-C
Amplifier
V−
A low value resistor in series with the B-input is
recommended. This resistor helps isolate trace parasitic from the inputs, reduces any tendency to oscillate, and controls frequency response peaking. Typical resistor values are from 25Ω to 200Ω.
(b) OTA Common−B Amplifier
Figure 33. Common-Base Transistor vs
Common-B OTA
Common-B Amplifier
Figure 33 shows the Common-B amplifier. This configuration produces an inverting gain and a low
impedance input. Equation 4 shows the gain for this
configuration.
RL
R
G
L
1
R
R E gm
E
(4)
This low impedance can be converted to a high
impedance by inserting the buffer amplifier in series.
15
OPA861
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SBOS338 – AUGUST 2005
Table 1. Current-Mode Analog Computation Using the OTA Section
FUNCTIONAL ELEMENT
TRANSFER FUNCTION
Current Amplifier
R
I OUT 1 I IN
R2
IMPLEMENTATION WITH THE OTA SECTION
IOUT
IIN
R1
R2
IOUT
1
I OUT Current Integrator
CR
IIN
I dt
C
IN
R
IOUT
n
I OUT 1 I j
j1
Current Summer
I1
I2
In
I OUT
Rj
I OUT 1 I j j1
R
n
Weighted Current Summer
R
R1
I1
R
Rn
In
OPA861 APPLICATIONS
Control-Loop Amplifier
DC-Restore Circuit
A new type of control loop amplifier for fast and
precise control circuits can be designed with the
OPA861. The circuit of Figure 34 illustrates a series
connection of two voltage control current sources that
have an integral (and at higher frequencies, a proportional) behavior versus frequency. The control
loop amplifiers show an integrator behavior from DC
to the frequency represented by the RC time constant
of the network from the C-output to GND. Above this
frequency, they operate as an amp with constant
gain. The series connection increases the overall gain
to about 110dB and thus minimizes the control loop
deviation. The differential configuration at the inputs
enables one to apply the measured output signal and
the reference voltage to two identical high-impedance
inputs. The output buffer decouples the C-output of
the second OTA in order to insure the AC performance and to drive subsequent output stages.
The OPA861 can be used advantageously with an
operational amplifier, here the OPA656, as a
DC-restore circuit. Figure 35 illustrates this design.
Depending on the collector current of the
transconductance amplifier (OTA) of the OPA861, a
switching function is realized with the diodes D1 and
D2.
16
When the C-output is sourcing current, the capacitor
C1 is being charged. When the C-output is sinking
current, D1 is turned off and D2 is turned on, letting
the voltage across C1 be discharged through R2.
The condition to charge C1 is set by the voltage
difference between VREF and VOUT. For the OTA
C-output to source current, VREF has to be greater
than VOUT. The rate of charge of C1 is set by both R1
and C1. The discharge rate is given by R2 and C1.
OPA861
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SBOS338 – AUGUST 2005
8
6
5
BUF602
VOUT
3
8
180Ω
2
10pF
VREF
10pF
3
2
10Ω
180Ω
33Ω
10Ω
33Ω
6
VIN
Figure 34. Control-Loop Amplifier Using Three OPA861s
C1
100pF
20Ω
JFET−Input, Wideband
VIN
D 1, D2 = 1N4148
RQ = 1kΩ
OPA656
R2
100kΩ
D1
VOUT
20Ω
D2
CCII
8 C
The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1.
R1 and C1 set the DC restoration time constant.
D1 adds a propagation delay to the DC restoration.
R2 and C1 set the decay time constant.
E 2
R1
40.2Ω
B
3
R2
100Ω
VREF
Figure 35. DC Restorer Circuit
Negative Impedance Converter Filter: Low-Pass
Filter
The OPA861 can be used as a negative impedance
converter to realize the low-pass filer shown in
Figure 36.
R
VIN
VOUT
C2
(5)
with:
0 R
C1
The transfer function is shown in Equation 5:
V OUT
1
V IN
1 sC 2R s 2C1C 2R 2
Q
1
C1C 2 R
C1
C2
Figure 36. Low-Pass Negative Impedance
Converter Filter
17
OPA861
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SBOS338 – AUGUST 2005
Differential Line Driver/Receiver
The input impedance is shown in Equation 6:
Z IN 1 R 1 sRC
2sC
1 2sRC
(6)
Figure 37 shows the frequency responses for
low-pass, Butterworth filters set at 20kHz and 10MHz.
For the 20kHz filter, set R to 1KΩ and
C 1 1 C 2 5.6F
2
. For the 10MHz filter, the
parasitic capacitance at the output pin needs to be
taken into consideration. In the example of Figure 37,
the parasitic is 3pF, which gives us the settings of R
= 1.13kΩ, C1 = 10pF, and C2 = 17pF.
The wide bandwidth and high slew rate of the
OPA861 current-mode amplifier make it an ideal line
driver. The circuit in Figure 39 makes use of two
OPA861s to realize a single-ended to differential
conversion. The high-impedance current source output of the OPA861 allows it to drive low-impedance or
capacitive loads without series resistances and
avoids any attenuation that would have otherwise
occured in the resistive network.
The OPA861 used as a differential receiver exhibits
excellent common-mode rejection ratio, as can be
seen in Figure 38.
Common−Mode Rejection Ratio (dBc)
0
−10
Gain (dB)
−20
−30
−40
−50
−60
−70
−80
1k
10k
100k
1M
10M
100M
1G
0
−10
−20
−30
−40
−50
−60
−70
−80
−90
−100
0.001
0.01
Frequency (Hz)
0.1
1
Figure 37. Small-Signal Frequency Response for a
Low-Pass Negative Impedance Converter Filter
50Ω
VIN
10Ω
50Ω
100Ω
50Ω
10Ω
50Ω
Figure 39. Twisted-Pair Differential Driver and Receiver with the OPA861
18
100
Figure 38. Differential Driver Common-Mode
Rejection Ratio for 2VPP Input Signals
To 50Ω Load
50Ω
10
Frequency (MHz)
OPA861
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SBOS338 – AUGUST 2005
ACTIVE FILTERS USING THE OPA861 IN
CURRENT CONVEYOR STRUCTURE
One further example of the versatility of the Diamond
Transistor and Buffer is the construction of
high-frequency (> 10MHz) active filters. Here, the
Current Conveyor structure, shown in Figure 40, is
used with the Diamond Transistor as a Current
Conveyor.
The method of converting RC circuit loops with
operational amplifiers in Current Conveyor structures
is based upon the adjoint network concept. A network
is reversible or reciprocal when the transfer function
does not change even when the input and output
have been exchanged. Most networks, of course, are
nonreciprocal. The networks of Figure 41, perform
interreciprocally when the input and output are
exchanged, while the original network, N, is
exchanged for a new network NA. In this case, the
transfer function remains the same, and NA is the
adjoing network. It is easy to construct an adjoint
network for any given circuit, and these networks are
the base for circuits in Current-Conveyor structure.
Individual elements can be interchanged according to
the list in Figure 42. Voltage sources at the input
become short circuits, and the current flowing there
becomes the output variable. In contrast, the voltage
output becomes the input, which is excitated by a
current source. The following equation describes the
interreciprocal features of the circuit: VOUT/VIN =
IOUT/IIN. Resistances and capacitances remain unchanged. In the final step, the operational amplifier
with infinite input impedance and 0Ω output impedance is transformed into a current amplifier with
0Ω input impedance and infinite output impedance. A
Diamond Transistor with the base at ground comes
quite close to an ideal current amplifier. The
well-known Sallen-Key low-pass filter with positive
feedback, is an example of conversion into Current-Conveyor structure, see Figure 43. The positive
gain of the operational amplifier becomes a negative
second type of Current Conveyor (CCII), as shown in
Figure 40. Both arrangements have identical transfer
functions and the same level of sensitivity to deviations. The most recent implementation of active
filters in a Current-Conveyor structure produced a
second-order Bi-Quad filter. The value of the resistance in the emitter of the Diamond Transistor controls
the filter characteristic. For more information, refer to
application note SBOS047, New Ultra High-Speed
Circuit Techniques with Analog ICs.
VOUT
+1
C
E
CCII− B
C
VIN
IOUT
C
R
R
R
R
IIN
C/2
T(s) =
C/2
VOUT
VIN
=
IOUT
IIN
=
4KQ2/R2C2
s2 + 2/RC[2Q(1 − K) + 1]s + 4KQ2/R2C2
Figure 40. Current Conveyor
19
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SBOS338 – AUGUST 2005
Reciprocal Networks
+
VIN
VOUT
N
I OUT
N
I IN
NA
I IN
−
VOUT
=
I OUT
VIN
IIN
Interreciprocal Networks
+
VIN
VOUT
N
I OUT
−
Figure 41. Networks
Element
VIN
1
Signal
Sources
1
− VOUT +
R
1
Passive
Elements
C
1
1
Controlled
Sources
1
Adjoint
+
V
−
2
IOUT
1
2
1
2
1
2
1
2
IIN
R
C
3
µV
2
2
3
µI
4
2
I
4
Figure 42. Individual Elements in the Current Conveyor
20
OPA861
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SBOS338 – AUGUST 2005
R3
R2
VIN
BUF602
C1
R1
RB1
RB2
R1S
C2
R1M
R2M
RB3
R2S
VOUT
R3S
Figure 43. Universal Active Filter
Transfer Function
The transfer function of the universal active filter of
Figure 43 is shown in Equation 7.
R
1
1M R
3S
1R
2S
R
(7)
1S
Filter Characteristics
Five filter types can be made with this structure:
• For a low-pass filter, set R2 = R3 = ∞,
• For a high-pass filter, set R1 = R2 = ∞,
• For a bandpass filter, set R1 = R3 = ∞,
• For a band rejection filter, set R2 = ∞; R1 = R3,
• For an all-pass filter, set R1 = R15; R2 = R25; and
R3 = R35.
A few designs for a low-pass filter are shown in
Figure 44 and Table 2.
Table 2. Component Values for Filters Shown In
Figure 44
fO
R
RO
CO
1MHz
150
100
2nF
20MHz
150
100
112.5pF
50MHz
150
100
55pF
Gain (dB)
R
s 2C1R 1M R2M sC 1 R1M R1
V OUT
3
2
1
F(p) R2M
R 1M
1
VIN
2
sCR
sC
3
0
50MHz Filter
−3
−6
−9
−12
−15
1MHz Filter
−18
20MHz Filter
−21
−24
−27
−30 For All Filters:
−33 R = R = ∞
2
3
−36
−39 R1 = R15 = R25 = 1/2 R35 = R
−42 R1M = R2M = R0
−45 C1 = C2 = C0
−48
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Figure 44. Butterworth Low-Pass Filter with the
Universal Active Filter
The advantages of building active filters using a
Current Conveyor structure are:
• The increase in output resistance of operational
amplifiers at high frequencies makes it difficult to
construct feedback filter structures (decrease in
stop-band attenuation).
• All filter coefficients are represented by resistances, making it possible to adjust the filter
frequency response without affecting the filter
coefficients.
21
OPA861
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SBOS338 – AUGUST 2005
•
The capacitors which determine the frequency
are located between the ground and the current
source outputs and are thus grounded on one
side. Therefore, all parasitic capacitances can be
viewed as part of these capacitors, making them
easier to comprehend.
The features which determine the frequency
characteristics are currents, which charge the
integration capacitors. This situation is similar to
the transfer characteristic of the Diamond Transistor.
6
5.6dB
3
Gain (dB)
•
OPA861
VIN1
600Ω
RE
600Ω
VCM
−9
1M
1G
75
Input−Referred
70
65
60
55
50
45
40
35
30
25
20
10k
100k
1M
10M
100M
Frequency (Hz)
600Ω
Figure 45. High CMRR, Moderate Precision, Differential I/O ADC Driver
22
100M
Figure 46. ADC Driver, Small-Signal Frequency
Response
1k
OPA861
VIN2
10M
Frequency (Hz)
Common−Mode Rejection Ratio(dB)
ADS5272
−3
−6
High-CMRR, Moderate Precision, Differential
I/O ADC Driver
The circuit shown in Figure 45 depicts an ADC driver
implemented with two OPA861s. Since the gain is set
here by the ratio of the internal 600Ω resistors and
RE, its accuracy will only be as good as the input
resistor of the ADS5272. The small-signal frequency
response for this circuit has 150MHz at –3dB
bandwidth for a gain of approximately 5.6dB, as
shown in Figure 46. The advantage of this circuit lies
in its high CMRR to 100kHz, as shown in Figure 47.
This circuit also has more than 10 bits of linearity.
0
Figure 47. CMRR of the ADC Driver
1G
OPA861
www.ti.com
SBOS338 – AUGUST 2005
DESIGN-IN TOOLS
DEMONSTRATION BOARDS
A printed circuit board (PCB) is available to assist in
the initial evaluation of circuit performance using the
OPA861. This module is available free, as an
unpopulated PCB delivered with descriptive documentation. The summary information for the board is
shown below:
PRODUCT
PACKAGE
BOARD PART
NUMBER
OPA861ID
SO-8
DEM-OPA86xD
LITERATURE
REQUEST
NUMBER
SBOU035
The board can be requested on Texas Instruments
web site (www.ti.com).
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using
SPICE is often useful when analyzing the performance of analog circuits and systems. This principle is
particularly true for Video and RF amplifier circuits
where parasitic capacitance and inductance can have
a major effect on circuit performance. A SPICE model
for the OPA861 is available through the Texas
Instruments web page (www.ti.com). These models
do a good job of predicting small-signal AC and
transient performance under a wide variety of
operating conditions. They do not do as well in
predicting the harmonic distortion. These models do
not attempt to distinguish between the package types
in their small-signal AC performance.
NOISE PERFORMANCE
The OTA noise model consists of three elements: a
voltage noise on the B-input; a current noise on the
B-input; and a current noise on the E-input. Figure 48
shows the OTA noise analysis model with all the
noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms
in either nV/√Hz or pA/√Hz.
en
VO
RL
RS
√4kTRS
ibn
RG
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 8 shows the
general form for the output noise voltage using the
terms shown in Figure 48.
eO 2
e RSi bn 4kTR S
2
n
RL
RG g1m
2
2
R
R Gibi 4kTR G 1L
gm
(8)
THERMAL ANALYSIS
Maximum desired junction temperature will set the
maximum allowed internal power dissipation as described below. In no case should the maximum
junction temperature be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by
TA + PD × θJA. The total internal power dissipation
(PD) is the sum of quiescent power (PDQ) and
additional power dissipated in the output stage (PDL)
to deliver output current. Quiescent power is simply
the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for the
OPA861 be at a maximum when the maximum IO is
being driven into a voltage source that puts the
maximum voltage across the output stage. Maximum
IO is 15mA times a 9V maximum across the output.
Note that it is the power in the output stage and not
into the load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ
using an OPA861IDBV in the circuit of Figure 29b
operating at the maximum specified ambient temperature of +85°C and driving a –1V voltage reference.
PD = 10V × 5.4mA + (15mA × 9V) = 185mW
Maximum TJ = +85°C + (0.19W × 150°C/W) = 114°C.
Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower tested junction temperatures. The highest possible internal dissipation
will occur if the load requires current to be forced into
the output for positive output voltages or sourced
from the output for negative output voltages. This
puts a high current through a large internal voltage
drop in the output transistors.
ibi
√4kTRS
Figure 48. OTA Noise Analysis Model
BOARD LAYOUT GUIDELINES
Achieving
optimum
performance
with
a
high-frequency amplifier like the OPA861 requires
careful attention to board layout parasitics and external component types. Recommendations that will
optimize performance include:
23
OPA861
www.ti.com
SBOS338 – AUGUST 2005
a) Minimize parasitic capacitance to any AC ground
for all of the signal I/O pins. Parasitic capacitance on
the inverting input pin can cause instability: on the
noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the
signal I/O pins should be opened in all of the ground
and power planes around those pins. Otherwise,
ground and power planes should be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25") from the
power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and
power-plane layout should not be in close proximity to
the signal I/O pins. Avoid narrow power and ground
traces to minimize inductance between the pins and
the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.1µF)
across the two power supplies (for bipolar operation)
will improve 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the
main supply pins. These may be placed somewhat
farther from the device and may be shared among
several devices in the same area of the PC board.
c) Careful selection and placement of external
components will preserve the high-frequency performance of the OPA861. Resistors should be a
very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal film
or carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
Again, keep their leads and PC board traces as short
as possible. Never use wirewound type resistors in a
high-frequency application.
d) Connections to other wideband devices on the
board may be made with short, direct traces or
through onboard transmission lines. For short con-
24
nections, consider the trace and the input to the next
device as a lumped capacitive load. Relatively wide
traces (50mils to 100mils) should be used, preferably
with ground and power planes opened up around
them.
e) Socketing a high-speed part like the OPA861 is
not recommended. The additional lead length and
pin-to-pin capacitance introduced by the socket can
create an extremely troublesome parasitic network
that makes it almost impossible to achieve a smooth,
stable frequency response. Best results are obtained
by soldering the OPA861 onto the board.
INPUT AND ESD PROTECTION
The OPA861 is built using a very high-speed complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD protection
diodes to the power supplies as shown in Figure 49.
+VCC
External
Pin
Internal
Circuitry
−VCC
Figure 49. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA continuous current. Where higher currents are possible (for
example, in systems with ±15V supply parts driving
into the OPA861), current-limiting series resistors
should be added into the two inputs. Keep these
resistor values as low as possible since high values
degrade both noise performance and frequency response.
PACKAGE OPTION ADDENDUM
www.ti.com
27-Sep-2005
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
Lead/Ball Finish
MSL Peak Temp (3)
OPA861ID
ACTIVE
SOIC
D
8
75
TBD
Call TI
Call TI
OPA861IDBVR
ACTIVE
SOT-23
DBV
6
3000
TBD
Call TI
Call TI
OPA861IDBVT
ACTIVE
SOT-23
DBV
6
250
TBD
Call TI
Call TI
OPA861IDR
ACTIVE
SOIC
D
8
2500
TBD
Call TI
Call TI
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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