MPS MP1517DR 3a, 25v, 1.1mhz step-up converter Datasheet

MP1517
3A, 25V, 1.1MHz
Step-Up Converter
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP1517 is a 3A, fixed frequency step-up
converter ideal for medium-to-high current
step-up, flyback and SEPIC applications. The
high 1.1MHz switching frequency allows for
smaller external components producing a
compact solution for size constrained cameras,
PDAs and cell phones. The low 0.7V feedback
voltage offers higher efficiency in white LED
driver applications including cell phone camera
flash. The MP1517 regulates the output voltage
up to 25V with efficiencies as high as 95%.
Soft-start, cycle-by-cycle current limiting, and
input
under
voltage
lockout
prevent
overstressing or damage to sensitive external
circuitry at startup and output short-circuit
conditions. Current-mode regulation and
external compensation components allow the
MP1517 control loop to be optimized over a
wide variety of input voltage, output voltage,
and load current conditions. The MP1517 is
available in the thermally enhanced QFN16
(4mm x 4mm) package.
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•
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•
•
•
•
•
•
•
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4A Peak Current Limit
Low 700mV Feedback Threshold
Internal 150mΩ Power Switch
Input Range of 2.6V to 25V
Up to 95% Efficiency
Zero Current Shutdown Mode
Under Voltage Lockout Protection
Open Load Protection
Soft-Start Operation
Thermal Shutdown
Tiny 4mm x 4mm 16-Pin QFN Package
APPLICATIONS
•
•
•
•
Boost and SEPIC Regulators
Handheld Computers
Cell Phone Camera Flash, PDAs
Digital Still and Video Cameras
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
EVALUATION BOARD REFERENCE
Board Number
Dimensions
EV0043
2.5”X x 2.0”Y x 0.4”Z
TYPICAL APPLICATION
Efficiency vs
Load Current
VIN
5V
100
IN
4
OFF ON
15
3
C4
10nF
C5
10nF
7
SW
EN
9, 10
95
D1
MBR320
VOUT
12V
SS
MP1517
FB
BP
OLS
COMP
SGND
5, 13
16
1
85
80
75
70
VOUT = 18V
65
60
PGND
11, 12
VOUT = 12V
90
EFFICIENCY (%)
8
C3
10nF
C6
OPTIONAL
50
MP1517_TAC_S01
VIN = 5V
55
1
10
100
LOAD CURRENT (mA)
1000
MP1517-EC01
MP1517 Rev. 1.4
4/28/2006
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1
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
ABSOLUTE MAXIMUM RATINGS (1)
PACKAGE REFERENCE
TOP VIEW
PIN 1
IDENTIFICATION
FB
SS
NC
SGND
16
15
14
13
EXPOSED PAD
CONNECT TO PIN 13
COMP
1
12
PGND
NC
2
11
PGND
BP
3
10
SW
EN
4
9
SW
5
6
7
8
SGND
NC
OLS
IN
Recommended Operating Conditions
(2)
IN Input Supply Voltage VIN ............. 2.6V to 25V
Output Voltage................................. 3.3V to 25V
Operating Temperature .............–40°C to +85°C
MP1517_PD01-QFN16
*
Input Supply Voltage VIN .............. –0.3V to +27V
SW Pin Voltage VSW .................... –0.3V to +27V
Voltage at All Other
Pins except OLS .................................–0.3V to +6V
Storage Temperature ..............–65°C to +150°C
Junction Temperature.............................+150°C
Part Number*
Package
Temperature
MP1517DR
QFN16
(4mm x 4mm)
–40°C to +85°C
For Tape & Reel, add suffix –Z (eg. MP1517DR–Z)
For Lead Free, add suffix –LF (eg. MP1517DR–LF–Z)
Thermal Resistance
(3)
θJA
θJC
QFN16 (4mm x 4mm) ............. 46 ...... 10... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The device is not guaranteed to function outside of its
operating conditions.
3) Measured on approximately 1” square of 1 oz. Copper.
ELECTRICAL CHARACTERISTICS
VIN = 5.0V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
IN Operating Supply Current
BP Voltage
IN Undervoltage Lockout
Threshold
IN Undervoltage Lockout
Hysteresis
EN Input Low Voltage
EN Input High Voltage
EN Input Hysteresis
EN Input Bias Current
SW Switching Frequency
SW Maximum Duty Cycle
Error Amplifier Voltage Gain (4)
Error Amplifier Transconductance
COMP Maximum Output Current
FB Regulation Threshold
FB Input Bias Current
SS Charging Current
MP1517 Rev. 1.4
4/28/2006
Min
VEN ≤ 0.3V
VEN > 2V, VFB = 0.8V
VIN = 2.6V to 25V
IN Shutdown Supply Current
VIN Rising
Typ
Max
Units
0.5
1.0
µA
0.9
2.4
1.2
mA
V
2.40
V
2.10
100
mV
0.4
1.5
100
fSW
VFB = 0.6V
0.9
85
AVEA
GEA
Sourcing and Sinking
679
VFB = 0.7V
During Soft-Start
1.1
93
400
350
30
700
2
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1
1.3
721
1
V
V
mV
µA
MHz
%
V/V
µA/V
µA
mV
µA
µA
2
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 5.0V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
VIN = 5V
VIN = 3V
SW On Resistance (4)
SW Current Limit (4)
SW Leakage Current
Thermal Shutdown
Min
3.0
VSW = 25V
(4)
Typ
150
225
4.0
0.5
Max
°C
V
160
Open Load Shutdown Threshold
Measured at OLS Pin
Units
mΩ
mΩ
A
µA
27
Note:
4) Guaranteed by design.
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs
Load Current
Feedback Voltage vs
Temperature
100
715
FEEDBACK VOLTAGE (mV)
VOUT = 18V
95
EFFICIENCY (%)
90
85
VOUT = 22V
80
75
70
65
60
VIN = 12V
55
50
1
10
100
1000
LOAD CURRENT (mA)
710
705
700
695
690
685
-50
10000
50
100
0
TEMPERATURE (°C)
MP1517-EC01
150
MP1517-TPC01
Frequency vs
Temperature
Maximum Duty Cycle vs
Temperature
1.15
91.8
DUTY CYCLE (%)
FREQUENCY (MHz)
91.7
1.10
1.05
91.6
91.5
91.4
91.3
91.2
1.00
-50
0
50
100
TEMPERATURE (°C)
150
91.1
-50
0
50
100
TEMPERATURE (°C)
MP1517-TPC02
MP1517 Rev. 1.4
4/28/2006
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150
MP1517-TPC03
3
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
BP Voltage vs
Temperature
2.43
920
2.42
BP VOLTAGE (V)
940
900
880
860
2.39
2.38
820
-50
2.36
-50
MP1517-TPC04
4.6
2.40
2.37
150
4.8
2.41
840
0
50
100
TEMPERATURE (°C)
Current Limit vs
Temperature
CURRENT LIMIT (A)
Operating Supply
Current vs Temperature
4.4
4.2
4.0
3.8
3.6
3.4
3.2
0
50
100
TEMPERATURE (°C)
150
3.0
-50
MP1517-TPC05
0
50
100
TEMPERATURE (°C)
150
MP1517-TPC06
PIN FUNCTIONS
Pin #
Name
1
COMP
2, 6, 14
NC
3
BP
4
EN
5, 13
7
SGND
OLS
8
IN
9, 10
SW
11, 12
PGND
15
SS
16
FB
MP1517 Rev. 1.4
4/28/2006
Description
Compensation: Error Amplifier Output. Connect to a series RC network to compensate
the regulator control loop.
No Connect
Output of the Internal 2.4V Low Dropout Regulator. Connect a 10nF bypass capacitor
between BP and SGND. Do not apply an external load to BP.
Regulator On/Off Control Input. A logic high input (VEN > 1.5V) turns on the regulator, a
logic low puts it into low current shutdown mode. The EN pin cannot be left floating.
Signal Ground
Open Load Shutdown Pin. OLS senses regulator output voltage to protect IC during
open load operation. When this pin’s voltage exceeds 27V, the output switch is shut off.
The device then restarts in soft-start mode until it is disabled.
Input Supply Pin. This pin can be connected to the regulator’s input supply or to the
output for boot-strapped operation.
Output Switch Node. SW is the drain of the internal N-Channel MOSFET. Connect the
inductor and rectifier to SW to complete the step-up converter.
Power Ground
Soft-Start Input. Connect a 10nF to 22nF capacitor from SS to SGND to set the soft-start
time. SS sources 2µA to an external soft-start capacitor during startup. As the voltage at
SS increases to 0.55V, the voltage at COMP is clamped to 0.7V above the voltage at SS
limiting the startup current. The external capacitor at SS is discharged to ground when
under voltage lockout, thermal shutdown occurs or open load shutdown occurs.
Regulation Feedback Input. The regulation threshold is 0.7V.
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MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
OPERATION
700mV reference voltage and the feedback
voltage. When these two voltages are equal,
the PWM comparator turns off the switch
forcing the inductor current to the output
capacitor through the external rectifier. This
causes the inductor current to decrease. The
peak inductor current is controlled by the
voltage at COMP, which in turn is controlled by
the output voltage. Thus the output voltage
controls the inductor current to satisfy the load.
The use of current-mode regulation improves
transient response and control loop stability.
The MP1517 uses a 1.1MHz fixed-frequency,
current-mode regulation architecture to regulate
the output. The operation of the MP1517 can be
understood by referring to the block diagram of
Figure 1. At the beginning of each cycle, the
N-Channel MOSFET switch is turned on,
forcing the inductor current to rise. The current
at the source of the switch is internally
measured and converted to a voltage by the
current sense amplifier. That voltage is
compared to the error voltage at COMP. The
voltage at the output of the error amplifier is an
amplified version of the difference between the
IN
8
BP
3
EN
4
OLS
7
2.4V
LDO
OSCILLATOR
16V
9, 10 SW
+
0.9V
PWM
CONTROL
LOGIC
-+
2.4V
--
CURRENT
SENSE
AMP
+
--
11, 12 PGND
SS 15
-SOFT-START
16 FB
GM
+
0.7V
1
COMP
5, 13
SGND
MP1517_BD01
Figure 1—Functional Block Diagram
MP1517 Rev. 1.4
4/28/2006
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MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
Internal Low-Dropout Regulator
The internal power to the MP1517 is supplied
from the IN pin through an internal 2.4V lowdropout linear regulator, whose output is BP.
Bypass BP to SGND with a 10nF or greater
capacitor to insure the MP1517 operates
properly. The internal regulator can not supply
any more current than is required to operate the
MP1517, therefore do not apply any external
load to BP.
Soft-Start
The MP1517 includes a soft-start timer that limits
the voltage at COMP during startup to prevent
excessive current at the input. This prevents
premature termination of the source voltage at
startup due to input current overshoot. When
power is applied to the MP1517, and enable is
asserted, a 2µA internal current source charges
the external capacitor at SS. As the capacitor
charges, the voltage at SS rises. The MP1517
internally clamps the voltage at COMP to 700mV
above the voltage at SS. This limits the inductor
current at startup, forcing the input current to rise
slowly to the current required to regulate the
output voltage.
Open Load Shutdown
The MP1517 includes an open load detect that
will stop the output from switching. In a fault
condition where the connection to the LED’s is
open, VOUT will rise up. Once VOUT exceeds
27V, the MP1517 will stop switching and the
output will stop rising. When the output falls
below 27V the MP1517 will restart in soft-start
mode and switches until the OLS threshold is
exceeded again. This will continue until the part
is disabled. To disable the open load shutdown
feature, connect the OLS pin to GND.
APPLICATION INFORMATION
GENERAL PURPOSE COMPONENT
SELECTION
Setting the Output Voltage
Set the output voltage by selecting the resistive
voltage divider ratio. Use 10kΩ to 50kΩ for the
low-side resistor R2 of the voltage divider.
Determine the high-side resistor R1 by the
equation:
R1 =
R2 × (VOUT - VFB )
VFB
where VOUT is the output voltage.
For R2 = 10kΩ and VFB = 0.7V, then
R1 (kΩ) = 14.29kΩ (VOUT – 0.7V).
Selecting the Inductor
The inductor is required to force the higher
output voltage while being driven by the input
voltage. A larger value inductor results in less
ripple current, resulting in lower peak inductor
current and reducing stress on the internal
N-Channel. MOSFET switch. However, the
larger value inductor has a larger physical size,
higher series resistance, and/or lower
saturation current.
Choose an inductor that does not saturate
under the worst-case load transient and startup
MP1517 Rev. 1.4
4/28/2006
conditions. A good rule for determining the
inductance is to allow the peak-to-peak ripple
current to be approximately 30% to 50% of the
maximum input current. Make sure that the
peak inductor current is below 3A to prevent
loss of regulation due to the current limit.
Calculate the required inductance value by the
equation:
L=
VIN × (VOUT - VIN )
VOUT × f SW × ∆I
IIN(MAX ) =
VOUT × ILOAD (MAX )
VIN × η
∆I = (30% − 50%) × IIN(MAX )
Where VIN is the input voltage, fSW is the
switching frequency, ILOAD(MAX) is the maximum
load current, ∆I is the peak-to-peak inductor ripple
current and η is the efficiency.
Selecting the Input Capacitor
An input capacitor is required to supply the AC
ripple current to the inductor, while limiting
noise at the input source. A low ESR capacitor
is required to keep the noise at the IC to a
minimum. Ceramic capacitors are preferred, but
tantalum or low-ESR electrolytic capacitors may
also suffice.
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MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
Use an input capacitor value greater than 10µF.
The capacitor can be electrolytic, tantalum or
ceramic. However since it absorbs the input
switching current it requires an adequate ripple
current rating. Use a capacitor with RMS current
rating greater than the inductor ripple current.
To insure stable operation place the input
capacitor as close to the IC as possible.
Alternately a smaller high quality ceramic 0.1µF
capacitor may be placed closer to the IC with the
larger capacitor placed further away. If using this
technique, it is recommended that the larger
capacitor be a tantalum or electrolytic type. All
ceramic capacitors should be placed close to the
MP1517.
Selecting the Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Low ESR capacitors are
preferred to keep the output voltage ripple to a
minimum. The characteristic of the output
capacitor also affects the stability of the regulation
control system. Ceramic, tantalum, or low ESR
electrolytic capacitors are recommended. In the
case of ceramic capacitors, the impedance of the
capacitor at the switching frequency is dominated
by the capacitance, and so the output voltage
ripple is mostly independent of the ESR. The
output voltage ripple is calculated as:
VRIPPLE =
ILOAD × (VO UT − VIN )
VO UT × C2 × f SW
Where VRIPPLE is the output ripple voltage, VIN and
VOUT are the DC input and output voltages
respectively, ILOAD is the load current, fSW is the
switching frequency, and C2 is the capacitance of
the output capacitor.
In the case of tantalum or low-ESR electrolytic
capacitors, the ESR dominates the impedance at
the switching frequency, and so the output ripple
is calculated as:
⎡ ( VOUT − VIN )
× VOUT ⎤
R
+ ESR
VRIPPLE = ILOAD × ⎢
⎥
VIN
⎣ VOUT × f SW × C2
⎦
Where RESR is the equivalent series resistance of
the output capacitors.
Choose an output capacitor to satisfy the output
ripple and load transient requirements of the
MP1517 Rev. 1.4
4/28/2006
design. Place the output capacitor close to SW
to minimize the AC loop and switching noise.
Selecting the Diode
The output rectifier diode supplies current to the
inductor when the internal MOSFET is off. To
reduce losses due to diode forward voltage and
recovery time, use a Schottky diode. Choose a
diode whose maximum reverse voltage rating is
greater than the maximum output voltage. The
rated average forward current needs to be
equal to or greater than the load current.
Selecting the Soft-Start Capacitor
The soft-start period is determined by the equation:
t SS = 0.275 × C4
Where CSS (in nF) is the soft-start capacitor
from SS to SGND, and tSS (in ms) is the
soft-start period.
Determine the capacitor required for a given
soft-start period by the equation:
C4 = 3.64 × t SS
It is recommended that values between 10nF and
22nF for CSS be used to set the soft-start period.
Compensation
The output of the transconductance error
amplifier (COMP) is used to compensate the
regulation control system. The system uses two
poles and one zero to stabilize the control loop.
The poles are fP1 set by the output capacitor
and load resistance and fP2 set by the
compensation capacitor C3. The zero fZ1 is set
by the compensation capacitor C3 and the
compensation resistor R3.
These are determined by the equations:
1
π × C2 × R LOAD
fP1 =
fP2 =
G EA
2 × π × C3 × A VEA
fZ1 =
1
2 × π × C3 × R3
Where RLOAD is the load resistance, GEA is the
error amplifier transconductance, and AVEA is
the error amplifier voltage gain.
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MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
The DC loop gain is:
AVDC =
7 × A VEA × VIN × R LOAD × VFB
V OUT
2
Where VFB is the feedback regulation threshold.
There is also a right-half-plane zero (fRHPZ) that
exists in the continuous conduction mode
(inductor current does not drop to zero on each
cycle) step-up converters. The frequency of the
right-half plane zero is:
2
fRHPZ =
VIN × R LOAD
2 × π × L × VOUT
resistance (ESR) is used, then a second
compensation capacitor (from COMP to SGND)
is required to compensate for the zero
introduced by the output capacitor ESR. The
extra capacitor is required if the ESR zero is
less than 4x the crossover frequency. The ESR
zero frequency is:
1
2 × π × C2 × R ESR
fZESR =
If this is the case, calculate the second
compensation capacitor by the equation:
2
To stabilize the regulation control loop, the
crossover frequency (The frequency where the
loop gain drops to 0dB or a gain of 1) should be
less than half of fRHPZ and should be at most
75KHz. fRHPZ is at its lowest frequency at
maximum output load current.
C6 =
C2 × R ESR
R3
For most applications C6 is not required.
Typical
values
for
the
compensation
components are:
C3 = 10nF
R3 = 2.2kΩ
In some cases, an output capacitor with a high
capacitance and high equivalent series
SEPIC CONVERTER COMPONENT
SELECTION
Selecting the Input Capacitor
An input capacitor is required to supply the AC
ripple current to the inductor, while limiting
noise at the input source. The input capacitor
selection is the same as that in the General
Purpose Component Selection section above.
Selecting the Inductors
The SEPIC converter inductors (refer to Figure
4) are required to store energy, and generate
an output voltage that is less than or greater
than the input voltage. If a coupled inductor is
used in a SEPIC converter, then the mutual
inductance of each winding forces each
inductor to become twice the original
inductance. Therefore smaller inductance can
be used with a coupled inductor. But the core
saturation of the coupled inductors is related to
the sum of both inductor currents.
There are two current paths to the internal
N-Channel MOSFET switch in a SEPIC
converter. One is from L1 and the other is from
L2.
MP1517 Rev. 1.4
4/28/2006
Each inductor’s ripple current can be defined
as:
L1 =
L2 =
VIN × D
f SW × ∆I
VOUT × (1 − D)
f SW × ∆I
VOUT + VD
VOUT + VD + VIN
D=
Where VD is the voltage drop on diode D1, and
∆I is the peak to peak inductor ripple current.
Set ∆I to approximately 20% of the maximum
switch current. Each inductor’s peak current is:
IL1(PEAK ) =
V
+ VD
∆I
+ ILOAD × OUT
2
VIN
IL 2(PEAK ) =
V + VD
∆I
+ ILOAD × IN
2
VIN
The total of these two currents is the total
switch current, and should be less than the
minimum device current limit of 3A.
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MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
Selecting the Output Diode
The output diode is typically a Schottky diode.
The Schottky diode is selected based upon
voltage requirement, current rating, and thermal
capability. The diode breakdown and power
switch breakdown voltage are set higher than
VOUT + VIN, since this is the voltage stress on
both of these devices. The current rating is set
based on the average load current in the diode.
The average diode current is equal to the load
current, but the peak current and power
consumption on D1 is:
ID1(PEAK ) =
V + VD
∆I
+ ILOAD × IN
2
VIN
PD1 = ILOAD × VD
Selecting the Coupling Capacitor
The steady state voltage across L1 and L2 is
equal to zero. Therefore the coupling capacitor
has VIN across it in steady state. The coupling
capacitor will need to be rated for an input
voltage plus some guard band. Also this
capacitor will need an IRMS ripple current rating:
I C 8(RMS ) = ILOAD ×
VOUT + VD
VIN
Application Examples
Figure 2 shows a typical application circuit
driving multiple strings of LEDs with the
MP1517. The 3 strings of 6 white LEDs can be
driven from a voltage supply range of 2.6V to
6V at an output current of 20mA. A 1µF output
capacitor is usually sufficient for this kind of
application. A 4.7µH inductor with low DCR
(inductor resistance) is recommended to
improve efficiency. A 10µF ceramic capacitor is
recommended for the input capacitance.
Schottky diodes have fast recovery and a low
forward voltage and are recommended. The
MP1517 soft-start helps to limit the amount of
current through VIN at startup and to also limit
the amount of overshoot on the output.
MP1517 Rev. 1.4
4/28/2006
Figure 3 shows a typical application running the
MP1517 in flash mode. During preview mode,
resistor R2 sets the current through the white
LEDs to 20mA. When a flash is required the
N-Channel MOSFET Q1 is turned on and
150mA flows through the LEDs. The
compensation capacitor has been increased to
47nF. This forces the compensation node to
slowly rise when Q1 is turned on which allows
the current through the inductor to slowly
increase without overshoot. By doing this the
inrush current on the input is minimized.
Figure 4 shows a SEPIC circuit using the
MP1517 to generate 3.3V output from a 3V to
4.2V input. A peak voltage detect circuit (D2
and C7) is added to the IN pin from the highest
voltage potential (SW node) in the entire circuit.
The average voltage on C7 is roughly the sum
of VIN and VOUT. This ensures the enhancement
of the internal MOSFET switch. A transformer
with two windings can be used to replace the
two separate inductors in Figure 4. In this case,
the effective inductance value is doubled due to
the mutual inductance of each winding. This
reduces the inductor ripple current and
improves efficiency. Figures 5 and 6 show
some other SEPIC application circuits.
Layout Consideration
High frequency switching regulators require
very careful layout for stable operation and low
noise. All components must be placed as close
to the IC as possible. Keep the path between
L1, D1, and C2 (also L2, C8 if applicable)
extremely short for minimal noise and ringing.
C1 must be placed close to the IN pin for best
decoupling. All feedback components must be
kept close to the FB pin to prevent noise
injection on the FB pin trace. The ground return
of C1 and C2 should be tied close to the PGND
pins. See the MP1517 demo board layout for
reference.
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9
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
D1
MBR0540
VIN
2.6V to 20V
8
UP TO 6
LEDs PER
STRING
IN
4
OFF ON
15
SW
SS
OLS
MP1517
3
C4
10nF
EN
FB
BP
C5
10nF
COMP
SGND
9, 10
7
16
1
PGND
5, 13
11, 12
C3
10nF
MP1517_F02
Figure 2—Driving 3 Strings of 6 White LEDs
MBR0540
VIN
2.6V to 6V
VOUT
8
SS
OFF ON
4
15
3
IN
SS
SW
EN
OLS
MP1517
FB
BP
COMP
SGND
5, 13
9, 10
7
16
1
PGND
11, 12
C3
47nF
Q1
NMOS
ZXMN6A08E6TA
GATE
MP1517_F03
Figure 3—Flash Circuit Driving 4 White LEDs (150mA Flash Current)
MP1517 Rev. 1.4
4/28/2006
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2006 MPS. All Rights Reserved.
10
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
VIN
3V to 4.2V
D2
1N4148 OR BAT54
8
D1
MBR320L
IN
4
OFF ON
15
3
C4
10nF
C5
10nF
7
SW
EN
9, 10
VOUT
3.3V/0.8A
SS
MP1517
FB
BP
OLS
COMP
SGND
16
1
R3
OPEN
C3
OPEN
PGND
5, 13
11, 12
C6
10nF
MP1517_F04
Figure 4—Typical SEPIC Application Circuit
VIN
3.3V
D2
IN4148 OR BAT54
8
D1
MBR320L
IN
4
OFF ON
15
3
C4
10nF
C5
10nF
7
SW
EN
9, 10
VOUT
5V/0.6A
L2
SS
MP1517
FB
BP
OLS
COMP
SGND
5, 13
16
1
Q1
MMBT3904
PGND
11, 12
C3
47nF
MP1517_F05
Figure 5—3.3V to 5V @ 600mA SEPIC Application Circuit
MP1517 Rev. 1.4
4/28/2006
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2006 MPS. All Rights Reserved.
11
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
VIN
5V
L1
VP1-0059
VERSA-PAC STANDARD
MULTI-WINDING XFMR
8
D1
MBRS130
IN
4
OFF ON
15
3
C4
10nF
C5
10nF
7
SW
EN
9, 10
VOUT1
12V/0.2A
L2
SS
MP1517
FB
BP
OLS
COMP
SGND
5, 13
16
1
PGND
L3
11, 12
C3
47nF
D2
MBRS130
VOUT2
-12V/0.2A
MP1517_F06
Figure 6—±12V SEPIC Application Circuit
MP1517 Rev. 1.4
4/28/2006
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2006 MPS. All Rights Reserved.
12
MP1517 – 3A, 25V, 1.1MHz STEP-UP CONVERTER
PACKAGE INFORMATION
QFN16 (4mm x 4mm)
3.90
4.10
2.15
2.45
0.50
0.70
PIN 1 ID
MARKING
0.25
0.35
3.90
4.10
PIN 1 ID
INDEX AREA
13
PIN 1 ID
SEE DETAIL A
16
1
12
2.15
2.45
0.65
BSC
9
4
8
TOP VIEW
5
BOTTOM VIEW
PIN 1 ID OPTION A
0.45x45º TYP.
PIN 1 ID OPTION B
R0.25 TYP.
0.80
1.00
0.20 REF
0.00
0.05
DETAIL A
SIDE VIEW
3.80
2.30
NOTE:
1) ALL DIMENSIONS ARE IN MILLIMETERS.
2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH.
3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX.
4) JEDEC REFERENCE IS MO-220, VARIATION VGGC.
5) DRAWING IS NOT TO SCALE.
1.00
0.35
0.65
RECOMMENDED LAND PATTERN
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP1517 Rev. 1.4
4/28/2006
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2006 MPS. All Rights Reserved.
13