Micro Linear ML4890 High efficiency, low ripple boost regulator Datasheet

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ML4890
High Efficiency, Low Ripple Boost Regulator
GENERAL DESCRIPTION
FEATURES
The ML4890 is a high efficiency, PFM (Pulse Frequency
Modulation), boost switching regulator connected in
series with an integrated LDO (Low Dropout Regulator)
that incorporates “Silent Switcher™” technology. This
technique incorporates a patented tracking scheme to
minimize the voltage drop across the LDO and increase
the total efficiency of the regulator beyond that which can
be obtained by using a discrete external LDO regulator.
■
■
The ML4890 is designed to convert single or multiple cell
battery inputs to regulated output voltages for integrated
circuits and is ideal for portable communications
equipment that cannot tolerate the output voltage ripple
normally associated with switching regulators.
Incorporates “Silent Switcher™” technology to deliver
very low output voltage ripple (typically 5mV)
Guaranteed full load start-up and operation at 1.0V
input and low operating quiescent current (<100µA)
for extended battery life
■
Pulse Frequency Modulation and internal synchronous
rectification for high efficiency
■
■
Minimum external components
Low ON resistance internal switching MOSFETs
■
5V, 3.3V, and 3V output versions
An integrated synchronous rectifier eliminates the need for
an external Schottky diode and provides a lower forward
voltage drop, resulting in higher conversion efficiency.
BLOCK DIAGRAM
L1
*CIN
VIN
C2
1
6
VL
VBAT
7
5
VBOOST
SHDN
VOUT
+
4
2
VREF
BOOST
CONTROL
FEEDBACK
3
GND
LDO
CONTROL
C1
PWR
GND
8
VOUT
–
FROM
POWER
MANAGEMENT
Patent Pending
*Optional
1
ML4890
PIN CONNECTION
ML4890-5/-3/-T
8-Pin SOIC (S08)
VIN
1
8
PWR GND
VREF
2
7
SHDN
GND
3
6
VL
VOUT
4
5
VBOOST
TOP VIEW
PIN DESCRIPTION
PIN
NO.
2
NAME
FUNCTION
1
VIN
Battery input voltage
2
VREF
200mV reference output
3
GND
Analog signal ground
4
VOUT
LDO linear regulator output
PIN
NO.
NAME
5
VBOOST
Boost regulator output for connection
of an output filter capacitor
6
VL
Boost inductor connection
7
SHDN
Pulling this pin high shuts down the
regulator, isolating the load from the
input
8
PWR GND Return for the NMOS boost transistor
FUNCTION
ML4890
ABSOLUTE MAXIMUM RATINGS
Absolute maximum ratings are those values beyond which
the device could be permanently damaged. Absolute
maximum ratings are stress ratings only and functional
device operation is not implied.
VBOOST ........................................................................ 7V
Voltage on Any Other Pin ... GND –0.3V to VBOOST +0.3V
Peak Switch Current (IPEAK) .......................................... 1A
Average Switch Current (IAVG) ............................... 500mA
LDO Output Current ............................................. 250mA
Junction Temperature .............................................. 150°C
Storage Temperature Range .................... –65°C to +150°C
Lead Temperature (Soldering 10s) .......................... +260°C
Thermal Resistance (θJA)
Plastic SOIC .................................................... 110°C/W
OPERATING CONDITIONS
Temperature Range
ML4890CS-X ............................................ 0°C to +70°C
ML4890ES-X ......................................... –20°C to +70°C
VIN Operating Range
ML4890CS-X ................................................ 1.0V to 6V
ML4890ES-X ................................................. 1.1V to 6V
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = Operating Voltage Range, TA = Operating Temperature Range. (Note 1)
PARAMETER
CONDITIONS
MIN
TYP.
MAX
UNITS
VIN = 6V
60
75
µA
SHDN = high
15
25
µA
VBOOST = VOUT + 0.5V
8
10
µA
1
µA
Supply
VIN Current
VOUT Quiescent Current
VL Quiescent Current
Reference
Output Voltage (VREF)
0 < IPIN2 < –5µA,
195
200
205
mV
4.5
5
5.5
µs
PFM Regulator
Pulse Width (TON)
LDO
DC Output Voltage (VOUT)
Load Regulation
Dropout Voltage
ML4890-5
VBOOST = VOUT + 0.5V, IOUT < 200mA
4.85
5.0
5.15
V
ML4890-3
VBOOST = VOUT + 0.5V, IOUT < 100mA
3.2
3.3
3.4
V
ML4890-T
VBOOST = VOUT + 0.5V, IOUT < 80mA
2.91
3.0
3.09
V
ML4890-5
See Figure 1
VIN = 1.2V, IOUT < 7mA
VIN = 2.4V, IOUT < 50mA
4.85
4.85
5.0
5.0
5.15
5.15
V
V
ML4890-3
VIN = 1.2V, IOUT < 14mA
VIN = 2.4V, IOUT < 75mA
3.2
3.2
3.3
3.3
3.4
3.4
V
V
ML4890-T
VIN = 1.2V, IOUT < 15mA
VIN = 2.4V, IOUT < 60mA
2.91
2.91
3.0
3.0
3.09
3.09
V
V
ML4890-5
See Figure 1
VIN = 1.2V, IOUT < 7mA
VIN = 2.4V, IOUT < 50mA
300
500
mV
mV
ML4890-3
VIN = 1.2V, IOUT < 14mA
VIN = 2.4V, IOUT < 75mA
300
500
mV
mV
ML4890-T
VIN = 1.2V, IOUT < 15mA
VIN = 2.4V, IOUT < 60mA
300
500
mV
mV
Output Ripple
5
mVP-P
Shutdown
SHDN Threshold
SHDN Bias Current
Note 1:
0.5
–100
0.8
1.0
V
100
nA
Limits are guaranteed by 100% testing, sampling, or correlation with worst case conditions.
3
ML4890
33µH
(Sumida CD54)
ML4890
VIN
VIN
100µF
1µF
PWR GND
VREF
SHDN
GND
VL
VOUT
VBOOST
33µF
IOUT
VOUT
100µF
Figure 1. Application Test Circuit
C2
L1
6
5
ILOAD
4
Q2
R1
Q3
+
+
A3
A2
–
R
S
5µs
ONE SHOT
–
Q1
A1
–
+
+ –
VOS = f (ILOAD)
Figure 2. PFM Regulator and LDO Block Diagram
4
VREF
R2
C1
ML4890
FUNCTIONAL DESCRIPTION
The ML4890 combines Pulse Frequency Modulation
(PFM) and synchronous rectification to create a boost
converter that is followed by a low dropout linear
regulator (LDO). This combination creates a low output
ripple boost converter that is both highly efficient and
simple to use.
The PFM regulator charges a single inductor for a fixed
period of time and then completely discharges before
another cycle begins, simplifying the design by
eliminating the need for conventional current limiting
circuitry. Synchronous rectification is accomplished by
replacing an external Schottky diode with an on-chip
PMOS device, reducing switching losses and external
component count.
The integrated LDO reduces the output ripple voltage to
less than 5mV peak-to-peak. Integrating the LDO along
with the PFM regulator allows the circuit to be optimized
for very high efficiency using a patented feedback
technique. It also allows the LDO to provide the
maximum ripple rejection over the operating frequency
range of the regulator.
A block diagram of the ML4890 is shown in Figure 2. The
PFM stage is comprised of Q1, Q2, A1, A2, the one shot,
the flip-flop, and externals L1 and C2. The LDO stage is
comprised of Q3, A3, R1, R2, the offset voltage control,
and external C1. Since the LDO actually controls the
operation of the PFM regulator, the operation of the LDO
stage will be covered first.
PFM REGULATOR OPERATION
When the output of the PFM stage, VBOOST (pin 5), is at or
above the dropout voltage, VOUT + VOS, the output of A1
stays low and the circuit remains idle. When VBOOST falls
below the required dropout voltage, the output of A1 goes
high, signaling the regulator to deliver charge to the
capacitor C2. Since the output of A2 is normally high, the
output of the flip-flop becomes SET. This triggers the one
shot to turn Q1 on and begins charging L1 for 5µs. When
the one shot times out, Q1 turns off, allowing L1 to
flyback and momentarily charge C2 through the body
diode of Q2. But, as the source voltage of Q2 rises above
the drain, the current sensing amplifier A2 drives the gate
of Q2 low, causing Q2 to short out the body diode. The
inductor then discharges into C2 through Q2. The output
of A2 going low also serves to RESET the flip-flop in
preparation for the next charging cycle. When the
inductor current in Q2 falls to zero, the output of A2 goes
high, releasing Q2‘s gate, allowing the flip-flop to be SET
again. If the voltage at VBOOST is still low, A1 will initiate
another pulse. Typical inductor current and voltage
waveforms are shown in Figure 4.
INDUCTOR
CURRENT
LDO OPERATION
The LDO stage operates as a linear regulator. A3 is the
error amplifier, which compares the output voltage
through the divider R1 and R2 to the reference, and Q3 is
the pass device. When the output voltage is lower than
desired, the output of A3 increases the gate drive of Q3,
which reduces the voltage drop across it and brings the
output back into regulation. Similarly, if the output voltage
is higher than desired, A3 adjusts the gate drive of Q3 for
more drop and the output is brought back into regulation.
Q(ONE SHOT)
Q1 ON
Q2
ON
Q1 ON
Q2
ON
Q1 & Q2 OFF
Figure 4. PFM Inductor Current Waveforms and Timing.
SHUTDOWN
The SHDN pin should be held low for normal operation.
Raising the voltage on SHDN above the threshold level
will release the gate of Q3, which effectively becomes an
open circuit. This also prevents the one shot from
triggering, which keeps switching from occurring.
450
400
VOS (mV)
Also included in the LDO stage is an offset voltage
control. This circuit monitors the output current and
adjusts the offset voltage according the general
characteristic shown in Figure 3. The offset control
ensures that the PFM stage provides just enough
“overhead” voltage for the LDO stage to operate properly.
350
300
250
200
150
100
0
10
20
30
40
50
60
70
80
90 100
IOUT (mA)
Figure 3. LDO VOS versus output current.
5
ML4890
where η is the efficiency, typically between 0.75 and
0.85, and VOS is the dropout voltage at IOUT(MAX) taken
from Figure 3. Note that this is the value of inductance
that just barely delivers the required output current under
worst case conditions. A lower value may be required to
cover inductor tolerance, the effect of lower peak inductor
currents caused by resistive losses, and minimum dead
time between pulses.
DESIGN CONSIDERATIONS
INDUCTOR
Selecting the proper inductor for a specific application
usually involves a trade-off between efficiency and
maximum output current. Choosing too high a value will
keep the regulator from delivering the required output
current under worst case conditions. Choosing too low a
value causes efficiency to suffer. It is necessary to know
the maximum required output current and the input
voltage range to select the proper inductor value. The
maximum inductor value can be estimated using the
following formula:
Another method of determining the appropriate inductor
value is to make an estimate based on the typical
performance curves given in Figures 5 and 6. Figure 5
shows maximum output current as a function of input
voltage for several inductor values. These are typical
performance curves and leave no margin for inductance
and ON-time variations. To accommodate worst case
conditions, it is necessary to derate these curves by at
least 10% in addition to inductor tolerance.
µH
=
L
15
70
40
30
22
L
=
33
µH
L
60
L
=
=
47
µH
µH
68
50
40
30
20
20
10
10
1.2
1.4
1.6
1.8
2.0
2.2
2.4
2.6
0
1.0
2.8
1.5
VIN (V)
2.0
VIN (V)
ML4890-5
µH
L=
22
180
L
=
33
160
µH
200
µH
15
120
60
µH
L=
µH
47
µH
68
10
80
L
L=
100
=
L=
IOUT MAX (mA)
140
40
20
0
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
VIN (V)
Figure 5. Output Current versus Input Voltage.
6
L
=
L=
=
IOUT MAX (mA)
L
50
80
µH
68
µH
10
µH
=
L
L=
60
0
1.0
ML4890-3
90
47
33
µH
µH
22
µH
100
L=
15
µH
L=
10
70
IOUT MAX (mA)
(1)
ML4890-T
L=
80
VIN(MIN) × TON(MIN) × η
2 × (VOUT + VOS ) × IOUT(MAX)
µH
2
LMAX =
2.5
3.0
ML4890
For example, a two cell to 5V application requires 40mA
of output current while using an inductor with 15%
tolerance. The output current should be derated by 25%
to 50mA to cover the combined inductor and ON-time
tolerances. Assuming that 2V is the end of life voltage of a
two cell input, Figure 5 shows that with a 2V input, the
ML4890-5 delivers 58mA with a 22µH inductor.
After the appropriate inductor value is chosen, it is
necessary to find the minimum inductor current rating
required. Peak inductor current is determined from the
following formula:
IL(PEAK ) =
Figure 6 shows efficiency under the conditions used to
create Figure 5. It can be seen that efficiency is mostly
independent of input voltage and is closely related to
inductor value. This illustrates the need to keep the
inductor value as high as possible to attain peak system
efficiency. As the inductor value goes down to 10µH, the
efficiency drops to between 70% and 75%. With 33µH,
the efficiency reaches approximately 85% and there is
little room for improvement. At values greater than 47µH,
the operation of the synchronous rectifier becomes
unreliable at low input voltages because the inductor
current is so small that it is difficult for the control circuitry
to detect. The data used to generate Figures 5 and 6 is
provided in Table 1.
Efficiency at IOUT MAX (%)
L = 33µH
L = 22µH
80
L = 15
75
µH
0µH
1.2
1.4
L = 68µH
L = 47µH
85 L = 33µH
80 L = 22µH
L = 15µH
75
70
70
65
1.0
90
L = 10µH
L=1
1.6
1.8
2.0
2.2
2.4
2.6
65
1.0
2.8
1.5
2.0
2.5
3.0
VIN (V)
VIN (V)
95
Efficiency at IOUT MAX (%)
Efficiency at IOUT MAX (%)
L = 47µH
85
ML4890-3
95
L = 68µH
90
(2)
It is important to note that for reliable operation, make
sure that IL(PEAK) does not exceed the 1A maximum switch
current rating. In the two cell application previously
described, a maximum input voltage of 3V would give a
peak current of 880mA. When comparing various
inductors, it is important to keep in mind that suppliers
use different criteria to determine their ratings. Many use a
conservative current level, where inductance has dropped
to 90% of its normal level. In any case, it is a good idea to
try inductors of various current ratings with the ML4890 to
determine which inductor is the best choice. Check
efficiency and maximum output current, and if a current
probe is available, look at the inductor current to see if it
looks like the waveform shown in Figure 4.
ML4890-T
95
TON(MAX) × VIN(MAX)
LMIN
ML4890-5
L = 68µH
90
L = 47µH
85
L = 15
80
µH
L = 33µH
L = 22µ
H
75
L = 10µH
70
65
60
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
Figure 6. Typical Efficiency as a Function of VIN.
7
ML4890
The DC resistance of the inductor should be kept to a
minimum to reduce losses. A good rule of thumb is to
allow 5 to 10mΩ of resistance for each µH of inductance.
Also, be aware that the DC resistance of an inductor
usually isn‘t specified tightly, so an inductor with a
maximum DC resistance spec of 150mΩ may actually
have 100mΩ of resistance.
Suitable inductors can be purchased from the following
suppliers:
AVX
(207) 282-5111
Sprague
(207) 324-4140
Coilcraft
(708) 639-6400
Coiltronics
(407) 241-7876
OUTPUT CAPACITOR
Dale
(605) 665-9301
Sumida
(708) 956-0666
The LDO stage output capacitor (C1) is required for
stability and to provide a high frequency filter. An output
capacitor with a capacitance of 100µF, an ESR of less than
100mΩ, and an ESL of less than 5nH is a good general
purpose choice.
BOOST CAPACITOR
The boost capacitor (C2) supplies current to the load
during the ON-time of Q1 and will limit the ripple the
LDO stage has to contend with. The ripple on C2 is
influenced by three capacitor parameters: capacitance,
ESL, and ESR. The contribution due to capacitance can be
determined by looking at the change in the capacitor
voltage required to store the energy delivered by the
inductor in a single charge-discharge cycle, as given by
the following formula:
2
2
TON × VIN
C2 ≥
(in Farads)
2 × L × ∆VBOOST × (VOUT – VIN)
(3)
For example, a 2.4V input, a 5V output, a 22µH inductor,
and an allowance of 100mV of ripple on the boost
capacitor results in a minimum C2 value of 15µF.
The boost capacitor‘s Equivalent Series Resistance (ESR)
and Equivalent Series Inductance (ESL), also contribute to
the ripple due to the inductor discharge current waveform.
Just after the NMOS transistor turns off, the output current
ramps quickly to match the peak inductor current. This
fast change in current through the boost capacitor‘s ESL
causes a high frequency (5ns) spike that can be over 1V in
magnitude. After the ESL spike settles, the boost voltage
still has a ripple component equal to the inductor
discharge current times the ESR. This component will have
a sawtooth waveshape and can be calculated using the
following formula:
ESR ≤
8
For example, a 2.4V input, a 22µH inductor, and an
allowance of 100mV of ripple on the boost capacitor
results in a maximum ESR of 200mΩ. Therefore, a boost
capacitor with a capacitance of 22µF or 33µF, an ESR of
less than 200mΩ, and an ESL of less than 5nH is a good
choice. Tantalum capacitors which meet these
requirements can be obtained from the following
suppliers:
∆VBOOST
(in Ω)
IL(PEAK )
(4)
INPUT CAPACITOR
Unless the input source is a very low impedance battery, it
will be necessary to decouple the input with a capacitor
with a value of between 47µF and 100µF. This provides
the benefits of preventing input ripple from affecting the
ML4890 control circuitry, and it also improves efficiency
by reducing I-squared R losses during the charge and
discharge cycles of the inductor. Again, a low ESR
capacitor (such as tantalum) is recommended.
REFERENCE CAPACITOR
Under some circumstances input ripple cannot be
reduced effectively. This occurs primarily in applications
where inductor currents are high, causing excess output
ripple due to “pulse grouping”, where the chargedischarge pulses are not evenly spaced in time. In such
cases it may be necessary to decouple the reference pin
(VREF) with a small 10nF to 100nF ceramic capacitor. This
is particularly true if the ripple voltage at VIN is greater
than 100mV.
ML4890
LAYOUT
Good PC board layout practices will ensure the proper
operation of the ML4890. Important layout considerations
include:
• Use adequate ground and power traces or planes
• Keep components as close as possible to the ML4890
• Use short trace lengths from the inductor to the VL pin
and from the output capacitor to the VBOOST pin.
• Use a single point ground for the ML4890 ground pins,
and the input and output capacitors
A sample PC board layout is shown in Figure 7.
Figure 7. Sample PC Board Layout.
9
ML4890
TABLE 1. MAXIMUM OUTPUT CURRENT AND EFFICIENCY
ML4890-T
VIN
L = 10µH
1.0
1.5
2.0
L = 15µH
1.0
1.5
2.0
2.5
2.8
L = 22µH
1.0
1.5
2.0
2.5
2.8
L = 33µH
1.0
1.5
2.0
2.5
2.8
L = 47µH
1.0
1.5
2.0
2.5
2.8
L = 68µH
1.0
1.5
2.0
2.5
2.8
10
ML4890-3
IOUT (mA)
EFFICIENCY PERCENTAGE
30.6
70.7
80.0
73.5
72.0
70.3
23.8
56.5
80.0
80.0
80.0
78.7
77.3
74.9
74.0
73.7
18.4
44.2
76.6
80.0
80.0
82.0
81.1
77.9
76.9
76.7
13.0
32.4
56.6
80.0
80.0
85.7
85.1
82.7
80.4
80.1
9.8
23.3
41.1
62.9
77.4
87.4
87.2
85.8
83.7
82.6
7.9
18.8
33.4
51.0
64.3
88.4
88.9
87.6
86.0
84.6
VIN
L = 10µH
1.0
1.5
2.0
L = 15µH
1.0
1.5
2.0
2.5
3.0
L = 22µH
1.0
1.5
2.0
2.5
3.0
L = 33µH
1.0
1.5
2.0
2.5
3.0
L = 47µH
1.0
1.5
2.0
2.5
3.0
L = 68µH
1.0
1.5
2.0
2.5
3.0
IOUT (mA)
EFFICIENCY PERCENTAGE
29.6
71.4
100.0
73.8
73.7
71.9
23.0
54.7
89.8
100.0
100.0
80.4
78.8
76.1
74.7
74.1
16.2
41.4
75.6
100.0
100.0
82.1
82.6
80.5
77.7
77.1
10.9
30.4
55.8
82.5
100.0
85.0
86.0
84.7
82.3
80.3
9.1
22.7
41.9
63.3
89.6
87.1
87.9
87.4
85.6
83.1
7.7
17.9
32.1
48.8
69.6
89.3
89.2
88.3
87.2
85.9
ML4890
TABLE 1. MAXIMUM OUTPUT CURRENT AND EFFICIENCY
(continued)
ML4890-5
VIN
L = 10µH
1.0
1.5
2.0
L = 15µH
1.0
1.5
2.0
2.5
3.0
L = 22µH
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
L = 33µH
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
L = 47µH
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
L = 68µH
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
IOUT (mA)
EFFICIENCY PERCENTAGE
13.6
42.7
82.2
67.9
77.2
78.1
9.3
31.7
60.6
95.7
137.9
69.6
80.9
81.6
80.5
79.4
7.4
23.1
46.1
73.9
108.9
145.1
184.5
200.0
73.6
83.6
84.6
84.0
83.0
82.4
81.3
80.1
6.0
18.3
34.2
57.0
82.3
106.0
137.1
169.3
76.9
85.8
86.8
86.6
86.2
85.3
84.6
84.0
14.2
25.7
41.4
59.4
82.9
105.5
131.3
87.2
88.3
88.3
88.0
87.3
86.6
86.2
17.9
31.7
46.2
63.2
82.5
99.7
88.9
89.8
89.7
89.6
89.2
88.0
11
ML4890
PHYSICAL DIMENSIONS inches (millimeters)
Package: S08
8-Pin SOIC
8
0.189 - 0.199
(4.80 - 5.06)
PIN 1 ID
0.017 - 0.027
(0.43 - 0.69)
(4 PLACES)
0.148 - 0.158 0.228 - 0.244
(3.76 - 4.01) (5.79 - 6.20)
1
0.050 BSC
(1.27 BSC)
0.059 - 0.069
(1.49 - 1.75)
0º - 8º
0.055 - 0.061
(1.40 - 1.55)
0.012 - 0.020
(0.30 - 0.51)
SEATING PLANE
0.004 - 0.010
(0.10 - 0.26)
0.015 - 0.035
(0.38 - 0.89)
0.006 - 0.010
(0.15 - 0.26)
ORDERING INFORMATION
PART NUMBER
OUTPUT VOLTAGE
TEMPERATURE RANGE
PACKAGE
ML4890CS-T
ML4890CS-3
ML4890CS-5
3.0V
3.3V
5.0V
0°C to +70°C
0°C to +70°C
0°C to +70°C
8-Pin SOIC (S08)
8-Pin SOIC (S08)
8-Pin SOIC (S08)
ML4890ES-T
ML4890ES-3
ML4890ES-5
3.0V
3.3V
5.0V
–20°C to +70°C
–20°C to +70°C
–20°C to +70°C
8-Pin SOIC (S08)
8-Pin SOIC (S08)
8-Pin SOIC (S08)
© Micro Linear 1997
is a registered trademark of Micro Linear Corporation
Products described in this document may be covered by one or more of the following patents, U.S.: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940;
5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; Japan: 2598946; 2619299. Other patents are pending.
Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design.
Micro Linear does not assume any liability arising out of the application or use of any product described herein,
neither does it convey any license under its patent right nor the rights of others. The circuits contained in this
data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to
whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility
or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel
before deciding on a particular application.
12
2092 Concourse Drive
San Jose, CA 95131
Tel: 408/433-5200
Fax: 408/432-0295
DS4890-01
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