AD AD712JR-REEL Dual precision, low cost, high speed bifet op amp Datasheet

Dual Precision, Low Cost,
High Speed BiFET Op Amp
AD712
Enhanced replacement for LF412 and TL082
AC performance
Settles to ±0.01% in 1.0 μs
16 V/μs minimum slew rate (AD712J)
3 MHz minimum unity-gain bandwidth (AD712J)
DC performance
200 V/mV minimum open-loop gain (AD712K)
Surface mount available in tape and reel in
accordance with the EIA-481A standard
MIL-STD-883B parts available
Single version available: AD711
Quad version: AD713
Available in PDIP, SOIC_N, and CERDIP packages
CONNECTION DIAGRAM
AMPLIFIER NO. 1
AMPLIFIER NO. 2
OUTPUT
1
8
V+
INVERTING
INPUT
OUTPUT
2
7
NONINVERTING
INPUT
3
INVERTING
6
INPUT
V–
4
AD712
5
NONINVERTING
INPUT
00823-001
FEATURES
Figure 1. 8-Lead PDIP (N-Suffix),
SOIC_N (R-Suffix), and CERDIP (Q-Suffix)
GENERAL DESCRIPTION
The AD712 is a high speed, precision, monolithic operational
amplifier offering high performance at very modest prices. Its
very low offset voltage and offset voltage drift are the results of
advanced laser wafer trimming technology. These performance
benefits allow the user to easily upgrade existing designs that
use older precision BiFETs and, in many cases, bipolar op amps.
military temperature range of −55°C to +125°C and is available
processed to MIL-STD-883B, Rev. C.
The superior ac and dc performance of this op amp makes it
suitable for active filter applications. With a slew rate of 16 V/μs
and a settling time of 1 μs to ±0.01%, the AD712 is ideal as a
buffer for 12-bit digital-to-analog and analog-to-digital
converters and as a high speed integrator. The settling time is
unmatched by any similar IC amplifier.
The AD712 is available in 8-lead PDIP, SOIC_N, and CERDIP
packages.
The combination of excellent noise performance and low input
current also make the AD712 useful for photo diode preamps.
Common-mode rejection of 88 dB and open-loop gain of
400 V/mV ensure 12-bit performance even in high speed
unity-gain buffer circuits.
The AD712 is pinned out in a standard op amp configuration
and is available in seven performance grades. The AD712J and
AD712K are rated over the commercial temperature range of
0°C to 70°C. The AD712A is rated over the industrial temperature range of −40°C to +85°C. The AD712S is rated over the
Extended reliability PLUS screening is available, specified over
the commercial and industrial temperature ranges. PLUS
screening includes 168-hour burn-in, in addition to other
environmental and physical tests.
PRODUCT HIGHLIGHTS
1. The AD712 offers excellent overall performance at very
competitive prices.
2. The Analog Devices, Inc. advanced processing technology
and 100% testing guarantee a low input offset voltage (3 mV
maximum, J grade). Input offset voltage is specified in the
warmed-up condition.
3. Together with precision dc performance, the AD712 offers
excellent dynamic response. It settles to ±0.01% in 1 μs and
has a minimum slew rate of 16 V/μs. Thus, this device is ideal
for applications such as DAC and ADC buffers that require a
combination of superior ac and dc performance.
Rev. G
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2006 Analog Devices, Inc. All rights reserved.
AD712
TABLE OF CONTENTS
Features .............................................................................................. 1
Guarding...................................................................................... 14
Connection Diagram ....................................................................... 1
Digital-to-Analog Converter Applications ............................. 14
General Description ......................................................................... 1
Noise Characteristics ................................................................. 15
Product Highlights ........................................................................... 1
Driving the Analog Input of an
Analog-to-Digital Converter .................................................... 15
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Typical Performance Characteristics ............................................. 6
Settling Time ................................................................................... 11
Optimizing Settling Time.......................................................... 11
Op Amp Settling Time—A Mathematical Model.................. 12
Driving a Large Capacitive Load.............................................. 16
Filters................................................................................................ 17
Active Filter Applications.......................................................... 17
Second-Order Low-Pass Filter.................................................. 17
9-Pole Chebychev Filter............................................................. 18
Outline Dimensions ....................................................................... 19
Ordering Guide .......................................................................... 20
Applications Information .............................................................. 14
REVISION HISTORY
8/06—Rev. F to Rev. G
7/02—Rev. D to Rev. E
Edits to Figure 1 ................................................................................ 1
Change to 9-Pole Chebychev Filter Section................................ 18
Edits to Features.................................................................................1
9/01—Rev. C to Rev. D
6/06—Rev. E to Rev. F
Updated Format..................................................................Universal
Deleted B, C, and T Models...............................................Universal
Changes to General Description .................................................... 1
Changes to Product Highlights....................................................... 1
Changes to Specifications Section.................................................. 3
Changes to Figure 43...................................................................... 15
Edits to Features.................................................................................1
Edits to General Description ...........................................................1
Edits to Connection Diagram..........................................................1
Edits to Ordering Guide ...................................................................3
Deleted Metalization Photograph ...................................................3
Edits to Absolute Maximum Ratings .............................................3
Edits to Figure 7.................................................................................9
Edits to Outline Dimensions......................................................... 15
Rev. G | Page 2 of 20
AD712
SPECIFICATIONS
VS = ±15 V @ TA = 25°C, unless otherwise noted. Specifications in boldface are tested on all production units at final electrical test.
Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed, although only those
shown in boldface are tested on all production units.
Table 1.
Parameter
INPUT OFFSET VOLTAGE 1
Initial Offset
TMIN to TMAX
vs. Temp
vs. Supply
TMIN to TMAX
Long-Term Offset Stability
INPUT BIAS CURRENT 2
VCM = 0 V
VCM = 0 V @ TMAX
VCM = ±10 V
INPUT OFFSET CURRENT
VCM = 0 V
VCM = 0 V @ TMAX
MATCHING CHARACTERISTICS
Input Offset Voltage
TMIN to TMAX
Input Offset Voltage Drift
Input Bias Current
Crosstalk
@ f = 1 kHz
@ f = 100 kHz
FREQUENCY RESPONSE
Small Signal Bandwidth
Full Power Response
Slew Rate
Settling Time to 0.01%
Total Harmonic Distortion
INPUT IMPEDANCE
Differential
Common Mode
INPUT VOLTAGE RANGE
Differential 3
Common-Mode Voltage 4
TMIN to TMAX
Common-Mode Rejection
Ratio
VCM = ±10 V
TMIN to TMAX
VCM = ±11 V
TMIN to TMAX
INPUT VOLTAGE NOISE
Min
AD712J/A/S
Typ
0.3
76
76/76/76
Max
Min
0.2
3/1/1
4/2/2
20/20/20
7
95
AD712K
Typ
80
80
15
7
100
mV
mV
μV/°C
dB
dB
μV/month
75
1.7/4.8/77
100
20
0.5
75
1.7
100
pA
nA
pA
10
0.3/0.7/11
25
0.6/1.6/26
5
0.1
25
0.6
pA
nA
1.0
2.0
10
25
mV
mV
μV/°C
pA
4.0
200
20
1.0
0.0003
3.4
18
1.2
120
90
dB
dB
4.0
200
20
1.0
0.0003
MHz
kHz
V/μs
μs
%
1.2
3×1012||5.5
3×1012||5.5
3×1012||5.5
3×1012||5.5
Ω||pF
Ω||pF
±20
+14.5, −11.5
±20
+14.5, −11.5
V
V
V
−VS + 4
76
76/76/76
70
70/70/70
1.0
2.0
10
25
0.6/1.6/26
120
90
16
Unit
15
3/1/1
4/2/2
20/20/20
25
3.0
Max
+VS − 2
88
84
84
80
2
45
22
18
16
−VS + 4
80
80
76
74
Rev. G | Page 3 of 20
+VS − 2
88
84
84
80
2
45
22
18
16
dB
dB
dB
dB
μV p-p
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
AD712
Parameter
INPUT CURRENT NOISE
OPEN-LOOP GAIN
OUTPUT CHARACTERISTICS
Voltage
Current
POWER SUPPLY
Rated Performance
Operating Range
Quiescent Current
Min
150
100/100/100
+13, −12.5
±12/±12/±12
AD712J/A/S
Typ
0.01
400
Max
Min
200
100
+13.9, −13.3
+13.8, −13.1
+25
+13, −12.5
±12
±15
±4.5
+5.0
AD712K
Typ
0.01
400
Max
+13.9, −13.3
+13.8, −13.1
+25
V
V
mA
±15
±18
+6.8
1
±4.5
+5.0
Unit
pA/√Hz
V/mV
V/mV
±18
+6.0
V
V
mA
Input offset voltage specifications are guaranteed after 5 minutes of operation at TA = 25°C.
Bias current specifications are guaranteed maximum at either input after 5 minutes of operation at TA = 25°C. For higher temperatures, the current doubles every 10°C.
Defined as voltage between inputs, such that neither exceeds ±10 V from ground.
4
Typically exceeding −14.1 V negative common-mode voltage on either input results in an output phase reversal.
2
3
Rev. G | Page 4 of 20
AD712
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage
Internal Power Dissipation1
Input Voltage2
Output Short-Circuit Duration
Differential Input Voltage
Storage Temperature Range
Q-Suffix
N-Suffix and R-Suffix
Operating Temperature Range
AD712J/K
AD712A
AD712S
Lead Temperature Range (Soldering 60 sec)
1
2
Thermal characteristics:
8-lead PDIP package:
8-lead CERDIP package:
8-lead SOIC package:
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rating
±18 V
±18 V
Indefinite
+VS and −VS
−65°C to +150°C
−65°C to +125°C
0°C to 70°C
−40°C to +85°C
−55°C to +125°C
300°C
θJA = 165°C/W
θJC = 22°C/W; θJA = 110°C/W
θJA = 100°C/W
For supply voltages less than ±18 V, the absolute maximum voltage is equal
to the supply voltage.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. G | Page 5 of 20
AD712
TYPICAL PERFORMANCE CHARACTERISTICS
15
10
RL = 2kΩ
25°C
5
0
5
10
SUPPLY VOLTAGE ± V
15
20
4
3
2
0
Figure 2. Input Voltage Swing vs. Supply Voltage
INPUT BIAS CURRENT (VCM = 0) (Amps)
15
+VOUT
–VOUT
10
RL = 2kΩ
25°C
5
0
5
10
SUPPLY VOLTAGE ± V
15
20
20
107
108
109
1010
1011
1012
–60
00823-003
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
120
140
Figure 6. Input Bias Current vs. Temperature
100
30
OUTPUT IMPEDANCE (Ω)
25
20
±15V SUPPLIES
15
10
10
1.0
0.1
5
100
1k
LOAD RESISTANCE (Ω)
10k
0.01
1k
00823-004
0
10
10k
100k
FREQUENCY (Hz)
1M
Figure 7. Output Impedance vs. Frequency
Figure 4. Output Voltage Swing vs. Load Resistance
Rev. G | Page 6 of 20
10M
00823-007
OUTPUT VOLTAGE SWING (V)
15
106
Figure 3. Output Voltage Swing vs. Supply Voltage
OUTPUT VOLTAGE SWING (V p-p)
10
SUPPLY VOLTAGE ± V
Figure 5. Quiescent Current vs. Supply Voltage
20
0
5
00823-006
0
5
00823-005
QUIESCENT CURRENT (mA)
6
00823-002
INPUT VOLTAGE SWING (V)
20
AD712
VS = 15V
25°C
50
25
100
80
80
60
60
40
40
GAIN
PHASE
2kΩ
100pF
LOAD
20
20
0
0
–5
0
5
COMMON MODE VOLTAGE (V)
10
–20
00823-008
0
–10
100
1k
10k
100k
1M
–20
10M
Figure 11. Open-Loop Gain and Phase Margin vs. Frequency
125
26
24
120
OPEN-LOOP GAIN (dB)
+ OUTPUT CURRENT
22
20
18
– OUTPUT CURRENT
16
14
115
RL = 2kΩ
25°C
110
105
10
–60
–40
–20
0
20
40
60
80
100
AMBIENT TEMPERATURE (°C)
120
140
95
0
Figure 9. Short-Circuit Current Limit vs. Temperature
5
10
SUPPLY VOLTAGE ± V
15
20
00823-012
100
12
00823-009
Figure 12. Open-Loop Gain vs. Supply Voltage
110
POWER SUPPLY REJECTION (dB)
5.0
4.5
4.0
3.5
100
+ SUPPLY
80
60
– SUPPLY
40
20
VS = ±15V SUPPLIES
WITH 1V p-p SINEWAVE 25°C
–40
–20
0
20
40
60
80
100
120
TEMPERATURE (°C)
140
0
10
00823-010
3.0
–60
100
1k
10k
100k
SUPPLY MODULATION FREQUENCY (Hz)
Figure 10. Unity-Gain Bandwidth vs. Temperature
Figure 13. Power Supply Rejection vs. Frequency
Rev. G | Page 7 of 20
1M
00823-013
SHORT-CIRCUIT CURRENT LIMIT (mA)
10
FREQUENCY (Hz)
Figure 8. Input Bias Current vs. Common-Mode Voltage
UNITY-GAIN BANDWIDTH (MHz)
PHASE MARGIN (Degrees)
OPEN-LOOP GAIN (dB)
INPUT BIAS CURRENT (pA)
MAX J GRADE LIMIT
75
100
00823-011
100
AD712
100
–70
VS = ±15V
VCM = 1V p-p
25°C
80
–80
3V rms
RL = 2kΩ
CL = 100pF
60
THD (dB)
CMR (dB)
–90
40
–100
–110
20
1k
10k
100k
1M
FREQUENCY (Hz)
–130
100
30
100k
1k
INPUT NOISE VOLTAGE (nV/√Hz)
RL = 2kΩ
25°C
VS = ±15V
25
20
15
10
0
100k
1M
FREQUENCY (Hz)
10M
10
1
00823-015
5
100
1
10
100
1k
FREQUENCY (Hz)
10k
100k
Figure 18. Input Noise Voltage Spectral Density
Figure 15. Large Signal Frequency Response
10
25
8
6
20
2
1% 0.1%
SLEW RATE (V/µs)
4
0.01%
0
–2
ERROR 1%
0.1%
0.01%
–4
–6
15
10
5
–8
0.6
0.8
0.7
SETTLING TIME (µs)
0.9
1.0
0
00823-016
–10
0.5
0
100
200
300
400
500
600
700
INPUT ERROR SIGNAL (mV)
(AT SUMMING JUNCTION)
Figure 19. Slew Rate vs. Input Error Signal
Figure 16. Output Swing and Error vs. Settling Time
Rev. G | Page 8 of 20
800
900
00823-019
OUTPUT VOLTAGE SWING (V p-p)
10k
FREQUENCY (Hz)
Figure 17. Total Harmonic Distortion vs. Frequency
Figure 14. Common-Mode Rejection vs. Frequency
OUTPUT SWING FROM 0V TO ±VOLTS
1k
00823-017
100
00823-018
10
00823-014
0
–120
AD712
25
+VS
–
8
VOUT
1/2
20
AD712
+
VIN
4
SQUARE
WAVE
INPUT
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
120
140
RL
2kΩ
CL
100pF
–VS
00823-020
15
–60
0.1µF
00823-023
SLEW RATE (V/µs)
0.1µF
Figure 23. Unity-Gain Follower
Figure 20. Slew Rate vs. Temperature
+VS
0.1µF
+
90
OUTPUT
100pF
2kΩ
0.1µF
00823-021
AD712
–
4
INPUT
100
8
1/2
–VS
10
5V
Figure 21. THD Test Circuit
Figure 24. Unity-Gain Follower Pulse Response (Large Signal)
VOUT
3
–
6
AD712
+
5
8
1/2
1
AD712
+
5kΩ
VIN
CROSSTALK = 20 log
VOUT
10V IN
7
5kΩ
1/2
4
–VS
10
0%
50mV
Figure 22. Crosstalk Test Circuit
100ns
00823-025
20V p-p
–
90
2.2kΩ
00823-022
2
100
20kΩ
+VS
1µs
00823-024
0%
Figure 25. Unity-Gain Follower Pulse Response (Small Signal)
Rev. G | Page 9 of 20
AD712
5kΩ
+VS
0.1µF
100
90
–
5kΩ
8
VOUT
1/2
SQUARE
WAVE
INPUT
4
0.1µF
RL
2kΩ
CL
100pF
00823-026
AD712
+
–VS
10
0%
50mV
Figure 28. Unity-Gain Inverter Pulse Response (Small Signal)
Figure 26. Unity-Gain Inverter
100
90
10
1µs
00823-027
0%
5V
200ns
00823-028
VIN
Figure 27. Unity-Gain Inverter Pulse Response (Large Signal)
Rev. G | Page 10 of 20
AD712
SETTLING TIME
OPTIMIZING SETTLING TIME
In addition to a significant improvement in settling time, the
low offset voltage, low offset voltage drift, and high open-loop
gain of the AD71x family assure 12-bit accuracy over the full
operating temperature range.
Most bipolar high speed digital-to-analog converters (DACs)
have current outputs; therefore, for most applications, an
external op amp is required for a current-to-voltage conversion.
The settling time of the converter/op amp combination depends
on the settling time of the DAC and output amplifier. A good
approximation is
t S Total =
The excellent high speed performance of the AD712 is shown in
the oscilloscope photos in Figure 30 and Figure 31. Measurements were taken using a low input capacitance amplifier
connected directly to the summing junction of the AD712 and
both figures show a worst-case situation: full-scale input
transition. The 4 kΩ [10 kΩ||8 kΩ = 4.4 kΩ] output impedance
of the DAC, together with a 10 kΩ feedback resistor, produce an
op amp noise gain of 3.25. The current output from the DAC
produces a 10 V step at the op amp output (0 to −10 V shown in
Figure 30, and −10 V to 0 V shown in Figure 31).
(tS DAC)2 + (t S AMP )2
The settling time of an op amp DAC buffer varies with the noise
gain of the circuit, the DAC output capacitance, and the amount
of external compensation capacitance across the DAC output
scaling resistor.
Settling time for a bipolar DAC is typically 100 ns to 500 ns.
Previously, conventional op amps have required much longer
settling times than have typical state-of-the-art DACs; therefore,
the amplifier settling time has been the major limitation to a
high speed, voltage output, digital-to-analog function. The
introduction of the AD71x family of op amps with their 1 μs (to
±0.01% of final value) settling time permits the full high speed
capabilities of most modern DACs to be realized.
Therefore, with an ideal op amp, settling to ±1/2 LSB (±0.01%)
requires that 375 μV or less appears at the summing junction.
This means that the error between the input and output (that
voltage which appears at the AD712 summing junction) must
be less than 375 μV. As shown in Figure 30, the total settling
time for the AD712/AD565A combination is 1.2 microseconds.
0.1µF
BIPOLAR
OFFSET ADJUST
R2
GAIN 100Ω
ADJUST
REF
OUT
VCC
BIPOLAR
OFF
20V
SPAN
+
10V
REF
IN
R1
100Ω
AD565A
–
19.95kΩ
5kΩ
9.95kΩ
10V
SPAN
0.5mA
5kΩ
IREF
DAC
REF
GND
20kΩ
IOUT = 4 ×
IREF × CODE
IO
10pF
DAC
OUT
8kΩ
+15V
0.1µF
–
8
1/2
4
POWER
GND
MSB
LSB
0.1µF
–15V
00823-029
–VEE
0.1µF
OUTPUT
–10V TO +10V
AD712
+
Figure 29. ±10 V Voltage Output Bipolar DAC
Rev. G | Page 11 of 20
AD712
Where
1mV
5V
ωO
= unity-gain frequency of the op amp.
2π
100
90
GN = noise gain of circuit ⎛⎜1 + R ⎞⎟ .
⎟
⎜
SUMMING
JUNCTION
⎝
RO ⎠
This equation can then be solved for Cf
0V
OUTPUT
10
CX =
–10V
500ns
00823-030
0%
(2)
In these equations, Capacitance CX is the total capacitance
appearing at the inverting terminal of the op amp. When
modeling a DAC buffer application, the Norton equivalent
circuit shown in Figure 32 can be used directly; Capacitance CX
is the total capacitance of the output of the DAC plus the input
capacitance of the op amp (because the two are in parallel).
Figure 30. Settling Characteristics for AD712 with AD565A,
Full-Scale Negative Transition
1mV
RC X ωO + (1 − GN )
2 − GN
+2
RωO
RωO
5V
100
90
+ 1/2
AD712
–
SUMMING
JUNCTION
0V
VOUT
CF
RL
CL
R
OUTPUT
IO
0%
RO
CX
00823-032
10
Figure 32. Simplified Model of the AD712 Used as a Current-Out DAC Buffer
Figure 31. Settling Characteristics for AD712 with AD565A,
Full-Scale Positive Transition
OP AMP SETTLING TIME—A MATHEMATICAL
MODEL
The design of the AD712 gives careful attention to optimizing
individual circuit components; in addition, a careful trade-off
was made: the gain bandwidth product (4 MHz) and slew rate
(20 V/μs) were chosen to be high enough to provide very fast
settling time but not too high to cause a significant reduction
in phase margin (and therefore, stability). Thus designed, the
AD712 settles to ±0.01%, with a 10 V output step, in under 1 μs,
while retaining the ability to drive a 250 pF load capacitance
when operating as a unity-gain follower.
If an op amp is modeled as an ideal integrator with a unity-gain
crossover frequency of ωO/2π, then Equation 1 accurately
describes the small signal behavior of the circuit of Figure 32,
consisting of an op amp connected as an I-to-V converter at the
output of a bipolar or CMOS DAC. This equation would completely describe the output of the system if not for the finite slew
rate and other nonlinear effects of the op amp.
VO
−R
=
I IN
R(C X ) 2 ⎛ G N
s + ⎜⎜
+ RC f
ωO
⎝ ωO
⎞
⎟ s +1
⎟
⎠
(1)
When RO and IO are replaced with their Thevenin VIN and RIN
equivalents, the general-purpose inverting amplifier shown in
Figure 33 is created. Note that when using this general model,
Capacitance CX is either the input capacitance of the op amp, if
a simple inverting op amp is being simulated, or the combined
capacitance of the DAC output and the op amp input if the
DAC buffer is being modeled.
+ 1/2
AD712
–
RIN
VIN
VOUT
CF
RL
CL
R
CX
00823-033
500ns
00823-031
–10V
Figure 33. Simplified Model of the AD712 Used as an Inverter
In either case, Capacitance CX causes the system to go from a
one-pole to a two-pole response; this additional pole increases
settling time by introducing peaking or ringing in the op amp
output. Because the value of CX can be estimated with reasonable
accuracy, Equation 2 can be used to choose a small capacitor
(CF) to cancel the input pole and optimize amplifier response.
Figure 34 is a graphical solution of Equation 2 for the AD712
with R = 4 kΩ.
Rev. G | Page 12 of 20
AD712
60
5V
50
100
90
GN = 4.0
30
GN = 3.0
GN = 2.0
20
10
GN = 1.5
10
GN = 1.0
0
10
20
30
CF
40
5mV
60
50
00823-034
0
0%
500ns
Figure 36. Settling Characteristics 0 V to −10 V Step
Upper Trace: Output of AD712 Under Test (5 V/Div)
Lower Trace: Amplified Error Voltage (0.01%/Div)
Figure 34. Value of Capacitor CF vs. Value of CX
The input of the settling time fixture is driven by a flat top pulse
generator. The error signal output from the false summing node
of A1 is clamped, amplified by A2, and then clamped again. The
error signal is thus clamped twice: once to prevent overloading
Amplifier A2 and then a second time to avoid overloading the
oscilloscope preamp. The Tektronix oscilloscope preamp type
7A26 was carefully chosen because it does not overload with
these input levels. Amplifier A2 needs to be a very high speed
FET-input op amp; it provides a gain of 10, amplifying the error
signal output of A1.
The photos of Figure 35 and Figure 36 show the dynamic
response of the AD712 in the settling test circuit of Figure 37.
5V
100
90
10
00823-035
0%
5mV
500ns
Figure 35. Settling Characteristics 0 V to +10 V Step
Upper Trace: Output of AD712 Under Test (5 V/Div)
Lower Trace: Amplified Error Voltage (0.01%/Div)
5pF
HP2835
+ 1/2
AD712
–
205Ω
VERROR × 5
TEKTRONIX 7A26
OSCILLOSCOPE
PREAMP
INPUT SECTION
1MΩ
20pF
HP2835
0.47µF
200Ω
DATA
DYNAMICS
5109
5 TO 18pF
10kΩ
VIN
10kΩ
0.47µF
4.99kΩ
4.99kΩ
–15V +15V
10kΩ
1.1kΩ
0.2 TO 0.6pF
–
1/2
AD712
+
0.1µF
VOUT
5kΩ
10pF
0.1µF
00823-037
(OR EQUIVALENT
FLAT TOP PULSE
GENERATION)
00823-036
CX
40
–15V +15V
Figure 37. Settling Time Test Circuit
Rev. G | Page 13 of 20
AD712
APPLICATIONS INFORMATION
GUARDING
Figure 39 and Figure 40 show the AD712 and AD7545 (12-bit
CMOS DAC) configured for unipolar binary (2-quadrant multiplication) or bipolar (4-quadrant multiplication) operation.
Capacitor C1 provides phase compensation to reduce overshoot
and ringing.
The low input bias current (15 pA) and low noise characteristics
of the AD712 BiFET op amp make it suitable for electrometer
applications such as photo diode preamplifiers and picoampere
current-to-voltage converters. The use of a guarding technique,
such as that shown in Figure 38, in printed circuit board layout
and construction is critical to minimize leakage currents. The
guard ring is connected to a low impedance potential at the
same level as the inputs. High impedance signal lines should
not be extended for any unnecessary length on the printed
circuit board.
VDD
R2A*
+15V
C1A
33pF
GAIN
ADJUST
VIN
R1A*
RFB
VDD
–
OUT1
VREF
0.1µF
1/2
AD7545
+AD712
AGND
DGND
PDIP (N), CERDIP (Q),
AND SOIC (R) PACKAGES.
ANALOG
COMMON
*REFER TO
TABLE 3
DB11 TO DB0
4
5
R2B*
VDD
6
C1B
33pF
00823-038
7
8
1
GAIN
ADJUST
VIN
Figure 38. Board Layout for Guarding Inputs
R1B*
RFB
VDD
–
OUT1
VREF
1/2
AD7545
+AD712
AGND
DGND
DIGITAL-TO-ANALOG CONVERTER APPLICATIONS
The AD712 is an excellent output amplifier for CMOS DACs. It
can be used to perform both 2-quadrant and 4-quadrant
operations. The output impedance of a DAC using an inverted
R-2R ladder approaches R for codes containing many 1s, and 3R
for codes containing a single 1. For codes containing all 0s, the
output impedance is infinite.
0.1µF
ANALOG
COMMON
*REFER TO
TABLE 3
–15V
DB11 TO DB0
Figure 39. Unipolar Binary Operation
R1 and R2 calibrate the zero offset and gain error of the DAC.
Specific values for these resistors depend upon the grade of
AD7545 and are listed in Table 3.
Table 3. Recommended Trim Resistor Values vs. Grades of
the AD7545 for VDD = 5 V
For example, the output resistance of the AD7545 modulates
between 11 kΩ and 33 kΩ. Therefore, with an 11 kΩ DAC
internal feedback resistance, the noise gain varies from 2 to 4/3.
This changing noise gain modulates the effect of the input offset
voltage of the amplifier, resulting in nonlinear DAC amplifier
performance.
Trim
Resistor
R1
R2
JN/AQ
500 Ω
150 Ω
KN/BQ
200 Ω
68 Ω
LN
100 Ω
33 Ω
The AD712K with guaranteed 700 μV offset voltage minimizes
this effect to achieve 12-bit performance.
R2*
+15V
C1
33pF
GAIN
ADJUST
R1*
VDD
RFB
OUT1
VREF AD7545
AGND
DB11 TO DB0
R4
20kΩ 1%
0.1µF
R5
20kΩ 1%
–
–
1/2
+AD712
DGND
R3
10kΩ 1%
1/2
AD712
+
0.1µF
12
DATA INPUT
*FOR VALUES OF
R1 AND R2 SEE TABLE 3
VOUT
ANALOG
COMMON
Figure 40. Bipolar Operation
Rev. G | Page 14 of 20
–15V
00823-040
VDD
VOUTB
00823-039
3
2
VIN
VOUTA
GLN
20 Ω
6.8 Ω
AD712
Figure 41 and Figure 42 show the settling time characteristics of
the AD712 when used as a DAC output buffer for the AD7545.
An op amp driving the analog input of an ADC, such as that
shown in Figure 43, must be capable of maintaining a constant
output voltage under dynamically changing load conditions. In
successive approximation converters, the input current is compared to a series of switched trial currents. The comparison
point is diode clamped, but can deviate several hundred millivolts
resulting in high frequency modulation of analog-to-digital
input current. The output impedance of a feedback amplifier
is made artificially low by the loop gain. At high frequencies,
where the loop gain is low, the amplifier output impedance
can approach its open-loop value. Most IC amplifiers exhibit a
minimum open-loop output impedance of 25 Ω due to currentlimiting resistors.
100
90
10
500ns
00823-041
0%
Figure 41. Positive Settling Characteristics for AD712 with AD7545
12/8
1mV
CS
AO
100
90
GAIN
ADJUST
+15V
10
0.1µF
OFFSET
ADJUST
–
0%
R2
100Ω
R1
100Ω
5V
500ns
00823-042
1/2
±10V
ANALOG
INPUT
REF OUT
BIP OFF
10VIN
AC
0.1µF
HIGH
BITS
R/C AD574A
MIDDLE
CE
BITS
REF IN
20VIN
AD712
+
STS
LOW
BITS
+5V
+15V
–15V
DC
Figure 42. Negative Settling Characteristics for AD712 with AD7545
–15V
NOISE CHARACTERISTICS
ANALOG COM
00823-043
1mV
5V
DRIVING THE ANALOG INPUT OF AN
ANALOG-TO-DIGITAL CONVERTER
Figure 43. AD712 as An ADC Unity-Gain Buffer
The random nature of noise, particularly in the flicker noise
region, makes it difficult to specify in practical terms. At the
same time, designers of precision instrumentation require
certain guaranteed maximum noise levels to realize the full
accuracy of their equipment. All grades of the AD712 are sample
tested on an AQL basis to a limit of 6 μV p-p, 0.1 Hz to 10 Hz.
A few hundred microamps reflected from the change in converter loading can introduce errors in instantaneous input
voltage. If the analog-to-digital conversion speed is not excessive
and the bandwidth of the amplifier is sufficient, the amplifier
output returns to the nominal value before the converter makes
its comparison. However, many amplifiers have relatively narrow
bandwidth yielding slow recovery from output transients. The
AD712 is ideally suited to drive high speed analog-to-digital
converters because it offers both wide bandwidth and high
open-loop gain.
Rev. G | Page 15 of 20
AD712
DRIVING A LARGE CAPACITIVE LOAD
PD711 BUFF
1mV
The circuit in Figure 46 uses a 100 Ω isolation resistor that
enables the amplifier to drive capacitive loads exceeding
1500 pF; the resistor effectively isolates the high frequency
feedback from the load and stabilizes the circuit. Low frequency
feedback is returned to the amplifier summing junction via the
low-pass filter formed by the 100 Ω series resistor and the Load
Capacitance CL. Figure 47 shows a typical transient response for
this connection.
100
90
10
0%
200ns
–10V ADC IN
4.99kΩ
00823-044
500mV
30pF
+VIN
Figure 44. ADC Input Unity Gain Buffer Recovery Times, −10 V ADC IN
0.1µF
+ –
–
4.99kΩ
1/2
INPUT
100
TYPICAL CAPACITANCE
LIMIT FOR VARIOUS
LOAD RESISTORS
90
R1
C1 UP TO
2kΩ
10kΩ
20Ω
1500pF
1500pF
1000pF
+AD712
100Ω
C1
OUTPUT
R1
0.1µF
– +
–VIN
00823-046
PD711 BUFF
1mV
Figure 46. Circuit for Driving a Large Capacitive Load
10
0%
–5V ADC IN
200ns
5V
00823-045
500mV
1µs
100
Figure 45. ADC Input Unity Gain Buffer Recovery Times, −5 V ADC IN
90
10
00823-047
0%
Figure 47. Transient Response RL = 2 kΩ, CL = 500 pF
Rev. G | Page 16 of 20
AD712
FILTERS
C1
560pF
ACTIVE FILTER APPLICATIONS
In active filter applications using op amps, the dc accuracy of
the amplifier is critical to optimal filter performance. The
amplifier offset voltage and bias current contribute to output
error. Offset voltage is passed by the filter and can be amplified
to produce excessive output offset. For low frequency
applications requiring large value input resistors, bias currents
flowing through these resistors also generate an offset voltage.
+15V
R1
20kΩ
VIN
R2
20kΩ
0.1µF
+
1/2
AD712
–
C2
280pF
VOUT
00823-048
0.1µF
–15V
In addition, at higher frequencies, the op amp dynamics must
be carefully considered. Here, slew rate, bandwidth, and openloop gain play a major role in op amp selection. The slew rate
must be fast as well as symmetrical to minimize distortion. The
amplifier bandwidth in conjunction with the filter gain dictates
the frequency response of the filter.
The use of a high performance amplifier such as the AD712
minimizes both dc and ac errors in all active filter applications.
SECOND-ORDER LOW-PASS FILTER
Figure 48 depicts the AD712 configured as a second-order,
Butterworth low-pass filter. With the values as shown, the
corner frequency is 20 kHz; however, the wide bandwidth of the
AD712 permits a corner frequency as high as several hundred
kilohertz. Equations for component selection are as follows:
Figure 48. Second-Order Low-Pass Filter
An important property of filters is their out-of-band rejection.
The simple 20 kHz low-pass filter shown in Figure 48, can be
used to condition a signal contaminated with clock pulses or
sampling glitches that have considerable energy content at high
frequencies.
The low output impedance and high bandwidth of the AD712
minimize high frequency feedthrough as shown in Figure 49.
The upper trace is that of another low cost BiFET op amp
showing 17 dB more feedthrough at 5 MHz.
REF 20.0 dBm
10dB/DIV
RANGE 15.0dBm
OFFSET .0 Hz
0dB
R1 = R2 = A user selected value (10 kΩ to 100 kΩ, typical)
C2 =
TYPICAL BIFET
1.414
(2π) f cutoff (R1)
(
)
AD712
0.707
(2π) f cutoff (R1)
(
)
CENTER 5 000 000.0Hz
RBW 30kHz
VBW 30kHz
SPAN 10 000 000.0Hz
ST .8 SEC
Figure 49. High Frequency Feedthrough
Rev. G | Page 17 of 20
00823-049
C1 (in farads) =
AD712
+15V
0.1µF
+
0.001µF
A1
AD711
–
2800Ω
4.9395E –15
0.1µF
A
–15V
100kΩ
*
6190Ω
6490Ω
6190Ω
5.9276E –15
5.9276E –15
4.9395E –15
+
B
+
*
C
2800Ω
D
+
*
+
A2
AD711
–
+
*
0.001µF
VOUT
0.1µF
4.99kΩ
–15V
124kΩ
4.99kΩ
*SEE TEXT
00823-050
VIN
+15V
0.1µF
Figure 50. 9-Pole Chebychev Filter
9-POLE CHEBYCHEV FILTER
Figure 50 and Figure 51 show the AD712 and its dual
counterpart, the AD711, as a 9-pole Chebychev filter using
active frequency dependent negative resistors (FDNRs). With a
cutoff frequency of 50 kHz and better than 90 dB rejection, it
can be used as an antialiasing filter for a 12-bit data acquisition
system with 100 kHz throughput.
0.001µF
R
–
+
1/2
AD712
–
0.1µF
1/2
AD712
+
0.001µF
–15V
1.0kΩ
24.9kΩ FOR 4.9395E –15
29.4kΩ FOR 5.9276E –15
4.99kΩ
00823-051
R:
Figure 51. FDNR for 9-Pole Chebychev Filter
REF 5.0dBm
10dB/DIV
RANGE –5.0dBm
START.0Hz
RBW 300Hz
VBW 30Hz
MARKER 96 800.0Hz
–90dBm
STOP 200 000.0Hz
ST 69.6 SEC
00823-052
As shown in Figure 50, the filter is comprised of four FDNRs
(A, B, C, D) having values of 4.9395 × 10−15 and 5.9276 × 10–15
farad-seconds. Each FDNR active network provides a two-pole
response for a total of eight poles. The ninth pole consists of a
0.001 μF capacitor and a 124 kΩ resistor at Pin 3 of Amplifier A2.
Figure 51 depicts the circuits for each FDNR with the proper
selection of R. To achieve optimal performance, the 0.001 μF
capacitors must be selected for 1% or better matching and all
resistors should have 1% or better tolerance.
+15V
0.1µF
+
Figure 52. High Frequency Response for 9-Pole Chebychev Filter
Rev. G | Page 18 of 20
AD712
OUTLINE DIMENSIONS
0.400 (10.16)
0.365 (9.27)
0.355 (9.02)
8
1
5
4
0.005 (0.13)
MIN
8
0.280 (7.11)
0.250 (6.35)
0.240 (6.10)
0.100 (2.54)
BSC
1
0.060 (1.52)
MAX
0.015 (0.38)
GAUGE
PLANE
SEATING
PLANE
0.005 (0.13)
MIN
0.060 (1.52)
0.015 (0.38)
0.200 (5.08)
MAX
0.150 (3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
0.070 (1.78)
0.060 (1.52)
0.045 (1.14)
0.070 (1.78)
0.030 (0.76)
5.00 (0.1968)
4.80 (0.1890)
5
6.20 (0.2440)
4 5.80 (0.2284)
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0040)
15°
0°
0.015 (0.38)
0.008 (0.20)
Figure 54. 8-Lead Ceramic Dual In-Line Package [CERDIP]
(Q-8)
Dimensions shown in inches and (millimeters)
Figure 53. 8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
Dimensions shown in inches and (millimeters)
8
SEATING
PLANE
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
COMPLIANT TO JEDEC STANDARDS MS-001-BA
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
4.00 (0.1574)
3.80 (0.1497) 1
0.320 (8.13)
0.290 (7.37)
0.405 (10.29) MAX
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
0.430 (10.92)
MAX
4
0.100 (2.54) BSC
0.195 (4.95)
0.130 (3.30)
0.115 (2.92)
0.015
(0.38)
MIN
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
5
0.310 (7.87)
0.220 (5.59)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
PIN 1
0.210
(5.33)
MAX
0.150 (3.81)
0.130 (3.30)
0.115 (2.92)
0.055 (1.40)
MAX
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
COPLANARITY
SEATING 0.31 (0.0122)
0.10
PLANE
0.50 (0.0196)
× 45°
0.25 (0.0099)
8°
0.25 (0.0098) 0° 1.27 (0.0500)
0.40 (0.0157)
0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012-AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 55. 8-Lead Standard Small Outline Package [SOIC_N]
Narrow Body
(R-8)
Dimensions shown in millimeters and (inches)
Rev. G | Page 19 of 20
AD712
ORDERING GUIDE
Model
AD712AQ
AD712JN
AD712JNZ1
AD712JR
AD712JR-REEL
AD712JR-REEL7
AD712JRZ1
AD712JRZ-REEL1
AD712JRZ-REEL71
AD712KN
AD712KNZ1
AD712KR
AD712KR-REEL
AD712KR-REEL7
AD712KRZ1
AD712KRZ-REEL1
AD712KRZ-REEL71
AD712SQ/883B
1
Temperature Range
−40°C to +85°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
0°C to 70°C
−55°C to +125°C
Package Description
8-Lead CERDIP
8-Lead PDIP
8-Lead PDIP
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead PDIP
8-Lead PDIP
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead CERDIP
Z = Pb-free part.
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00823-0-8/06(G)
Rev. G | Page 20 of 20
Package Option
Q-8
N-8
N-8
R-8
R-8
R-8
R-8
R-8
R-8
N-8
N-8
R-8
R-8
R-8
R-8
R-8
R-8
Q-8