LINER LT1725 General purpose isolated flyback controller Datasheet

LT1725
General Purpose
Isolated Flyback Controller
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FEATURES
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DESCRIPTIO
Drives External Power MOSFET with External
ISENSE Resistor
Application Input Voltage Limited Only by
External Power Components
Senses Output Voltage Directly from Primary Side
Winding—No Optoisolator Required
Accurate Regulation Without User Trims
Regulation Maintained Well into Discontinuous Mode
Switching Frequency from 50kHz to 250kHz with
External Capacitor
Optional Load Compensation
Optional Undervoltage Lockout
Available in 16-Pin SO and SSOP Packages
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APPLICATIO S
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Telecom Isolated Converters
Offline Isolated Power Supplies
Instrumentation Power Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
The LT®1725 is a monolithic switching regulator controller specifically designed for the isolated flyback topology.
It drives the gate of an external MOSFET and is generally
powered from a third transformer winding. These features
allow for an application input voltage limited only by
external power path components. The third transformer
winding also provides output voltage feedback information, such that an optoisolator is not required. Its gate
drive capability coupled with a suitable external MOSFET
can deliver load power up to tens of watts.
The LT1725 has a number of features not found on most
other switching regulator ICs. By utilizing current mode
switching techniques, it provides excellent AC and DC line
regulation. Its unique control circuitry can maintain regulation well into discontinuous mode in most applications.
Optional load compensation circuitry allows for improved
load regulation. An optional undervoltage lockout pin
halts operation when the application input voltage is too
low. An optional external capacitor implements a softstart function. A 3V output is available at up to several mA
for powering primary side application circuitry.
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TYPICAL APPLICATIO
48V to Isolated 5V Converter
Output Load Regulation
CTX02-14989
6
LT1725
BAS16
3VOUT
FB
3.01k
1%
18Ω
VIN 36V TO 72V
VIN = 36V
22Ω
47k
1nF
47pF
SFST
VCC
+
2
15µF
VC
1µF
UVLO
51k
33k
1.5µF
12
51k
2.7k
0.1µF
ROCMP
GATE
RCMPC
ISENSE
SGND
PGND
51Ω
1W
VIN = 72V
5.00
+
150µF
4
MENAB
12CWQ06
10
100pF
ENDLY
VOUT = 5V
IOUT = 0 to 2A
VIN = 48V
11
150pF
820k
51k
9
68Ω
OSCAP
tON
5.25
1
VOUT (V)
35.7k
1%
470pF
IRF620
4.75
0
0.5
1.0
ILOAD (A)
1.5
2.0
1725 F10b
0.18Ω
1725 TA01a
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LT1725
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AXI U
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
VCC Supply Voltage ................................................. 22V
UVLO Pin Voltage .................................................... VCC
ISENSE Pin Voltage .................................................... 2V
FB Pin Current ..................................................... ±2mA
Operating Junction
Temperature Range
LT1725C .............................................. 0°C to 100°C
LT1725I ........................................... –40°C TO 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................ 300°C
ORDER PART
NUMBER
TOP VIEW
PGND 1
16 GATE
ISENSE 2
15 VCC
SFST 3
14 tON
ROCMP 4
13 ENDLY
RCMPC 5
12 MINENAB
OSCAP 6
11 SGND
VC 7
10 UVLO
FB 8
9
LT1725CGN
LT1725CS
LT1725IGN
LT1725IS
GN PART
MARKING
3VOUT
GN PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
1725
1725I
TJMAX = 125°C, θJA = 110°C/W (GN)
TJMAX = 125°C, θJA = 100°C/W (SO)
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 14V, GATE open, VC = 1.4V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
●
●
●
14.0
8
4.0
15.1
9.7
5.4
16.0
11
6.5
V
V
V
●
●
6
10
120
15
280
mA
µA
1.230
1.220
1.245
●
1.260
1.270
V
V
●
400
1000
1800
µmho
●
30
50
80
µA
0.05
%/V
Power Supply
VCC
VCC Turn-On Voltage
VCC Turn-Off Voltage
VCC Hysteresis (Note 3)
ICC
Supply Current
Start-Up Current
(VTURN-ON – VTURN-OFF)
VC = Open
Feedback Amplifier
VFB
Feedback Voltage
IFB
Feedback Pin Input Current
gm
Feedback Amplifier Transconductance
ISRC, ISNK
Feedback Amplifier Source or Sink Current
VCL
Feedback Amplifier Clamp Voltage
500
∆lC = ±10µA
nA
2.5
Reference Voltage/Current Line Regulation
12V ≤ VIN ≤ 18V
Voltage Gain
VC = 1V to 2V
0.01
●
V
2000
V/V
50
µA
Soft-Start Charging Current
VSFST = 0V
25
40
Soft-Start Discharge Current
VSFST = 1.5V, VUVLO = 0V
0.8
1.5
mA
Output High Level
IGATE = 100mA
IGATE = 500mA
●
●
11.5
11.0
12.1
11.8
V
V
Output Low Level
IGATE = 100mA
IGATE = 500mA
●
●
●
Gate Output
VGATE
0.3
0.6
1.2
0.45
1.0
V
V
IGATE
Output Sink Current in Shutdown, VUVLO = 0V
VGATE = 2V
2.5
mA
tr
Rise Time
CL = 1000pF
30
ns
tf
Fall Time
CL = 1000pF
30
ns
1725f
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LT1725
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 14V, GATE open, VC = 1.4V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.90
0.80
1.12
●
1.25
1.35
V
V
220
200
250
●
270
280
220
mV
mV
mV
0.30
mV
Current Amplifier
VC
VISENSE
Control Pin Threshold
Switch Current Limit
Duty Cycle = Min
Duty Cycle ≤ 30%
Duty Cycle ≤ 30%
Duty Cycle = 80%
∆VISENSE/∆VC
Timing
f
Switching Frequency
COSCAP = 100pF
●
90
80
100
kHz
kHz
200
pF
COSCAP
Oscillator Capacitor Value
(Note 2)
tON
Minimum Switch On Time
RtON = 50k
tED
Flyback Enable Delay Time
RENDLY = 50k
200
ns
tEN
Minimum Flyback Enable Time
RMENAB = 50k
200
ns
Rt
Timing Resistor Value
(Note 2)
Maximum Switch Duty Cycle
33
115
125
200
24
●
85
ns
200
90
kΩ
%
Load Compensation
Sense Offset Voltage
Current Gain Factor
2
5
mV
0.80
0.95
1.05
mV
1.21
1.25
1.29
V
– 0.25
– 4.50
+ 0.1
– 3.5
+ 0.25
– 2.50
µA
µA
2.8
3.0
3.2
UVLO Function
VUVLO
UVLO Pin Threshold
IUVLO
UVLO Pin Bias Current
●
VUVLO = 1.2V
VUVLO = 1.3V
3V Output Function
VREF
Reference Output Voltage
ILOAD = 1mA
●
Output Impedance
Current Limit
●
8
V
10
Ω
15
mA
Note 1: Absolute Maximum Ratings are those values beyond which the life of
a device may be impaired.
Note 2: Component value range guaranteed by design.
Note 3: The VCC turn-on/turn-off voltages and hysteresis voltage are
proportional in magnitude to each other-guaranteed by design.
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LT1725
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TYPICAL PERFOR A CE CHARACTERISTICS
VCC Hysteresis Voltage vs
Temperature
6.50
15.75
6.25
VCC TURN-ON VOLTAGE (V)
VCC HYSTERESIS VOLTAGE (V)
16.00
15.50
15.25
15.00
14.75
6.00
5.75
5.50
5.25
14.25
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
4.75
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
150
100
50
100
0
–50
125
13
115
50
25
0
75
TEMPERATURE (°C)
–1
–2
–4
–5
–6
–50 –25
125
100
VUVLO = 1.3V
–3
50
25
75
0
TEMPERATURE (°C)
1725 G04
100
105
100
95
90
–0.5
VCC-VGATE (V)
–1.0
0.6
TA = 25°C
TA = 125°C
TA = 25°C
–1.5
0.4
–2.0
TA = –55°C
0.2
10
100
1000
ISINK (mA)
1725 G07
TA = –55°C
–2.5
–3.0
0
1
125
VC Clamp Voltage, Switching
Threshold vs Temperature
0
1.0
TA = 125°C
100
1725 G06
VCC-VGATE vs ISOURCE
0.8
50
25
75
0
TEMPERATURE (°C)
1725 G05
VGATE vs ISINK
1
110
85
–50 –25
125
10
100
ISOURCE (mA)
1000
1725 G08
VC CLAMP VOLTAGE, SWITCHING THRESHOLD (V)
8
–50 –25
OSCILLATOR FREQUENCY (kHz)
9
125
VUVLO = 1.2V
0
UVLO PIN INPUT CURRENT (µA)
10
100
Oscillator Frequency vs
Temperature
1
11
50
0
75
25
TEMPERATURE (°C)
1725 G03
UVLO Pin Input Current vs
Temperature
Supply Current vs Temperature
12
–25
1725 G02
1725 G01
SUPPLY CURRENT (mA)
200
5.00
14.50
VGATE (V)
Start-Up Current vs Temperature
250
START-UP CURRENT (µA)
VCC Turn-On Voltage vs
Temperature
3.0
CLAMP VOLTAGE
2.5
2.0
1.5
1.0
SWITCHING THRESHOLD
0.5
0
–50
–25
50
25
75
0
TEMPERATURE (°C)
100
125
1725 G09
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LT1725
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TYPICAL PERFOR A CE CHARACTERISTICS
Minimum Switch-On Time vs
Temperature
275
MINIMUM ENABLE TIME (ns)
250
225
200
175
150
250
250
225
200
175
50
25
75
0
TEMPERATURE (°C)
100
125
–50
125
–25
50
25
75
0
TEMPERATURE (°C)
100
1725 G10
80
60
40
TA = 25°C
TA = –55°C
0
–20
TA = 125°C
–40
–60
–80
1.05 1.10
1.15 1.20 1.25 1.30 1.35 1.40
FB PIN VOLTAGE (V)
FEEDBACK AMPLIFIER TRANSCONDUCTANCE (µmho)
Feedback Amplifier Output Current
vs FB Pin Voltage
20
200
175
125
125
–50
–25
50
25
75
0
TEMPERATURE (°C)
125
Feedback Amplifier
Transconductance vs Temperature
1600
1400
1200
1000
800
600
400
200
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1725 G14
1725 G13
Soft-Start Charging Current vs
Temperature
Soft-Start Sink Current vs
Temperature
2.5
60
V(SFST) = 0V
50
40
30
20
10
0
–50
100
1725 G12
1725 G11
V(SFST) = 1.5V
SOFT-START SINK CURRENT (mA)
–25
225
150
150
FEEDBACK AMPLIFIER OUTPUT CURRENT (µA)
125
–50
275
RMINENAB = 50k
ENABLE DELAY TIME (ns)
RTON = 50k
SOFT-START CHARGING CURRENT (µA)
MINIMUM SWITCH-ON TIME (ns)
275
Enable Delay Time vs
Temperature
Minimum Enable Time vs
Temperature
–25
50
25
75
0
TEMPERATURE (°C)
100
125
1725 G15
2.0
1.5
1.0
0.5
0
–50
–25
50
0
75
25
TEMPERATURE (°C)
100
125
1725 G16
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LT1725
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PI FU CTIO S
PGND (Pin 1): The power ground pin carries the GATE
node discharge current. This is typically a current spike of
several hundred mA with a duration of tens of nanoseconds. It should be connected directly to a good quality
ground plane.
ISENSE (Pin 2): Pin to measure switch current with external sense resistor. The sense resistor should be of a
noninductive construction as high speed performance is
essential. Proper grounding technique is also required to
avoid distortion of the high speed current waveform. A
preset internal limit of nominally 250mV at this pin effects
a switch current limit.
SFST (Pin 3): Pin for optional external capacitor to effect
soft-start function. See Applications Information for details.
ROCMP (Pin 4): Input pin for optional external load compensation resistor. Use of this pin allows nominal compensation for nonzero output impedance in the power transformer
secondary circuit, including secondary winding impedance,
output Schottky diode impedance and output capacitor
ESR. In less demanding applications, this resistor is not
needed. See Applications Information for more details.
RCMPC (Pin 5): Pin for external filter capacitor for optional
load compensation function. A common 0.1µF ceramic
capacitor will suffice for most applications. See Applications Information for further details.
OSCAP (Pin 6): Pin for external timing capacitor to set
oscillator switching frequency. See Applications Information for details.
VC (pin 7): This is the control voltage pin which is the
output of the feedback amplifier and the input of the
current comparator. Frequency compensation of the
overall loop is effected in most cases by placing a capacitor between this node and ground.
FB (Pin 8): Input pin for external “feedback” resistor
divider. The ratio of this divider, times the internal
bandgap (VBG) reference, times the effective output-to-
third winding transformer turns ratio is the primary determinant of the output voltage. The Thevenin equivalent
resistance of the feedback divider should be roughly 3k.
See Applications Information for more details.
3VOUT (Pin 9): Output pin for nominal 3V reference. This
facilitates various user applications. This node is internally
current limited for protection and is intended to drive
either moderate capacitive loads of several hundred pF or
less, or, very large capacitive loads of 0.1µF or more. See
Applications Information for more details.
UVLO (Pin 10): This pin allows the use of an optional
external resistor divider to set an undervoltage lockout
based upon VIN (not VCC) level. (Note: If the VCC voltage is
sufficient to allow the part to start up, but the UVLO pin is
held below its threshold, output switching action will be
disabled, but the part will draw its normal quiescent
current from VCC. This typically causes a benign relaxation
oscillation action on the VCC pin in the conventional
“trickle-charge” bootstrapped configuration.)
The bias current on this pin is a function of the state of the
UVLO comparator; as the threshold is exceeded, the bias
current increases. This creates a hysteresis band equal to
the change in bias current times the Thevenin impedance
of the user’s resistive divider. The user may thereby adjust
the impedance of the UVLO divider to achieve a desired
degree of hysteresis. A 100pF capacitor to ground is
recommended on this pin. See Applications Information
for details.
SGND (Pin 11): The signal ground pin is a clean ground.
The internal reference, oscillator and feedback amplifier
are referred to it. Keep the ground path connection to the
FB pin, OSCAP capacitor and the VC compensation capacitor free of large ground currents.
MINENAB (Pin 12): Pin for external programming resistor
to set minimum enable time. See Applications Information
for details.
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LT1725
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ENDLY (Pin 13): Pin for external programming resistor to
set enable delay time. See Applications Information for
details.
tON (Pin 14): Pin for external programming resistor to set
switch minimum on time. See Applications Information
for details.
VCC (Pin 15): Supply voltage for the LT1725. Bypass this
pin to ground with 1µF or more.
GATE (Pin 16): This is the gate drive to the external power
MOSFET switch and has large dynamic currents flowing
through it. Keep the trace to the MOSFET as short as
possible to minimize electromagnetic radiation and voltage spikes. A series resistance of 5Ω or more may help to
dampen ringing in less than ideal layouts.
W
BLOCK DIAGRA
VCC
3VOUT
UVLO
BIAS
3V REG
(INTERNAL)
tON
MINENAB
ENDLY
OSCAP
GATE
MOSFET
DRIVER
LOGIC
OSC
PGND
ISENSE
COMP
IAMP
SGND
FB
FDBK
SOFT-START
LOAD
COMPENSATION
1725 BD
VC
SFST
ROCMP
RCMPC
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LT1725
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TI I G DIAGRA
VSW
VOLTAGE
COLLAPSE
DETECT
VFLBK
0.80×
VFLBK
VIN
GND
SWITCH
STATE
OFF
ON
MINIMUM tON
OFF
ON
ENABLE DELAY
FLYBACK AMP
STATE
DISABLED
ENABLED
DISABLED
MINIMUM ENABLE TIME
1725 TD
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FLYBACK ERROR A PLIFIER
T1
D1
•
+
+
ISOLATED
VOUT
C1
•
VIN
–
•
M1
IM
IFXD
VC
R1
ENAB
FB
Q1 Q2
C2
VBG
R2
I
IM
1725 EA
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LT1725
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OPERATIO
The LT1725 is a current mode switcher controller IC
designed specifically for the isolated flyback topology. The
Block Diagram shows an overall view of the system. Many
of the blocks are similar to those found in traditional
designs, including: Internal Bias Regulator, Oscillator,
Logic, Current Amplifier and Comparator, Driver and Output Switch. The novel sections include a special Flyback
Error Amplifier and a Load Compensation mechanism.
Also, due to the special dynamic requirements of flyback
control, the Logic system contains additional functionality
not found in conventional designs.
The LT1725 operates much the same as traditional current
mode switchers, the major difference being a different
type of error amplifier that derives its feedback information from the flyback pulse. Due to space constraints, this
discussion will not reiterate the basics of current mode
switcher/controllers and isolated flyback converters. A
good source of information on these topics is Application
Note AN19.
ERROR AMPLIFIER—PSEUDO DC THEORY
Please refer to the simplified diagram of the Flyback Error
Amplifier. Operation is as follows: when MOSFET output
switch M1 turns off, its drain voltage rises above the VIN
rail. The amplitude of this flyback pulse as seen on the third
winding is given as:
VFLBK =
( VOUT + VF + ISEC • ESR)
NST
VF = D1 forward voltage
ISEC = transformer secondary current
ESR = total impedance of secondary circuit
NST = transformer effective secondary-to-third
winding turns ratio
The flyback voltage is then scaled by external resistor
divider R1/R2 and presented at the FB pin. This is then
compared to the internal bandgap reference by the differential transistor pair Q1/Q2. The collector current from Q1
is mirrored around and subtracted from fixed current
source IFXD at the VC pin. An external capacitor integrates
this net current to provide the control voltage to set the
current mode trip point.
The relatively high gain in the overall loop will then cause
the voltage at the FB pin to be nearly equal to the bandgap
reference VBG. The relationship between VFLBK and VBG
may then be expressed as:
VFLBK =
(R1+ R2) V
BG
R2
Combination with the previous VFLBK expression yields an
expression for VOUT in terms of the internal reference,
programming resistors, transformer turns ratio and diode
forward voltage drop:
VOUT = VBG
(R1+ R2) 
R2
1 

 – VF – ISEC • ESR
 NST 
Additionally, it includes the effect of nonzero secondary
output impedance, which is discussed below in further
detail, see Load Compensation Theory. The practical aspects of applying this equation for VOUT are found in the
Applications Information section.
So far, this has been a pseudo-DC treatment of flyback
error amplifier operation. But the flyback signal is a pulse,
not a DC level. Provision must be made to enable the
flyback amplifier only when the flyback pulse is present.
This is accomplished by the dotted line connections to the
block labeled “ENAB”. Timing signals are then required to
enable and disable the flyback amplifier.
ERROR AMPLIFIER—DYNAMIC THEORY
There are several timing signals which are required for
proper LT1725 operation. Please refer to the Timing
Diagram.
Minimum Output Switch On Time
The LT1725 effects output voltage regulation via flyback
pulse action. If the output switch is not turned on at all,
there will be no flyback pulse and output voltage information is no longer available. This would cause irregular loop
response and start-up/latchup problems. The solution chosen is to require the output switch to be on for an absolute
minimum time per each oscillator cycle. This in turn establishes a minimum load requirement to maintain regulation. See Applications Information for further details.
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LT1725
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OPERATIO
Enable Delay
Minimum Enable Time
When the output switch shuts off, the flyback pulse
appears. However, it takes a finite time until the transformer primary side voltage waveform approximately represents the output voltage. This is partly due to rise time
on the MOSFET drain node, but more importantly, due to
transformer leakage inductance. The latter causes a voltage spike on the primary side not directly related to output
voltage. (Some time is also required for internal settling of
the feedback amplifier circuitry.)
The feedback amplifier, once enabled, stays enabled for a
fixed minimum time period termed “minimum enable
time.” This prevents lockup, especially when the output
voltage is abnormally low, e.g., during start-up. The minimum enable time period ensures that the VC node is able
to “pump up” and increase the current mode trip point to
the level where the collapse detect system exhibits proper
operation. The “minimum enable time” often determines
the low load level at which output voltage regulation is lost.
See Applications Information for details.
In order to maintain immunity to these phenomena, a
fixed delay is introduced between the switch turnoff
command and the enabling of the feedback amplifier. This
is termed “enable delay”. In certain cases where the
leakage spike is not sufficiently settled by the end of the
enable delay period, regulation error may result. See
Applications Information for further details.
Collapse Detect
Once the feedback amplifier is enabled, some mechanism
is then required to disable it. This is accomplished by a
collapse detect comparator, which compares the flyback
voltage (FB referred) to a fixed reference, nominally 80%
of VBG. When the flyback waveform drops below this
level, the feedback amplifier is disabled. This action
accommodates both continuous and discontinuous mode
operation.
Effects of Variable Enable Period
It should now be clear that the flyback amplifier is enabled
during only a portion of the cycle time. This can vary from
the fixed “minimum enable time” described to a maximum
of roughly the “off” switch time minus the enable delay
time. Certain parameters of flyback amp behavior will then
be directly affected by the variable enable period. These
include effective transconductance and VC node slew rate.
LOAD COMPENSATION THEORY
The LT1725 uses the flyback pulse to obtain information
about the isolated output voltage. A potential error source
is caused by transformer secondary current flow through
the real life nonzero impedances of the output rectifier,
T1
VIN
IM
R1
M1
+
FB
Q1 Q2
R2
VBG
Q3
A1
–
LOAD
COMP I
IM
ROCMP
R3
50k
RCMPC
ISENSE
RSENSE
1725 F01
Figure 1. Load Compensation Diagram
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LT1725
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OPERATIO
transformer secondary and output capacitor. This has
been represented previously by the expression “ISEC •
ESR.” However, it is generally more useful to convert this
expression to an effective output impedance. Because the
secondary current only flows during the off portion of the
duty cycle, the effective output impedance equals the
lumped secondary impedance times the inverse of the OFF
duty cycle. That is:
 1 
ROUT = ESR 
 where
 DCOFF 
ROUT = effective supply output impedance
ESR = lumped secondary impedance
DCOFF = OFF duty cycle
Expressing this in terms of the ON duty cycle, remembering DCOFF = 1 – DC,
 1 
ROUT = ESR 

 1– DC 
DC = ON duty cycle
In less critical applications, or if output load current
remains relatively constant, this output impedance error
may be judged acceptable and the external FB resistor
divider adjusted to compensate for nominal expected
error. In more demanding applications, output impedance
error may be minimized by the use of the load compensation function.
To implement the load compensation function, a voltage is
developed that is proportional to average output switch
current. This voltage is then impressed across the external
ROCMP resistor, and the resulting current acts to increase
the VBG reference used by the flyback error amplifier. As
output loading increases, average switch current increases
to maintain rough output voltage regulation. This causes
an increase in ROCMP resistor current which effects a
corresponding increase in target output voltage.
Assuming a relatively fixed power supply efficiency, Eff,
Power Out = Eff • Power In
VOUT • IOUT = Eff • VIN • IIN
Average primary side current may be expressed in terms
of output current as follows:
 V

IIN =  OUT  • IOUT
 VIN • EFF 
combining the efficiency and voltage terms in a single
variable:
IIN = K1 • IOUT, where
 V

K1=  OUT 
 VIN • EFF 
Switch current is converted to voltage by the external
sense resistor and averaged/lowpass filtered by R3 and
the external capacitor on RCMPC. This voltage is then
impressed across the external ROCMP resistor by op amp
A1 and transistor Q3. This produces a current at the
collector of Q3 which is then mirrored around and then
subtracted from the FB node. This action effectively increases the voltage required at the top of the R1/R2
feedback divider to achieve equilibrium. So the effective
change in VOUT target is:
R

∆VOUT = K1 • ∆IOUT  SENSE  • (R1|| R2) or
 ROCMP 
R

∆VOUT
= K1 SENSE  • (R1|| R2)
∆IOUT
 ROCMP 
(
)
Nominal output impedance cancellation is obtained by
equating this expression with ROUT:
R

ROUT = K1 SENSE  • (R1|| R2) and
 ROCMP 
R

ROCMP = K1 SENSE  • (R1|| R2) where
 ROUT 
K1 = dimensionless variable related to VIN, VOUT and
efficiency as above
RSENSE = external sense resistor
ROUT = uncompensated output impedance
(R1||R2) = impedance of R1 and R2 in parallel
The practical aspects of applying this equation to determine an appropriate value for the ROCMP resistor are found
in the Applications Information section.
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TRANSFORMER DESIGN CONSIDERATIONS
Transformer specification and design is perhaps the most
critical part of applying the LT1725 successfully. In addition to the usual list of caveats dealing with high frequency
isolated power supply transformer design, the following
information should prove useful.
Turns Ratios
Note that due to the use of the external feedback resistor
divider ratio to set output voltage, the user has relative
freedom in selecting transformer turns ratio to suit a given
application. In other words, “screwball” turns ratios like
“1.736:1.0” can scrupulously be avoided! In contrast,
simpler ratios of small integers, e.g., 1:1, 2:1, 3:2, etc. can
be employed which yield more freedom in setting total
turns and mutual inductance. Turns ratio can then be
chosen on the basis of desired duty cycle. However,
remember that the input supply voltage plus the secondary-to-primary referred version of the flyback pulse (including leakage spike) must not exceed the allowed external
MOSFET breakdown rating.
Leakage Inductance
Transformer leakage inductance (on either the primary or
secondary) causes a spike after output switch turnoff. This
is increasingly prominent at higher load currents, where
more stored energy must be dissipated. In many cases a
“snubber” circuit will be required to avoid overvoltage
breakdown at the output switch node. Application Note
AN19 is a good reference on snubber design.
In situations where the flyback pulse extends beyond the
enable delay time, the output voltage regulation will be
affected to some degree. It is important to realize that the
feedback system has a deliberately limited input range,
roughly ±50mV referred to the FB node, and this works to
the user’s advantage in rejecting large, i.e., higher voltage,
leakage spikes. In other words, once a leakage spike is
several volts in amplitude, a further increase in amplitude
has little effect on the feedback system. So the user is
generally advised to arrange the snubber circuit to clamp
at as high a voltage as comfortably possible, observing
MOSFET breakdown, such that leakage spike duration is
as short as possible.
As a rough guide, total leakage inductances of several
percent (of mutual inductance) or less may require a
snubber, but exhibit little to no regulation error due to
leakage spike behavior. Inductances from several percent
up to perhaps ten percent cause increasing regulation
error.
Severe leakage inductances in the double digit percentage
range should be avoided if at all possible as there is a
potential for abrupt loss of control at high load current.
This curious condition potentially occurs when the leakage spike becomes such a large portion of the flyback
waveform that the processing circuitry is fooled into
thinking that the leakage spike itself is the real flyback
signal! It then reverts to a potentially stable state whereby
the top of the leakage spike is the control point, and the
trailing edge of the leakage spike triggers the collapse
detect circuitry. This will typically reduce the output voltage abruptly to a fraction, perhaps between one-third to
two-thirds of its correct value. If load current is reduced
sufficiently, the system will snap back to normal operation. When using transformers with considerable leakage
inductance, it is important to exercise this worst-case
check for potential bistability:
1. Operate the prototype supply at maximum expected
load current.
2. Temporarily short circuit the output.
3. Observe that normal operation is restored.
If the output voltage is found to hang up at a abnormally
low value, the system has a problem. This will usually be
evident by simultaneously monitoring the VSW waveform
on an oscilloscope to observe leakage spike behavior
firsthand. A final note—the susceptibility of the system to
bistable behavior is somewhat a function of the load I/V
characteristics. A load with resistive, i.e., I = V/R behavior
is the most susceptible to bistability. Loads which exhibit
“CMOSsy”, i.e., I = V2 /R behavior are less susceptible.
Secondary Leakage Inductance
In addition to the previously described effects of leakage
inductance in general, leakage inductance on the secondary in particular exhibits an additional phenomenon. It
forms an inductive divider on the transformer secondary,
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which reduces the size of the primary-referred flyback
pulse used for feedback. This will increase the output
voltage target by a similar percentage. Note that unlike
leakage spike behavior, this phenomena is load independent. To the extent that the secondary leakage inductance
is a constant percentage of mutual inductance (over
manufacturing variations), this can be accommodated by
adjusting the feedback resistor divider ratio.
Winding Resistance Effects
Resistance in either the primary or secondary will act to
reduce overall efficiency (POUT/PIN). Resistance in the
secondary increases effective output impedance which
degrades load regulation, (at least before load compensation is employed).
Bifilar Winding
A bifilar or similar winding technique is a good way to
minimize troublesome leakage inductances. However remember that this will increase primary-to-secondary capacitance and limit the primary-to-secondary breakdown
voltage, so bifilar winding is not always practical.
Finally, the LTC Applications group is available to assist
in the choice and/or design of the transformer. Happy
Winding!
SELECTING FEEDBACK RESISTOR DIVIDER VALUES
The expression for VOUT developed in the Operation section can be rearranged to yield the following expression for
the R1/R2 ratio:
(R1+ R2) = ( VOUT + VF + ISEC • ESR) N
R2
VBG
ST
where:
VOUT = desired output voltage
VF = switching diode forward voltage
ISEC • ESR = secondary resistive losses
VBG = data sheet reference voltage value
NST = effective secondary-to-third winding turns ratio
The above equation defines only the ratio of R1 to R2, not
their individual values. However, a “second equation for
two unknowns” is obtained from noting that the Thevenin
impedance of the resistor divider should be roughly 3k for
bias current cancellation and other reasons.
SELECTING ROCMP RESISTOR VALUE
The Operation section previously derived the following
expressions for ROUT, i.e., effective output impedance and
ROCMP, the external resistor value required for its nominal
compensation:
 1 
ROUT = ESR 

 1 – DC 
R

ROCMP = K1  SENSE  R1|| R2
 ROUT 
(
)
While the value for ROCMP may therefore be theoretically
determined, it is usually better in practice to employ
empirical methods. This is because several of the required
input variables are difficult to estimate precisely. For
instance, the ESR term above includes that of the transformer secondary, but its effective ESR value depends on
high frequency behavior, not simply DC winding resistance. Similarly, K1 appears to be a simple ratio of VIN to
VOUT times (differential) efficiency, but theoretically estimating efficiency is not a simple calculation. The suggested empirical method is as follows:
Build a prototype of the desired supply using the eventual
secondary components. Temporarily ground the RCMPC
pin to disable the load compensation function. Operate the
supply over the expected range of output current loading
while measuring the output voltage deviation. Approximate this variation as a single value of ROUT (straight line
approximation). Calculate a value for the K1 constant
based on VIN, VOUT and the measured (differential) efficiency. These are then combined with RSENSE as indicated
to yield a value for ROCMP.
Verify this result by connecting a resistor of roughly this
value from the ROCMP pin to ground. (Disconnect the
ground short to RCMPC and connect the requisite 0.1µF
filter capacitor to ground.) Measure the output impedance
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1000
with the new compensation in place. Modify the original
ROCMP value if necessary to increase or decrease the
effective compensation.
The switching frequency of the LT1725 is set by an
external capacitor connected between the OSCAP pin and
ground. Recommended values are between 200pF and
33pF, yielding switching frequencies between 50kHz and
250kHz. Figure 2 shows the nominal relationship between
external capacitance and switching frequency. To minimize stray capacitance and potential noise pickup, this
capacitor should be placed as close as possible to the IC
and the OSCAP node length/area minimized.
200
fOSC (kHz)
100
20
100
250
RT (kΩ)
1725 F03
Figure 3. “One Shot” Times vs Programming Resistor
Minimum On Time
This time defines a period whereby the normal switch
current limit is ignored. This feature provides immunity to
the leading edge current spike often seen at the source
node of the external power MOSFET, due to rapid charging
of its gate/source capacitance. This current spike is not
indicative of actual current level in the transformer primary, and may cause irregular current mode switching
action, especially at light load.
300
100
50
30
TIME (ns)
SELECTING OSCILLATOR CAPACITOR VALUE
500
100
COSCAP (pF)
200
1725 F02
Figure 2. fOSC vs OSCAP Value
SELECTING TIMING RESISTOR VALUES
There are three internal “one-shot” times that are programmed by external application resistors: minimum on
time, enable delay time and minimum enable time. These
are all part of the isolated flyback control technique, and
their functions have been previously outlined in the Theory
of Operation section. Figures 3 shows nominal observed
time versus external resistor value for these functions.
The following information should help in selecting and/or
optimizing these timing values.
However, the user must remember that the LT1725 does
not “skip cycles” at light loads. Therefore, minimum on
time will set a limit on minimum delivered power and consequently a minimum load requirement to maintain regulation (see Minimum Load Considerations). Similarly,
minimum on time has a direct effect on short-circuit behavior (see Maximum Load/Short-Circuit Considerations).
The user is normally tempted to set the minimum on time
to be short to minimize these load related consequences.
(After all, a smaller minimum on time approaches the ideal
case of zero, or no minimum.) However, a longer time may
be required in certain applications based on MOSFET
switching current spike considerations.
Enable Delay Time
This function provides a programmed delay between
turnoff of the gate drive node and the subsequent enabling
of the feedback amplifier. At high loads, a primary side
voltage spike after MOSFET turnoff may be observed due
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to transformer leakage inductance. This spike is not indicative of actual output voltage (see Figure 4B). Delaying
the enabling of the feedback amplifier allows this system
to effectively ignore most or all of the voltage spike and
maintain proper output voltage regulation. The enable
delay time should therefore be set to the maximum expected duration of the leakage spike. This may have
implications regarding output voltage regulation at minimum load (see Minimum Load Considerations).
A second benefit of the enable delay time function occurs
at light load. Under such conditions the amount of energy
stored in the transformer is small. The flyback waveform
becomes “lazy” and some time elapses before it indicates
the actual secondary output voltage (see Figure 4C). So
the enable delay time should also be set long enough to
ignore the “irrelevant” portion of the flyback waveform at
light load.
Additionally, there are cases wherein the gate output is
called upon to drive a large geometry MOSFET such that
the turnoff transition is slowed significantly. Under such
circumstances, the enable delay time may be increased to
accommodate for the lengthy transition.
MOSFET GATE DRIVE
IDEALIZED FLYBACK
WAVEFORM
A
FLYBACK WAVEFORM
WITH LARGE LEAKAGE
SPIKE AT HEAVY LOAD
B
ENABLE
DELAY
TIME
NEEDED
DISCONTINUOUS
MODE
RINGING
“SLOW” FLYBACK
WAVEFORM AT
LIGHT LOAD
C
ENABLE DELAY
TIME NEEDED
1725 F04
Figure 4
Minimum Enable Time
This function sets a minimum duration for the expected
flyback pulse. Its primary purpose is to provide a minimum source current at the VC node to avoid start-up
problems.
Average “start-up” VC current =
Minimum Enable Time
• ISRC
Switching Frequency
Minimum enable time can also have implications at light
load (see Minimum Load Considerations). The temptation
is to set the minimum enable time to be fairly short, as this
is the least restrictive in terms of minimum load behavior.
However, to provide a “reliable” minimum start-up current
of say, nominally 1µA, the user should set the minimum
enable time at no less that 2% of the switching period
(= 1/switching frequency).
CURRENT SENSE RESISTOR CONSIDERATIONS
The external current sense resistor allows the user to
optimize the current limit behavior for the particular application under consideration. As the current sense resistor
is varied from several ohms down to tens of milliohms,
peak switch current goes from a fraction of an ampere to
tens of amperes. Care must be taken to ensure proper
circuit operation, especially with small current sense
resistor values.
For example, a peak switch current of 10A requires a sense
resistor of 0.025Ω. Note that the instantaneous peak power
in the sense resistor is 2.5W, and it must be rated accordingly. The LT1725 has only a single sense line to this resistor. Therefore, any parasitic resistance in the ground side
connection of the sense resistor will increase its apparent
value. In the case of a 0.025Ω sense resistor, one milliohm
of parasitic resistance will cause a 4% reduction in peak
switch current. So resistance of printed circuit copper
traces and vias cannot necessarily be ignored.
An additional consideration is parasitic inductance. Inductance in series with the current sense resistor will accentuate the high frequency components of the current
waveform. In particular, the gate switching spike and
multimegahertz ringing at the MOSFET can be considerably
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amplified. If severe enough, this can cause erratic operation. For example, assume 3nH of parasitic inductance
(equivalent to about 0.1 inch of wire in free space) is in series
with an ideal 0.025Ω sense resistor. A “zero” will be formed
at f = R/(2πL), or 1.3MHz. Above this frequency the sense
resistor will behave like an inductor.
Several techniques can be used to tame this potential
parasitic inductance problem. First, any resistor used for
current sensing purposes must be of an inherently noninductive construction. Mounting this resistor directly
above an unbroken ground plane and minimizing its
ground side connection will serve to absolutely minimize
parasitic inductance. In the case of low valued sense
resistors, these may be implemented as a parallel combination of several resistors for the thermal considerations
cited above. The parallel combination will help to lower the
parasitic inductance. Finally, it may be necessary to place
a “pole” between the current sense resistor and the
LT1725 ISENSE pin to undo the action of the inductive zero
(see Figure 5). A value of 51Ω is suggested for the resistor,
while the capacitor is selected empirically for the particular
application and layout. Using good high frequency measurement techniques, the ISENSE pin waveform may be
observed directly with an oscilloscope while the capacitor
value is varied.
SENSE RESISTOR ZERO AT:
R
f = SENSE
2πLP
GATE
51Ω
ISENSE
SGND PGND
CCOMP
RSENSE
LP
PARASITIC
INDUCTANCE
1725 F05
COMPENSATING POLE AT:
1
f=
2π(51Ω)CCOMP
FOR CANCELLATION:
LP
CCOMP =
RSENSE(51Ω)
Figure 5
SOFT-START FUNCTION
The LT1725 contains an optional soft-start function that is
enabled by connecting an explicit external capacitor between the SFST pin and ground. Internal circuitry prevents
the control voltage at the VC pin from exceeding that on the
SFST pin.
The soft-start function is enagaged whenever VCC power
is removed, or as a result of either undervoltage lockout
or thermal (overtemperature) shutdown. The SFST node
is then discharged to roughly a VBE above ground.
(Remember that the VC pin control node switching threshold is deliberately set at a VBE plus several hundred
millivolts.) When this condition is removed, a nominal
40µA current acts to charge up the SFST node towards
roughly 3V. So, for example, a 0.1µF soft-start capacitor
will place a 0.4V/ms limit on the ramp rate at the VC node.
UVLO PIN FUNCTION
The UVLO pin effects an undervoltage lockout function
with at threshold of roughly 1.25V. An external resistor
divider between the input supply and ground can then be
used to achieve a user-programmable undervoltage lockout (see Figure 6a).
An additional feature of this pin is that there is a change in
the input bias current at this pin as a function of the state
of the internal UVLO comparator. As the pin is brought
above the UVLO threshold, the bias current sourced by the
part increases. This positive feedback effects a hysteresis
band for reliable switching action. Note that the size of the
hysteresis is proportional to the Thevenin impedance of
the external UVLO resistor divider network, which makes
it user programmable. As a rough rule of thumb, each 4k
or so of impedance generates about 1% of hysteresis.
(This is based on roughly 1.25V for the threshold and 3µA
for the bias current shift.)
Even in good quality ground plane layouts, it is common
for the switching node (MOSFET drain) to couple to the
UVLO pin with a stray capacitance of several thousandths
of a pF. To ensure proper UVLO action, a 100pF capacitor
is recommended from this pin to ground as shown in
Figure 6b. This will typically reduce the coupled noise to
a few millivolts. The UVLO filter capacitor should not be
made much larger than a few hundred pF, however, as the
hysteresis action will become too slow. In cases where
further filtering is required, e.g., to attenuate high speed
supply ripple, the topology in Figure 6c is recommended.
Resistor R1 has been split into two equal parts. This
provides a node for effecting capacitor filtering of high
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VIN
R1/2
VIN
VIN
R1
C2
R1
UVLO
R2
R1/2
UVLO
C1
100pF
UVLO
C1
100pF
R2
R2
1725 F06
(6a) “Standard” UVLO
Divider Topology
(6b) Filter Capacitor
Directly On UVLO Node
(6c) Recommended Topology to
Filter High Frequency Ripple
Figure 6
speed supply ripple, while leaving the UVLO pin node
impedance relatively unchanged at high frequency.
VIN
R1
INTERNAL WIDE HYSTERESIS
UNDERVOLTAGE LOCKOUT
The LT1725 is designed to implement isolated DC/DC
converters operating from input voltages of typically 48V
or more. The standard operating topology utilizes a third
transformer winding on the primary side that provides
both feedback information and local power for the LT1725
via its VCC pin. However, this arrangement is not inherently
self-starting. Start-up is effected by the use of an external
“trickle-charge” resistor and the presence of an internal
wide hysteresis undervoltage lockout circuit that monitors
VCC pin voltage (see Figure 7). Operation is as follows:
“Trickle charge” resistor R1 is connected to VIN and
supplies a small current, typically on the order of a single
mA, to charge C1. At first, the LT1725 is off and draws only
its start-up current. After some time, the voltage on C1
(VCC) reaches the VCC turn-on threshold. The LT1725 then
turns on abruptly and draws its normal supply current.
Switching action commences at the GATE pin and the
MOSFET begins to deliver power. The voltage on C1
begins to decline as the LT1725 draws its normal supply
current, which greatly exceeds that delivered by R1. After
some time, typically tens of milliseconds, the output
voltage approaches its desired value. By this time, the
third transformer winding is providing virtually all the
supply current required by the LT1725.
One potential design pitfall is undersizing the value of
capacitor C1. In this case, the normal supply current
VIN
+
IVCC
C1
VCC
LT1725
PGND
GATE
SGND
1725 F07
VON THRESHOLD
VVCC
IVCC
0
VGATE
Figure 7
drawn by the LT1725 will discharge C1 too rapidly; before
the third winding drive becomes effective, the VCC turn-off
threshold will be reached. The LT1725 turns off, and the
VCC node begins to charge via R1 back up to the turn-on
threshold. Depending upon the particular situation, this
may result in either several on-off cycles before proper
operation is reached, or, permanent relaxation oscillation
at the VCC node.
Component selection is as follows:
Resistor R1 should be selected to yield a worst-case
minimum charging current greater than the maximum
rated LT1725 start-up current, and a worst-case maximum charging current less than the minimum rated
LT1725 supply current.
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Capacitor C1 should then be made large enough to avoid
the relaxation oscillatory behavior described above. This
is complicated to determine theoretically as it depends on
the particulars of the secondary circuit and load behavior.
Empirical testing is recommended. (Use of the optional
soft-start function will lengthen the power-up timing and
require a correspondingly larger value for C1.)
A further note—certain users may wish to utilize the
general functionality of the LT1725, but may have an
available input voltage significantly lower than, say, 48V.
If this input voltage is within the allowable VCC range, i.e.,
perhaps 20V maximum, the internal wide hysteresis range
UVLO function becomes counterproductive. In such cases
it is simply better to operate the LT1725 directly from the
available DC input supply. The LT1737 is identical to the
LT1725, with the exception that it lacks the internal wide
hysteresis UVLO function. It is therefore designed to
operate directly from DC input supplies in the range of
4.5V to 20V. See the LT1737 data sheet for further
information.
FREQUENCY COMPENSATION
Loop frequency compensation is performed by connecting a capacitor from the output of the error amplifier (VC
pin) to ground. An additional series resistor, often required in traditional current mode switcher controllers, is
usually not required and can even prove detrimental. The
phase margin improvement traditionally offered by this
extra resistor will usually be already accomplished by the
nonzero secondary circuit impedance, which adds a “zero”
to the loop response.
In further contrast to traditional current mode switchers,
VC pin ripple is generally not an issue with the LT1725. The
dynamic nature of the clamped feedback amplifier forms
an effective track/hold type response, whereby the VC
voltage changes during the flyback pulse, but is then “held”
during the subsequent “switch on” portion of the next
cycle. This action naturally holds the VC voltage stable
during the current comparator sense action (current mode
switching).
OUTPUT VOLTAGE ERROR SOURCES
Conventional nonisolated switching power supply ICs
typically have only two substantial sources of output
voltage error: the internal or external resistor divider
network that connects to VOUT and the internal IC reference. The LT1725, which senses the output voltage in both
a dynamic and an isolated manner, exhibits additional
potential error sources to contend with. Some of these
errors are proportional to output voltage, others are fixed
in an absolute millivolt sense. Here is a list of possible
error sources and their effective contribution.
Internal Voltage Reference
The internal bandgap voltage reference is, of course,
imperfect. Its error, both at 25°C and over temperature is
already included in the specifications.
User Programming Resistors
Output voltage is controlled by the user-supplied feedback
resistor divider ratio. To the extent that the resistor ratio
differs from the ideal value, the output voltage will be
proportionally affected. Highest accuracy systems will
demand 1% components.
Schottky Diode Drop
The LT1725 senses the output voltage from the transformer primary side during the flyback portion of the cycle.
This sensed voltage therefore includes the forward drop,
VF, of the rectifier (usually a Schottky diode). The nominal
VF of this diode should therefore be included in feedback
resistor divider calculations. Lot to lot and ambient temperature variations will show up as output voltage shift/
drift.
Secondary Leakage Inductance
Leakage inductance on the transformer secondary reduces the effective secondary-to-third winding turns ratio
(NS/NT) from its ideal value. This will increase the output
voltage target by a similar percentage. To the extent that
secondary leakage inductance is constant from part to
part, this can be accommodated by adjusting the feedback
resistor ratio.
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Output Impedance Error
An additional error source is caused by transformer secondary current flow through the real life nonzero impedances of the output rectifier, transformer secondary and
output capacitor. Because the secondary current only
flows during the off portion of the duty cycle, the effective
output impedance equals the “DC” lumped secondary
impedance times the inverse of the off duty cycle. If the
output load current remains relatively constant, or, in less
critical applications, the error may be judged acceptable
and the feedback resistor divider ratio adjusted for nominal expected error. In more demanding applications, output impedance error may be minimized by the use of the
load compensation function (see Load Compensation).
MINIMUM LOAD CONSIDERATIONS
The LT1725 generally provides better low load performance than previous generation switcher/controllers utilizing indirect output voltage sensing techniques.
Specifically, it contains circuitry to detect flyback pulse
“collapse,” thereby supporting operation well into discontinuous mode. Nevertheless, there still remain constraints
to ultimate low load operation. These relate to the minimum switch on time and the minimum enable time.
Discontinuous mode operation will be assumed in the
following theoretical derivations.
As outlined in the Operation section, the LT1725 utilizes a
minimum output switch on time, tON. This value can be
combined with expected VIN and switching frequency to
yield an expression for minimum delivered power.
1 f 
2
Minimum Power = 
 ( VIN • tON )
2  LPRI 
= VOUT • IOUT
This expression then yields a minimum output current
constraint:

f
1
IOUT(MIN) = 
 VIN • tON
2  LPRI • VOUT 
(
)
2
where
f = switching frequency
LPRI = transformer primary side inductance
VIN = input voltage
VOUT = output voltage
tON = output switch minimum on time
An additional constraint has to do with the minimum
enable time. The LT1725 derives its output voltage information from the flyback pulse. If the internal minimum
enable time pulse extends beyond the flyback pulse, loss
of regulation will occur. The onset of this condition can be
determined by setting the width of the flyback pulse equal
to the sum of the flyback enable delay, tED, plus the
minimum enable time, tEN. Minimum power delivered to
the load is then:
[
1 f 
Minimum Power = 
 VOUT • tEN + tED
2  LSEC 
= VOUT • IOUT
(
)]
2
Which yields a minimum output constraint:
1  f • VOUT 
IOUT(MIN) = 
 tED + tEN
2  LSEC 
(
)
2
where
f = switching frequency
LSEC = transformer secondary side inductance
VOUT = output voltage
tED = enable delay time
tEN = minimum enable time
Note that generally, depending on the particulars of input
and output voltages and transformer inductance, one of
the above constraints will prove more restrictive. In other
words, the minimum load current in a particular application will be either “output switch minimum on time”
constrained, or “minimum flyback pulse time” constrained.
(A final note—LPRI and LSEC refer to transformer inductance as seen from the primary or secondary side respectively. This general treatment allows these expressions to
be used when the transformer turns ratio is nonunity.)
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MAXIMUM LOAD/SHORT-CIRCUIT CONSIDERATIONS
VIN = input voltage
The LT1725 is a current mode controller. It uses the VC
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the VC
node, nominally 2.5V, then acts as an output switch peak
current limit.
NSP = secondary-to-primary turns ratio ( NSEC /NPRI)
This 2.5V at the VC pin corresponds to a value of 250mV
at the ISENSE pin, when the (ON) switch duty cycle is less
than 40%. For a duty cycle above 40%, the internal slope
compensation mechanism lowers the effective ISENSE
voltage limit. For example, at a duty cycle of 80%, the
nominal ISENSE voltage limit is 220mV. This action becomes the switch current limit specification. Maximum
available output power is then determined by the switch
current limit, which is somewhat duty cycle dependent
due to internal slope compensation action.
Overcurrent conditions are handled by the same mechanism. The output switch turns on, the peak current is
quickly reached and the switch is turned off. Because the
output switch is only on for a small fraction of the available
period, power dissipation is controlled.
Loss of current limit is possible under certain conditions.
Remember that the LT1725 normally exhibits a minimum
switch on time, irrespective of current trip point. If the duty
cycle exhibited by this minimum on time is greater than the
ratio of secondary winding voltage (referred-to-primary)
divided by input voltage, then peak current will not be
controlled at the nominal value, and will cycle-by-cycle
ratchet up to some higher level. Expressed mathematically, the requirement to maintain short-circuit control is:
tON • f <
(VF + ISC • RSEC ) where
VIN • NSP
tON = output switch minimum on time
f = switching frequency
ISC = short-circuit output current
VF = output diode forward voltage at ISC
RSEC = resistance of transformer secondary
Trouble is typically only encountered in applications with
a relatively high product of input voltage times secondaryto-primary turns ratio and/or a relatively long minimum
switch on time. (Additionally, several real world effects such
as transformer leakage inductance, AC winding losses, and
output switch voltage drop combine to make this simple
theoretical calculation a conservative estimate.)
THERMAL CONSIDERATIONS
Care should be taken to ensure that the worst-case input
voltage condition does not cause excessive die temperatures. The 16-lead SO package is rated at 100°C/W, and
the 16-lead GN at 110°C/W.
Average supply current is simply the sum of quiescent
current given in the specifications section plus gate drive
current. Gate drive current can be computed as:
IG = f • QG where
QG = total gate charge
f = switching frequency
(Note: Total gate charge is more complicated than CGS • VG
as it is frequently dominated by Miller effect of the CGD.
Furthermore, both capacitances are nonlinear in practice.
Fortunately, most MOSFET data sheets provide figures
and graphs which yield the total gate charge directly per
operating conditions.) Nearly all gate drive power is dissipated in the IC, except for a small amount in the external
gate series resistor, so total IC dissipation may be computed as:
PD(TOTAL) = VCC (IQ + • f • QG ), where
IQ = quiescent current (from specifications)
QG = total gate charge
f = switching frequency
VCC = LT1725 supply voltage
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20
LT1725
U
W
U U
APPLICATIO S I FOR ATIO
SWITCH NODE CONSIDERATIONS
GATE DRIVE RESISTOR CONSIDERATIONS
For maximum efficiency, gate drive rise and fall times are
made as short as practical. To prevent radiation and high
frequency resonance problems, proper layout of the
components connected to the IC is essential, especially
the power paths (primary and secondary). B field (magnetic) radiation is minimized by keeping MOSFET leads,
output diode, and output bypass capacitor leads as short
as possible. E field radiation is kept low by minimizing the
length and area of all similar traces. A ground plane
should always be used under the switcher circuitry to
prevent interplane coupling.
The gate drive circuitry internal to the LT1725 has been
designed to have as low an output impedance as practically possible—only a few ohms. A strong L/C resonance
is potentially presented by the inductance of the path
leading to the gate of the power MOSFET and its overall
gate capacitance. For this reason the path from the GATE
package pin to the physical MOSFET gate should be kept
as short as possible, and good layout/ground plane practice used to minimize the parasitic inductance.
An explicit series gate drive resistor may be useful in some
applications to damp out this potential L/C resonance
(typically tens of MHz). A minimum value of perhaps
several ohms is suggested, and higher values (typically a
few tens of ohms) will offer increased damping. However,
as this resistor value becomes too large, gate voltage rise
time will increase to unacceptable levels, and efficiency
will suffer due to the sluggish switching action.
The high speed switching current paths are shown schematically in Figure 8. Minimum lead length in these paths
are essential to ensure clean switching and minimal EMI.
The path containing the input capacitor, transformer primary and MOSFET, and the path containing the transformer secondary, output diode and output capacitor
contain “nanosecond” rise and fall times. Keep these
paths as short as possible.
VCC
VIN
+
+
+
VCC
GATE
SECONDARY
POWER
PATH
PRIMARY
POWER
PATH
PGND
GATE
DISCHARGE
PATH
1725 F08
Figure 8. High Speed Current Switching Paths
1725f
21
LT1725
U
TYPICAL APPLICATIO S
TELECOM 48V TO ISOLATED 15V APPLICATION
ground-referred version of the flyback voltage waveform
for both feedback information and providing power to the
LT1725 itself.
The design in Figure 9 accepts an input voltage in the
range of 36V to 72V and outputs an isolated 15V at up to
2A. Transformer T1 is an off-the-shelf VERSA-PAKTM
#VP5-0155, produced by Coiltronics. As manufactured, it
consists of six ideally identical independent windings. In
this application, three windings are stacked in series on
the primary side and two are placed in parallel on the
secondary side. This arrangement provides a 3:1 primaryto-secondary turns ratio while maximizing overall efficiency. The remaining winding provides a primary-side
Capacitor C7 sets the switching frequency at approximately 200kHz. Optimal load compensation for the transformer and secondary circuit components is set by resistor
R8. Output voltage regulation and overall efficiency are
shown in the accompanying graphs. The resistor divider
formed by R14 and R15 sets the undervoltage lockout
threshold at about 32V, with a hysteresis band of about 2V.
The soft-start and 3VOUT features are unused as shown.
VERSA-PAK is a trademark of Coiltronics, Inc
T1
VP5-0155
7
D5
BAS16
6•
VIN
C2
1.5µF
×3
R14
820k
R1
24k
+
C10
100pF
R3
34.0k
1%
8
7
R4
3.01k
1%
D4
1N5257
C1
22µF
D2
MBRS1100
15
10
3VOUT UVLO
VCC
FB
GATE
LT1725
ISENSE
VC
OSCAP SFST tON
6
3
ENDLY MENAB ROCMP
14
13
R5
51k
R6
51k
12
4
RCMPC
SGND PGND
5
11
•
10
R11
150Ω 4•
R10
18Ω
R15
33k
9
3
D3
1N5257
C3
100pF
12
11
+
•1
•2
C5
1µF
C4
150µF
R13
750Ω
1W
VOUT
15V
•
8
R12
16 5.1Ω
2
9
5
D1
MBRD660
R9
51Ω
C9
470pF
1
M1
IRF620
R2
0.1Ω
1725 F09a
C6
1nF
C7
47pF
R7
51k
R8
6.2k
C8
0.1µF
Figure 9. 48V to Isolated 15V Converter
1725f
22
LT1725
U
TYPICAL APPLICATIO S
Application Regulation
Application Efficiency
90
15.5
80
VIN = 48V
VOUT = 15V
EFFICIENCY (%)
VOUT (V)
70
VIN = 48V
VIN = 36V
15.0
VIN = 72V
60
50
40
30
14.5
0
0.5
1.5
1.0
ILOAD (A)
2.0
2.5
20
0.01
0.1
1
10
ILOAD (A)
1725 F09c
1725 F09b
48V to Isolated 15V Application Parts List
T1: Coiltronics VP5-0155 VERSA-PAK
C8: 0.1µF, 25V, Z5U ceramic capacitor
M1: International Rectifier IRF620. 200V, 0.8Ω N-channel
MOSFET
C9: 470pF, 25V, X7R ceramic capacitor
D1: Motorola MBRD660. 6A, 60V Schottky diode
R1: 24k, 1/4W, 5% resistor
D2: Motorola MBRS1100. 1A, 100V Schottky diode
R2: IRC LR2010. 0.1Ω, 1/2W current sense resistor
D3, D4: 1N5257. 33V, 500mW Zener diode
R3: 34.0k, 1% resistor
D5: BAS16. 75V rectifier diode
R4: 3.01k, 1% resistor
C1: AVX TPSD226M025R0200. 22µF, 25V tantalum
capacitor
R5, R6, R7: 51k, 5% resistor
C2a, C2b, C2c: Vishay/Vitramon VJ1825Y155MXB. 1.5µF,
100V X7R ceramic capacitor
C10: 100pF, 25V, X7R ceramic capacitor
R8: 6.2k, 5% resistor
R9: 51Ω, 5% resistor
C3: 100pF, 100V, X7R ceramic capacitor
R10: 18Ω, 5% resistor
C4: Sanyo 20SV150M. 150µF, 20V, OS-CON electrolytic
capacitor
R11: 150Ω, 1/4W, 5% resistor
R12: 5.1Ω, 5% resistor
C5: 1µF, 25V, Z5U ceramic capacitor
R13a, R13b: 1.5k, 1/2W, 5% resistor
C6: 1nF, 25V, X7R ceramic capacitor
R14: 820k, 5% resistor
C7: 47pF, 25V NPO/COG ceramic capacitor
R15: 33k, 5% resistor
1725f
23
LT1725
U
TYPICAL APPLICATIO S
TELECOM 48V TO ISOLATED 5V APPLICATION
The design in Figure 10 accepts an input voltage in the
range of 36V to 72V and outputs an isolated 5V at up to 2A.
Transformer T1 is available as a Coiltronics CTX02-14989.
Capacitor C7 sets the switching frequency at approximately 275kHz. Optimal load compensation for the transformer and secondary circuit components is set by resistor
R8. Output voltage regulation and overall efficiency are
shown in the accompanying graphs. Efficiency is shown
both with and without the R11 preload. The resistor divider
formed by R13 and R14 sets the undervoltage lockout
threshold at about 32V, with a hysteresis band of about 2V.
The soft-start and 3VOUT features are unused as shown.
T1
CTX02-14989
6
D2
BAS16
1
VIN
R13
820k
R10
22Ω
C5
470pF
VOUT
5V
9
2
C2
1.5µF
R1
47k
R9
18Ω
D1
12CWQ06
R12
68Ω
R11
51Ω
1W
11
C9
100pF
R3
35.7k
1%
+
R14
33k
C1
15µF
C4
150pF
C10
1µF
10
8
7
R4
3.01k
1%
15
10
VCC
3VOUT UVLO
FB
GATE
LT1725
VC
ISENSE
OSCAP SFST tON
6
3
C3
150µF
12
4
9
+
ENDLY MENAB ROCMP
14
13
12
R5
51k
R6
51k
4
RCMPC
16
M1
IRF620
2
R2
0.18Ω
SGND PGND
5
11
1
1725 F10a
C6
1nF
C7
47pF
R7
51k
R8
2.7k
C8
0.1µF
Figure 10. 48V to Isolated 5V Converter
Application Regulation
Application Efficiency
90
5.25
VIN = 36V
VIN = 48V
80
VIN = 48V
EFFICIENCY (%)
VOUT (V)
70
VIN = 72V
5.00
WITHOUT R11
PRELOAD
60
WITH R11
PRELOAD
50
40
30
4.75
0
0.5
1.0
1.5
2.0
20
0.01
0.1
1
10
ILOAD (A)
ILOAD (A)
1725 F10b
1725 F10c
1725f
24
LT1725
U
TYPICAL APPLICATIO S
48V to Isolated 5V Application Parts List
T1: Coiltronics CTX02-14989
C9: 100pF, 25V, X7R ceramic capacitor
M1: International Rectifier IRF620. 200V, 0.8Ω N-channel
MOSFET
C10: 1µF, 25V, Z5U ceramic capacitor
D1: International Rectifier 12CWQ06FN. 12A, 60V Schottky
diode
R2: Panasonic type ERJ-14RSJ. 0.18Ω, 1/4W, 5%
resistor
D2: BAS16. 75V switching diode
R3: 35.7k, 1% resistor
C1: AVX TPSD156M035R0300. 15µF, 35V tantalum
capacitor
R4: 3.01k, 1% resistor
C2: Vishay/Vitramon VJ1825Y155MXB. 1.5µF, 100V, X7R
ceramic capacitor
R1: 47k, 1/4W, 5% resistor
R5, R6, R7: 51k, 5% resistor
R8: 2.7k, 5% resistor
C3: Sanyo 6SA150M. 150µF, 6.3V, OS-CON electrolytic
capacitor
R9: 18Ω, 5% resistor
C4: 150pF, 100V, X7R ceramic capacitor
R11: 51Ω, 1W, 5% resistor
C5: 470pF, 50V, X7R ceramic capacitor
R12: 68Ω, 5% resistor
C6: 1nF, 25V X7R ceramic capacitor
R13: 820k, 5% resistor
C7: 47pF, 25V, NPO ceramic capacitor
R14: 33k, 5% resistor
R10: 22Ω, 5% resistor
C8: 0.1µF, 25V, Z5U ceramic capacitor
1725f
25
LT1725
U
PACKAGE DESCRIPTIO
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
0.189 – 0.196*
(4.801 – 4.978)
16 15 14 13 12 11 10 9
0.229 – 0.244
(5.817 – 6.198)
0.150 – 0.157**
(3.810 – 3.988)
1
0.015 ± 0.004
× 45°
(0.38 ± 0.10)
0.007 – 0.0098
(0.178 – 0.249)
0.009
(0.229)
REF
0.053 – 0.068
(1.351 – 1.727)
2 3
4
5 6
7
8
0.004 – 0.0098
(0.102 – 0.249)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
* DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.008 – 0.012
(0.203 – 0.305)
0.0250
(0.635)
BSC
GN16 (SSOP) 1098
1725f
26
LT1725
U
PACKAGE DESCRIPTIO
S Package
16-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
2
3
4
5
6
0.053 – 0.069
(1.346 – 1.752)
0.008 – 0.010
(0.203 – 0.254)
0.014 – 0.019
(0.355 – 0.483)
TYP
8
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
7
0.050
(1.270)
BSC
S16 1098
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1725f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT1725
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1424-5
Isolated Flyback Switching Regulator
5V Output Voltage, No Optoisolator Required
LT1424-9
Isolated Flyback Switching Regulator
9V Output , Regulation Maintained Under Light Loads
LT1425
Isolated Flyback Switching Regulator
No Third Winding or Optoisolator Required
LT1533
Ultralow Noise 1A Switching Regulator
Low Switching Harmonics and Reduced EMI, VIN = 2.7V to 23V
LT1681/LTC1698
Isolated DC/DC Controller Chip-Set in Ouarter
and Half-Brick Footprint
36V ≤ VIN ≤ 72V; VOUT: 3.3V, 5V; POUT ≤ 100W;
Half the cost of a DC/DC Module; Low Profile, High Efficiency
LT1737
High Power Isolated Flyback Controller
Powered from a DC Supply Voltage
1725f
28
Linear Technology Corporation
LT/TP 1201 2K • PRINTED IN THE USA
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