NCP5392Q 2/3/4-Phase Controller for CPU Applications The NCP5392Q provides up to a four−phase buck solution which combines differential voltage sensing, differential phase current sensing, and adaptive voltage positioning to provide accurately regulated power for Intel processors. Dual−edge pulse−width modulation (PWM) combined with inductor current sensing reduces system cost by providing the fastest initial response to dynamic load events. Dual−edge multiphase modulation reduces the total bulk and ceramic output capacitance required to meet transient regulation specifications. A high performance operational error amplifier is provided to simplify compensation of the system. Dynamic Reference Injection further simplifies loop compensation by eliminating the need to compromise between closed−loop transient response and Dynamic VID performance. In addition, NCP5392Q provides an automatic power saving feature (Auto−PSI). When Auto−PSI function is enabled, NCP5392Q will automatically detect the VID transitions and direct the Vcore regulator in or out of low power states. As a result, the best efficiency scheme is always chosen. Features • • • • • • • • • • • • • • • • • • • • • • Meets Intel’s VR11.1 Specifications Dual−edge PWM for Fastest Initial Response to Transient Loading High Performance Operational Error Amplifier Internal Soft Start Dynamic Reference Injection (Patent #US07057381) DAC Range from 0.375 V to 1.6 V DAC Feed Forward Function (Patent Pending) ±0.5% DAC Voltage Accuracy from 1.0 V to 1.6 V True Differential Remote Voltage Sensing Amplifier Phase−to−Phase Current Balancing “Lossless” Differential Inductor Current Sensing Differential Current Sense Amplifiers for each Phase Adaptive Voltage Positioning (AVP) Oscillator Frequency Range of 100 kHz – 1 MHz Latched Over Voltage Protection (OVP) Guaranteed Startup into Pre−Charged Loads Threshold Sensitive Enable Pin for VTT Sensing Power Good Output with Internal Delays Thermally Compensated Current Monitoring Automatic Power Saving (AUTO PSI Mode) Compatible to PSI Power Saving Requirements This is a Pb−Free Device http://onsemi.com MARKING DIAGRAM 1 1 40 NCP5392Q AWLYYWWG 40 PIN QFN, 6x6 MN SUFFIX CASE 488AR NCP5392Q = Specific Device Code A = Assembly Location WL = Wafer Lot YY = Year WW = Work Week G = Pb−Free Package *Pin 41 is the thermal pad on the bottom of the device. ORDERING INFORMATION Device NCP5392QMNR2G* Package Shipping† QFN−40 2500/Tape & Reel (Pb−Free) *Temperature Range: 0°C to 85°C †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specification Brochure, BRD8011/D. Applications • Desktop Processors © Semiconductor Components Industries, LLC, 2009 June, 2009 − Rev. 0 1 Publication Order Number: NCP5392Q/D NCP5392Q 31 G2 33 34 32 G3 VCC 12VMON 35 36 DAC 37 PSI 39 38 APSI_EN G4 CS3 NCP5392Q VID4 CS3N 2/3/4−Phase Buck Controller (QFN40) VID5 CS2 COMP CS1N 30 29 28 27 26 25 24 23 22 21 20 19 18 17 11 ILIM ROSC CSSUM CS1 VDFB VID7 VDRP CS2N VFB VID6 16 9 10 VID3 DIFFOUT 8 CS4N VSN 7 VID2 15 6 CS4 14 5 VID1 VSP 4 DRVON 13 3 G1 VID0 IMON 2 EN 12 1 VR_RDY NC 40 PIN CONNECTIONS Figure 1. NCP5392Q QFN40 Pin Connections (Top View) http://onsemi.com 2 NCP5392Q VID0 VID1 VID2 VID3 VID4 VID5 VID6 VID7 Flexible DAC Overvoltage Protection − DAC + VSN − VSP + + G1 − Diff Amp DIFFOUT + VFB Error Amp + − 1.3 V + VDRP G2 − COMP + − Droop Amp VDFB CSSUM −2/3 + CS1P CS1N + − CS2P CS2N + − CS3P CS3N + − CS4P CS4N + − + + G3 − Gain = 6 + Gain = 6 + Gain = 6 + + G4 − + Gain = 6 Oscillator IMON ROSC + − ILIM EN VCC ILimit + 4.25 V + − UVLO GND (FLAG) Figure 2. NCP5392Q Block Diagram http://onsemi.com 3 Control, Fault Logic and Monitor Circuits DRVON PSI APSI_EN 12VMON VR_RDY NCP5392Q 12V_FILTER 12V_FILTER +5V 12V_FILTER VTT D1 C4 C3 VTT C1 5 6 7 8 9 1 39 37 PSI 38 APSI_EN 40 NC VID1 VID2 VID3 IMON 12 VID4 VID5 G1 VID6 VID7 CS1P CS1N EN G2 VR_RDY CS2P 14 CS2N VSN 13 VSP NCP5392Q G3 CS3P RFB CS3N 15 RF 16 17 CH 18 RDRP 19 RISO1 36 CS4P COMP CS4N PSI APSI_EN DRL IN L1 Q2 R2 RS1 PGND IMON C2 30 CS1 22 21 12V_FILTER 12V_FILTER 31 24 23 32 BST 26 VCC DRH NCP5359 SW OD 25 33 DRL 28 IN 27 PGND VFB VDRP DRVON 29 VDFB 12V_FILTER 12V_FILTER CSSUM + DAC GND CDFB R6 20 CDNI RNOR G4 DIFFOUT Q1 DRH NCP5359 SW OD RT2 RISO2 41 11 ROSC CF RFB1 ILIM CFB1 BST VCC 10 CPU GND 4 VID0 VCC 3 35 12VMON 34 2 U2 BST RDNP VCC RLIM1 DRH NCP5359 SW OD RLIM2 IN DRL PGND 12V_FILTER 12V_FILTER BST VCC DRH NCP5359 SW OD IN DRL PGND VCCP VSSN Figure 3. Application Schematic for Four Phases http://onsemi.com 4 NCP5392Q 12V_FILTER 12V_FILTER +5V 12V_FILTER VTT D1 C4 C3 VTT 5 6 7 8 9 1 39 PSI VID1 APSI_EN VID2 NC VID3 VID5 G1 VID6 VID7 CS1P CS1N EN G2 VR_RDY CS2N 14 VSN NCP5392Q 13 VSP G3 CS3P RFB CS3N RF 16 17 CH 18 RDRP 19 RISO1 20 36 CS4P COMP CS4N APSI_EN DRL IN Q2 R2 RS1 C2 CS1 PGND IMON 30 22 21 12V_FILTER 12V_FILTER 31 24 23 32 BST 26 VCC DRH NCP5359 SW OD 25 33 DRL 28 IN 27 PGND VFB VDRP DRVON 29 VDFB 12V_FILTER 12V_FILTER CSSUM + DAC GND CDFB R6 CDNI RNOR G4 DIFFOUT L1 RT2 RISO2 41 11 ROSC 15 ILIM CF RFB1 Q1 DRH NCP5359 SW OD PSI 38 40 IMON 12 VID4 CS2P CFB1 37 BST VCC 10 BST RDNP VCC RLIM1 DRH NCP5359 SW OD RLIM2 IN DRL PGND VCCP VSSN Figure 4. Application Schematic for Three Phases http://onsemi.com 5 CPU GND 4 VID0 VCC 3 35 12VMON 34 2 C1 U2 NCP5392Q 12V_FILTER 12V_FILTER +5V 12V_FILTER VTT D1 C4 C3 VTT 5 6 7 8 9 1 39 PSI VID1 APSI_EN VID2 NC VID3 30 G1 VID7 CS1P CS1N EN G2 VR_RDY CS2N VSN 13 VSP NCP5392Q G3 CS3P RFB CS3N 15 RF 16 17 CH 18 RDRP 19 RISO1 20 36 CS4P COMP CS4N L1 DRL IN Q2 R2 RS1 C2 CS1 PGND IMON 22 21 31 24 23 32 26 25 33 28 27 VFB VDRP DRVON 29 VDFB 12V_FILTER 12V_FILTER CSSUM + DAC GND CDFB R6 CDNI RNOR G4 DIFFOUT Q1 RT2 RISO2 41 11 ROSC CF RFB1 ILIM CFB1 BST VCC DRH NCP5359 SW OD APSI_EN 40 VID5 C1 PSI 38 IMON 12 CS2P 14 37 VID4 VID6 RNTC1 10 BST RDNP VCC RLIM1 DRH NCP5359 SW OD RLIM2 IN DRL PGND VCCP VSSN Figure 5. Application Schematic for Two Phases http://onsemi.com 6 CPU GND 4 VID0 VCC 3 35 12VMON 34 2 U2 NCP5392Q PIN DESCRIPTIONS Pin No. Symbol 1 EN Description 2 VID0 Voltage ID DAC input 3 VID1 Voltage ID DAC input 4 VID2 Voltage ID DAC input 5 VID3 Voltage ID DAC input 6 VID4 Voltage ID DAC input 7 VID5 Voltage ID DAC input 8 VID6 Voltage ID DAC input 9 VID7 Voltage ID DAC input 10 ROSC 11 ILIM 12 IMON 13 VSP Non−inverting input to the internal differential remote sense amplifier 14 VSN Inverting input to the internal differential remote sense amplifier 15 DIFFOUT 16 COMP Threshold sensitive input. High = startup, Low = shutdown. A resistance from this pin to ground programs the oscillator frequency according to fSW. This pin supplies a trimmed output voltage of 2 V. Overcurrent shutdown threshold setting. Connect this pin to the ROSC pin via a resistor divider as shown in the Application Schematics. To disable the overcurrent feature, connect this pin directly to the ROSC pin. To guarantee correct operation, this pin should only be connected to the voltage generated by the ROSC pin; do not connect this pin to any externally generated voltages. 0 mV to 900 mV analog signal proportional to the output load current. VSN referenced Output of the differential remote sense amplifier Output of the error amplifier 17 VFB 18 VDRP Compensation Amplifier Voltage feedback Voltage output signal proportional to current used for current limit and output voltage droop 19 VDFB Droop Amplifier Voltage Feedback 20 CSSUM 21 CS1N Inverted Sum of the Differential Current Sense inputs. Av=CSSUM/CSx = −4 Inverting input to current sense amplifier #1 22 CS1 23 CS2N 24 CS2 25 CS3N 26 CS3 27 CS4N 28 CS4 29 DRVON 30 G1 PWM output pulse to gate driver. 3−level output: Low = LSFET Enabled, Mid = Diode Emulation Enabled, High = HSFET Enabled 31 G2 PWM output pulse to gate driver. 3−level output (see G1) 32 G3 PWM output pulse to gate driver. 3−level output (see G1) 33 G4 PWM output pulse to gate driver. 3−level output (see G1) 34 12VMON 35 VCC Power for the internal control circuits. 36 DAC DAC Feed Forward Output 37 PSI Power Saving Control. Low = power saving operation, High = normal operation. PSI signal has higher priority over APSI_EN signal. 38 APSI_EN APSI_EN High: Enable AUTO PSI function. When PSI = low, system will be forced into PSI mode, unconditionally. When PSI = high, APSI_EN will determine if the system needs to be in AUTO PSI mode. Once in AUTO PSI mode, system switches on/off PSI functions automatically based on VID change status. 39 VR_RDY Open collector output. High indicates that the output is regulating 40 NC FLAG GND Non−inverting input to current sense amplifier #1 Inverting input to current sense amplifier #2 Non−inverting input to current sense amplifier #2 Inverting input to current sense amplifier #3 Non−inverting input to current sense amplifier #3 Inverting input to current sense amplifier #4 Non−inverting input to current sense amplifier #4 Bidirectional Gate Drive Enable Monitor a 12 V input through a resistor divider. Not Connected Power supply return (QFN Flag) http://onsemi.com 7 NCP5392Q PIN CONNECTIONS VS. PHASE COUNT Number of Phases G4 G3 G2 G1 CS4−CS4N CS3−CS3N CS2−CS2N CS1−CS1N 4 Phase 4 Out Phase 3 Out Phase 2 Out Phase 1 Out Phase 4 CS input Phase 3 CS input Phase 2 CS input Phase 1 CS input 3 Tie to GND Phase 3 Out Phase 2 Out Phase 1 Out Tie to CSN pin used Phase 3 CS input Phase 2 CS input Phase 1 CS input 2 Tie to GND Phase 2 Out Tie to GND Phase 1 Out Tie to CSN pin used Phase 2 CS input Tie to CSN pin used Phase 1 CS input MAXIMUM RATINGS ELECTRICAL INFORMATION Pin Symbol VMAX VMIN ISOURCE ISINK COMP 5.5 V −0.3 V 10 mA 10 mA VDRP 5.5 V −0.3 V 5 mA 5 mA V– GND + 300 mV GND – 300 mV 1 mA 1 mA DIFFOUT 5.5 V −0.3 V 20 mA 20 mA VR_RDY 5.5 V −0.3 V N/A 20 mA VCC 7.0 V −0.3 V N/A 10 mA ROSC 5.5 V −0.3 V 1 mA N/A IMON Output 1.1 V All Other Pins 5.5 V −0.3 V *All signals referenced to AGND unless otherwise noted. THERMAL INFORMATION Rating Symbol Value Unit Thermal Characteristic, QFN Package (Note 1) RqJA 34 °C/W Operating Junction Temperature Range (Note 2) TJ 0 to 125 °C Operating Ambient Temperature Range TA 0 to +85 °C Maximum Storage Temperature Range TSTG −55 to +150 °C Moisture Sensitivity Level, QFN Package MSL 1 Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. *The maximum package power dissipation must be observed. 1. JESD 51−5 (1S2P Direct−Attach Method) with 0 LFM. 2. JESD 51−7 (1S2P Direct−Attach Method) with 0 LFM. http://onsemi.com 8 NCP5392Q ELECTRICAL CHARACTERISTICS (Unless otherwise stated: 0°C < TA < 85°C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF) Parameter Test Conditions Min Typ Max Unit ERROR AMPLIFIER Input Bias Current (Note 3) −200 Noninverting Voltage Range (Note 3) 0 200 nA 1.3 3 V 1.0 mV Input Offset Voltage (Note 3) V+ = V− = 1.1 V −1.0 − Open Loop DC Gain CL = 60 pF to GND, RL = 10 KW to GND − 100 Open Loop Unity Gain Bandwidth CL = 60 pF to GND, RL = 10 KW to GND − 10 − MHz Open Loop Phase Margin CL = 60 pF to GND, RL = 10 KW to GND − 80 − ° Slew Rate DVin = 100 mV, G = − 10 V/V, DVout = 1.5 V – 2.5 V, CL = 60 pF to GND, DC Load = ±125 mA to GND − 5 − V/ms Maximum Output Voltage ISOURCE = 2.0 mA 3.5 − − V Minimum Output Voltage ISINK = 0.2 mA − − 50 mV Output source current (Note 3) Vout = 3.5 V 2 − − mA Output sink current (Note 3) Vout = 1.0 V 2 − − mA dB DIFFERENTIAL SUMMING AMPLIFIER VSN Input Bias Current VSN Voltage = 0 V 30 mA VSP Input Resistance DRVON = Low DRVON = High 1.5 17 kW VSP Input Bias Voltage DRVON = Low DRVON = High 0.09 0.66 V Input Voltage Range (Note 3) −0.3 − 3.0 V − 10 − MHz 1.025 V/V −3 dB Bandwidth CL = 80 pF to GND, RL = 10 KW to GND Closed Loop DC Gain VS to Diffout VS+ to VS− = 0.5 to 1.6 V 0.98 1.0 Maximum Output Voltage ISOURCE = 2 mA 3.0 − − V Minimum Output Voltage ISINK = 2 mA − − 0.5 V Output source current (Note 3) Vout = 3 V 2.0 − − mA Output sink current (Note 3) Vout = 0.5 V 2.0 − − mA − 1.30 − V INTERNAL OFFSET VOLTAGE Offset Voltage to the (+) Pin of the Error Amp and the VDRP pin VDROOP AMPLIFIER Input Bias Current (Note 3) −200 Non−inverting Voltage Range (Note 3) 0 200 nA 1.3 3 V 4.0 mV Input Offset Voltage (Note 3) V+ = V− = 1.1 V −4.0 − Open Loop DC Gain CL = 20 pF to GND including ESD, RL = 1 kW to GND − 100 Open Loop Unity Gain Bandwidth CL = 20 pF to GND including ESD, RL = 1 kW to GND − 10 − MHz Slew Rate CL = 20 pF to GND including ESD, RL = 1 kW to GND − 5 − V/ms Maximum Output Voltage ISOURCE = 4.0 mA 3 − − V Minimum Output Voltage ISINK = 1.0 mA − − 1 V Output source current (Note 3) Vout = 3.0 V 4 − − mA Output sink current (Note 3) Vout = 1.0 V 1 − − mA 3. Guaranteed by design, not tested in production. http://onsemi.com 9 dB NCP5392Q ELECTRICAL CHARACTERISTICS (Unless otherwise stated: 0°C < TA < 85°C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF) Parameter Test Conditions Min Typ Max Unit CSSUM AMPLIFIER Current Sense Input to CSSUM Gain −60 mV < CS < 60 mV −4.00 −3.88 −3.76 V/V Current Sense Input to CSSUM −3 dB Bandwidth CL = 10 pF to GND, RL = 10 kW to GND − 4 − MHz Current Sense Input to CSSUM Output Slew Rate DVin = 25 mV, CL = 10 pF to GND, Load = 1 k to 1.3 V − 4 − V/s Current Summing Amp Output Offset Voltage CSx – CSNx = 0, CSx = 1.1 V −15 − +15 mV Maximum CSSUM Output Voltage CSx – CSxN = −0.15 V (All Phases) ISOURCE = 1 mA 3.0 − − V Minimum CSSUM Output Voltage CSx – CSxN = 0.066 V (All Phases) ISINK = 1 mA − − 0.3 V Output source current (Note 3) Vout = 3.0 V 1 − − mA Output sink current (Note 3) Vout = 0.3 V 1 − − mA Enable High Input Leakage Current External 1 K Pullup to 3.3 V − − 1.0 mA Upper Threshold VUPPER − 650 770 mV Lower Threshold VLOWER 450 550 − mV Hysteresis VUPPER − VLOWER − 100 − mV − − 1.0 mA PSI (Power Saving Control, Active Low) APSI_EN (AUTO PSI Function Enable, Active High) Enable High Input Leakage Current External 1k Pullup to 3.3 V Upper Threshold VUPPER − 650 770 mV Lower Threshold VLOWER 450 550 − mV Hysteresis VUPPER − VLOWER − 100 − mV 3.0 − − V DRVON Output High Voltage Sourcing 500 mA Sourcing Current for Output High VCC = 5 V − 2.5 4.0 mA Output Low Voltage Sinking 500 mA − − 0.7 V Sinking Current for Output Low 2.5 − − mA Delay Time Propagation Delay from EN Low to DRVON − 10 − ns Rise Time CL (PCB) = 20 pF, DVo = 10% to 90% − 130 − ns Fall Time CL (PCB) = 20 pF, DVo = 10% to 90% − 10 − ns Internal Pulldown Resistance 35 70 140 kW VCC Voltage when DRVON Output Valid − − 2.0 V − 0 − nA Common Mode Input Voltage Range (Note 3) −0.3 − 2.0 V Differential Mode Input Voltage Range (Note 3) −120 − 120 mV CURRENT SENSE AMPLIFIERS Input Bias Current (Note 3) CSx = CSxN = 1.4 V Input Offset Voltage CSx = CSxN = 1.1 V, −1.0 − 1.0 mV Current Sense Input to PWM Gain (Note 3) 0 V < CSx − CSxN < 0.1 V, 5.7 6.0 6.3 V/V Current Sharing Offset CS1 to CSx All VID codes −2.5 − 2.5 mV 3. Guaranteed by design, not tested in production. http://onsemi.com 10 NCP5392Q ELECTRICAL CHARACTERISTICS (Unless otherwise stated: 0°C < TA < 85°C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF) Parameter Test Conditions Min Typ Max 2.02 Unit IMON VDRP to IMON Gain 1.325 V< VDRP < 1.8 V 1.98 2 VDRP to IMON −3 dB Bandwidth CL = 30 pF to GND, RL = 100 kW to GND − 4 V/V Output Referred Offset Voltage VDRP = 1.6 V, ISOURCE = 0 mA 81 90 99 mV Minimum Output Voltage VDRP = 1.2 V, ISINK = 100 mA − − 0.11 V Output source current (Note 3) Vout = 1 V 300 − − mA Output sink current (Note 3) Vout = 0.3 V 300 − − mA Maximum Clamp Voltage VDRP Voltage = 2 V, RLOAD = 100 k − − 1.15 V MHz OSCILLATOR 100 − 1000 kHz ROSC = 49.9 kW 200 − 224 kHz ROSC = 24.9 kW 374 − 414 ROSC = 10 kW 800 − 978 ROSC = 49.9 kW 191 − 234 ROSC = 24.9 kW 354 − 434 ROSC = 10 kW 755 − 1000 1.95 2.01 2.065 V Switching Frequency Range (Note 3) Switching Frequency Accuracy 2− or 4−Phase Switching Frequency Accuracy 3−Phase ROSC Output Voltage kHz MODULATORS (PWM Comparators) Minimum Pulse Width FSW = 800 KHz − 30 − ns Propagation Delay ±20 mV of Overdrive − 10 − ns 0% Duty Cycle COMP Voltage when the PWM Outputs Remain LO − 1.3 − V 100% Duty Cycle COMP Voltage when the PWM Outputs Remain HI − 2.3 − V PWM Ramp Duty Cycle Matching Between Any Two Phases − 90 − % PWM Phase Angle Error (Note 3) Between Adjacent Phases 15 − 15 ° VR_RDY Output Saturation Voltage IPGD = 10 mA, − − 0.4 V VR_RDY Rise Time (Note 3) External Pullup of 1 kW to 1.25 V, CTOT = 45 pF, DVo = 10% to 90% − 100 150 ns VR_RDY Output Voltage at Powerup (Note 3) VR_RDY Pulled up to 5 V via 2 kW, tR(VCC) ≤ 3 x tR(5V) 100 ms ≤ tR(VCC) ≤ 20 ms − − 1.0 V VR_RDY High – Output Leakage Current (Note 3) VR_RDY = 5.5 V via 1 K − − 0.2 mA VR_RDY Upper Threshold Voltage VCore Increasing, DAC = 1.3 V − 310 270 mV Below DAC VR_RDY Lower Threshold Voltage VCore Decreasing DAC = 1.3 V 410 370 VR_RDY Rising Delay VCore Increasing − 500 − ms VR_RDY Falling Delay VCore Decreasing − 5 − ms 3.0 − − V 1.4 1.5 1.6 V VR_RDY (POWER GOOD) OUTPUT mV Below DAC PWM OUTPUTS Output High Voltage Sourcing 500 mA Mid Output Voltage 3. Guaranteed by design, not tested in production. http://onsemi.com 11 NCP5392Q ELECTRICAL CHARACTERISTICS (Unless otherwise stated: 0°C < TA < 85°C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF) Parameter Test Conditions Min Typ Max Unit PWM OUTPUTS Output Low Voltage Sinking 500 mA − − 0.7 V Delay + Fall Time (Note 3) CL (PCB) = 50 pF, DVo = VCC to GND − 10 15 ns Delay + Rise Time (Note 3) CL (PCB) = 50 pF, DVo = GND to VCC − 10 15 ns Output Impedance – HI or LO State Resistance to VCC (HI) or GND (LO) − 75 − W Gate Pin Source Current 60 80 150 mA Gate Pin Threshold Voltage 210 240 265 mV Phase Detect Timer 15 20 27 ms 1.0 − 1.5 ms 400 500 600 ms 2/3/4−PhASE DETECTION DIGITAL SOFT−START Soft−Start Ramp Time DAC = 0 to DAC = 1.1 V VR11 Vboot time VID INPUT VID Upper Threshold VUPPER − 650 770 mV VID Lower Threshold VLOWER 450 550 − mV VID Hysteresis VUPPER − VLOWER − 100 VR11 Input Bias Current (Note 3) Delay before Latching VID Change (VID De−Skewing) (Note 3) Measured from the edge of the 1st VID change 200 − VID7 Valid Range − mV 200 nA 300 ns 3.33 V ENABLE INPUT Enable High Input Leakage Current (Note 3) Pullup to 1.3 V VR11 Rising Threshold VR11 Falling Threshold − − 200 nA − 650 770 mV 450 550 − mV − 100 − mV 5.0 ms VR11 Total Hysteresis Rising− Falling Threshold Enable Delay Time Measure Time from Enable Transitioning HI to when Output Begins 2.5 ILIM to VDRP Gain Between VDRP − VDFB = 450 mV and VDRP − VDFB = 650 mV 0.95 1 1.05 V/V ILIM to VDRP Gain in PSI 4 phase Between VDRP − VDFB = 450 mV and VDRP − VDFB = 650 mV − 0.25 − V/V ILIM to VDRP Gain in PSI 3 phase Between VDRP − VDFB = 450 mV and VDRP − VDFB = 650 mV − 0.33 − V/V ILIM to VDRP Gain in PSI 2 phase Between VDRP − VDFB = 450 mV and VDRP − VDFB = 650 mV − 0.5 − V/V ILIM Offset VDRP − VDFB = 520 mV −50 0 50 mV − 100 − ns DAC +150 DAC +185 DAC +200 mV (1.6 V DAC) +200 mV CURRENT LIMIT Delay OVERVOLTAGE PROTECTION VR11 Overvoltage Threshold VR11 PSI Overvoltage Threshold (Note 3) (1.6 V DAC) +150 Delay 100 3. Guaranteed by design, not tested in production. http://onsemi.com 12 ns NCP5392Q ELECTRICAL CHARACTERISTICS (Unless otherwise stated: 0°C < TA < 85°C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF) Parameter Test Conditions Min Typ Max Unit VCC UVLO Start Threshold 4 4.25 4.5 V VCC UVLO Stop Threshold 3.8 4.05 4.3 V UNDERVOLTAGE PROTECTION VCC UVLO Hysteresis 200 mV 12VMON UVLO 12VMON (High Threshold) VCC Valid 0.73 0.77 0.82 V 12VMON (Low Threshold) VCC Valid 0.64 0.68 0.73 V Output Source Current VOUT = 3 V 0.25 mA Output Sink Current VOUT = 0.3 V 1.5 mA Max Output Voltage (Note 3) Isource = 2 mA 3 V Min Output Voltage (Note 3) Isink = 2 mA DAC (FEED FORWARD FUNCTION) 0.5 V VRM 11 DAC 11 − 16.5 mV/ms 1.0 V < DAC < 1.6 V 0.8 V < DAC < 1.0 V 0.5 V < DAC < 0.8 V − − − − − − ±0.5 ±5 ±8 % mV mV EN Low, No PWM − 15 30 mA Positive DAC Slew Rate System Voltage Accuracy (DAC Value has a 19 mV Offset Over the Output Value) VCC VCC Operating Current 3. Guaranteed by design, not tested in production. http://onsemi.com 13 NCP5392Q Table 1. VRM11 VID Codes VID7 800 mV VID6 400 mV VID5 200 mV VID4 100 mV VID3 50 mV VID2 25 mV VID1 12.5 mV VID0 6.25 mV 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 1.60000 02 0 0 0 0 0 0 1 1 1.59375 03 0 0 0 0 0 1 0 0 1.58750 04 0 0 0 0 0 1 0 1 1.58125 05 0 0 0 0 0 1 1 0 1.57500 06 0 0 0 0 0 1 1 1 1.56875 07 0 0 0 0 1 0 0 0 1.56250 08 0 0 0 0 1 0 0 1 1.55625 09 0 0 0 0 1 0 1 0 1.55000 0A 0 0 0 0 1 0 1 1 1.54375 0B 0 0 0 0 1 1 0 0 1.53750 0C 0 0 0 0 1 1 0 1 1.53125 0D 0 0 0 0 1 1 1 0 1.52500 0E 0 0 0 0 1 1 1 1 1.51875 0F 0 0 0 1 0 0 0 0 1.51250 10 0 0 0 1 0 0 0 1 1.50625 11 0 0 0 1 0 0 1 0 1.50000 12 0 0 0 1 0 0 1 1 1.49375 13 0 0 0 1 0 1 0 0 1.48750 14 0 0 0 1 0 1 0 1 1.48125 15 0 0 0 1 0 1 1 0 1.47500 16 0 0 0 1 0 1 1 1 1.46875 17 0 0 0 1 1 0 0 0 1.46250 18 0 0 0 1 1 0 0 1 1.45625 19 0 0 0 1 1 0 1 0 1.45000 1A 0 0 0 1 1 0 1 1 1.44375 1B 0 0 0 1 1 1 0 0 1.43750 1C 0 0 0 1 1 1 0 1 1.43125 1D 0 0 0 1 1 1 1 0 1.42500 1E 0 0 0 1 1 1 1 1 1.41875 1F 0 0 1 0 0 0 0 0 1.41250 20 0 0 1 0 0 0 0 1 1.40625 21 0 0 1 0 0 0 1 0 1.40000 22 0 0 1 0 0 0 1 1 1.39375 23 0 0 1 0 0 1 0 0 1.38750 24 0 0 1 0 0 1 0 1 1.38125 25 0 0 1 0 0 1 1 0 1.37500 26 0 0 1 0 0 1 1 1 1.36875 27 0 0 1 0 1 0 0 0 1.36250 28 0 0 1 0 1 0 0 1 1.35625 29 0 0 1 0 1 0 1 0 1.35000 2A 0 0 1 0 1 0 1 1 1.34375 2B 0 0 1 0 1 1 0 0 1.33750 2C 0 0 1 0 1 1 0 1 1.33125 2D 0 0 1 0 1 1 1 0 1.32500 2E 0 0 1 0 1 1 1 1 1.31875 2F http://onsemi.com 14 Voltage (V) HEX 00 01 NCP5392Q Table 1. VRM11 VID Codes VID7 800 mV VID6 400 mV VID5 200 mV VID4 100 mV VID3 50 mV VID2 25 mV VID1 12.5 mV VID0 6.25 mV Voltage (V) HEX 0 0 1 1 0 0 0 0 1.31250 30 0 0 1 1 0 0 0 1 1.30625 31 0 0 1 1 0 0 1 0 1.30000 32 0 0 1 1 0 0 1 1 1.29375 33 0 0 1 1 0 1 0 0 1.28750 34 0 0 1 1 0 1 0 1 1.28125 35 0 0 1 1 0 1 1 0 1.27500 36 0 0 1 1 0 1 1 1 1.26875 37 0 0 1 1 1 0 0 0 1.26250 38 0 0 1 1 1 0 0 1 1.25625 39 0 0 1 1 1 0 1 0 1.25000 3A 0 0 1 1 1 0 1 1 1.24375 3B 0 0 1 1 1 1 0 0 1.23750 3C 0 0 1 1 1 1 0 1 1.23125 3D 0 0 1 1 1 1 1 0 1.22500 3E 0 0 1 1 1 1 1 1 1.21875 3F 0 1 0 0 0 0 0 0 1.21250 40 0 1 0 0 0 0 0 1 1.20625 41 0 1 0 0 0 0 1 0 1.20000 42 0 1 0 0 0 0 1 1 1.19375 43 0 1 0 0 0 1 0 0 1.18750 44 0 1 0 0 0 1 0 1 1.18125 45 0 1 0 0 0 1 1 0 1.17500 46 0 1 0 0 0 1 1 1 1.16875 47 0 1 0 0 1 0 0 0 1.16250 48 0 1 0 0 1 0 0 1 1.15625 49 0 1 0 0 1 0 1 0 1.15000 4A 0 1 0 0 1 0 1 1 1.14375 4B 0 1 0 0 1 1 0 0 1.13750 4C 0 1 0 0 1 1 0 1 1.13125 4D 0 1 0 0 1 1 1 0 1.12500 4E 0 1 0 0 1 1 1 1 1.11875 4F 0 1 0 1 0 0 0 0 1.11250 50 0 1 0 1 0 0 0 1 1.10625 51 0 1 0 1 0 0 1 0 1.10000 52 0 1 0 1 0 0 1 1 1.09375 53 0 1 0 1 0 1 0 0 1.08750 54 0 1 0 1 0 1 0 1 1.08125 55 0 1 0 1 0 1 1 0 1.07500 56 0 1 0 1 0 1 1 1 1.06875 57 0 1 0 1 1 0 0 0 1.06250 58 0 1 0 1 1 0 0 1 1.05625 59 0 1 0 1 1 0 1 0 1.05000 5A 0 1 0 1 1 0 1 1 1.04375 5B 0 1 0 1 1 1 0 0 1.03750 5C 0 1 0 1 1 1 0 1 1.03125 5D 0 1 0 1 1 1 1 0 1.02500 5E 0 1 0 1 1 1 1 1 1.01875 5F http://onsemi.com 15 NCP5392Q Table 1. VRM11 VID Codes VID7 800 mV VID6 400 mV VID5 200 mV VID4 100 mV VID3 50 mV VID2 25 mV VID1 12.5 mV VID0 6.25 mV Voltage (V) HEX 0 1 1 0 0 0 0 0 1.01250 60 0 1 1 0 0 0 0 1 1.00625 61 0 1 1 0 0 0 1 0 1.00000 62 0 1 1 0 0 0 1 1 0.99375 63 0 1 1 0 0 1 0 0 0.98750 64 0 1 1 0 0 1 0 1 0.98125 65 0 1 1 0 0 1 1 0 0.97500 66 0 1 1 0 0 1 1 1 0.96875 67 0 1 1 0 1 0 0 0 0.96250 68 0 1 1 0 1 0 0 1 0.95625 69 0 1 1 0 1 0 1 0 0.95000 6A 0 1 1 0 1 0 1 1 0.94375 6B 0 1 1 0 1 1 0 0 0.93750 6C 0 1 1 0 1 1 0 1 0.93125 6D 0 1 1 0 1 1 1 0 0.92500 6E 0 1 1 0 1 1 1 1 0.91875 6F 0 1 1 1 0 0 0 0 0.91250 70 0 1 1 1 0 0 0 1 0.90625 71 0 1 1 1 0 0 1 0 0.90000 72 0 1 1 1 0 0 1 1 0.89375 73 0 1 1 1 0 1 0 0 0.88750 74 0 1 1 1 0 1 0 1 0.88125 75 0 1 1 1 0 1 1 0 0.87500 76 0 1 1 1 0 1 1 1 0.86875 77 0 1 1 1 1 0 0 0 0.86250 78 0 1 1 1 1 0 0 1 0.85625 79 0 1 1 1 1 0 1 0 0.85000 7A 0 1 1 1 1 0 1 1 0.84375 7B 0 1 1 1 1 1 0 0 0.83750 7C 0 1 1 1 1 1 0 1 0.83125 7D 0 1 1 1 1 1 1 0 0.82500 7E 0 1 1 1 1 1 1 1 0.81875 7F 1 0 0 0 0 0 0 0 0.81250 80 1 0 0 0 0 0 0 1 0.80625 81 1 0 0 0 0 0 1 0 0.80000 82 1 0 0 0 0 0 1 1 0.79375 83 1 0 0 0 0 1 0 0 0.78750 84 1 0 0 0 0 1 0 1 0.78125 85 1 0 0 0 0 1 1 0 0.77500 86 1 0 0 0 0 1 1 1 0.76875 87 1 0 0 0 1 0 0 0 0.76250 88 1 0 0 0 1 0 0 1 0.75625 89 1 0 0 0 1 0 1 0 0.75000 8A 1 0 0 0 1 0 1 1 0.74375 8B 1 0 0 0 1 1 0 0 0.73750 8C 1 0 0 0 1 1 0 1 0.73125 8D 1 0 0 0 1 1 1 0 0.72500 8E 1 0 0 0 1 1 1 1 0.71875 8F http://onsemi.com 16 NCP5392Q Table 1. VRM11 VID Codes VID7 800 mV VID6 400 mV VID5 200 mV VID4 100 mV VID3 50 mV VID2 25 mV VID1 12.5 mV VID0 6.25 mV Voltage (V) HEX 1 0 0 1 0 0 0 0 0.71250 90 1 0 0 1 0 0 0 1 0.70625 91 1 0 0 1 0 0 1 0 0.70000 92 1 0 0 1 0 0 1 1 0.69375 93 1 0 0 1 0 1 0 0 0.68750 94 1 0 0 1 0 1 0 1 0.68125 95 1 0 0 1 0 1 1 0 0.67500 96 1 0 0 1 0 1 1 1 0.66875 97 1 0 0 1 1 0 0 0 0.66250 98 1 0 0 1 1 0 0 1 0.65625 99 1 0 0 1 1 0 1 0 0.65000 9A 1 0 0 1 1 0 1 1 0.64375 9B 1 0 0 1 1 1 0 0 0.63750 9C 1 0 0 1 1 1 0 1 0.63125 9D 1 0 0 1 1 1 1 0 0.62500 9E 1 0 0 1 1 1 1 1 0.61875 9F 1 0 1 0 0 0 0 0 0.61250 A0 1 0 1 0 0 0 0 1 0.60625 A1 1 0 1 0 0 0 1 0 0.60000 A2 1 0 1 0 0 0 1 1 0.59375 A3 1 0 1 0 0 1 0 0 0.58750 A4 1 0 1 0 0 1 0 1 0.58125 A5 1 0 1 0 0 1 1 0 0.57500 A6 1 0 1 0 0 1 1 1 0.56875 A7 1 0 1 0 1 0 0 0 0.56250 A8 1 0 1 0 1 0 0 1 0.55625 A9 1 0 1 0 1 0 1 0 0.55000 AA 1 0 1 0 1 0 1 1 0.54375 AB 1 0 1 0 1 1 0 0 0.53750 AC 1 0 1 0 1 1 0 1 0.53125 AD 1 0 1 0 1 1 1 0 0.52500 AE 1 0 1 0 1 1 1 1 0.51875 AF 1 0 1 1 0 0 0 0 0.51250 B0 1 0 1 1 0 0 0 1 0.50625 B1 1 0 1 1 0 0 1 0 0.50000 B2 1 1 1 1 1 1 1 0 OFF FE 1 1 1 1 1 1 1 1 OFF FF http://onsemi.com 17 NCP5392Q FUNCTIONAL DESCRIPTION General High Performance Voltage Error Amplifier The NCP5392Q provides up to four−phase buck solution which combines differential voltage sensing, differential phase current sensing, and adaptive voltage positioning to provide accurately regulated power necessary for Intel VR11.1 CPU power system. NCP5392Q has been designed to work with the NCP5359 driver. The error amplifier is designed to provide high slew rate and bandwidth. Although not required when operating as the controller of a voltage regulator, a capacitor from COMP to VFB is required for stable unity gain test configurations. AUTO−PSI Function The part can be configured to run in 2−, 3−, or 4−phase mode. In 2−phase mode, phases 1 and 3 should be used to drive the external gate drivers as shown in the 2−phase Applications Schematic, G2 and G4 must be grounded. In 3−phase mode, gate output G4 must be grounded as shown in the 3−phase Applications Schematic. In 4−phase mode all 4 gate outputs are used as shown in the 4−phase Applications Schematic. The Current Sense inputs of unused channels should be connected to VCCP shown in the Application Schematics. Please refer to table “PIN CONNECTIONS vs. PHASE COUNTS” for details. Gate Driver Outputs and 2/3/4 Phase Operation NCP5392Q makes energy saving possible without receiving PSI signal from the CPU by wisely introducing Auto−PSI feature. The device will monitor VID lines for transition into/out−of Low Power States. When the VID drops (An indication of entering power saving state), the Auot−PSI logic will detect the transition and enable PSI mode. On the other hand, when the VID rises (exiting power saving mode), the Auto−PSI logic detects the transition and exit PSI mode automatically. Auto−PSI uses the dynamic VID(DVID) transitions of VR11.0 and VR11.1 to shed phases. The phase shedding improves the efficiency of the Vcore regulator eventually. In PSI mode, the total current limit is reduced by the ratio of the phase count left after phase shedding. Auto−PSI function can be activated and deactivated by toggling APSI_EN (PIN38), but with lower priority compared to PSI signal. When PSI (PIN37) is pulled to low, the system will be forced into PSI mode unconditionally, and APSI_EN signal will be shielded. NCP5392Q can be operated up to four phases. It operates at one phase mode when the system enter PSI mode automatically (for example, VID down from 1.2 V to 1.1 V). Differential Current Sense Amplifiers and Summing Amplifier Four differential amplifiers are provided to sense the output current of each phase. The inputs of each current sense amplifier must be connected across the current sensing element of the phase controlled by the corresponding gate output (G1, G2, G3, or G4). If a phase is unused, the differential inputs to that phase’s current sense amplifier must be shorted together and connected to the output as shown in the 2− and 3−phase Application Schematics. The current signals sensed from inductor DCR are fed into a summing amplifier to have a summed−up output (CSSUM). Signal of CSSUM combines information of total current of all phases in operation. The outputs of current sense amplifiers control three functions. First, the summing current signal (CCSUM) of all phases will go through DROOP amplifier and join the voltage feedback loop for output voltage positioning. Second, the output signal from DROOP amplifier also goes to ILIM amplifier to monitor the output current limit. Finally, the individual phase current contributes to the current balance of all phases by offsetting their ramp signals of PWM comparators. Remote Output Sensing Amplifier(RSA) A true differential amplifier allows the NCP5392Q to measure Vcore voltage feedback with respect to the Vcore ground reference point by connecting the Vcore reference point to VSP, and the Vcore ground reference point to VSN. This configuration keeps ground potential differences between the local controller ground and the Vcore ground reference point from affecting regulation of Vcore between Vcore and Vcore ground reference points. The RSA also subtracts the DAC (minus VID offset) voltage, thereby producing an unamplified output error voltage at the DIFFOUT pin. This output also has a 1.3 V bias voltage as the floating ground to allow both positive and negative error voltages. Thermal Compensation Amplifier with VDRP and VDFB Pins Thermal compensation amplifier is an internal amplifier in the path of droop current feedback for additional adjustment of the gain of summing current and temperature compensation. The way thermal compensation is implemented separately ensures minimum interference to the voltage loop compensation network. Precision Programmable DAC A precision programmable DAC is provided and system trimmed. This DAC has 0.5% accuracy over the entire operating temperature range of the part. The DAC can be programmed to support either Intel VR11 VID code specifications. http://onsemi.com 18 NCP5392Q Oscillator and Triangle Wave Generator divider. During power-up, the 12VMON is monitored and the PWM outputs and soft-start circuit are disabled until the voltage exceeds the threshold of its UVLO comparator. The UVLO comparator incorporates hysteresis to avoid chattering. A programmable precision oscillator is provided. The oscillator ’s frequency is programmed by the resistance connected from the ROSC pin to ground. The user will usually form this resistance from two resistors in order to create a voltage divider that uses the ROSC output voltage as the reference for creating the current limit setpoint voltage. The oscillator frequency range is 100 kHz per phase to 1.0 MHz per phase. The oscillator generates up to 4 symmetrical triangle waveforms with amplitude between 1.3 V and 2.3 V. The triangle waves have a phase delay between them such that for 2−, 3− and 4−phase operation the PWM outputs are separated by 180, 120, and 90 angular degrees, respectively. Overcurrent Shutdown A programmable overcurrent function is incorporated within the IC. A comparator and latch make up this function. The inverting input of the comparator is connected to the ILIM pin. The voltage at this pin sets the maximum output current the converter can produce. The ROSC pin provides a convenient and accurate reference voltage from which a resistor divider can create the overcurrent setpoint voltage. Although not actually disabled, tying the ILIM pin directly to the ROSC pin sets the limit above useful levels − effectively disabling overcurrent shutdown. The comparator noninverting input is the summed current information from the VDRP minus offset voltage. The overcurrent latch is set when the current information exceeds the voltage at the ILIM pin. The outputs are pulled low, and the soft−start is pulled low. The outputs will remain disabled until the VCC voltage is removed and re−applied, or the ENABLE input is brought low and then high. PWM Comparators with Hysteresis Four PWM comparators receive an error signal at their noninverting input. Each comparator receives one of the triangle waves at its inverting output. The output of each comparator generates the PWM outputs G1, G2, G3, and G4. During steady state operation, the duty cycle will center on the valley of the triangle waveform, with steady state duty cycle calculated by Vout/Vin. During a transient event, both high and low comparator output transitions shift phase to the points where the error signal intersects the down and up ramp of the triangle wave. Output Overvoltage and Undervoltage Protection and Power Good Monitor PROTECTION FEATURES An output voltage monitor is incorporated. During normal operation, if the output voltage is 180 mV (typical) over the DAC voltage, the VR_RDY goes low, the DRVON signal remains high, the PWM outputs are set low. The outputs will remain disabled until the VCC voltage is removed and reapplied. During normal operation, if the output voltage falls more than 350 mV below the DAC setting, the VR_RDY pin will be set low until the output voltage rises. Power Saving Mode Upon receiving PSI low command, or VID down with Auto−PSI enabled, the NCP5392Q enters power saving mode with only single phase running. The device operates in power saving mode to maintain a high power efficiency and good transient performance. Undervoltage Lockout An undervoltage lockout (UVLO) senses the VCC input. During power−up, the input voltage to the controller is monitored, and the PWM outputs and the soft−start circuit are disabled until the input voltage exceeds the threshold voltage of the UVLO comparator. The UVLO comparator incorporates hysteresis to avoid chattering. Soft−Start The VR11 mode ramps Vcore to 1.1 V boot voltage at a fixed rate of 0.8 mV/mS, pauses at 1.1 V for around 500 mS, reads the VID pins to determine the DAC setting. Then ramps Vcore to the final DAC setting at the Dynamic VID slew rate of up to 12.5 mV/mS. Typical VR11 soft−start sequences are shown in the following graphs (Figure 9 and 10). 12VMON UVLO and VIN Information 12V UVLO senses the 12V power supply by connecting it to the 12VMON pin through an appropriate resistor http://onsemi.com 19 NCP5392Q APPLICATION INFORMATION The NCP5392Q demo board for the NCP5392Q is available by request. It is configured as a four phase solution with decoupling designed to provide a 1 mW load line under a 100 A step load. the lowest VID selection, then enable the test. See Figures 7 and 8. Startup Procedure Start by installing the test tool software. It is best to power the test tool from a separate ATX power supply. The test tool should be set to a valid VID code of 0.5 V or above in order for the controller to start. Consult the VTT help manual for more detailed instruction. Step Load Testing The VTT tool is used to generate the di/dt step load. Select the dynamic loading option in the VTT test tool software. Set the desired step load size, frequency, duty, and slew rate. See Figure 6. Figure 7. 1.6 V to 0.5 V Dynamic VID response Figure 6. Typical Load Step Response (full load, 35 A − 100 A) Dynamic VID Testing The VTT tool provides for VID stepping based on the Intel Requirements. Select the Dynamic VID option. Before enabling the test set the lowest VID to 0.5 V or greater and set the highest VID to a value that is greater than Figure 8. Dynamic VID Settling Time Rising (CH1: VID1, CH2: DAC, CH3:VCCP) http://onsemi.com 20 NCP5392Q DESIGN METHODOLOGY Programming the Current Limit and the Oscillator Frequency Decoupling the VCC Pin on the IC The demo board is set for an operating frequency of approximately 330 kHz. The ROSC pin provides a 2.0 V reference voltage which is divided down with a resistor divider and fed into the current limit pin ILIM. Then calculate the individual RLIM1 and RLIM2 values for the divider. The series resistors RLIM1 and RLIM2 sink current from the ILIM pin to ground. This current is internally mirrored into a capacitor to create an oscillator. The period is proportional to the resistance and frequency is inversely proportional to the total resistance. The total resistance may be estimated by Equation 1. This equation is valid for the individual phase frequency in both three and four phase mode. An RC input filter is required as shown in the VCC pin to minimize supply noise on the IC. The resistor should be sized such that it does not generate a large voltage drop between 5 V supply and the IC. Understanding Soft−Start The controller supports typical VR11 startup routines. Vcore voltage ramps to the 1.1 V boot voltage, with a pause to capture the VID code then resume ramping to target value based on internal slew rate limit. The initial ramp rate was set to be 0.8 mV/mS. Rosc ^ 20947 FSW *1.1262 30.5 kW ^ 20947 330 (eq. 1) *1.1262 60 Rosc−kohm 50 40 30 20 10 Figure 9. VR11.1 Startup 0 100 Calculation Real 1000 Freq−kHz Figure 11. ROSC vs. Frequency The current limit function is based on the total sensed current of all phases multiplied by a controlled gain (Acssum*Adrp). DCR sensed inductor current is a function of the winding temperature. The best approach is to set the maximum current limit based on expected average maximum temperature of the inductor windings, DCRTmax + DCR 25C(1 ) 0.00393 @ (Tmax * 25)) (eq. 2) For multiphase controller, the ripple current can be calculated as, Ipp + (Vin * N @ V out) @ Vout L @ F SW @ Vin (eq. 3) Therefore calculate the current limit voltage as below, Figure 10. VR11.1 Biased Startup VLIMIT ^ A CSSUM @ A DRP @ DCR Tmax @ (I MIN_OCP @ ) 0.5 @ Ipp) ǒ VLIMIT ^ A CSSUM @ A DRP @ DCR Tmax @ I MIN_OCP @ ) 0.5 @ 21 Ǔ (Vin * N @ V out) @ V out L @ F SW @ V in In Equation 4, ACSSUM and ADRP are the gain of current summing amplifier and droop amplifier. http://onsemi.com (eq. 4) NCP5392Q Acssum I1 I2 I3 I4 Adrp RNOR RISO1 + RSUM RT2 − + RISO1 and RISO2 are in series with RT2, the NTC temperature sense resistor placed near inductor. RSUM is the resistor connecting between pin VDFB and pin CSSUM. If PSI = 1, PSI function is off, the current limit follows the Equation 7; if PSI = 0, the power saving mode will be enabled, COEpsi is a coefficient for the current limiting related with power saving function (PSI), the current limit can be calculated from Equation 8. COEpsi value is one over the original phase count N. Refer to the PSI and phase shedding section for more details. RISO2 + − OCP event Ilim Figure 12. ACSSUM and ADRP As introduced before, VLIMIT comes from a resistor divider connected to Rosc pin, thus, VLIMIT + 2 V @ ACSSUM + *4 ADRP + * RLIM2 @ COEpsi RLIM1 ) RLIM2 (eq. 5) RNOR @ (RISO1 ) RISO2 ) RT2) (eq. 6) (R NOR ) RISO1 ) RISO2 ) RT2) @ RSUM Final Equations for the Current Limit Threshold Final equations are described based on two conditions: normal mode and PSI mode. ILIMIT(normal) ^ ILIMIT(PSI) ^ 2V@R LIM2 R LIM1)R LIM2 4@ R @(R )R )R T2) NOR ISO1 ISO2 (R )R )R )R T2)@R NOR ISO1 ISO2 SUM @ DCR 25C(1 ) 0.00393 @ (Tinductor * 25)) 2V@R LIM2 R LIM1)R LIM2 4@ R @(R )R )R T2) NOR ISO1 ISO2 (R )R )R )R T2)@R NOR ISO1 ISO2 SUM @ COEpsi @ DCR 25C(1 ) 0.00393 @ (Tinductor * 25)) * 0.5 @ * 0.5 @ (V in * N @ V out) @ V out L @ F SW @ V in (eq. 7) (V in * V out) @ Vout L @ FSW @ Vin (eq. 8) Inductor Current Sensing Compensation N is the number of phases involved in the circuit. The inductors on the demo board have a DCR at 25°C of 0.6 mW. Selecting the closest available values of 21.3 kW for RLIM1 and 9.28 kW for RLIM2 yields a nominal operating frequency of 330 kHz. Select RISO1 = 1 k, RISO2 = 1 k, RT2 = 10 K (25°C), RNOR/RSUM = 2, (refer to application diagram). That results to an approximate current limit of 133 A at 100°C for a four phase operation and 131 A at 25°C. The total sensed current can be observed as a scaled voltage at the VDRP with a positive no−load offset of approximately 1.3 V. The NCP5392Q uses the inductor current sensing method. An RC filter is selected to cancel out the impedance from inductor and recover the current information through the inductor’s DCR. This is done by matching the RC time constant of the sensing filter to the L/DCR time constant. The first cut approach is to use a 0.1 mF capacitor for C and then solve for R. (eq. 9) L Rsense(T) + 0.1 @ mF @ DCR 25C @ (1 ) 0.00393(T * 25)) Because the inductor value is a function of load and inductor temperature final selection of R is best done experimentally on the bench by monitoring the Vdroop pin and performing a step load test on the actual solution. Inductor Selection When using inductor current sensing it is recommended that the inductor does not saturate by more than 10% at maximum load. The inductor also must not go into hard saturation before current limit trips. The demo board includes a four phase output filter using the T44−8 core from Micrometals with 3 turns and a DCR target of 0.6 mW @ 25°C. Smaller DCR values can be used, however, current sharing accuracy and droop accuracy decrease as DCR decreases. Use the NCP5392Q design aide for regulation accuracy calculations for specific value of DCR. http://onsemi.com 22 NCP5392Q Simple Average SPICE Model A simple state average model shown in Figure 13 can be used to determine a stable solution and provide insight into the control system. GAIN = 1 {−2/3*4} Voff VRamp_min 1.3V V3 12V 0 0 0 12 2k R8 1k CH RF 22p 2.2k Unity Gain BW=15MHz R12 5.11k CF 1.8n R6 ESLBulk {3.5e−9/6} CCer {22e−6*18} ESRCer {1.5e−3/18} 0Aac 2 0Adc ESLCer {1.5e−9/18} 1 1 CBulk {560e−6*6} ESRBulk {7e−3/6} 2 CDFB 1E3 C5 10.6p 0 1E3 L LBRD RBRD 2 DCR 1 1 2 {185e−9/4} {0.6E−3/4} 100p 0.75m RSUM 1k RDFB 22p Voff + E1 + − − E GAIN = {6} Vdrp RDAC CDAC 50 CFB1 680P 12n RFB1 RFB 69.8 I1 = 50 I2 = 110 I1 TD = 100u TR = 50n TF = 50n Vout PW = 100u PER = 200u 0 VDAC DC = 1.2V AC = 0 TRAN = PULSE (0 0.05 400u 5u 5u 500u 1000u) 0 1k Voff R11 Voffset 1.3V C4 1k 10.6p 1k 0 0 Vdrp R10 2k R9 C6 1k 10.6p Figure 13. NCP5392Q Average SPICE Model 1E3 Voff IMON 0 Compensation and Output Filter Design If the required output filter and switching frequency are significantly different, it’s best to use the available PSPICE models to design the compensation and output filter from scratch. The design target for this demo board was 1.0 mW up to 2.0 MHz. The phase switching frequency is currently set to 330 kHz. It can easily be seen that the board impedance of 0.75 mW between the load and the bulk capacitance has a large effect on the output filter. In this case the six 560 mF bulk capacitors have an ESR of 7.0 mW. Thus the bulk ESR plus the board impedance is 1.15 mW + 0.75 mW or 1.9 mW. The actual output filter impedance does not drop to 1.0 mW until the ceramic breaks in at over 375 kHz. The controller must provide some loop gain slightly less than one out to a frequency in excess 300 kHz. At frequencies below where the bulk capacitance ESR breaks with the bulk capacitance, the DC−DC converter must have sufficiently high gain to control the output impedance completely. Standard Type−3 compensation works well with the NCP5392Q. http://onsemi.com 23 NCP5392Q Zout Open Loop Zout Closed Loop Open Loop Gain with Current Loop Closed Voltage Loop Compensation Gain 80 60 40 20 dB 0 −20 −40 −60 1mOhm −80 −100 100 1000 10000 100000 1000000 10000000 Frequency Figure 14. NCP5392Q Circuit Frequency Response The goal is to compensate the system such that the resulting gain generates constant output impedance from DC up to the frequency where the ceramic takes over holding the impedance below 1.0 mW. See the example of the locations of the poles and zeros that were set to optimize the model above. By matching the following equations a good set of starting compensation values can be found for a typical mixed bulk and ceramic capacitor type output filter. CH RFB1 CFB1 I Bias RDRP RISO2 RT RFB Gain = 4 − + CSSUM Amp RSx 1.3 V (eq. 11) 1 1 + 2p @ CFB1 @ (RFB1 ) RFB) 2p @ CCer @ (RBRD ) ESR Bulk) RL RFB should be set to provide optimal thermal compensation in conjunction with thermistor RT2, RISO1 and RISO2. With RFB set to 1.0 kW, RFB1 is usually set to 100 W for maximum phase boost, and the value of RF is typically set to 3.0 kW. CSx + − PWM Comparator Error Amp 1.3 V RISO1 RSUM 1 1 + (eq. 10) 2p @ CF @ RF 2p @ (RBRD ) ESR Bulk) @ C Bulk − + 1.3 V Droop Amp + − RNOR RF CF + + − + Gain = 1 Figure 15. Droop Injection and Thermal Compensation RDRP determines the target output impedance by the basic equation: Droop Injection and Thermal Compensation The VDRP signal is generated by summing the sensed output currents for each phase. A droop amplifier is added to adjust the total gain to approximately eight. VDRP is externally summed into the feedback network by the resistor RDRP. This introduces an offset which is proportional to the output current thereby forcing a controlled, resistive output impedance. R @ DCR @ A CSSUM @ A DRP Vout + Z out + FB (eq. 12) Iout R DRP RDRP + R FB @ DCR @ ACSSUM @ ADRP Zout (eq. 13) The value of the inductor’s DCR is a function of temperature according to the Equation 14: DCR (T) + DCR25C @ (1 ) 0.00393 @ (T * 25)) (eq. 14) http://onsemi.com 24 NCP5392Q temperature. The series resistor is split and inserted on both sides of the NTC to reduce noise injection into the feedback loop. The recommended total value for RISO1 plus RISO2 is approximately 1.0 kW. The output impedance varies with inductor temperature by the equation: Actual DCR increases by temperature, the system can be thermally compensated to cancel this effect to a great degree by adding an NTC in parallel with RNOR to reduce the droop gain as the temperature increases. The NTC device is nonlinear. Putting a resistor in series with the NTC helps make the device appear more linear with Zout(T) + R FB @ DCR 25C @ (1 ) 0.00393 @ (T * 25)) @ ACSSUM @ ADRP RDRP (eq. 15) By including the NTC RT2 and the series isolation resistors the new equation becomes: Zout(T) + R FB @ DCR 25C @ (1 ) 0.00393 @ (T * 25)) @ ACSSUM @ (R RDRP Acssum The typical equation of an NTC is based on a curve fit Equation 17 RT2(T) + RT2 25C @ e b 1 Ǔ ǒ 1 Ǔƫ ƪǒ273)T * 298 R @(R )R )R T2) NOR ISO1 ISO2 )R )R )R T2)@R NOR ISO1 ISO2 SUM The demo board use a 10 kW NTC with a b value of 3740. Figure 16 shows the comparison of the compensated output impedance and uncompensated output impedance varying with temperature. Adrp RNOR RISO1 I1 I2 I3 I4 (eq. 17) + RSUM RT2 − + RISO2 + − OCP event Ilim Imon + − Gain = 2 0.0013 Zout Zout(uncomp) 0.0012 (eq. 16) 0.001 Figure 17. IMON Circuit 0.0009 0.0008 1.05 0.0007 0.84 0.0006 25 45 65 85 105 Vimon−V Ohm 0.0011 Celsius Figure 16. Zout vs. Temperature Vimon vs. Iout 0.63 0.42 0.21 IMON for Current Monitor Since VDRP signal reflects the current information of all phases. It can be fed into the IMON amplifier for current monitoring as shown in Figure 17. IMON amplifier has a fixed gain of 2 with an offset when VDRP is equal to 1.3 V, the internal floating reference voltage. The IMON amplifier will be saturated at an maximum output of 1.09 V therefore the total gain of current should be carefully considered to make the maximum load current indicated by the IMON output. Figure 18 shows a typical of the relation between IMON output and the load current. 0 0 10 20 30 40 50 60 Iout−A 70 80 90 100 Figure 18. IMON Output vs. Output Current Power Saving Indicator (PSI) and Phase Shedding VR11.1 requires the processor to provide an output signal to the VR controller to indicate when the processor is in a low power state. NCP5392Q use the status of PSI pin to decide if there is a need to change its operating state to maximize efficiency at light loads. When PSI = 0, the PSI function will be enabled, and VR system will be running at a single phase power saving mode. The PSI signal will de−assert 1 ms prior to moving to a normal power state. http://onsemi.com 25 NCP5392Q Auto−PSI Function: When system switches on PSI function, a phase shedding will be presented. Only one is active in the emulation mode while other phases are shed. Figure 19 indicates a PSI−on transition from a 3−phase mode to a single phase mode. While staying stable in PSI mode, the PWM signal of phase 1 will vary from a low−state level to high level while other phases all go to mid−state level (1.5 V typical). Vice verse, when PSI signal goes high, the system will go back to the original phase mode such as shown in Figure 20. In Auto−PSI mode (APSI_EN=1, PSI=1), the device will monitor VID lines for transition into/out−of Low Power States. Figures 21 and 22 describe the Auto−PSI function during VID transitions. Figure 21. 10 A Load, VID Down, into PSI, CH1: PWM1, CH2: PWM2, CH3: PWM3, CH4: VOUT Figure 19. PSI turns on, CH1: PWM1, CH2: PWM2, CH3: PWM3, CH4: PSI Figure 22. 10 A Load, VID Up, Out of PSI, CH1: PWM1, CH2: PWM2, CH3: PWM3, CH4: VOUT Figure 20. PSI turns off, CH1: PWM1, CH2: PWM2, CH3: PWM3, CH4: PSI http://onsemi.com 26 NCP5392Q OVP Improved Performance Gate Driver and MOSFET Selection The overvoltage protection threshold is not adjustable. OVP protection is enabled as soon as soft−start begins and is disabled when part is disabled. When OVP is tripped, the controller commands all four gate drivers to enable their low side MOSFETs and VR_RDY transitions low. In order to recover from an OVP condition, VCC must fall below the UVLO threshold. See the state diagram for further details. The OVP circuit monitors the output of DIFFOUT. If the DIFFOUT signal reaches 180 mV (typical) above the nominal 1.3 V offset the OVP will trip and VRRDY will be pulled low, after eight consecutive OVP events are detected, all PWMs will be latched. The DIFFOUT signal is the difference between the output voltage and the DAC voltage (minus 19 mV if in VR11.1 modes) plus the 1.3 V internal offset. This results in the OVP tracking on the DAC voltage even during a dynamic change in the VID setting during operation. ON Semiconductor provides the NCP5359 as a companion gate driver IC. The NCP5359 driver is optimized to work with a range of MOSFETs commonly used in CPU applications. The NCP5359 provides special functionality including power saving mode operation and is required for high performance dynamic VID operation. Contact your local ON Semiconductor applications engineer for MOSFET recommendations. Board Stackup and Board Layout Close attention should be paid to the routing of the sense traces and control lines that propagate away from the controller IC. Routing should follow the demo board example. For further information or layout review contact ON Semiconductor. Figure 23. VR11.1, 1.6 V OVP Event http://onsemi.com 27 NCP5392Q SYSTEM TIMING DIAGRAM 12 V (Gate Driver) UVLO 5 V (Controller) UVLO EN 3.5 ms VID Valid VID DRVON 1 ms min 1.5 ms 500 ms VSP−VSN 500 ms VR_RDY Figure 24. Normal Startup UVLO UVLO EN 12 V (Gate Driver) 5 V (Controller) POR 3.5 ms DRVON VID Valid VID 1 ms min 1.5 ms VSP−VSN 1 ms 500 ms 500 ms VR_RDY Figure 25. Driver UVLO Limited Startup http://onsemi.com 28 NCP5392Q Diffout ~ 1.3 V 1 2 3 4 5 6 7 8 1 2 3 4 5 6 7 8 185 mV VR_RDY DRVON = High VSP = VID − 19 mV 185 mV Figure 26. OVP Shutdown Ilimit + 1.3 VDRP VR_RDY DRVON Figure 27. Non−PSI Current Limit http://onsemi.com 29 NCP5392Q PACKAGE DIMENSIONS QFN40 6x6, 0.5P CASE 488AR−01 ISSUE A ÉÉÉÉ ÉÉÉÉ ÉÉÉÉ ÉÉÉÉ D PIN ONE LOCATION 2X 0.15 C 2X NOTES: 1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994. 2. CONTROLLING DIMENSIONS: MILLIMETERS. 3. DIMENSION b APPLIES TO PLATED TERMINAL AND IS MEASURED BETWEEN 0.25 AND 0.30mm FROM TERMINAL 4. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS THE TERMINALS. A B E DIM A A1 A3 b D D2 E E2 e L K TOP VIEW 0.15 C (A3) 0.10 C A 40X SIDE VIEW A1 0.08 C C D2 L 40X 11 6.30 4.20 40X 40X 21 10 SOLDERING FOOTPRINT* SEATING PLANE K 20 MILLIMETERS MIN MAX 0.80 1.00 0.00 0.05 0.20 REF 0.18 0.30 6.00 BSC 4.00 4.20 6.00 BSC 4.00 4.20 0.50 BSC 0.30 0.50 0.20 −−− 0.65 EXPOSED PAD 1 E2 b 0.10 C A B 40X 0.05 C 4.20 6.30 1 30 40 31 e 36X BOTTOM VIEW 40X 0.30 36X 0.50 PITCH DIMENSIONS: MILLIMETERS *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. The products described herein (NCP5392Q), is covered by one or more of the following U.S. patent; US07057381. There may be other patents pending. ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. 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