AD AD8042ARZ-REEL Dual 160 mhz rail-to-rail amplifier Datasheet

Dual 160 MHz
Rail-to-Rail Amplifier
AD8042
FEATURES
Single AD8041 and Quad AD8044 also Available
Fully Specified at +3 V, +5 V, and ⴞ5 V Supplies
Output Swings to Within 30 mV of Either Rail
Input Voltage Range Extends 200 mV Below Ground
No Phase Reversal with Inputs 0.5 V Beyond Supplies
Low Power of 5.2 mA per Amplifier
High Speed and Fast Settling on 5 V:
160 MHz –3 dB Bandwidth (G = +1)
200 V/␮s Slew Rate
39 ns Settling Time to 0.1%
Good Video Specifications (RL = 150 ⍀, G = +2)
Gain Flatness of 0.1 dB to 14 MHz
0.02% Differential Gain Error
0.04ⴗ Differential Phase Error
Low Distortion
–64 dBc Worst Harmonic @ 10 MHz
Drives 50 mA 0.5 V from Supply Rails
APPLICATIONS
Video Switchers
Distribution Amplifiers
A/D Driver
Professional Cameras
CCD Imaging Systems
Ultrasound Equipment (Multichannel)
CONNECTION DIAGRAM
8-Lead Plastic DIP and SOIC
OUT1
1
8
+VS
–IN1
2
7
OUT2
+IN1
3
6
–IN2
–VS
4
5
+IN2
AD8042
The output voltage swing extends to within 30 mV of each rail,
providing the maximum output dynamic range. Additionally, it
features gain flatness of 0.1 dB to 14 MHz while offering differential gain and phase error of 0.04% and 0.06∞ on a single 5 V
supply. This makes the AD8042 useful for professional video
electronics such as cameras, video switchers, or any high speed
portable equipment. The AD8042’s low distortion and fast
settling make it ideal for buffering single supply, high speed
A/D converters.
The AD8042 offers low power supply current of 12 mA max
and can run on a single 3.3 V power supply. These features are
ideally suited for portable and battery-powered applications
where size and power are critical.
PRODUCT DESCRIPTION
The AD8042 is a low power voltage feedback, high speed amplifier designed to operate on +3 V, +5 V, or ± 5 V supplies. It
has true single supply capability with an input voltage range
extending 200 mV below the negative rail and within 1 V of the
positive rail.
The wide bandwidth of 160 MHz along with 200 V/ms of slew rate
on a single 5 V supply make the AD8042 useful in many generalpurpose, high speed applications where single supplies from
3.3 V to 12 V and dual power supplies of up to ± 6 V are needed.
The AD8042 is available in 8-lead plastic DIP and SOIC.
15
VS = 5V
12
G = +1
CL = 5pF
RL = 2k⍀ TO 2.5V
5V
2.5V
0V
CLOSED–LOOP GAIN (dB)
9
G=1
RL = 2k⍀ TO +2.5V
6
3
0
–3
–6
–9
–12
1V
1␮s
–15
1
Figure 1. Output Swing: Gain = –1, VS = +5 V
10
FREQUENCY (MHz)
100
500
Figure 2. Frequency Response
REV. C
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© 2004 Analog Devices, Inc. All rights reserved.
AD8042–SPECIFICATIONS (@T = 25ⴗC, V = 5 V, R = 2 k⍀ to 2.5 V, unless otherwise noted.)
A
S
L
AD8042
Parameter
Conditions
Min
Typ
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, VO < 0.5 V p-p
Bandwidth for 0.1 dB Flatness
Slew Rate
Full Power Response
Settling Time to 1%
Settling Time to 0.1%
G = +1
G = +2, RL = 150 W, RF = 200 W
G = –1, VO = 2 V Step
VO = 2 V p-p
G = –1, VO = 2 V Step
125
160
14
200
30
26
39
MHz
MHz
V/ms
MHz
ns
ns
–73
15
700
0.04
0.04
0.06
0.24
–63
dB
nV/÷Hz
fA/÷Hz
%
%
Degrees
Degrees
dB
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion
Input Voltage Noise
Input Current Noise
Differential Gain Error (NTSC, 100 IRE)
Differential Phase Error (NTSC, 100 IRE)
Worst-Case Crosstalk
130
fC = 5 MHz, VO = 2 V p-p, G = +2, RL = 1 kW
f = 10 kHz
f = 10 kHz
G = +2, RL = 150 W to 2.5 V
G = +2, RL = 75 W to 2.5 V
G = +2, RL = 150 W to 2.5 V
G = +2, RL = 75 W to 2.5 V
f = 5 MHz, RL = 150 W to 2.5 V
DC PERFORMANCE
Input Offset Voltage
3
TMIN to TMAX
Offset Drift
Input Bias Current
12
1.2
TMIN to TMAX
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Voltage Swing:
Output Voltage Swing:
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current (Per Amplifier)
Power Supply Rejection Ratio
RL = 1 kW
TMIN to TMAX
90
0.2
100
90
68
RL = 10 kW to 2.5 V
RL = 1 kW to 2.5 V
RL = 50 W to 2.5 V
TMIN to TMAX, VOUT = 0.5 V to 4.5 V
Sourcing
Sinking
G = +1
0.03 to 4.97
0.10 to 4.9 0.05 to 4.95
0.4 to 4.4 0.36 to 4.45
50
90
100
20
3
OPERATING TEMPERATURE RANGE
72
–40
0.06
0.12
9
12
3.2
4.8
0.5
300
1.5
–0.2 to +4
74
VCM = 0 V to 3.5 V
VS– = 0 V to –1 V, or VS+ = +5 V to +6 V
Max
5.5
80
Unit
mV
mV
mV/∞C
mA
mA
mA
dB
dB
kW
pF
V
dB
V
V
V
mA
mA
mA
pF
12
6.4
V
mA
dB
+85
∞C
Specifications subject to change without notice.
–2–
REV. C
AD8042
SPECIFICATIONS (@T = 25ⴗC, V = 5 V, R = 2 k⍀ to 1.5 V, unless otherwise noted.)
A
S
L
Parameter
Conditions
Min
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, VO < 0.5 V p-p
Bandwidth for 0.1 dB Flatness
Slew Rate
Full Power Response
Settling Time to 1%
Settling Time to 0.1%
G = +1
G = +2, RL = 150 W, RF = 200 W
G = –1, VO = 2 V Step
VO = 2 V p-p
G = –1, VO = 1 V Step
120
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion
Input Voltage Noise
Input Current Noise
Differential Gain Error (NTSC, 100 IRE)
Differential Phase Error (NTSC, 100 IRE)
Worst-Case Crosstalk
120
fC = 5 MHz, VO = 2 V p-p, G = –1, RL = 100 W
f = 10 kHz
f = 10 kHz
G = +2, RL = 150 W to 1.5 V, Input VCM = 1 V
RL = 75 W to 1.5 V, Input VCM = 1 V
G = +2, RL = 150 W to 1.5 V, Input VCM = 1 V
RL = 75 W to 1.5 V, Input VCM = 1 V
f = 5 MHz, RL = 1 kW to 1.5 V
DC PERFORMANCE
Input Offset Voltage
AD8042
Typ
MHz
MHz
V/ms
MHz
ns
ns
–56
16
500
0.10
0.10
0.12
0.27
–68
dB
nV/÷Hz
fA/÷Hz
%
%
Degrees
Degrees
dB
3
12
1.2
TMIN to TMAX
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Voltage Swing:
Output Voltage Swing:
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current (Per Amplifier)
Power Supply Rejection Ratio
RL = 1 kW
TMIN to TMAX
90
VCM = 0 V to 1.5 V
RL = 10 kW to 1.5 V
RL = 1 kW to 1.5 V
RL = 50 W to 1.5 V
TMIN to TMAX, VOUT = 0.5 V to 2.5 V
Sourcing
Sinking
G = +1
0.1 to 2.9
0.3 to 2.6
0.2
100
90
VS– = 0 V to –1 V, or VS+ = +3 V to +4 V
68
0
Specifications subject to change without notice.
–3–
9
12
3.2
4.8
0.6
mV
mV
mV/∞C
mA
mA
mA
dB
dB
300
1.5
–0.2 to +2
74
kW
pF
V
dB
0.03 to 2.97
0.05 to 2.95
0.25 to 2.65
50
50
70
17
V
V
V
mA
mA
mA
pF
3
OPERATING TEMPERATURE RANGE
REV. C
66
Unit
140
11
170
25
30
45
TMIN to TMAX
Offset Drift
Input Bias Current
Max
5.5
80
12
6.4
V
mA
dB
70
∞C
AD8042
SPECIFICATIONS (@T = 25ⴗC, V = ⴞ5 V, R = 2 k⍀ to 0 V, unless otherwise noted.)
A
S
L
Parameter
Conditions
Min
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth, VO < 0.5 V p-p
Bandwidth for 0.1 dB Flatness
Slew Rate
Full Power Response
Settling Time to 1%
Settling Time to 0.1%
G = +1
G = +2, RL = 150 W, RF = 200 W
G = –1, VO = 2 V Step
VO = 2 V p-p
G = –1, VO = 2 V Step
125
NOISE/DISTORTION PERFORMANCE
Total Harmonic Distortion
Input Voltage Noise
Input Current Noise
Differential Gain Error (NTSC, 100 IRE)
Differential Phase Error (NTSC, 100 IRE)
Worst-Case Crosstalk
145
fC = 5 MHz, VO = 2 V p-p, G = +2, RL = 1 kW
f = 10 kHz
f = 10 kHz
G = +2, RL = 150 W
G = +2, RL = 75 W
G = +2, RL = 150 W
G = +2, RL = 75 W
f = 5 MHz, RL = 150 W
DC PERFORMANCE
Input Offset Voltage
AD8042
Typ
Max
Unit
170
18
225
35
22
32
MHz
MHz
V/ms
MHz
ns
ns
–78
15
700
0.02
0.02
0.04
0.12
–63
dB
nV/÷Hz
fA/÷Hz
%
%
Degrees
Degrees
dB
3
TMIN to TMAX
Offset Drift
Input Bias Current
12
1.2
TMIN to TMAX
Input Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio
OUTPUT CHARACTERISTICS
Output Voltage Swing
Output Voltage Swing
Output Voltage Swing
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current (Per Amplifier)
Power Supply Rejection Ratio
RL = 1 kW
TMIN to TMAX
90
0.2
94
86
0.05
0.10
9.8
14
3.2
4.8
0.6
300
1.5
–5.2 to +4
74
mV
mV
mV/∞C
mA
mA
mA
dB
dB
kW
pF
V
dB
VCM = –5 V to +3.5 V
66
RL = 10 kW
RL = 1 kW
RL = 50 W
TMIN to TMAX, VOUT = –4.5 V to +4.5 V
Sourcing
Sinking
G = +1
–4.97 to +4.97
–4.8 to +4.8 –4.9 to +4.9
–4 to +3.2 –4.2 to +3.5
50
100
100
25
V
V
V
mA
mA
mA
pF
3
12
7
V
mA
dB
+85
∞C
VS– = –5 V to –6 V, or VS+ = +5 V to +6 V
OPERATING TEMPERATURE RANGE
68
–40
6
80
Specifications subject to change without notice.
–4–
REV. C
AD8042
ABSOLUTE MAXIMUM RATINGS 1
While the AD8042 is internally short-circuit protected, this may
not be sufficient to guarantee that the maximum junction
temperature (150∞C) is not exceeded under all conditions. To
ensure proper operation, it is necessary to observe the maximum
power derating curves.
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12.6 V
Internal Power Dissipation2
Plastic DIP Package (N) . . . . . . . . . . . . . . . . . . . . . . 1.3 W
Small Outline Package (R) . . . . . . . . . . . . . . . . . . . . . 0.9 W
Input Voltage (Common Mode) . . . . . . . . . . . . . ± VS ± 0.5 V
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . ± 3.4 V
Output Short-Circuit Duration
Observe Power Derating Curves
Storage Temperature Range (N, R) . . . . . . . –65∞C to +125∞C
Lead Temperature Range (Soldering 10 sec) . . . . . . . . . 300∞C
2.0
MAXIMUM POWER DISSIPATION (W)
8-LEAD PLASTIC-DIP PACKAGE
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for the device in free air:
8-Lead Plastic DIP Package: qJA = 90∞C/W
8-Lead SOIC Package: qJA = 155∞C/W
TJ = 150ⴗC
1.5
1.0
8-LEAD SOIC PACKAGE
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60
AMBIENT TEMPERATURE (ⴗC)
MAXIMUM POWER DISSIPATION
Figure 3. Maximum Power Dissipation vs. Temperature
The maximum power that can be safely dissipated by the AD8042
is limited by the associated rise in junction temperature. The
maximum safe junction temperature for plastic encapsulated
devices is determined by the glass transition temperature of the
plastic—approximately 150∞C. Exceeding this limit temporarily
may cause a shift in parametric performance due to a change in
the stresses exerted on the die by the package.
Exceeding a junction temperature of 175∞C for an extended
period can result in device failure.
ORDERING GUIDE
Model
AD8042AN
AD8042AR
AD8042AR-REEL
AD8042AR-REEL7
AD8042ARZ*
AD8042ARZ-REEL*
AD8042ARZ-REEL7*
AD8042ACHIPS
Temperature
Range
Package
Description
Package
Option
–40∞C to +85∞C
–40∞C to +85∞C
–40∞C to +85∞C
–40∞C to +85∞C
–40∞C to +85∞C
–40∞C to +85∞C
–40∞C to +85∞C
8-Lead DIP
8-Lead SOIC
8-Lead SOIC 13" REEL
8-Lead SOIC 7" REEL
8-Lead SOIC
8-Lead SOIC 13" REEL
8-Lead SOIC 7" REEL
DIE
N-8
SO-8
SO-8
SO-8
SO-8
SO-8
SO-8
*Z = Pb-free part
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4,000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD8042 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
REV. C
70 80 90
–5–
AD8042–Typical Performance Characteristics
100
100
VS = 5V
T = 25ⴗC
140 PARTS, SIDE A & B
MEAN = –1.52mV
STD DEVIATION = 1.15
SAMPLE SIZE = 280
(140 AD8042S)
80
FREQUENCY
70
60
95
OPEN-LOOP GAIN (dB)
90
50
40
30
20
90
85
VS = 5V
T = 25ⴗC
80
75
10
0
–6
–5
–4
–3
–2
–1 0
1
VOS (mV)
2
3
4
5
70
6
TPC 1. Typical Distribution of VOS
250
500
750
1000 1250
1500
LOAD RESISTANCE (⍀)
1750
2000
TPC 4. Open-Loop Gain vs. RL to 2.5 V
30
100
VS = 5V
MEAN = –12.6␮V/ⴗC
STD DEV = 2.02␮V/ⴗC
SAMPLE SIZE = 60
98
OPEN-LOOP GAIN (dB)
25
20
FREQUENCY
0
15
10
VS = 5V
RL = 1k⍀
96
94
92
90
5
88
0
–18
–16
–14
–12 –10
–8
–6
VOS DRIFT(␮V/ⴗC)
–2
–4
0
86
–40
TPC 2. VOS Drift Over –40∞C to +85∞C
–20
0
20
40
TEMPERATURE (ⴗC)
60
80
TPC 5. Open-Loop Gain vs. Temperature
0
100
VS = 5V
VCM = 0V
RL = 500⍀ TO 2.5V
VS = 5V
90
–0.4
–0.6
OPEN-LOOP GAIN (dB)
INPUT BIAS CURRENT (␮A)
–0.2
–0.8
–1
–1.2
–1.4
80
RL = 50⍀ TO 2.5V
70
60
–1.6
50
–1.8
–2
–40 –30 –20 –10
0
10
20
30
40
50
60
70
80
40
90
TEMPERATURE(ⴗC)
0
0.5
1
1.5
2
2.5
3
3.5
OUTPUT VOLTAGE (V)
4
4.5
5
TPC 6. Open-Loop Gain vs. Output Voltage
TPC 3. IB vs. Temperature
–6–
REV. C
AD8042
0.04
NTSC Subcarrier (3.579 MHz)
DIFFERENTIAL
GAIN ERROR (%)
100
30
0.02
3
1
100
1k
10k
100k
1M
10M
100M
FREQUENCY (Hz)
0.00
0.03
0.02
VS = ⴞ5V
G = +2
RL = 150⍀
0.01
0
20
30
40
50
60
70
80
MODULATING RAMP LEVEL (IRE)
90
100
0.6
0.4
NORMALIZED GAIN (dB)
VS = 5V, AV = 2,
RL = 100⍀ TO 2.5V
–50
VS = 5V, AV = 1,
RL = 100⍀ TO 2.5V
–60
–70
–80
VS = 5V, AV = 2,
RL = 1k⍀ TO 2.5V
0.3
0.2
0.1
0
14MHz
–0.1
–0.2
VS = 5V, AV = 1,
RL = 1k⍀ TO 2.5V
–90
VS = 5V
G = +2
RF = 200⍀
RL = 150⍀ TO 2.5V
0.5
VS = 3V, AV = –1,
RL = 100⍀ TO 1.5V
–40
–0.3
–0.4
3
2
4
5
6
FUNDAMENTAL FREQUENCY (MHz)
1
7 8 9 10
100
500
TPC 11. 0.1 dB Gain Flatness
–30
120
–40
100
VS = 5V
G = +2
RF = 200⍀
RL = 150⍀ TO 2.5V
80
OPEN-LOOP GAIN (dB)
10MHz
–50
10
FREQUENCY (MHz)
–60
5MHz
–70
–80
1MHz
GAIN
60
45
40
0
20
–45
–90
0
PHASE
–20
–135
–40
–180
–60
–225
–90
VS = 5V, G = +2,
RL = 1k⍀ TO 2.5V
–100
–110
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
–80
0.01
5.0
OUTPUT VOLTAGE (V p-p)
TPC 9. Worst Harmonic vs. Output Voltage
REV. C
0.1
1
10
FREQUENCY (MHz)
100
TPC 12. Open-Loop Gain and Phase
vs. Frequency
–7–
–270
500
PHASE (Degrees)
1
TPC 8. Total Harmonic Distortion
WORST HARMONIC (dBc)
10
TPC 10. Differential Gain and Phase Errors
–30
TOTAL HARMONIC DISTORTION (dBc)
VS = 5V
G = +2
RL = 150⍀ TO 2.5V
0.04
TPC 7. Input Voltage Noise vs. Frequency
–100
VS = ⴞ5V
G = +2
RL = 150⍀
–0.01
0
1G
VS = 5V
G = +2
RL = 150⍀ TO 2.5V
0.01
–0.01
0.05
10
10
0.03
DIFFERENTIAL
PHASE ERROR (deg)
INPUT VOLTAGE NOISE (nV/ Hz)
300
AD8042
60
10
VS = 5V
G = +1
CL = 5pF
RL = 2k⍀ TO 2.5V
8
T = +85ⴗC
VS = 3V, 0.1%
50
4
SETTLING TIME (ns)
CLOSED-LOOP GAIN (dB)
6
G = –1
RL = 2k⍀ TO MIDPOINT
CL = 5pF
55
T = +25ⴗC
2
T = –40ⴗC
0
–2
–4
45
VS = 3V, 1%
40
VS = +5V, 0.1%
35
VS = ⴞ5V, 0.1%
30
–6
VS = +5V, 1%
25
–8
–10
1
10
FREQUENCY (MHz)
100
VS = ⴞ5V, 1%
20
0.5
500
1
1.5
BIPOLAR INPUT STEP (V)
2
TPC 16. Settling Time
TPC 13. Closed-Loop Frequency Response
vs. Temperature
12
Vs = 5V
1.02k⍀
1.02k⍀
VS = 5V
RL AND CL TO 2.5V
6
TEST CIRCUIT:
0
COMMON-MODE REJECTION (dB)
8
CLOSED-LOOP GAIN (dB)
VS = 3V
RL AND CL TO 1.5V
G = +1
CL = 5pF
RL = 2k⍀
10
VS = ⴞ5V
4
2
0
–2
–4
–6
–10
INCM
OUT
1.02k⍀ 1.02k⍀
–20
–30
–40
–50
–60
–70
–80
–8
1
10
FREQUENCY (MHz)
100
–90
10k
500
1M
10M
FREQUENCY (Hz)
100k
500M
100M
TPC 17. CMRR vs. Frequency
TPC 14. Closed-Loop Frequency Response vs. Supply
0.80
VS = +5V
OUTPUT SATURATION VOLTAGE (V)
OUTPUT RESISTANCE (⍀)
100
RBT = 50⍀
10
VS = 5V
G = +1
RBT = 0
RBT
1
VOUT
0.1
0.01
0.70
+5V – VOH (+125ⴗC)
+5V – VOH (+25ⴗC)
0.60
+5V – VOH (–55ⴗC)
0.50
0.40
0.30
0.20
+VOL (+125ⴗC)
+VOL (+25ⴗC)
0.10
+VOL (–55ⴗC)
0
0.01
0.1
1
10
FREQUENCY (MHz)
100
0
500
5
10
15
20
25
30
35
LOAD CURRENT (mA)
40
45
50
TPC 18. Output Saturation Voltage vs. Load Current
TPC 15. Output Resistance vs. Frequency
–8–
REV. C
AD8042
50
12.0
VS = ⴞ5V
VS = +5V
VOUT = 100mV STEP
11.5
10.5
VS = +5V
10.0
VS = +3V
% OVERSHOOT
SUPPLY CURRENT (mA)
40
11.0
9.5
G = +2
30
20
G = +3
9.0
10
8.5
8.0
–40 –30 –20 –10
0
0
10
20
30
40
50
60
70
90
80
0
20
40
60
80
100
140
120
160
180
200
LOAD CAPACITANCE (pF)
TEMPERATURE (ⴗC)
TPC 19. Supply Current vs. Temperature
TPC 22. % Overshoot vs. Load Capacitance
6
VS = +5V
–10
4
–20
3
–30
–40
–PSRR
–50
–60
+PSRR
G = +2
2
1
0
G = +10
–2
–80
–3
G = +5
–4
10k
100k
1M
10M
FREQUENCY (Hz)
100M
1
500M
TPC 20. PSRR vs. Frequency
100
500
–10
9
–20
VS = ⴞ5V
RL = 2k⍀
G = –1
8
–30
7
CROSSTALK (dB)
OUTPUT VOLTAGE (V p-p)
10
FREQUENCY (MHz)
TPC 23. Frequency Response vs. Closed-Loop Gain
10
6
5
4
VOUT 1
–50
VOUT 2
VOUT 1
–60
2
–90
1
–100
0
0.1
–110
0.1
100.0
VOUT 2
, RL = 1k⍀ TO +2.5V
, RL = 150⍀ TO +2.5V
–70
–80
1.0
10.0
FREQUENCY (MHz)
VS = +5V
VIN = 0.6V p-p
G = +2
RF = 1k⍀
–40
3
TPC 21. Output Voltage Swing vs. Frequency
REV. C
G = +2
RF = 200⍀
–1
–70
–90
VS = +5V
RF = 2k⍀
RL = 2k⍀ TO +2.5V
5
NORMALIZED GAIN (dB)
PSRR (dB)
0
VOUT2
VOUT1
VOUT2
VOUT1
1
, RL = 150⍀ TO +2.5V
, RL = 1k⍀ TO +2.5V
10
FREQUENCY (MHz)
100
200
TPC 24. Crosstalk (Output-to-Output) vs. Frequency
–9–
AD8042
5V
4.770V
4V
VS = +5V
G = –1
RL = 150⍀ TO +2.5V
AV = +1
+2.6V
VS = +5V
VIN = 100mV p-p
RL = 1k⍀ TO 2.5V
CL = 5pF
3V
+2.5V
2V
1V
0.160V
+2.4V
200␮s
0.5V
25mV
10ns
0V
TPC 27. 100 mV Pulse Response, VS = +5 V
TPC 25a. Output Swing with Load Reference to
Supply Midpoint
5V
4.59V
4V
G = –1
RL = 2k⍀ TO +1.5V
VS= +5V
G = –1
RL= 150⍀ TO GND
3V
3V
1.5V
2V
0V
1V
0.035V
0.5V
200␮s
1␮s
0.5V
0V
TPC 28. Rail-to-Rail Output Swing, VS = +3 V
TPC 25b. Output Swing with Load Reference to
Negative to Supply
4.5V
AV = +2
VS = +5V
CL = 5pF
RL = 1k⍀ TO +2.5V
VIN = 1V p-p
3.5V
VIN = 100mV p-p
+1.6V
RL = 1k⍀ TO 1.5V
VS = +3V
CL = 5pF
AV = +1V
+1.5V
2.5V
1.5V
+1.4V
0.5V
10ns
25mV
10ns
0.5V
TPC 29. 100 mV Pulse Response, VS = +3 V
TPC 26. One Volt Pulse Response, VS = +5 V
–10–
REV. C
AD8042
Biasing of Q8 and Q36 is accomplished by I8 and I5, along with
a common-mode feedback loop (not shown). This circuit topology allows the AD8042 to drive 40 mA of output current with
the outputs within 0.5 V of the supply rails.
OVERDRIVE RECOVERY
Overdrive of an amplifier occurs when the output and/or input
range are exceeded. The amplifier must recover from this overdrive condition. As shown in Figure 4, the AD8042 recovers
within 30 ns from negative overdrive and within 25 ns from
positive overdrive.
On the input side, the device can handle voltages from 0.2 V
below the negative rail to within 1.2 V of the positive rail.
Exceeding these values will not cause phase reversal; however,
the input ESD devices will begin to conduct if the input voltages exceed the rails by greater than 0.5 V.
+5V
DRIVING CAPACITIVE LOADS
The capacitive load drive of the AD8042 can be increased by
adding a low valued resistor in series with the load. Figure 6
shows the effects of a series resistor on capacitive drive for varying voltage gains. As the closed-loop gain is increased, the larger
phase margin allows for larger capacitive loads with less overshoot. Adding a series resistor with lower closed-loop gains
accomplishes this same effect. For large capacitive loads, the
frequency response of the amplifier will be dominated by the
roll-off of the series resistor and capacitive load.
+2.5V
VS = +5V
VIN= +5V p-p
G = +2
RL = 1k⍀ TO +2.5V
0V
1V
50ns
Figure 4. Overdrive Recovery
1000
VS = 5V
200mV STEP WITH 30% OVERSHOOT
The AD8042 is fabricated on Analog Devices’ proprietary
eXtra-Fast Complementary Bipolar (XFCB) process which
enables the construction of PNP and NPN transistors with
similar fTs in the 2 GHz to 4 GHz region. The process is dielectrically isolated to eliminate the parasitic and latch-up problems
caused by junction isolation. These features allow the construction of high frequency, low distortion amplifiers with low supply
currents. This design uses a differential output input stage to
maximize bandwidth and headroom (see Figure 5). The smaller
signal swings required on the first stage outputs (nodes S1P,
S1N) reduce the effect of nonlinear currents due to junction
capacitances and improve the distortion performance. With this
design harmonic distortion of better than –77 dB @ 1 MHz into
100 W with VOUT = 2 V p-p (Gain = +2) on a single 5 V supply
is achieved.
I10
R26
I2
R39
Q4
Q36
Q5
Q51
VEE
C3
Q31
Q21
SIP
VEE
R23 R27
Q7
VINN
VOUT
Q27
C9
SIN
Q2
Q8
Q11
Q3
C7
I5
Q39
Q23
Q22
Q17
Q13
I9
Q50
Q40
R15 R2
VINP
I 3 Q25
R5
Q24
R21
R3
I7
I8
Q47
VCC
VEE
Figure 5. AD8042 Simplified Schematic
The AD8042’s rail-to-rail output range is provided by a
complementary common-emitter output stage. High output drive
capability is provided by injecting all output stage predriver currents directly into the bases of the output devices Q8 and Q36.
REV. C
RS = 5⍀
RS
CL
RS = 0
100
RS = 20⍀
10
1
2
3
4
CLOSED-LOOP GAIN (V/V)
5
Figure 6. Capacitive Load Drive vs. Closed-Loop Gain
Single-Supply Composite Video Line Driver
The two op amps of an AD8042 can be configured as a singlesupply dual line driver for composite video. The wide signal
swing of the AD8042 enables this function to be performed
without using any type of clamping or dc restore circuit which
can cause signal distortion.
VCC
I1
CAPACITIVE LOAD (pF)
CIRCUIT DESCRIPTION
Figure 7 shows a schematic for a circuit that is driven by a
single composite video source that is ac coupled, level shifted
and applied to both + inputs of the two amplifiers. Each op amp
provides a separate 75 W composite video output. To obtain
single-supply operation, ac coupling is used throughout. The
large capacitor values are required to ensure that there is minimal
tilting of the video signals due to their low frequency (30 Hz)
signal content. The circuit shown was measured to have a differential gain of 0.06% and a differential phase of 0.06∞.
The input is terminated in 75 W and ac coupled via CIN to a
voltage divider that provides the dc bias point to the input.
Setting the optimal bias point requires some understanding of
the nature of composite video signals and the video performance
of the AD8042.
–11–
AD8042
To test this, the differential gain and differential phase were
measured for the AD8042 while the supplies were varied. As the
lower supply is raised to approach the video signal, the first
effect to be observed is that the sync tips become compressed
before the differential gain and differential phase are adversely
affected. Thus, there must be adequate swing in the negative
direction to pass the sync tips without compression.
+5V
4.99k⍀
0.1µF
4.99k⍀
10␮F
10␮F
3
COMPOSITE
VIDEO
IN
8
1
2
RF
1k⍀
75⍀
10k⍀
75⍀
COAX
1000␮F
RT
75⍀
0.1␮F
VOUT
RL
75⍀
RG
1k⍀
As the upper supply is lowered to approach the video, the differential gain and differential phase were not significantly adversely
affected until the difference between the peak video output and
the supply reached 0.6 V. Thus, the highest video level should
be kept at least 0.6 V below the positive supply rail.
220␮F
5
1000␮F
7
6
RT
75⍀
4
0.1␮F
RG
1k⍀
VOUT
RL
75⍀
Taking the above into account, it was found that the optimal
point to bias the noninverting input is at 2.2 V dc. Operating at
this point, the worst-case differential gain is measured at 0.06%
and the worst-case differential phase is 0.06∞.
RF
1k⍀
220␮F
The ac-coupling capacitors used in the circuit at first glance
appear quite large. A composite video signal has a lower frequency band edge of 30 Hz. The resistances at the various ac
coupling points—especially at the output—are quite small. In
order to minimize phase shifts and baseline tilt, the large value
capacitors are required. For video system performance that is
not to be of the highest quality, the value of these capacitors can
be reduced by a factor of up to five with only a slightly observable change in the picture quality.
Figure 7. Single-Supply Composite Video Line
Driver Using AD8042
Signals of bounded peak-to-peak amplitude that vary in duty
cycle require larger dynamic swing capability than their peak-topeak amplitude after ac coupling. As a worst case, the dynamic
signal swing required will approach twice the peak-to-peak value.
The two bounding cases are for a duty cycle that is mostly low,
but occasionally goes high at a fraction of a percent duty cycle
and vice versa.
Single-Ended-to-Differential Driver
Composite video is not quite this demanding. One bounding
extreme is for a signal that is mostly black for an entire frame,
but has a white (full intensity), minimum width spike at least
once per frame.
Using a cross-coupled single-ended-to-differential converter, the
AD8042 makes a good general purpose differential line driver.
This can be used for applications such as driving category 5 twisted
pair wire which is becoming common for data communications
in buildings. Figure 8 shows a configuration for a circuit that
performs this function that can be used for video transmission
over a differential pair or various data communication purposes.
The other extreme is for a video signal that is full white everywhere. The blanking intervals and sync tips of such a signal will
have negative going excursions in compliance with composite
video specifications. The combination of horizontal and vertical
blanking intervals limit such a signal to being at its highest level
(white) for only about 75% of the time.
+5V
As a result of the duty cycle variations between the two extremes presented above, a 1 V p-p composite video signal that is
multiplied by a gain of two requires about 3.2 V p-p of dynamic
voltage swing at the output for an op amp to pass a composite
video signal of arbitrary duty cycle without distortion.
Some circuits use a sync tip clamp along with ac coupling to
hold the sync tips at a relatively constant level in order to lower
the amount of dynamic signal swing required. However, these
circuits can have artifacts like sync tip compression unless they
are driven by sources with very low output impedance.
VIN
RIN
1k⍀
49.9⍀
3
8
2 AMP1
10␮F
RF
1 1k⍀
60.4⍀
RA
1k⍀
AD8042
RB
1k⍀
6
50m
RB
1k⍀
121⍀
VOUT
RA
1k⍀
5 AMP2
4
The AD8042 not only has ample signal swing capability to
handle the dynamic range required without using a sync tip
clamp, but also has good video specifications like differential
gain and differential phase when buffering these signals in an
ac-coupled configuration.
0.1␮F
7
60.4⍀
100⍀
0.1␮F
10␮F
–5V
Figure 8. Single-Ended-to-Differential Twisted
Pair Line Driver
–12–
REV. C
AD8042
Each of the AD8042’s op amps is configured as a unity gain
follower by the feedback resistors (RA). Each op amp output
also drives the other as a unity gain inverter via the two RBS,
creating a totally symmetrical circuit.
+5V
+5V
0.1␮F
If the + input to Amp 2 is grounded and a small positive signal
is applied to the + input of Amp 1, the output of Amp 1 will be
driven to saturation in the positive direction and the input of
Amp 2 driven to saturation in the negative direction. This is
similar to the way a conventional op amp behaves without any
feedback.
0.1␮F
1k⍀
VIN
3
1
2
+5V
1k⍀
AD8042
1k⍀
1k⍀
6
7
2.49k⍀
2.49k⍀
0.1␮F
100
90
DVDD
AVDD
AVDD
14
OTR
13
BIT1
12
BIT2
11
BIT3
10
BIT4
9
BIT5
8
BIT6
7
BIT7
6
BIT8
5
BIT9
4
BIT10
3
BIT11
2
BIT12
CAPT
0.1␮F
10/16
0.1␮F
AD9220
CAPB
18
0.1␮F
17
VREF
SENSE
22
CML
0.1␮F
1
CLOCK
CLK
REFCOM
19
0.1␮F
26
15
4
DVSS AVSS AVSS
27
25
16
Figure 10. AD8042 Differential Driver for
the AD9220 12-Bit, 10-MSPS A/D Converter
The scope photo in Figure 9 shows a 10 MHz, 2 V p-p input signal
driving the circuit with 50 m of category 5 twisted pair wire.
VIN
0.1␮F
28
VINB
5
The cable has a characteristic impedance of about 120 W. Each
driver output is back terminated with a pair of 60.4 W resistors
to make the source look like 120 W. The receive end is terminated with 121 W, and the signal is measured differentially with
a pair of scope probes. One channel on the oscilloscope is
inverted and then the signals are added.
50ns
+5V
VINA
1k⍀
The gain of this circuit from input to either output will be ± RF/
RI. Or the single-ended-to-differential gain will be 2 ¥ RF/RI.
This gives the circuit the advantage of being able to adjust its
gain by changing a single resistor.
200mV
+5V
0.1␮F
+5V
If a resistor (RF) is connected from the output of Amp 2 to the
+ input of Amp 1, negative feedback is provided which closes
the loop. An input resistor (RI) will make the circuit look like a
conventional inverting op amp configuration with differential
outputs.
1V
1k⍀
8
The circuit was tested with a 1 MHz input signal and clocked at
10 MHz. An FFT response of the digital output is shown in
Figure 11.
Pin 5 is biased at 2.5 V by the voltage divider and bypassed. This
biases each output at 2.5 V. VIN is ac coupled such that VIN going
positive makes VINA go positive and VINB go in the negative
direction. The opposite happens for a negative going VIN.
1
VERTICAL SCALE (15dB/DIV)
VOUT
10
0%
200mV
Figure 9. Differential Driver Frequency Response
Single-Supply Differential A/D Driver
The single-ended-to-differential converter circuit is also useful
as a differential driver for video speed, single-ended, differential
input A/D converters. Figure 10 is a schematic that shows such
a circuit differentially driving an AD9220, a 12-bit, 10-MSPS
A/D converter.
9
FUND FRQ 1000977
SMPL FRQ 10000000
3
7
2
8
THD
SNR
SINAD
SFDR
–82.00
71.13
70.79
–86.74
6
4
5
HARMONICS (dBc)
2nd –88.34 6th –99.47
3rd –86.74 7th –91.16
4th –99.26 8th –97.25
5th –90.67 9th –91.61
Figure 11. FFT of AD9220 Output When Driven by AD8042
REV. C
–13–
AD8042
HDSL Line Driver
LAYOUT CONSIDERATIONS
HDSL or high-bit-rate digital subscriber line is becoming popular as a means to provide data communication at DS1 rates
(1.544 MBPS) over moderate distances via conventional telephone twisted pair wires. In these systems, the transceiver at the
customer’s end is sometimes powered via the twisted pair from
a power source at the central office. It is sometimes required to
raise the dc voltage of the power source to compensate for IR
drops in long lines or lines with narrow gauge wires.
The specified high speed performance of the AD8042 requires
careful attention to board layout and component selection.
Proper RF design techniques and low-pass parasitic component
selection are necessary.
Because of this, it is highly desirable to keep the power consumption of the customer’s transceiver as low as possible. One means
to realize significant power savings is to run the transceiver from
a ± 5 V supply instead of the more conventional ± 12 V.
Chip capacitors should be used for the supply bypassing. One
end should be connected to the ground plane and the other
within 1/8 inch of each power pin. An additional large
(0.47 mF–10 mF) tantalum electrolytic capacitor should be
connected in parallel, but not necessarily so close, to supply
current for fast, large signal changes at the output.
The high output swing and current drive capability of the
AD8042 make it ideally suited to this application. Figure 12
shows a circuit for the analog portion of an HDSL transceiver
using the AD8042 as the line driver.
2k⍀
3k⍀
6
232⍀
ATT
2718AF
93DJ39
7
VIN
1/2
AD8042
2k⍀
1
4
10
5
2
7
9
6
3k⍀
2
The feedback resistor should be located close to the inverting
input pin in order to keep the stray capacitance at this node to a
minimum. Capacitance variations of less than 1 pF at the
inverting input will significantly affect high speed performance.
Stripline design techniques should be used for long signal traces
(greater than about one inch). These should be designed with a
characteristic impedance of 50 W or 75 W and be properly terminated at each end.
VOUT
5
The PCB should have a ground plane covering all unused
portions of the component side of the board to provide a low
impedance path. The ground plane should be removed from the
area near the input pins to reduce the stray capacitance.
1
3
0.001␮F
1/2
AD8042
912⍀
0.0027␮F
34⍀
2k⍀
2k⍀
2
2k⍀
2k⍀
1
3
2k⍀
249⍀
VREC
1/4
AD8044
0.001␮F
Figure 12. HDSL Line Driver
–14–
REV. C
AD8042
OUTLINE DIMENSIONS
8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
Dimensions shown in inches and (millimeters)
0.375 (9.53)
0.365 (9.27)
0.355 (9.02)
8
5
1
4
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.100 (2.54)
BSC
0.015
(0.38)
MIN
0.180
(4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79)
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
SEATING
PLANE
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
COMPLIANT TO JEDEC STANDARDS MO-095AA
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
8
5
1
4
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0040)
COPLANARITY
SEATING
0.10
PLANE
6.20 (0.2440)
5.80 (0.2284)
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
0.31 (0.0122)
0.50 (0.0196)
ⴛ 45ⴗ
0.25 (0.0099)
8ⴗ
0.25 (0.0098) 0ⴗ 1.27 (0.0500)
0.40 (0.0157)
0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Revision History
Location
Page
8/04—Data Sheet changed from REV. B to REV. C.
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Changes to OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
7/02—Data Sheet changed from REV. A to REV. B.
Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
REV. C
–15–
–16–
C01059–0–8/04(C)