LINER LT1223 100mhz current feedback amplifier Datasheet

LT1223
100MHz Current
Feedback Amplifier
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FEATURES
DESCRIPTIO
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The LT®1223 is a 100MHz current feedback amplifier with
very good DC characteristics. The LT1223’s high slew
rate, 1000V/µs, wide supply range, ±15V, and large output
drive, ±50mA, make it ideal for driving analog signals over
double-terminated cables. The current feedback amplifier
has high gain bandwidth at high gains, unlike conventional
op amps.
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100MHz Bandwidth at AV = 1
1000V/µs Slew Rate
Wide Supply Range: ±5V to ±15V
1mV Input Offset Voltage
1µA Input Bias Current
5MΩ Input Resistance
75ns Settling Time to 0.1%
50mA Output Current
6mA Quiescent Current
Available in 8-Lead Plastic DIP and SO Packages
The LT1223 comes in the industry standard pinout and
can upgrade the performance of many older products.
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APPLICATIO S
The LT1223 is manufactured on Linear Technology’s
proprietary complementary bipolar process.
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, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
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Video Amplifiers
Buffers
IF and RF Amplification
Cable Drivers
8-, 10-, 12-Bit Data Acquisition Systems
TYPICAL APPLICATIO
Video Cable Driver
Voltage Gain vs Frequency
60
LT1223
–
RF
1k
40
75Ω
CABLE
VOUT
RG
1k
100MHz GAIN
BANDWIDTH
50
75Ω
VOLTAGE GAIN (dB)
+
V IN
30
20
10
0
75Ω
RG = 10
–
RG = 33
RG
1k
RG = 110
RG = 470
RG = ∞
–10
–20
100k
R
AV = 1 + F
RG
AT AMPLIFIER OUTPUT
6dB LESS AT VOUT
+
1M
10M
100M
1G
FREQUENCY (Hz)
LT1223 • TPC01
LT1223 • TA02
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LT1223
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AXI U RATI GS
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ABSOLUTE
PACKAGE/ORDER I FOR ATIO
(Note 1)
ORDER PART
NUMBER
TOP VIEW
Supply Voltage ...................................................... ±18V
Differential Input Voltage ......................................... ±5V
Input Voltage ............................ Equal to Supply Voltage
Output Short Circuit Duration (Note 2) ......... Continuous
Operating Temperature Range
LT1223M (OBSOLETE) .............. –55°C to 125°C
LT1223C ................................................ 0°C to 70°C
Storage Temperature Range ..................–65°C to 150°C
Junction Temperature Plastic Package ........... 150°C
Junction Temperature Ceramic Package
(OBSOLETE) ..................................... 175°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
NULL 1
8
SHUTDOWN
–IN 2
7
V+
+IN 3
6
OUT
V– 4
5
NULL
N8 PACKAGE
8-LEAD PLASTIC DIP
LT1223CN8
LT1223CS8
S8 PART MARKING
S8 PACKAGE
8-LEAD PLASTIC SO
1223
TJ MAX = 150°C, θJA = 100°C/W(N8)
TJ MAX = 150°C, θJA = 150°C/W(S8)
J8 PACKAGE
8-LEAD CERAMIC DIP
TJMAX = 175°C, θJA = 100°CW (J8)
LT1223CJ8
LT1223MJ8
OBSOLETE PACKAGE
Consider the N8 or S8 for Alternative Source
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
VS = ± 15V, TA = 25°C, unless otherwise noted.
LT1223M/C
TYP
MAX
VCM = 0V
±1
±3
mV
Noninverting Input Current
VCM = 0V
±1
±3
µA
Inverting Input Current
VCM = 0V
±1
±3
µA
en
Input Noise Voltage Density
f = 1kHz, RF = 1k, RG = 10Ω
3.3
nV/√Hz
in
Input Noise Current Density
f = 1kHz, RF = 1k, RG = 10Ω
2.2
pA/√Hz
RIN
Input Resistance
VIN = ±10V
10
MΩ
CIN
Input Capacitance
1.5
pF
CMRR
Common Mode Rejection Ratio
VCM = ±10V
Inverting Input Current Common Mode Rejection
VCM = ±10V
Power Supply Rejection Ratio
VS = ±4.5V to ±18V
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
IIN+
IIN–
Input Voltage Range
PSRR
MIN
1
UNITS
±10
±12
V
56
63
dB
30
68
100
80
nA/V
dB
12
100
nA/V
60
500
nA/V
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
AV
Large Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
70
89
dB
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
1.5
5
MΩ
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
±10
±12
V
IOUT
Maximum Output Current
RLOAD = 200Ω
50
60
mA
SR
Slew Rate
RF = 1.5k, RG = 1.5k (Note 3)
800
1300
V/µs
BW
Bandwidth
RF = 1k, RG = 1k, VOUT = 100mV
100
MHz
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LT1223
ELECTRICAL CHARACTERISTICS
VS = ± 15V, TA = 25°C, unless otherwise noted.
PARAMETER
CONDITIONS
tr
Rise Time
RF = 1.5k, RG = 1.5k, VOUT = 1V
6.0
ns
tPD
Propagation Delay
RF = 1.5k, RG = 1.5k, VOUT = 1V
6.0
ns
Overshoot
RF = 1.5k, RG = 1.5k, VOUT = 1V
5
%
Settling Time, 0.1%
RF = 1k, RG = 1k, VOUT = 10V
75
ns
Differential Gain
RF = 1k, RG = 1k, RL = 150Ω
0.02
%
Differential Phase
RF = 1k, RG = 1k, RL = 150Ω
0.12
Deg
ROUT
Open-Loop Output Resistance
VOUT = 0, IOUT = 0
IS
Supply Current
VIN = 0V
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
2
4
mA
ts
MIN
LT1223M/C
TYP
SYMBOL
MAX
UNITS
Ω
35
The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at VS = ± 15V,
VCM = 0V, 0°C ≤ TA ≤ 70°C, unless otherwise noted.
LT1223C
TYP
MAX
●
±1
±3
mV
VCM = 0V
●
±1
±3
µA
±1
±3
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
VCM = 0V
IIN+
Noninverting Input Current
MIN
UNITS
µA
IIN–
Inverting Input Current
VCM = 0V
●
RIN
Input Resistance
VIN = ±10V
●
1
10
●
±10
±12
V
CMRR
Common Mode Rejection Ratio
VCM = ±10V
●
56
63
dB
Inverting Input Current Common Mode Rejection
VCM = ±10V
●
Power Supply Rejection Ratio
VS = ±4.5V to ±18V
●
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
●
Input Voltage Range
PSRR
30
68
MΩ
100
nA/V
12
100
nA/V
60
500
nA/ V
80
dB
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
●
AV
Large-Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
●
70
89
dB
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
●
1.5
5
MΩ
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
●
±10
±12
IOUT
Maximum Output Current
RLOAD = 200Ω
●
50
60
IS
Supply Current
VIN = 0V
●
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
●
2
4
mA
V
mA
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LT1223
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at VS = ± 15V, VCM = 0V, – 55°C ≤ TA ≤ 125°C, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
VCM = 0V
LT1223M
TYP
MAX
●
±1
±5
MIN
UNITS
mV
Noninverting Input Current
VCM = 0V
●
±1
±5
µA
IIN–
Inverting Input Current
VCM = 0V
●
±1
±10
µA
RIN
Input Resistance
VIN = ±10V
●
1
10
MΩ
●
±10
±12
V
VCM = ±10V
●
56
63
dB
IIN
+
Input Voltage Range
CMRR
Common Mode Rejection Ratio
Inverting Input Current Common Mode Rejection
VCM = ±10V
●
PSRR
Power Supply Rejection Ratio
VS = ±4.5V to ±15V
●
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±15V
●
12
200
nA/V
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±15V
●
60
500
nA/V
AV
Large-Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
●
70
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
●
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
●
IOUT
Maximum Output Current
RLOAD = 200Ω
●
IS
Supply Current
VIN = 0V
●
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
●
2
4
mA
30
68
100
80
nA/V
dB
89
dB
1.5
5
MΩ
±7
±12
35
60
V
mA
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: A heat sink may be required.
Note 3: Noninverting operation, VOUT = ±10V, measured at ±5V.
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LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Supply Voltage
(Shutdown)
4
10
100
PIN 8 = 0V
125°C
SUPPLY CURRENT (mA)
SUPPLY CURRENT (mA)
8
25°C
6
–55°C
4
3
25°C
125°C
2
–55°C
1
2
2
4
6
8
0
10 12 14 16 18 20
2
4
SUPPLY VOLTAGE (±V)
6
8
70
60
50
40
30
20
10
5
–1
4
V S = 15V
IB (µA)
3
1
V–
–50
VS = ±5V
0
25
50
75
100
–55°C
0
125°C
–1
–2
–3
–6
–10
–5
0
5
OUTPUT VOLTAGE SWING (V)
10
5
125°C
0
25°C
–55°C
5
10
15
COMMON MODE VOLTAGE (V)
LT1223 • TPC08
0
5
10
10
Output Voltage Swing vs
Supply Voltage
15
125°C
25°C, –55°C
5
0
–5
–10
–20
100
15
20
25°C, –55°C
125°C
–15
–15
–5
LT1223 ∑ TPC07
VS = ±15V
15
0
–10
COMMON MODE VOLTAGE (V)
Output Voltage Swing vs
Load Resistor
20
–5
15
LT1223 • TPC06
15
–10
10
–10
–15
COMMON MODE VOLTAGE (V)
V S = ±15V
–20
–15
25°C
–8
LT1223 • TPC05
–10
–55°C
0
–4
TEMPERATURE (°C)
–5
V S = ±15V
2
–2
VOS vs Common Mode Voltage
125
4
25°C
–5
–15
125
100
125°C
–4
–25
75
6
1
VS = ±15V
8
–IB (µA)
VS = 5V
4
50
–IB vs Common Mode Voltage
10
3
–4
25
LT1223 • TPC04
VS = ±15V
2
2
0
CASE TEMPERATURE (°C)
+IB vs Common Mode Voltage
V+
–3
–25
LT1223 • TPC03
Input Common Mode Limit vs
Temperature
–2
0
–50
SUPPLY VOLTAGE (±V)
LT1223 • TPC02
COMMON MODE RANGE (V)
80
10 12 14 16 18 20
OUTPUT VOLTAGE SWING (V)
0
VOS (mV)
90
0
0
20
Output Short Circuit-Current vs
Temperature
OUTPUT SHORT CIRCUIT CURRENT (mA)
Supply Current vs Supply Voltage,
VIN = 0 (Operating)
125°C
10
–55°C
25°C
5
0
25°C
–5
–10
125°C
–55°C
–15
–20
1000
10000
LOAD RESISTOR (Ω)
0
2
4
6
8
10 12 14 16 18 20
SUPPLY VOLTAGE (±V)
LT1223 • TPC09
LT1223 • TPC10
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LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
80
70
60
50
40
30
80
70
40
RF = 1.5k
30
10
10
0
0
1
2
3
FEEDBACK RESISTOR (k Ω)
RF = 1k
50
20
RF = 2k
5
10
SUPPLY VOLTAGE (± V)
100
3
VS = ± 15V
9 V = ± 10V
O
80
125°C
60
8
7
25°C
6
125°C
5
4
–55°C
3
2
50
VS = ±15V
VO = ± 10V
1000
1
0
100
10000
1000
Power Supply Rejection vs
Frequency
POWER SUPPLY REJECTION (dB)
en
60
Output Impedance vs Frequency
VS = ±15V
RF = 1k
POSITIVE
40
NEGATIVE
20
+i n
1
10k
FREQUENCY (Hz)
0
10k
100k
1M
10M
100M
FREQUENCY (Hz)
LT1223 • TPC17
LT1223 • TPC16
LT1223 • TPC15
80
–i n
10000
LOAD RESISTOR (Ω)
LOAD RESISTOR (Ω)
1000
SPOT NOISE (nV/√Hz OR pA/√Hz)
Transimpedance vs Load Resistor
–55°C
70
60
50
LT1223 • TPC13
90
Spot Noise Voltage and Current vs
Frequency
1k
20
40
30
VOLTAGE GAIN (V/V)
10
10
LT1223 • TPC14
100
300
0
25°C
40
100
10
10
2dB PEAKING
400
15
TRANSIMPEDANCE (MΩ)
OPEN LOOP VOLTAGE GAIN (dB)
CAPACITIVE LOAD (pF)
1k
10
500
Open-Loop Voltage Gain vs
Load Resistor
A V = 2; RF = RG
R L = 100; VS = ± 15V
PEAKING < 5dB
100
0dB PEAKING
600
LT1223 • TPC12
100
1
2
FEEDBACK RESISTOR (kΩ)
700
100
0
Maximum Capacitive Load vs
Feedback Resistor
0
800
200
LT1223 • TPC11
10k
VS = ±15V
R L = 100
900
RF = 750
60
20
0
1000
RF = R G
AV = 2
RL = 100 Ω
TA = 25°C
90
–3dB BANDWIDTH (MHz)
–3dB BANDWIDTH (MHz)
100
A V = 2; RF = RG
R L = 100 Ω ; VS = ±15V
NO CAPACITIVE LOAD
90
Minimum Feedback Resistor vs
Voltage Gain
100
MAGNITUDE OF OUTPUT IMPEDANCE (Ω)
100
–3dB Bandwidth vs
Supply Voltage
FEEDBACK RESISTOR (Ω)
–3dB Bandwidth vs
Feedback Resistor
VS = ±15V
10
1
RF = RG = 3k
RF = RG = 1k
0.1
0.01
10k
100k
1M
10M
100M
FREQUENCY (Hz)
LT1223 • TPC18
LT1223 • TPC19
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LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
225
VS = ±15V
180
RF = RG = 1k
10
135
GAIN
5
RL = 100Ω
90
RL ≥ 1k
0
45
PHASE
–5
–10
RL = 100Ω
0
RL ≥ 1k
–45
0.1
TOTAL HARMONIC DISTORTION (%)
15
PHASE SHIFT (DEGREES)
VOLTAGE GAIN (dB)
20
Total Harmonic Distortion vs
Frequency
–15
–90
–20
–135
–25
–180
–225
–30
1M
10M
100M
–20
VS = ±15V
VO = 7VRMS
RL = 400 Ω
RF = RG =1k
0.01
THD
0.001
1G
100
1k
10k
100k
10
1
6
LT1223 • TPC22
2
0
–2
–4
6
TO 1mV
4
2
0
–2
–4
TO 1mV
0
–2
–4
–8
–8
–10
0
1
2
SETTLING TIME (µs)
0
20
40
60
80
100
SETTLING TIME (ns)
LT1223 • TPC24
LT1223 • TPC23
TO 1mV
TO 10mV
–6
–10
100
TO 1mV
2
–10
80
TO 10mV
4
–8
SETTLING TIME (ns)
A V = –1
RF = 1k
VS = ± 15V
RL = 1k
8
–6
TO 10mV
Inverting Amplifier Settling
Time vs Output Step
OUTPUT STEP (V)
TO 10mV
60
100
FREQUENCY (MHz)
10
A V = +1
R F = 1k
VS = ± 15V
RL = 1k
8
OUTPUT STEP (V)
OUTPUT STEP (V)
–50
Noninverting Amplifier Settling
Time to 1mV vs Output Step
10
40
3RD
LT1223 • TPC21
10
20
2ND
–40
FREQUENCY (Hz)
Noninverting Amplifier Settling
Time to 10mV vs Output Step
0
= ± 15V
= 2VP-P
= 100
= 1k
= 10dB
–70
10
LT1223 • TPC20
–6
VS
VO
R
–30 RL
F
AV
–60
FREQUENCY (Hz)
A V = +1
8 RF = 1k
VS = ± 15V
6
RL = 1k
4
2nd and 3rd Harmonic
Distortion vs Frequency
DISTORTION (dBc)
Voltage Gain and Phase vs
Frequency
LT1223 • TPC25
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APPLICATIO S I FOR ATIO
Current Feedback Basics
The small-signal bandwidth of the LT1223, like all current
feedback amplifiers, isn’t a straight inverse function of the
closed-loop gain. This is because the feedback resistors
determine the amount of current driving the amplifier’s
internal compensation capacitor. In fact, the amplifier’s
feedback resistor (RF) from output to inverting input
works with internal junction capacitances of the LT1223 to
set the closed-loop bandwidth.
Even though the gain set resistor (RG) from inverting input
to ground works with RF to set the voltage gain just like it
does in a voltage feedback op amp, the closed-loop
bandwidth does not change. This is because the equivalent gain bandwidth product of the current feedback amplifier is set by the Thevenin equivalent resistance at the
inverting input and the internal compensation capacitor.
By keeping RF constant and changing the gain with RG, the
Thevenin resistance changes by the same amount as the
change in gain. As a result, the net closed-loop bandwidth
of the LT1223 remains the same for various closed-loop
gains.
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LT1223
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APPLICATIO S I FOR ATIO
The curve on the first page shows the LT1223 voltage gain
versus frequency while driving 100Ω, for five gain settings
from 1 to 100. The feedback resistor is a constant 1k and
the gain resistor is varied from infinity to 10Ω. Shown for
comparison is a plot of the fixed 100MHz gain bandwidth
limitation that a voltage feedback amplifier would have. It
is obvious that for gains greater than one, the LT1223
provides 3 to 20 times more bandwidth. It is also evident
that second order effects reduce the bandwidth somewhat
at the higher gain settings.
Feedback Resistor Selection
Because the feedback resistor determines the compensation of the LT1223, bandwidth and transient response can
be optimized for almost every application. To increase the
bandwidth when using higher gains, the feedback resistor
(and gain resistor) can be reduced from the nominal 1k
value. The Minimum Feedback Resistor versus Voltage
Gain curve shows the values to use for ±15V supplies.
Larger feedback resistors can also be used to slow down
the LT1223 as shown in the –3dB Bandwidth versus
Feedback Resistor curve.
Capacitive Loads
The LT1223 can be isolated from capacitive loads with a
small resistor (10Ω to 20Ω) or it can drive the capacitive
load directly if the feedback resistor is increased. Both
techniques lower the amplifier’s bandwidth about the
same amount. The advantage of resistive isolation is that
the bandwidth is only reduced when the capacitive load is
present. The disadvantage of resistor isolation is that
resistive loading causes gain errors. Because the DC
accuracy is not degraded with resistive loading, the desired way of driving capacitive loads, such as flash converters, is to increase the feedback resistor. The Maximum
Capacitive Load versus Feedback Resistor curve shows
the value of feedback resistor and capacitive load that
gives 5dB of peaking. For less peaking, use a larger
feedback resistor.
Power Supplies
The LT1223 may be operated with single or split supplies
as low as ±4V (8V total) to as high as ±18V (36V total). It
is not necessary to use equal value split supplies, however, the offset voltage will degrade about 350µV per volt
of mismatch. The internal compensation capacitor decreases with increasing supply voltage. The –3dB Bandwidth versus Supply Voltage curve shows how this affects
the bandwidth for various feedback resistors. Generally,
the bandwidth at ±5V supplies is about half the value it is
at ±15V supplies for a given feedback resistor.
The LT1223 is very stable even with minimal supply
bypassing, however, the transient response will suffer if
the supply rings. It is recommended for good slew rate and
settling time that 4.7µF tantalum capacitors be placed
within 0.5 inches of the supply pins.
Input Range
The noninverting input of the LT1223 looks like a 10M
resistor in parallel with a 3pF capacitor until the common
mode range is exceeded. The input impedance drops
somewhat and the input current rises to about 10µA when
the input comes too close to the supplies. Eventually,
when the input exceeds the supply by one diode drop, the
base collector junction of the input transistor forward
biases and the input current rises dramatically. The input
current should be limited to 10mA when exceeding the
supplies. The amplifier will recover quickly when the input
is returned to its normal common mode range unless the
input was over 500mV beyond the supplies, then it will
take an extra 100ns.
Offset Adjust
Output offset voltage is equal to the input offset voltage
times the gain plus the inverting input bias current times
the feedback resistor. For low gain applications (3 or less)
a 10kΩ pot connected to Pins 1 and 5 with wiper to V+ will
trim the inverting input current (±10µA) to null the output;
it does not change the offset voltage very much. If the
LT1223 is used in a high gain application, where input
offset voltage is the dominate error, it can be nulled by
pulling approximately 100µA from Pin 1 or 5. The easy
way to do this is to use a 10kΩ pot between Pin 1 and 5 with
a 150k resistor from the wiper to ground for 15V supply
applications. Use a 47k resistor when operating on a 5V
supply.
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LT1223
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APPLICATIO S I FOR ATIO
Shutdown
Output Slew Rate of 500V/µs
Pin 8 activates a shutdown control function. Pulling more
than 200mA from Pin 8 drops the supply current to less
than 3mA, and puts the output into a high impedance state.
The easy way to force shutdown is to ground Pin 8, using
an open collector (drain) logic stage. An internal resistor
limits current, allowing direct interfacing with no additional parts. When Pin 8 is open, the LT1223 operates
normally.
Slew Rate
The slew rate of a current feedback amplifier is not independent of the amplifier gain configuration the way it is in
a traditional op amp. This is because the input stage and
the output stage both have slew rate limitations. Inverting
amplifiers do not slew the input and are therefore limited
only by the output stage. High gain, noninverting amplifiers are similar. The input stage slew rate of the LT1223 is
about 350V/µs before it becomes nonlinear and is enhanced by the normally reverse-biased emitters on the
input transistors. The output slew rate depends on the size
of the feedback resistors. The peak output slew rate is
about 2000V/µs with a 1k feedback resistor and drops
proportionally for larger values. At an output slew rate of
1000V/µs or more, the transistors in the “mirror circuits”
will begin to saturate due to the large feedback currents.
This causes the output to have slew induced overshoot and
is somewhat unusual looking; it is in no way harmful or
dangerous to the device. The photos show the LT1223 in
a noninverting gain of three (RF = 1k, RG = 500Ω) with a
20V peak-to-peak output slewing at 500V/µs, 1000V/µs
and 2000V/µs.
1223 A01
Output Slew Rate of 1000V/µs
1223 A02
Output Slew Rate at 2000V/µs Shows Aberrations (See Text)
Settling Time
The Inverting Amplifier Settling Time versus Output Step
curve shows that the LT1223 will settle to within 1mV of
final value in less than 100ns for all output changes of 10V
or less. When operated as an inverting amplifier there is
less than 500µV of thermal settling in the amplifier.
However, when operating the LT1223 as a noninverting
amplifier, there is an additional thermal settling component that is about 200µV for every volt of input common
mode change. So a noninverting gain of one amplifier will
1223 A03
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9
LT1223
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APPLICATIO S I FOR ATIO
have about 2.5mV thermal tail on a 10V step. Unfortunately, reducing the input signal and increasing the gain
always results in a thermal tail of about the same amount
for a given output step. For this reason we show separate
graphs of 10mV and 1mV non-inverting amplifier settling
times. Just as the bandwidth of the LT1223 is fairly
constant for various closed-loop gains, the settling time
remains constant as well.
Adjustable Gain Amplifier
To make a variable gain amplifier with the LT1223, vary the
value of RG. The implementation of RG can be a pot, a light
controlled resistor, a FET, or any other low capacitance
variable resistor. The value of RF should not be varied to
change the gain. If RF is changed, then the bandwidth will
be reduced at maximum gain and the circuit will oscillate
when RF is very small.
Accurate Bandwidth Limiting the LT1223
It is very common to limit the bandwidth of an op amp by
putting a small capacitor in parallel with RF. DO NOT PUT
A SMALL CAPACITOR FROM THE INVERTING INPUT OF
A CURRENT FEEDBACK AMPLIFIER TO ANYWHERE ELSE,
ESPECIALLY NOT TO THE OUTPUT. The capacitor on the
inverting input will cause peaking or oscillations. If you
need to limit the bandwidth of a current feedback amplifier,
use a resistor and capacitor at the noninverting input (R1
and C1). This technique will also cancel (to a degree) the
peaking caused by stray capacitance at the inverting input.
Unfortunately, this will not limit the output noise the way
it does for the op amp.
V IN
R1
+
LT1223
C1
VOUT
–
+
V IN
R1 = 300Ω
C1 = 100pF
BW = 5MHz
VOUT
LT1223
–
RF
RG
RF
LT1223 • TA05
RG
LT1223 • TA03
Current Feedback Amplifier Integrator
Adjustable Bandwidth Amplifier
Because the resistance at the inverting input determines
the bandwidth of the LT1223, an adjustable bandwidth
circuit can be made easily. The gain is set as before with
RF and RG; the bandwidth is maximum when the variable
resistor is at a minimum.
Since we remember that the inverting input wants to see
a resistor, we can add one to the standard integrator
circuit. This generates a new summing node where we can
apply capacitive feedback. The LT1223 integrator has
excellent large signal capability and accurate phase shift at
high frequencies.
+
+
V IN
LT1223
VOUT
–
LT1223
VOUT
= 1
sC1R1
VIN
5k
RG
R1
RF
VOUT
–
RF
1k
C1
VIN
LT1223 • TA06
LT1223 • TA04
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10
LT1223
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APPLICATIO S I FOR ATIO
Summing Amplifier (DC Accurate)
The summing amplifier is easily made by adding additional
inputs to the basic inverting amplifier configuration. The
LT1223 has no IOS spec because there is no correlation
between the two input bias currents. Therefore, we will not
improve the DC accuracy of the inverting amplifier by
putting in the extra resistor in the noninverting input.
+
VOUT
LT1223
R 1
G
–
V I1
R 2
R
G
V I2
•
•
•
R n
G
F
VOUT = –RF
(
VI1
V
V
+ I2 + In
R G1 R G2 R Gn
VIn
)
inverting input (A1) senses the shield and the non-inverting input (A2) senses the center conductor. Since this
amplifier does not load the cable (take care to minimize
stray capacitance) and it rejects common mode hum and
noise, several amplifiers can sense the signal with only
one termination at the end of the cable. The design
equations are simple. Just select the gain you need (it
should be two or more) and the value of the feedback
resistor (typically 1k) and calculate RG1 and RG2. The gain
can be tweaked with RG2 and the CMRR with RG1 if needed.
The bandwidth of the noninverting input signal is not
reduced by the presence of the other amplifier, however,
the inverting input signal bandwidth is reduced since it
passes two amplifiers. The CMRR is good at high frequencies because the bandwidth of the amplifiers are about the
same even though they do not necessarily operate at the
same gain.
LT1223 • TA07
RG1
1k
RF1
1k
RG2
1k
RF2
1k
Difference Amplifier
The LT1223 difference amplifier delivers excellent
performance if the source impedance is very low. This is
because the common mode input resistance is only equal
to RF + RG.
A1
LT1223
RG
(RF – 50)
100
OPTIONAL TRIM
FOR CMRR
A2
LT1223
VOUT = G (VIN+ – VIN–)
R
RF1 = RF2; RG1 = (G – 1) RF2; RG2 = F2
G–1
TRIM GAIN (G) WITH RG2; TRIM CMRR WITH RG1
VOUT
+
+
VIN –
V1
–
–
VIN +
LT1223 • TA09
+
RG
V2
VOUT =
RF
(V1 – V2)
RG
LT1223
VOUT
–
RF
LT1223 • TA08
Video Instrumentation Amplifier
This instrumentation amplifier uses two LT1223s to increase the input resistance to well over 1M. This makes an
excellent “loop through” or cable sensing amplifier if the
Cable Driver
The cable driver circuit is shown on the front page. When
driving a cable it is important to properly terminate both
ends if even modest high frequency performance is
required. The additional advantage of this is that it isolates
the capacitive load of the cable from the amplifier so it can
operate at maximum bandwidth.
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11
LT1223
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TYPICAL APPLICATIO
150mA Output Current Video Amp
V+
V+
LT1223
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
20 Ω
+
V IN
75Ω
IN
LT1010
BIAS
OUT
–
V–
V–
2k
2k
R f = 2k TO STABILIZE CIRCUIT
DIFFERENTIAL GAIN = 1%
DIFFERENTIAL PHASE = 1°
LT1223 • TA10
W
W
SI PLIFIED SCHE ATIC
7
15k
1
5
BIAS
10k
8
3
2
6
BIAS
4
LT1223 • TA01
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LT1223
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PACKAGE DESCRIPTIO
J8 Package
8-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
CORNER LEADS OPTION
(4 PLCS)
.023 – .045
(0.584 – 1.143)
HALF LEAD
OPTION
.045 – .068
(1.143 – 1.650)
FULL LEAD
OPTION
.005
(0.127)
MIN
.405
(10.287)
MAX
8
7
6
5
.025
(0.635)
RAD TYP
.220 – .310
(5.588 – 7.874)
1
.300 BSC
(7.62 BSC)
2
3
4
.200
(5.080)
MAX
.015 – .060
(0.381 – 1.524)
.008 – .018
(0.203 – 0.457)
0° – 15°
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE
OR TIN PLATE LEADS
.045 – .065
(1.143 – 1.651)
.014 – .026
(0.360 – 0.660)
.100
(2.54)
BSC
.125
3.175
MIN
J8 0801
OBSOLETE PACKAGE
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LT1223
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PACKAGE DESCRIPTIO
N8 Package
8-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
.400*
(10.160)
MAX
8
7
6
5
1
2
3
4
.255 ± .015*
(6.477 ± 0.381)
.300 – .325
(7.620 – 8.255)
.008 – .015
(0.203 – 0.381)
(
+.035
.325 –.015
8.255
+0.889
–0.381
)
.045 – .065
(1.143 – 1.651)
.130 ± .005
(3.302 ± 0.127)
.065
(1.651)
TYP
.100
(2.54)
BSC
.120
(3.048) .020
MIN (0.508)
MIN
.018 ± .003
(0.457 ± 0.076)
N8 1002
NOTE:
1. DIMENSIONS ARE
INCHES
MILLIMETERS
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010 INCH (0.254mm)
1223fb
14
LT1223
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PACKAGE DESCRIPTIO
S8 Package
8-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
.189 – .197
(4.801 – 5.004)
NOTE 3
.045 ±.005
.050 BSC
8
.245
MIN
7
6
5
.160 ±.005
.150 – .157
(3.810 – 3.988)
NOTE 3
.228 – .244
(5.791 – 6.197)
.030 ±.005
TYP
1
RECOMMENDED SOLDER PAD LAYOUT
.010 – .020
× 45°
(0.254 – 0.508)
.008 – .010
(0.203 – 0.254)
3
4
.053 – .069
(1.346 – 1.752)
.004 – .010
(0.101 – 0.254)
0°– 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. DIMENSIONS IN
2
.014 – .019
(0.355 – 0.483)
TYP
INCHES
(MILLIMETERS)
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
.050
(1.270)
BSC
SO8 0303
1223fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1223
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1206
250mA/60MHz Current Feedback Amplifier
900V/µs, Shutdown
LT1395
400MHz Current Feedback Amplifier
800V/µs, 4.6mA Supply Current, SOT-23 Package
LT1497
Dual 125mA, 50MHz Current Feedback Amplifier
900V/µs, 7mA Supply Current
LT6210/LT6211
Single/Dual Programmable Supply Current, Rail-to-Rail
Output
C-LoadTM Stable, 200MHz, 700V/µs
C-Load is a trademark of Linear Technology Corporation.
1223fb
16
Linear Technology Corporation
LT/LT 0605 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 1992