LINER LT1111CS8-5 Micropower dc/dc converter adjustable and fixed 5v, 12v Datasheet

LT1111
Micropower
DC/DC Converter
Adjustable and Fixed 5V, 12V
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DESCRIPTIO
FEATURES
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Operates at Supply Voltages from 2V to 30V
72kHz Oscillator
Works with Surface Mount Inductors
Only Three External Components Required
Step-Up or Step-Down Mode
Low-Battery Detector Comparator On-Chip
User Adjustable Current Limit
Internal 1A Power Switch
Fixed or Adjustable Output Voltage Versions
Space Saving 8-Pin MiniDIP or SO-8 Package
The LT1111 is a versatile micropower DC/DC converter.
The device requires only three external components to
deliver a fixed output of 5V or 12V. Supply voltage ranges
from 2V to 12V in step-up mode and to 30V in step-down
mode. The LT1111 functions equally well in step-up, stepdown, or inverting applications.
The LT1111 oscillator is set at 72kHz, optimizing the
device to work with off-the-shelf surface mount inductors.
The device can deliver 5V at 100mA from a 3V input in
step-up mode or 5V at 200mA from a 12V input in stepdown mode.
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APPLICATI
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S
Switch current limit can be programmed with a single
resistor. An auxiliary open-collector gain block can be
configured as a low-battery detector, linear post regulator,
undervoltage lock-out circuit, or error amplifier.
3V to 5V, 5V to 12V Converters
9V to 5V, 12V to 5V Converters
Remote Controls
Peripherals and Add-On Cards
Battery Backup Supplies
Uninterruptible Supplies
Laptop and Palmtop Computers
Cellular Telephones
Portable Instruments
Flash Memory VPP Generators
For input sources of less than 2V use the LT1110.
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TYPICAL APPLICATI
Typical Load Regulation
All Surface Mount 3V to 5V Step-Up Converter
SUMIDA
CD54-220M
MBRS120T3
22µH
5V
100mA
+
10 µ F*
V IN
SW1
LT1111CS8-5
+
33 µ F
SENSE
GND
OUTPUT VOLTAGE (V)
5
3V INPUT
I LIM
6
VIN = 2V 2.2 2.4
4
2.6
2.8 3V
3
2
1
SW2
0
0
25
50
75
100 125
150 175 200
LOAD CURRENT (mA)
*OPTIONAL
LT1111 • TA01
LT1111 • TA02
1
LT1111
W W
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AXI U
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ABSOLUTE
RATI GS
Supply Voltage (VIN) ............................................... 36V
SW1 Pin Voltage (VSW1) ......................................... 50V
SW2 Pin Voltage (VSW2) ............................ – 0.5V to VIN
Feedback Pin Voltage (LT1111) ............................. 5.5V
Switch Current ....................................................... 1.5A
Maximum Power Dissipation ............................ 500mW
Operating Temperature Range
LT1111C ............................................... 0°C to 70°C
LT1111I ......................................... – 40°C to 105°C
LT1111M ....................................... – 55°C to 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
TOP VIEW
ILIM 1
8
FB (SENSE)*
VIN 2
7
SET
SW1 3
6
A0
SW2 4
5
GND
J8 PACKAGE
8-LEAD CERAMIC DIP
N8 PACKAGE
8-LEAD PLASTIC DIP
*FIXED VERSIONS
ORDER PART
NUMBER
LT1111CN8
LT1111CN8-5
LT1111CN8-12
LT1111MJ8
LT1111MJ8-5
LT1111MJ8-12
ORDER PART
NUMBER
LT1111CS8
LT1111CS8-5
LT1111CS8-12
TOP VIEW
ILIM 1
8
FB (SENSE)*
VIN 2
7
SET
SW1 3
6
A0
SW2 4
5
GND
S8 PART MARKING
1111
11115
11111
S8 PACKAGE
8-LEAD PLASTIC SO
*FIXED VERSION
TJMAX = 90°C, θJA = 150°C/W
TJMAX = 150°C, θJA = 120°C/W (J)
TJMAX = 90°C, θJA = 130°C/W (N)
Consult factory for Industrial grade parts
ELECTRICAL CHARACTERISTICS
VIN = 3V, Military or Commercial Version
SYMBOL
PARAMETER
CONDITIONS
IQ
Quiescent Current
Switch OFF
VIN
Input Voltage
Step-Up Mode
Step-Down Mode
●
●
2.0
Comparator Trip Point Voltage
LT1111 (Note 1)
●
1.20
1.25
1.30
V
Output Sense Voltage
LT1111-5 (Note 2)
LT1111-12 (Note 2)
●
●
4.75
11.40
5.00
12.00
5.25
12.60
V
V
Comparator Hysteresis
LT1111
●
8
12.5
mV
Output Hysteresis
LT1111-5
LT1111-12
●
●
32
75
50
120
mV
mV
54
72
88
kHz
VOUT
MIN
TYP
MAX
UNITS
300
400
µA
12.6
30.0
V
V
fOSC
Oscillator Frequency
DC
Duty Cycle: Step-Up Mode
Step-Down Mode
Full Load
43
24
50
34
59
50
%
%
tON
Switch ON Time: Step-Up Mode
Step-Down Mode
ILIM Tied to VIN
VOUT, = 5V, VIN = 12V
5
3.3
7
5
9
7.8
µs
µs
VSAT
SW Saturation Voltage, Step-Up Mode
VIN = 3.0V, ISW = 650mA
VIN = 5.0V, ISW = 1A
0.5
0.8
0.65
1.0
V
V
SW Saturation Voltage, Step-Down Mode
VIN = 12V, ISW = 650mA
1.1
1.5
V
IFB
Feedback Pin Bias Current
LT1111, VFB = 0V
●
70
120
nA
ISET
Set Pin Bias Current
VSET = VREF
●
70
300
nA
VOL
Gain Block Output Low
ISINK = 300µA, VSET = 1.00V
●
0.15
0.4
V
2
LT1111
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
Reference Line Regulation
VIN = 3V, Military or Commercial Version
CONDITIONS
MIN
5V ≤ VIN ≤ 30V
2V ≤ VIN ≤ 5V
●
●
AV
Gain Block Gain
RL = 100k (Note 3)
ILIM
Current Limit
220Ω from ILIM to VIN
Current Limit Temperature Coefficient
1000
●
Switch OFF Leakage Current
Measured at SW1 Pin, VSW1 = 12V
Maximum Excursion Below GND
ISW1≤ 10µA, Switch OFF
TYP
MAX
UNITS
0.02
0.20
0.075
0.400
%/V
%/V
6000
V/V
400
mA
– 0.3
%/°C
1
10
µA
– 400
– 350
mV
LT1111M
TYP
MAX
UNITS
300
500
µA
VIN = 3V, – 55°C ≤ TA ≤ 125°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
IQ
Quiescent Current
Switch OFF
fOSC
Oscillator Frequency
DC
Duty Cycle: Step-Up Mode
Step-Down Mode
tON
VSAT
MIN
●
●
45
72
100
kHz
Full Load
●
●
40
20
50
62
55
%
%
Switch ON Time: Step-Up Mode
Step-Down Mode
ILIM Tied to VIN
VOUT = 5V, VIN = 12V
●
●
5
3
7
11
9
µs
µs
Reference Line Regulation
2V ≤ VIN ≤ 5V, 25°C ≤ TA ≤ 125°C
2.4V ≤ VIN ≤ 5V, TA = – 55°C
0.2
0.4
0.8
%/V
%/V
SW Saturation Voltage, Step-Up Mode
0°C ≤ TA ≤ 125°C, ISW = 500mA,
TA = – 55°C, ISW = 400mA
0.5
0.65
V
SW Saturation Voltage, Step-Down Mode
VIN = 12V,
0°C ≤ TA ≤ 125°C
1.5
V
ISW = 500mA
TA = – 55°C
2.0
V
VIN = 3V, 0°C ≤ TA ≤ 70°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
IQ
Quiescent Current
Switch OFF
MIN
●
300
LT1111C
TYP
MAX
UNITS
450
µA
fOSC
Oscillator Frequency
●
54
72
95
kH
DC
Duty Cycle: Step-Up Mode
Step-Down Mode
Full Load
●
●
43
24
50
34
59
50
%
%
tON
Switch ON Time: Step-Up Mode
Step-Down Mode
ILIM Tied to VIN
VOUT = 5V, VIN = 12V
●
●
5.0
3.3
7
5
9.0
7.8
µs
µs
Reference Line Regulation
2V ≤ VIN ≤ 5V
●
0.2
0.7
%/V
SW Saturation Voltage, Step-Up Mode
SW Saturation Voltage, Step-Down Mode
VIN = 3V, ISW = 650mA
VIN = 12V, ISW = 650mA
●
●
0.5
1.1
0.65
1.50
V
V
VSAT
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: This specification guarantees that both the high and low trip points
of the comparator fall within the 1.20V to 1.30V range.
Note 2: The output voltage waveform will exhibit a sawtooth shape due to
the comparator hysteresis. The output voltage on the fixed output versions
will always be within the specified range.
Note 3: 100k resistor connected between a 5V source and the A0 pin.
3
LT1111
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
Oscillator Frequency
90
10
74
9.5
9.0
70
60
8.5
72
ON TIME (µs)
80
71
70
7.0
5.5
67
–25
25
75
0
50
TEMPERATURE (°C)
100
125
0
3
6
5.0
–50
9 12 15 18 21 24 27 30
INPUT VOLTAGE (V)
0.9
56
0.8
SATURATION VOLTAGE (V)
58
48
46
44
VIN = 3V
VIN = 3V
ISW = 650mA
1.2
0.7
0.6
0.5
0.4
0.3
0.2
40
–50
–25
50
75
0
25
TEMPERATURE (°C)
100
0
–50
125
VIN = 2V
1.0
0.8
VIN = 5V
0.6
0.4
0.2
0.1
42
– 25
50
75
0
25
TEMPERATURE (°C)
LT1111 • TPC04
100
0
125
0
0.2
0.4 0.6 0.8 1.0 1.2
SWITCH CURRENT (A)
LT1111 • TPC05
Switch ON Voltage
Step-Down Mode
1.6
Minimum/Maximum Frequency
vs ON Time
100
1.4
VIN = 12V
ISW = 650mA
1.4
LT1111 • TPC06
Switch ON Voltage
Step-Down Mode
2.00
125
1.4
1.0
50
100
Saturation Voltage
Step-Up Mode
SATURATION VOLTAGE (V)
60
52
50
75
0
25
TEMPERATURE (°C)
LT111 • TPC03
Saturation Voltage
Step-Up Mode
54
–25
LT1111 • TPC02
Duty Cycle
DUTY CYCLE (%)
7.5
6.0
68
LT1111 • TPC01
VIN = 12V
1.2
OSCILLATOR FREQUENCY (KHz)
1.75
8.0
6.5
69
50
40
–50
1.0
1.50
ON VOLTAGE (V)
ON VOLTAGE (V)
Switch ON Time
75
73
FREQUENCY (KHz)
OSCILLATOR FREQUENCY (KHz)
100
1.25
1.00
0.8
0.6
0.4
0.75
0.2
90
0°C ≤ TA ≤ 70°C
80
70
60
50
–55°C ≤ TA ≤ 125°C
0.50
–50 –25
0
50
75
0
25
TEMPERATURE (°C)
100
125
LT1111 • TPC07
4
0
0.2
0.8
0.4
0.6
SWITCH CURRENT (A)
1.0
LT1111 • TPC08
40
4
5
8
9
6
7
10
SWITCH ON TIME (µs)
11
12
LT1111 • TPC09
LT1111
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TYPICAL PERFOR A CE CHARACTERISTICS
Quiescent Current
Quiescent Current
500
400
450
340
320
300
280
260
240
400
SWITCH CURRENT (A)
360
QUIESCENT CURRENT (µA)
QUIESCENT CURRENT (µA)
380
350
300
250
200
150
220
200
0
3
6
9 12 15 18 21 24 27 30
INPUT VOLTAGE (V)
100
–50
–25
0
25
50
75
TEMPERATURE (°C)
LT1111 • TPC10
100
125
Set Pin Bias Current
80
70
70
BIAS CURRENT (nA)
90
80
BIAS CURRENT (nA)
90
60
50
40
30
10
100
RLIM (Ω)
1000
LT1111 • TPC12
60
50
40
30
20
20
10
10
0
25
50
75
TEMPERATURE (°C)
STEP-DOWN
VIN = 12V
Feedback Bias Current
100
–25
STEP-UP
2V ≤ VIN ≤ 5V
LT1111 • TPC11
100
0
–50
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
Maximum Switch Current
vs RLIM
100
125
LT1111 • TPC13
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
LT1111 • TPC14
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ILIM (Pin 1): Connect this pin to VIN for normal use. Where
lower current limit is desired, connect a resistor between
ILIM and VIN. A 220Ω resistor will limit the switch current
to approximately 400mA.
VIN (Pin 2): Input Supply Voltage.
SW1 (Pin 3): Collector of Power Transistor. For step-up
mode connect to inductor/diode. For step-down mode
connect to VIN.
SW2 (Pin 4): Emitter of Power Transistor. For step-up
mode connect to ground. For step-down mode connect to
inductor/diode. This pin must never be allowed to go more
than a Schottky diode drop below ground.
GND (Pin 5): Ground.
A0 (Pin 6): Auxiliary Gain Block (GB) Output. Open collector,
can sink 300µA.
SET (Pin 7): GB Input. GB is an op amp with positive input
connected to SET pin and negative input connected to
1.25V reference.
FB/SENSE (Pin 8): On the LT1111 (adjustable) this pin
goes to the comparator input. On the LT1111-5 and
LT1111-12, this pin goes to the internal application resistor
that sets output voltage.
5
LT1111
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BLOCK DIAGRA S
LT1111
LT1111-5/LT1111-12
+
SET
SET
A2
A2
–
V IN
I LIM
GAIN BLOCK/
ERROR AMP
1.25V
REFERENCE
A1
–
DRIVER
SW2
FB
OSCILLATOR
DRIVER
COMPARATOR
COMPARATOR
GND
SW1
+
OSCILLATOR
–
I LIM
SW1
+
A1
A0
–
V IN
GAIN BLOCK/
ERROR AMP
1.25V
REFERENCE
+
A0
R1
LT1111 • BD01
SW2
R2
220k
SENSE
GND
LT1111-5: R1 = 73.5k
LT1111-12: R1 = 25.5k
LT1111 • BD02
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LT1111 OPERATI
The LT1111 is a gated oscillator switcher. This type
architecture has very low supply current because the
switch is cycled when the feedback pin voltage drops
below the reference voltage. Circuit operation can best be
understood by referring to the LT1111 block diagram.
Comparator A1 compares the feedback (FB) pin voltage
with the 1.25V reference signal. When FB drops below
1.25V, A1 switches on the 72kHz oscillator. The driver
amplifier boosts the signal level to drive the output NPN
power switch. The switch cycling action raises the output
voltage and FB pin voltage. When the FB voltage is sufficient to trip A1, the oscillator is gated off. A small amount
of hysteresis built into A1 ensures loop stability without
external frequency compensation. When the comparator
output is low, the oscillator and all high current circuitry is
turned off, lowering device quiescent current to just 300µA.
The oscillator is set internally for 7µs ON time and 7µs OFF
time, optimizing the device for circuits where VOUT and VIN
differ by roughly a factor of 2. Examples include a 3V to 5V
step-up converter or a 9V to 5V step-down converter.
6
Gain block A2 can serve as a low-battery detector. The
negative input of A2 is the 1.25V reference. A resistor
divider from VIN to GND, with the mid-point connected to
the SET pin provides the trip voltage in a low-battery
detector application. AO can sink 300µA (use a 22k
resistor pull-up to 5V).
A resistor connected between the ILIM pin and VIN sets
maximum switch current. When the switch current exceeds the set value, the switch cycle is prematurely
terminated. If current limit is not used, ILIM should be tied
directly to VIN. Propagation delay through the current limit
circuitry is approximately 1µs.
In step-up mode the switch emitter (SW2) is connected to
ground and the switch collector (SW1) drives the inductor; in step-down mode the collector is connected to VIN
and the emitter drives the inductor.
The LT1111-5 and LT1111-12 are functionally identical to
the LT1111. The -5 and -12 versions have on-chip voltage
setting resistors for fixed 5V or 12V outputs. Pin 8 on the
fixed versions should be connected to the output. No
external resistors are needed.
LT1111
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APPLICATI
S I FOR ATIO
Inductor Selection — General
PL / f OSC
A DC/DC converter operates by storing energy as magnetic flux in an inductor core, and then switching this
energy into the load. Since it is flux, not charge, that is
stored, the output voltage can be higher, lower, or opposite in polarity to the input voltage by choosing an
appropriate switching topology. To operate as an efficient
energy transfer element, the inductor must fulfill three
requirements. First, the inductance must be low enough
for the inductor to store adequate energy under the worst
case condition of minimum input voltage and switch-on
time. The inductance must also be high enough so maximum current ratings of the LT1111 and inductor are not
exceeded at the other worst case condition of maximum
input voltage and ON time. Additionally, the inductor core
must be able to store the required flux; i.e., it must not
saturate. At power levels generally encountered with
LT1111 based designs, small surface mount ferrite core
units with saturation current ratings in the 300mA to 1A
range and DCR less than 0.4Ω (depending on application)
are adequate. Lastly, the inductor must have sufficiently
low DC resistance so excessive power is not lost as heat
in the windings. An additional consideration is ElectroMagnetic Interference (EMI). Toroid and pot core type
inductors are recommended in applications where EMI
must be kept to a minimum; for example, where there are
sensitive analog circuitry or transducers nearby. Rod core
types are a less expensive choice where EMI is not a
problem. Minimum and maximum input voltage, output
voltage and output current must be established before an
inductor can be selected.
Inductor Selection — Step-Up Converter
In a step-up, or boost converter (Figure 4), power generated by the inductor makes up the difference between
input and output. Power required from the inductor is
determined by:
(
)(
PL = VOUT + V D – VIN MIN IOUT
)
(01)
where VD is the diode drop (0.5V for a 1N5818 Schottky).
Energy required by the inductor per cycle must be equal or
greater than:
(02)
in order for the converter to regulate the output.
When the switch is closed, current in the inductor builds
according to:
–R ′t 
V 
IL ( t) = IN  1– e L 
R′ 

(03)
where R′ is the sum of the switch equivalent resistance
(0.8Ω typical at 25°C) and the inductor DC resistance.
When the drop across the switch is small compared to VIN,
the simple lossless equation:
()
V
I L t = IN t
L
(04)
can be used. These equations assume that at t = 0,
inductor current is zero. This situation is called “discontinuous mode operation” in switching regulator parlance.
Setting “t” to the switch-on time from the LT1111 specification table (typically 7µs) will yield IPEAK for a specific
“L” and VIN. Once IPEAK is known, energy in the inductor
at the end of the switch-on time can be calculated as:
EL =
1 2
LI
2 PEAK
(05)
EL must be greater than PL/fOSC for the converter to deliver
the required power. For best efficiency IPEAK should be
kept to 1A or less. Higher switch currents will cause
excessive drop across the switch resulting in reduced
efficiency. In general, switch current should be held to as
low a value as possible in order to keep switch, diode and
inductor losses at a minimum.
As an example, suppose 12V at 60mA is to be generated
from a 4.5V to 8V input. Recalling equation (01),
(
)(
)
PL = 12 V + 0.5 V – 4.5 V 60mA = 480mW
(06)
Energy required from the inductor is
PL
f OSC
=
480mW
= 6.7µJ
72kHz
(07)
7
LT1111
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APPLICATI
S I FOR ATIO
Picking an inductor value of 47µH with 0.2Ω DCR results
in a peak switch current of:
I PEAK =
–1.0Ω × 7µs 
4.5 V 
1 – e 47µH  = 623mA .

1.0Ω 

(08)
Substituting IPEAK into Equation 04 results in:
(
)(
)
1
E L = 47µH 0.623 A 2 = 9.1µJ
2
(09)
L=
A resistor can be added in series with the ILIM pin to invoke
switch current limit. The resistor should be picked so the
calculated IPEAK at minimum VIN is equal to the Maximum
Switch Current (from Typical Performance Characteristic
curves). Then, as VIN increases, switch current is held
constant, resulting in increasing efficiency.
The step-down case (Figure 5) differs from the step-up in
that the inductor current flows through the load during
both the charge and discharge periods of the inductor.
Current through the switch should be limited to ~650mA
in this mode. Higher current can be obtained by using an
external switch (see Figure 6). The ILIM pin is the key to
successful operation over varying inputs.
After establishing output voltage, output current and input
voltage range, peak switch current can be calculated by the
formula:
 V OUT + V D 
V – V

SW + V D 
 IN
where DC = duty cycle (0.50)
VSW = switch drop in step-down mode
VD = diode drop (0.5V for a 1N5818)
8
VIN MIN − V SW − V OUT
× t ON
I PEAK
(11)
where tON = switch-on time (7µs).
Next, the current limit resistor RLIM is selected to give
IPEAK from the RLIM Step-Down Mode curve. The addition
of this resistor keeps maximum switch current constant as
the input voltage is increased.
As an example, suppose 5V at 300mA is to be generated
from a 12V to 24V input. Recalling Equation (10),
IPEAK =
(
)
2 300mA  5 + 0.5 
 12 – 1.5 + 0.5  = 600mA
0.50


(12)
Next, inductor value is calculated using Equation (11):
Inductor Selection — Step-Down Converter
2 I OUT
DC
VSW is actually a function of switch current which is in turn
a function of VIN, L, time, and VOUT. To simplify, 1.5V can
be used for VSW as a very conservative value.
Once IPEAK is known, inductor value can be derived from:
Since 9.1µJ > 6.7µJ, the 47µH inductor will work. This
trial-and-error approach can be used to select the optimum inductor. Keep in mind the switch current maximum
rating of 1.5A. If the calculated peak current exceeds this,
consider using the LT1110. The 70% duty cycle of the
LT1110 allows more energy per cycle to be stored in the
inductor, resulting in more output power.
IPEAK =
IOUT = output current
VOUT = output voltage
VIN = minimum input voltage
(10)
L=
12 – 1.5 – 5
7µs = 64µH.
600mA
(13)
Use the next lowest standard value (56µH).
Then pick RLIM from the curve. For IPEAK = 600mA, RLIM
= 56Ω.
Inductor Selection — Positive-to-Negative Converter
Figure 7 shows hookup for positive-to-negative conversion. All of the output power must come from the inductor.
In this case,
PL = (VOUT+ VD)(IOUT)
(14)
In this mode the switch is arranged in common collector
or step-down mode. The switch drop can be modeled as
a 0.75V source in series with a 0.65Ω resistor. When the
LT1111
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APPLICATI
S I FOR ATIO
()
IL t =
VL
R′
–R ′t 

1
–
e
L 



(15)
where R′ = 0.65Ω + DCRL
VL = VIN – 0.75V
As an example, suppose –5V at 50mA is to be generated
from a 4.5V to 5.5V input. Recalling Equation (14),
PL = (-5V+0.5V)(50mA) = 275mW
(16)
Energy required from the inductor is:
275mW
PL
=
= 3.8µJ.
72kHz
fOSC
(17)
Picking an inductor value of 56µH with 0.2Ω DCR results
in a peak switch current of:
IPEAK =
(4.5V – 0.75V) 1 – e–0.85Ω × 7µs  = 445mA .
56µH


(0.65Ω + 0.2Ω) 
(18)
capacitors provide still better performance at more expense. We recommend OS-CON capacitors from Sanyo
Corporation (San Diego, CA). These units are physically
quite small and have extremely low ESR. To illustrate,
Figures 1, 2, and 3 show the output voltage of an LT1111
based converter with three 100µF capacitors. The peak
switch current is 500mA in all cases. Figure 1 shows a
Sprague 501D, 25V aluminum capacitor. VOUT jumps by
over 120mV when the switch turns off, followed by a drop
in voltage as the inductor dumps into the capacitor. This
works out to be an ESR of over 0.24Ω. Figure 2 shows the
same circuit, but with a Sprague 150D, 20V tantalum
capacitor replacing the aluminum unit. Output jump is
now about 35mV, corresponding to an ESR of 0.07Ω.
Figure 3 shows the circuit with a 16V OS-CON unit. ESR
is now only 0.02Ω.
50mV/DIV
switch closes, current in the inductor builds according to
Substituting IPEAK into Equation (04) results in:
(
)(
)
1
E L = 56µH 0.445 A 2 = 5.54µJ.
2
5µs/DIV
LT1111 • F01
Figure 1. Aluminum
(19)
With this relatively small input range, RLIM is not usually
necessary and the ILIM pin can be tied directly to VIN. As in
the step-down case, peak switch current should be limited
to ~650mA.
50mV/DIV
Since 5.54µJ > 3.82µJ, the 56µH inductor will work.
5µs/DIV
Capacitor Selection
Figure 2. Tantalum
50mV/DIV
Selecting the right output capacitor is almost as important
as selecting the right inductor. A poor choice for a filter
capacitor can result in poor efficiency and/or high output
ripple. Ordinary aluminum electrolytics, while inexpensive
and readily available, may have unacceptably poor Equivalent Series Resistance (ESR) and ESL (inductance). There
are low ESR aluminum capacitors on the market specifically designed for switch mode DC/DC converters which
work much better than general-purpose units. Tantalum
LT1111 • F02
5µs/DIV
LT1111 • F01
Figure 3. OS-CON
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At the end of the switch ON time the current in L1 is1:
Diode Selection
Speed, forward drop, and leakage current are the three
main considerations in selecting a catch diode for LT1111
converters. General purpose rectifiers such as the 1N4001
are unsuitable for use in any switching regulator application. Although they are rated at 1A, the switching time of
a 1N4001 is in the 10µs to 50µs range. At best, efficiency
will be severely compromised when these diodes are
used; at worst, the circuit may not work at all. Most
LT1111 circuits will be well served by a 1N5818 Schottky
diode, or its surface mount equivalent, the MBRS130T3.
The combination of 500mV forward drop at 1A current,
fast turn ON and turn OFF time, and 4µA to 10µA leakage
current fit nicely with LT1111 requirements. At peak
switch currents of 100mA or less, a 1N4148 signal diode
may be used. This diode has leakage current in the 1nA to
5nA range at 25°C and lower cost than a 1N5818. (You can
also use them to get your circuit up and running, but
beware of destroying the diode at 1A switch currents.)
Step-Up (Boost Mode) Operation
A step-up DC/DC converter delivers an output voltage
higher than the input voltage. Step-up converters are not
short-circuit protected since there is a DC path from input
to output.
IPEAK =
VIN
L
t ON
(20)
Immediately after switch turn-off, the SW1 voltage pin
starts to rise because current cannot instantaneously stop
flowing in L1. When the voltage reaches VOUT + VD, the
inductor current flows through D1 into C1, increasing
VOUT. This action is repeated as needed by the LT1111 to
keep VFB at the internal reference voltage of 1.25V. R1 and
R2 set the output voltage according to the formula
 R2 
VOUT =  1 +  1.25 V
R1

(
)
(21)
Step-Down (Buck Mode) Operation
A step-down DC/DC converter converts a higher voltage
to a lower voltage. The usual hookup for an LT1111 based
step-down converter is shown in Figure 5.
VIN
R3
100 Ω
+
C2
I LIM
V IN
SW1
FB
The usual step-up configuration for the LT1111 is shown
in Figure 4. The LT1111 first pulls SW1 low causing VIN –
VCESAT to appear across L1. A current then builds up in L1.
L1
LT1111
L1
VOUT
SW2
GND
R2
D1
1N5818
D1
+
C1
R1
V IN
V OUT
R3*
I LIM
LT1111 • F05
V IN
SW1
LT1111
GND
R2
Figure 5. Step-Down Mode Hookup
+
C1
FB
SW2
R1
*OPTIONAL
When the switch turns on, SW2 pulls up to VIN – VSW. This
puts a voltage across L1 equal to VIN – VSW – VOUT,
causing a current to build up in L1. At the end of the switch
ON time, the current in L1 is equal to:
LT1111 • F04
Figure 4. Step-Up Mode Hookup.
Refer to Table 1 for Component Values.
IPEAK =
VIN − VSW − VOUT
L
t ON
(22)
Note 1: This simple expression neglects the effect of switch and coil
resistance. This is taken into account in the “Inductor Selection” section.
10
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When the switch turns off, the SW2 pin falls rapidly and
actually goes below ground. D1 turns on when SW2
reaches 0.4V below ground. D1 MUST BE A SCHOTTKY
DIODE. The voltage at SW2 must never be allowed to go
below –0.5V. A silicon diode such as the 1N4933 will allow
SW2 to go to –0.8V, causing potentially destructive power
dissipation inside the LT1111. Output voltage is determined by:
Q1
MJE210 OR
ZETEX ZTX749
R1
0.3Ω
VIN
30V
MAX
L1
VOUT
R2
220
VIN
+
D1
1N5821
R3
330
IL
+
SW1
C2
C1
LT1111
R4
FB
 R2 
VOUT =  1 +  1.25 V
R1

(
)
(
R5
(23)
R4
VOUT = 1.25V 1 + R5
)
LT1111 • TA08
R3 programs switch current limit. This is especially important in applications where the input varies over a wide
range. Without R3, the switch stays on for a fixed time each
cycle. Under certain conditions the current in L1 can build
up to excessive levels, exceeding the switch rating and/or
saturating the inductor. The 100Ω resistor programs the
switch to turn off when the current reaches approximately
700mA. When using the LT1111 in step-down mode,
output voltage should be limited to 6.2V or less. Higher
output voltages can be accommodated by inserting a
1N5818 diode in series with the SW2 pin (anode connected to SW2).
Higher Current Step-Down Operation
Figure 6. Q1 Permits Higher Current Switching.
LT1111 Functions as Controller.
Inverting Configurations
The LT1111 can be configured as a positive-to-negative
converter (Figure 7), or a negative-to-positive converter
(Figure 8). In Figure 7, the arrangement is very similar to
a step-down, except that the high side of the feedback is
referred to ground. This level shifts the output negative. As
in the step-down mode, D1 must be a Schottky diode,
and VOUTshould be less than 6.2V. More negative output voltages can be accommodated as in the prior section.
VIN
Output current can be increased by using a discrete PNP
pass transistor as shown in Figure 6. R1 serves as a
current limit sense. When the voltage drop across R1
equals a VBE, the switch turns off. For temperature compensation a Schottky diode can be inserted in series with
the ILIM pin. This also lowers the maximum drop across R1
to VBE – VD, increasing efficiency. As shown, switch
current is limited to 2A. Inductor value can be calculated
based on formulas in the “Inductor Selection — StepDown Converter” section with the following conservative
expression for VSW:
VSW = V R1 + V Q1SAT ≈ 1.0 V
SW2
GND
(24)
R2 provides a current path to turn off Q1. R3 provides base
drive to Q1. R4 and R5 set output voltage. A PMOS FET can
be used in place of Q1 when VIN is between 10V and 20V.
R3
I LIM
V IN
SW1
FB
+
C2
LT1111
L1
SW2
GND
R1
D1
1N5818
+
C1
R2
–VOUT
LT1111 • F07
Figure 7. Positive-to-Negative Converter
In Figure 8, the input is negative while the output is
positive. In this configuration, the magnitude of the input
voltage can be higher or lower than the output voltage. A
level shift, provided by the PNP transistor, supplies proper
polarity feedback information to the regulator.
11
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D1
L1
VOUT
+
C1
I LIM
+
C2
VIN
SW1
R1
IL
2N3906
LT1111
A0
GND
FB
SW2
R2
–VIN
SWITCH
LT1111 • F08
ON
OFF
( )
VOUT = R1 1.25V + 0.6V
R2
LT1111 • F09
Figure 9. No Current Limit Causes Large Inductor
Current Build-Up
Figure 8. Negative-to-Positive Converter
PROGRAMMED CURRENT LIMIT
Using the ILIM Pin
The LT1111 switch can be programmed to turn off at a set
switch current, a feature not found on competing devices.
This enables the input to vary over a wide range without
exceeding the maximum switch rating or saturating the
inductor. Consider the case wh ere analysis shows the
LT1111 must operate at an 800mA peak switch current
with a 2V input. If VIN rises to 4V, the peak switch current
will rise to 1.6A, exceeding the maximum switch current
rating. With the proper resistor selected (see the “Maximum Switch Current vs ILIM” characteristic), the switch
current will be limited to 800mA, even if the input voltage
increases.
Another situation where the ILIM feature is useful occurs
when the device goes into continuous mode operation.
This occurs in step-up mode when:
VOUT + VDIODE
1
<
VIN − VSW
1 − DC
(25)
When the input and output voltages satisfy this relationship, inductor current does not go to zero during the
switch OFF time. When the switch turns on again, the
current ramp starts from the non-zero current level in the
inductor just prior to switch turn-on. As shown in Figure
9, the inductor current increases to a high level before the
comparator turns off the oscillator. This high current can
cause excessive output ripple and requires oversizing the
output capacitor and inductor. With the ILIM feature,
however, the switch current turns off at a programmed
level as shown in Figure 10, keeping output ripple to a
minimum.
12
IL
SWITCH
ON
OFF
LT1111 • F10
Figure 10. Current Limit Keeps Inductor Current Under Control
Figure 11 details current limit circuitry. Sense transistor
Q1, whose base and emitter are paralleled with power
switch Q2, is ratioed such that approximately 0.5% of
Q2’s collector current flows in Q1’s collector. This current
is passed through internal 80Ω resistor R1 and out
through the ILIM pin. The value of the external resistor
connected between ILIM and VIN sets the current limit.
When sufficient switch current flows to develop a VBE
across R1 + RLIM, Q3 turns on and injects current into the
oscillator, turning off the switch. Delay through this circuitry is approximately 1µs. The current trip point becomes less accurate for switch ON times less than 3µs.
Resistor values programming switch ON time for 1µs or
less will cause spurious response in the switch circuitry
although the device will still maintain output regulation.
RLIM
(EXTERNAL)
VIN
ILIM
R1
80Ω
(INTERNAL)
Q3
SW1
DRIVER
OSCILLATOR
Q1
Q2
SW2
LT1111 • F11
Figure 11. LT1111 Current Limit Circuitry
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Using the Gain Block
The gain block (GB) on the LT1111 can be used as an error
amplifier, low-battery detector or linear post regulator.
The gain block itself is a very simple PNP input op amp with
an open collector NPN output. The negative input of the
gain block is tied internally to the 1.25V reference. The
positive input comes out on the SET pin.
Arrangement of the gain block as a low-battery detector
is straightforward. Figure 12 shows hookup. R1 and R2
need only be low enough in value so that the bias current
of the SET input does not cause large errors. 33k for R2
is adequate. R3 can be added to introduce a small amount
of hysteresis. This will cause the gain block to “snap”
when the trip point is reached. Values in the 1M to 10M
range are optimal. However, the addition of R3 will
change the trip point.
5V
V IN
LT1111
R1
VBAT
1.25V
REF
–
SET
+
47k
A0
VLB – 1.25V
35.1µA
VLB = BATTERY TRIP POINT
R2 = 33k
R3 = 1.6M
R1 =
GND
R2
TO
PROCESSOR
R3
LT1111 • F12
Figure 12. Setting Low-Battery Detector Trip Point
Table 1. Component Selection for Common Converters
INPUT
VOLTAGE
OUTPUT
VOLTAGE
OUTPUT
CURRENT (MIN)
CIRCUIT
FIGURE
INDUCTOR
VALUE
INDUCTOR
PART NUMBER
CAPACITOR
VALUE
2 to 3.1
2 to 3.1
2 to 3.1
2 to 3.1
5
5
6.5 to 11
12 to 20
20 to 30
5
12
5
5
12
12
12
12
5
5
5
–5
–5
90mA
10mA
30mA
10mA
90mA
30mA
50mA
300mA
300mA
75mA
250mA
4
4
4
4
4
4
5
5
5
6
6
15µH
47µH
15µH
47µH
33µH
47µH
15µH
56µH
120µH
56µH
120µH
S CD75-750K
S CD54-470K, C CTX50-1
S CD75-150K
S CD54-470K, C CTX50-1
S CD75-330K
S CD75-470K, C CTX50-1
S CD54-150K
S CD105-560K, C CTX50-4
S CD105-121K, C CTX100-4
S CD75-560K, C CTX50-4
S CD105-121K, C CTX100-4
33µF
10µF
22µF
10µF
22µF
15µF
47µF
47µF
47µF
47µF
100µF
S = Sumida
C = Coiltronics
NOTES
*
**
**
**
**
* Add 47Ω from ILIM to VIN
** Add 220Ω from ILIM to VIN
Table 3. Capacitor Manufacturers
Table 2. Inductor Manufacturers
MANUFACTURER
PART NUMBERS
MANUFACTURER
PART NUMBERS
Coiltronics Incorporated
6000 Park of Commerce Blvd.
Boca Raton, FL 33487
407-241-7876
CTX100-4 Series
Surface Mount
Sanyo Video Components
1201 Sanyo Avenue
San Diego, CA 92073
619-661-6322
OS-CON Series
Toko America Incorporated
1250 Feehanville Drive
Mount Prospect, IL 60056
312-297-0070
Type 8RBS
Nichicon America Corporation
927 East State Parkway
Schaumberg, IL 60173
708-843-7500
PL Series
Sumida Electric Co. USA
708-956-0666
CD54
CDR74
CDR105
Surface Mount
Sprague Electric Company
Lower Main Street
Sanford, ME 04073
207-324-4140
150D Solid Tantalums
550D Tantalex
Matsuo
714-969-2491
267 Series
Surface Mount
13
LT1111
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S
3V to – 22V LCD Bias Generator
L1*
27µH
1N4148
R1
100Ω
I LIM
732k
1%
V IN
SW1
2 × 1.5V
CELLS
3V
LT1111
0.1µF
FB
GND
SW2
+
4.7µF
39.2k
1%
MBRS130T3
MBRS130T3
+
22µF
220k
* L1 = SUMIDA CD54-270K
FOR 5V INPUT CHANGE R1 TO 47Ω.
CONVERTER WILL DELIVER –22V AT 40mA.
–22V OUTPUT
7mA AT 2V INPUT
LT1111 • TA03
20V to 5V Step-Down Converter
9V to 5V Step-Down Converter
VIN
12V TO 28V
100 Ω
ILIM
100 Ω
V IN
ILIM
SW1
9V
BATTERY
SW1
LT1111-5
LT1111-5
SENSE
GND
SW2
V IN
SENSE
L1*
15µH
MBRS130T3
5V OUTPUT
150mA AT 9V INPUT
50mA AT 6.5V INPUT
+
GND
SW2
L1*
68µH
+
22µF
MBRS130T3
5V OUTPUT
300mA
47µF
* L1 = SUMIDA CD54-150K
LT1111 • TA04
* L1 = SUMIDA CD74-680M
14
LT1111 • TA06
LT1111
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TYPICAL APPLICATI
S
5V to –5V Converter
VIN
5V INPUT
100 Ω
I LIM
V IN
SW1
+
22µF
LT1111-5
SENSE
GND
SW2
L1*
33µH
MBRS130T3
+
33µF
–5V OUTPUT
75mA
* L1 = SUMIDA CD54-330K
LT1111 • TA05
Voltage Controlled Positive-to-Negative Converter
VIN
5V TO 12V
L1*
20µH, 3A
ZETEX†
ZTX788A
0.22Ω
+
BAT54
V IN
ILIM
220Ω
–VOUT = –5.13 × VC
2W MAXIMUM OUTPUT
220Ω
V IN
SW1
200k
–
LT1111
39k
VC (0V TO 5V)
LT1006
FB
GND
47µF
MBRD320
SW2
+
* L1 = COILTRONICS CTX20-4
†
ZETEX INC. 516-543-7100
LT1111 • TA07
High Power, Low Quiescent Current Step-Down Converter
0.22Ω
VIN
8V TO 18V
MTM20P08
BAT54
2k
51Ω
L1*
10µH, 3A
MBRD320
5V
500mA
+
220µF
2N3904
V IN
ILIM
SW1
1N4148
LT1111
121k
FB
GND
SW2
40.2k
* L1 = SUMIDA CDR105-100M
OPERATE STANDBY
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
LT1111 • TA20
15
LT1111
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PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
J8 Package
8-Lead Ceramic DIP
CORNER LEADS OPTION
(4 PLCS)
0.200
(5.080)
MAX
0.290 – 0.320
(7.366 – 8.128)
0.023 – 0.045
(0.584 – 1.143)
HALF LEAD
OPTION
0.008 – 0.018
(0.203 – 0.457)
0° – 15°
0.015 – 0.060
(0.381 – 1.524)
0.405
(10.287)
MAX
0.005
(0.127)
MIN
8
0.220 – 0.310
(5.588 – 7.874)
1
0.385 ± 0.025
(9.779 ± 0.635)
5
0.025
(0.635)
RAD TYP
0.045 – 0.068
(1.143 – 1.727)
FULL LEAD
OPTION
0.045 – 0.068
(1.143 – 1.727)
6
7
2
3
4
0.125
3.175
0.100 ± 0.010 MIN
(2.540 ± 0.254)
0.014 – 0.026
(0.360 – 0.660)
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP OR TIN PLATE LEADS.
N8 Package
8-Lead Plastic DIP
0.300 – 0.320
(7.620 – 8.128)
0.009 – 0.015
(0.229 – 0.381)
(
+0.025
0.325 –0.015
8.255
+0.635
–0.381
)
0.130 ± 0.005
(3.302 ± 0.127)
0.045 – 0.065
(1.143 – 1.651)
0.400
(10.160)
MAX
8
7
6
5
0.065
(1.651)
TYP
0.250 ± 0.010
(6.350 ± 0.254)
0.125
(3.175)
MIN
0.045 ± 0.015
(1.143 ± 0.381)
0.020
(0.508)
MIN
1
2
4
3
0.018 ± 0.003
(0.457 ± 0.076)
0.100 ± 0.010
(2.540 ± 0.254)
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197
(4.801 – 5.004)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
7
6
5
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm).
16
8
0.004 – 0.010
(0.101 – 0.254)
Linear Technology Corporation
0.050
(1.270)
BSC
0.150 – 0.157
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
2
3
4
SO8 0294
LT/GP 0594 5K REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
 LINEAR TECHNOLOGY CORPORATION 1994