AD AD9220AR Complete 12-bit 1.5/3.0/10.0 msps olithic a/d converter Datasheet

Complete 12-Bit 1.5/3.0/10.0 MSPS
Monolithic A/D Converters
AD9221/AD9223/AD9220
FEATURES
Monolithic 12-Bit A/D Converter Product Family
Family Members Are: AD9221, AD9223, and AD9220
Flexible Sampling Rates: 1.5 MSPS, 3.0 MSPS, and
10.0 MSPS
Low Power Dissipation: 59 mW, 100 mW, and 250 mW
Single 5 V Supply
Integral Nonlinearity Error: 0.5 LSB
Differential Nonlinearity Error: 0.3 LSB
Input Referred Noise: 0.09 LSB
Complete On-Chip Sample-and-Hold Amplifier and
Voltage Reference
Signal-to-Noise and Distortion Ratio: 70 dB
Spurious-Free Dynamic Range: 86 dB
Out-of-Range Indicator
Straight Binary Output Data
28-Lead SOIC and 28-Lead SSOP
GENERAL DESCRIPTION
The AD9221, AD9223, and AD9220 are a generation of high
performance, single supply 12-bit analog-to-digital converters.
Each device exhibits true 12-bit linearity and temperature drift
performance1 as well as 11.5-bit or better ac performance.2 The
AD9221/AD9223/AD9220 share the same interface options,
package, and pinout. Thus, the product family provides an upward
or downward component selection path based on performance,
sample rate and power. The devices differ with respect to their
specified sampling rate, and power consumption, which is reflected
in their dynamic performance over frequency.
The AD9221/AD9223/AD9220 combine a low cost, high speed
CMOS process and a novel architecture to achieve the resolution
and speed of existing hybrid and monolithic implementations at
a fraction of the power consumption and cost. Each device is a
complete, monolithic ADC with an on-chip, high performance,
low noise sample-and-hold amplifier and programmable voltage
reference. An external reference can also be chosen to suit the
dc accuracy and temperature drift requirements of the application.
The devices use a multistage differential pipelined architecture
with digital output error correction logic to provide 12-bit accuracy at the specified data rates and to guarantee no missing
codes over the full operating temperature range.
The input of the AD9221/AD9223/AD9220 is highly flexible,
allowing for easy interfacing to imaging, communications, medical, and data-acquisition systems. A truly differential input
structure allows for both single-ended and differential input
interfaces of varying input spans. The sample-and-hold
NOTES
1
Excluding internal voltage reference.
2
Depends on the analog input configuration.
FUNCTIONAL BLOCK DIAGRAM
AVDD
CLK
DVDD
SHA
VINA
MDAC1
GAIN = 16
VINB
A/D
CAPT
5
MDAC2
GAIN = 8
A/D
5
4
MDAC3
GAIN = 4
A/D
3
A/D
4
CAPB
3
DIGITAL CORRECTION LOGIC
VREF
12
OUTPUT BUFFERS
3
OTR
SENSE
MODE
SELECT
1V
AD9221/AD9223/AD9220
REFCOM
AVSS
DVSS
BIT 1
(MSB)
BIT 12
(LSB)
CML
amplifier (SHA) is equally suited for both multiplexed systems that switch full-scale voltage levels in successive channels
as well as sampling single-channel inputs at frequencies up to
and beyond the Nyquist rate. Also, the AD9221/AD9223/AD9220
is well suited for communication systems employing DirectIF down conversion since the SHA in the differential input
mode can achieve excellent dynamic performance far beyond its
specified Nyquist frequency.2
A single clock input is used to control all internal conversion
cycles. The digital output data is presented in straight binary
output format. An out-of-range (OTR) signal indicates an overflow condition that can be used with the most significant bit to
determine low or high overflow.
PRODUCT HIGHLIGHTS
The AD9221/AD9223/AD9220 family offers a complete singlechip sampling 12-bit, analog-to-digital conversion function in
pin compatible 28-lead SOIC and SSOP packages.
Flexible Sampling Rates—The AD9221, AD9223, and AD9220
offer sampling rates of 1.5 MSPS, 3.0 MSPS, and 10.0 MSPS,
respectively.
Low Power and Single Supply—The AD9221, AD9223, and
AD9220 consume only 59 mW, 100 mW, and 250 mW, respectively, on a single 5 V power supply.
Excellent DC Performance Over Temperature—The AD9221/
AD9223/AD9220 provide 12-bit linearity and temperature drift
performance.1
Excellent AC Performance and Low Noise—The AD9221/
AD9223/AD9220 provide better than 11.3 ENOB performance
and have an input referred noise of 0.09 LSB rms.2
Flexible Analog Input Range—The versatile on-board sampleand-hold (SHA) can be configured for either single-ended or
differential inputs of varying input spans.
REV. E
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective companies.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© 2003 Analog Devices, Inc. All rights reserved.
AD9221/AD9223/AD9220–SPECIFICATIONS
DC SPECIFICATIONS
(AVDD = 5 V, DVDD = 5 V, fSAMPLE = Max Conversion Rate, VREF = 2.5 V, VINB = 2.5 V, TMIN to TMAX, unless
otherwise noted.)
Parameter
AD9221
AD9223
AD9220
Unit
RESOLUTION
12
12
12
Bits min
MAX CONVERSION RATE
1.5
3
10
MHz min
INPUT REFERRED NOISE (TYP)
VREF = 1 V
VREF = 2.5 V
0.23
0.09
0.23
0.09
0.23
0.09
LSB rms typ
LSB rms typ
INL1
DNL1
No Missing Codes
Zero Error (@ 25°C)
Gain Error (@ 25°C)2
Gain Error (@ 25°C)3
± 0.4
± 1.25
± 0.3
± 0.75
± 0.6
± 0.3
12
± 0.3
± 1.5
± 0.75
± 0.5
± 1.25
± 0.3
± 0.75
± 0.6
± 0.3
12
± 0.3
± 1.5
± 0.75
± 0.5
± 1.25
± 0.3
± 0.75
± 0.7
± 0.35
12
± 0.3
± 1.5
± 0.75
LSB typ
LSB max
LSB typ
LSB max
LSB typ
LSB typ
Bits Guaranteed
% FSR max
% FSR max
% FSR max
TEMPERATURE DRIFT
Zero Error
Gain Error2
Gain Error3
±2
± 26
± 0.4
±2
± 26
± 0.4
±2
± 26
± 0.4
ppm/°C typ
ppm/°C typ
ppm/°C typ
POWER SUPPLY REJECTION
AVDD, DVDD (+5 V ± 0.25 V)
± 0.06
± 0.06
± 0.06
% FSR max
2
5
0
AVDD
16
2
5
0
AVDD
16
2
5
0
AVDD
16
V p-p min
V p-p max
V min
V max
pF typ
INTERNAL VOLTAGE REFERENCE
Output Voltage (1 V Mode)
Output Voltage Tolerance (1 V Mode)
Output Voltage (2.5 V Mode)
Output Voltage Tolerance (2.5 V Mode)
Load Regulation4
1
± 14
2.5
± 35
2.0
1
± 14
2.5
± 35
2.0
1
± 14
2.5
± 35
2.0
V typ
mV max
V typ
mV max
mV max
REFERENCE INPUT RESISTANCE
5
5
5
kΩ typ
POWER SUPPLIES
Supply Voltages
AVDD
DVDD
Supply Current
IAVDD
5
2.7 to 5.25
5
2.7 to 5.25
5
2.7 to 5.25
V (± 5% AVDD Operating)
V
14.0
11.8
0.5
0.02
26
20
0.5
0.02
58
51
4.0
<1.0
mA max
mA typ
mA max
mA typ
59.0
70.0
100
130
254
310
mW typ
mW max
ACCURACY
Integral Nonlinearity (INL)
Differential Nonlinearity (DNL)
ANALOG INPUT
Input Span (with VREF = 1.0 V)
Input Span (with VREF = 2.5 V)
Input (VINA or VINB) Range
Input Capacitance
IDVDD
POWER CONSUMPTION
NOTES
1
VREF = 1 V.
2
Including internal reference.
3
Excluding internal reference.
4
Load regulation with 1 mA load current (in addition to that required by the AD9221/AD9223/AD9220).
Specification subject to change without notice.
–2–
REV. E
AD9221/AD9223/AD9220
AC SPECIFICATIONS
(AVDD = 5 V, DVDD= 5 V, fSAMPLE = Max Conversion Rate, VREF = 1.0 V, VINB = 2.5 V, DC Coupled/SingleEnded Input TMIN to TMAX, unless otherwise noted.)
Parameter
AD9221
AD9223
AD9220
Unit
MAX CONVERSION RATE
1.5
3.0
10.0
MHz min
100
70.0
69.0
11.3
11.2
70.2
69.0
–83.4
–77.5
86.0
79.0
0.50
69.9
69.0
11.3
11.2
70.1
69.0
–83.4
–77.5
86.0
79.0
25
25
1
4
125
500
70.0
68.5
11.3
11.1
70.0
68.5
–83.4
–76.0
87.5
77.5
1.50
69.4
68.0
11.2
11.1
69.7
68.5
–82.9
–75.0
85.7
76.0
40
40
1
4
43
1000
70
68.5
11.3
11.1
70.2
69.0
–83.7
–76.0
88.0
77.5
5.0
67.0
65.0
10.8
10.5
68.8
67.5
–72.0
–68.0
75.0
69.0
60
60
1
4
30
kHz
dB typ
dB min
dB typ
dB min
dB typ
dB min
dB typ
dB max
dB typ
dB max
MHz
dB typ
dB min
dB typ
dB min
dB typ
dB min
dB typ
dB max
dB typ
dB max
MHz typ
MHz typ
ns typ
ps rms typ
ns typ
DYNAMIC PERFORMANCE
Input Test Frequency 1 (VINA = –0.5 dBFS)
Signal-to-Noise and Distortion (SINAD)
Effective Number of Bits (ENOBs)
Signal-to-Noise Ratio (SNR)
Total Harmonic Distortion (THD)
Spurious Free Dynamic Range (SFDR)
Input Test Frequency 2 (VINA = –0.5 dBFS)
Signal-to-Noise and Distortion (SINAD)
Effective Number of Bits (ENOBs)
Signal-to-Noise Ratio (SNR)
Total Harmonic Distortion (THD)
Spurious Free Dynamic Range (SFDR)
Full Power Bandwidth
Small Signal Bandwidth
Aperture Delay
Aperture Jitter
Acquisition to Full-Scale Step
Specifications subject to change without notice.
DIGITAL SPECIFICATIONS
(AVDD = 5 V, DVDD = 5 V, TMIN to TMAX, unless otherwise noted.)
Parameter
Symbol
CLOCK INPUT
High Level Input Voltage
Low Level Input Voltage
High Level Input Current (VIN = DVDD)
Low Level Input Current (VIN = 0 V)
Input Capacitance
VIH
VIL
IIH
IIL
CIN
3.5
1.0
± 10
± 10
5
V min
V max
µA max
µA max
pF typ
VOH
VOH
VOL
VOL
4.5
2.4
0.4
0.1
V min
V min
V max
V max
VOH
VOH
VOL
VOL
COUT
2.95
2.80
0.4
0.05
5
V min
V min
V max
V max
pF typ
LOGIC OUTPUTS
DVDD = 5 V
High Level Output Voltage (IOH = 50 µA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOL = 1.6 mA)
Low Level Output Voltage (IOL = 50 µA)
DVDD = 3 V
High Level Output Voltage (IOH = 50 µA)
High Level Output Voltage (IOH = 0.5 mA)
Low Level Output Voltage (IOL = 1.6 mA)
Low Level Output Voltage (IOL = 50 µA)
Output Capacitance
Specifications subject to change without notice.
REV. E
–3–
Unit
AD9221/AD9223/AD9220
SWITCHING SPECIFICATIONS
(TMIN to TMAX with AVDD = 5 V, DVDD = 5 V, CL = 20 pF)
Parameter
Symbol
AD9221
AD9223
AD9220
Unit
Clock Period*
CLOCK Pulsewidth High
CLOCK Pulsewidth Low
Output Delay
tC
tCH
tCL
tOD
667
300
300
8
13
19
3
333
150
150
8
13
19
3
100
45
45
8
13
19
3
ns min
ns min
ns min
ns min
ns typ
ns max
Clock Cycles
Pipeline Delay (Latency)
*The clock period may be extended to 1 ms without degradation in specified performance @ 25 °C.
Specifications subject to change without notice.
S1
ANALOG
INPUT
S2
S4
tC
tCH
S3
tCL
INPUT
CLOCK
tOD
DATA
OUTPUT
DATA 1
Figure 1. Timing Diagram
ABSOLUTE MAXIMUM RATINGS*
Parameter
AVDD
DVDD
AVSS
AVDD
REFCOM
CLK
Digital Outputs
VINA, VINB
VREF
SENSE
CAPB, CAPT
Junction Temperature
Storage Temperature
Lead Temperature
(10 sec)
THERMAL CHARACTERISTICS
With
Respect
to
Min
Max
Unit
AVSS
DVSS
DVSS
DVDD
AVSS
AVSS
DVSS
AVSS
AVSS
AVSS
AVSS
–0.3
–0.3
–0.3
–6.5
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
+6.5
+6.5
+0.3
+6.5
+0.3
AVDD + 0.3
DVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
150
+150
V
V
V
V
V
V
V
V
V
V
V
°C
°C
300
°C
–65
Thermal Resistance
28-Lead SOIC
␪JA = 71.4°C/W
␪JC = 23°C/W
28-Lead SSOP
␪JA = 63.3°C/W
␪JC = 23°C/W
ORDERING GUIDE
Model
AD9221AR
AD9223AR
AD9220AR
AD9221ARS
AD9223ARS
AD9220ARS
AD9221-EB
AD9223-EB
AD9220-EB
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum ratings
for extended periods may effect device reliability.
Temperature
Range
Package
Description
Package
Option
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
28-Lead SOIC
28-Lead SOIC
28-Lead SOIC
28-Lead SSOP
28-Lead SSOP
28-Lead SSOP
Evaluation Board
Evaluation Board
Evaluation Board
R-28
R-28
R-28
RS-28
RS-28
RS-28
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD9221/AD9223/AD9220 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions
are recommended to avoid performance degradation or loss of functionality.
–4–
REV. E
AD9221/AD9223/AD9220
PIN CONFIGURATION
Zero Error
The major carry transition should occur for an analog value 1/2
LSB below VINA = VINB. Zero error is defined as the deviation of the actual transition from that point.
CLK 1
28 DVDD
(LSB) BIT 12 2
27 DVSS
BIT 11 3
26 AVDD
Gain Error
BIT 10 4
25 AVSS
The first code transition should occur at an analog value 1/2 LSB
above negative full scale. The last transition should occur at an
analog value 1 1/2 LSB below the nominal full scale. Gain error
is the deviation of the actual difference between first and last
code transitions and the ideal difference between first and last
code transitions.
BIT 9 5
BIT 8 6
AD9221/
AD9223/
AD9220
24 VINB
23 VINA
BIT 7 7
TOP VIEW 22 CML
(Not to Scale)
BIT 6 8
21 CAPT
BIT 5 9
20 CAPB
BIT 4 10
19 REFCOM
BIT 3 11
18 VREF
BIT 2 12
17 SENSE
(MSB) BIT 1 13
16 AVSS
OTR 14
15 AVDD
Temperature Drift
The temperature drift for zero error and gain error specifies the
maximum change from the initial (25°C) value to the value at
TMIN or TMAX.
Power Supply Rejection
The specification shows the maximum change in full scale from
the value with the supply at the minimum limit to the value with
the supply at its maximum limit.
PIN FUNCTION DESCRIPTIONS
Aperture Jitter
Pin
Number
Mnemonic
Description
1
2
3–12
13
14
15, 26
16, 25
17
18
19
20
21
22
23
24
27
28
CLK
BIT 12
BITS 11–2
BIT 1
OTR
AVDD
AVSS
SENSE
VREF
REFCOM
CAPB
CAPT
CML
VINA
VINB
DVSS
DVDD
Clock Input Pin
Least Significant Data Bit (LSB)
Data Output Bit
Most Significant Data Bit (MSB)
Out of Range
5 V Analog Supply
Analog Ground
Reference Select
Reference I/O
Reference Common
Noise Reduction Pin
Noise Reduction Pin
Common-Mode Level (Midsupply)
Analog Input Pin (+)
Analog Input Pin (–)
Digital Ground
3 V to 5 V Digital Supply
Aperture jitter is the variation in aperture delay for successive
samples and is manifested as noise on the input to the A/D.
Aperture Delay
Aperture delay is a measure of the sample-and-hold amplifier
(SHA) performance and is measured from the rising edge of the
clock input to when the input signal is held for conversion.
Signal-to-Noise and Distortion (S/N+D, SINAD) Ratio
S/N+D is the ratio of the rms value of the measured input signal
to the rms sum of all other spectral components below the Nyquist
frequency, including harmonics but excluding dc. The value for
S/N+D is expressed in decibels.
Effective Number of Bits (ENOB)
For a sine wave, SINAD can be expressed in terms of the number of bits. Using the following formula,
N = (SINAD – 1.76) / 6.02
it is possible to get a measure of performance expressed as N,
the effective number of bits.
Thus, effective number of bits for a device for sine wave inputs
at a given input frequency can be calculated directly from its
measured SINAD.
DEFINITIONS OF SPECIFICATIONS
Integral Nonlinearity (INL)
Total Harmonic Distortion (THD)
INL refers to the deviation of each individual code from a line
drawn from “negative full scale” through “positive full scale.”
The point used as negative full scale occurs 1/2 LSB before the
first code transition. Positive full scale is defined as a level 1 1/2
LSB beyond the last code transition. The deviation is measured
from the middle of each particular code to the true straight line.
THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is
expressed as a percentage or in decibels.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured input signal to
the rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in decibels.
Differential Nonlinearity (DNL, No Missing Codes)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 12-bit resolution indicates that all 4096
codes, respectively, must be present over all operating ranges.
REV. E
Spurious Free Dynamic Range (SFDR)
SFDR is the difference in dB between the rms amplitude of the
input signal and the peak spurious signal.
–5–
AD9221/AD9223/AD9220
1.0
0.8
0.6
0.6
0.4
0.4
0.2
0.2
0.0
–0.2
8,180,388
0.0
–0.2
–0.4
–0.4
–0.6
–0.6
–0.8
–0.8
–1.0
0
121,764
85,895
–1.0
0
4095
TPC 2. Typical INL
TPC 1. Typical DNL
80
N–1
4095
CODE
CODE
–60
70
75
70
–65
55
–20.0dB
–70
–75
–6.0dB
–80
–85
50
SINAD – dB
THD – dB
60
–0.5dB
–100
0.1
1.0
FREQUENCY – MHz
TPC 4. SINAD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
–55
0.1
TPC 6. SINAD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
100
–65
90
–75
–0.5dB
–75
80
–80
SNR/SFDR – dB
–65
THD – dB
THD– dB
–70
5V p-p
–85
2V p-p
–6.0dB
70
SFDR
60
50
–90
30
–85
–95
20
–90
–100
0.2
1.0
FREQUENCY – MHz
TPC 7. THD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
2
0.6 0.8 1
0.3 0.4
SAMPLE RATE – MSPS
TPC 8. THD vs. Sample Rate
(AIN = –0.5 dB, fIN = 500 kHz,
VCM = 2.5 V)
–6–
3
SNR
40
–80
0.1
1.0
FREQUENCY – MHz
–60
–60
–20.0dB
40
–20.0dB
–70
55
1.0
FREQUENCY – MHz
TPC 5. THD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
–50
–6.0dB
60
45
–95
40
0.1
65
50
–90
45
–0.5dB
–20.0dB
–6.0dB
65
N+1
80
–55
–0.5dB
N
CODE
TPC 3. “Grounded-Input”
Histogram (Input Span = 2 V p-p)
–50
75
SINAD – dB
(AVDD = 5 V, DVDD = 5 V, fSAMPLE = 1.5 MSPS, TA = 25C)
HITS
1.0
0.8
INL – LSBs
DNL – LSBs
AD9221–Typical Performance Characteristics
10
–60
–50
–40
–20
–30
A IN – dBFS
–10
0
TPC 9. SNR/SFDR vs. AIN (Input
Amplitude) (fIN = 500 kHz, Input
Span = 2 V p-p, VCM = 2.5 V)
REV. E
AD9221/AD9223/AD9220
1.0
0.8
0.6
0.6
0.4
0.4
0.2
0.2
0.0
–0.2
SAMPLE
= 3.0 MSPS, TA = 25C)
8,123,672
HITS
1.0
0.8
INL – LSBs
DNL – LSBs
AD9223–Typical Performance Characteristics (AVDD = 5 V, DVDD = 5 V, f
0.0
–0.2
–0.4
–0.4
–0.6
–0.6
–0.8
–0.8
130,323
96,830
–1.0
–1.0
0
0
4095
CODE
TPC 10. Typical DNL
TPC 11. Typical INL
75
–60
70
–20.0dB
–65
–6.0dB
THD – dB
65
60
–70
–75
–0.5dB
–80
55
–6.0dB
–20.0dB
45
40
1.0
FREQUENCY – MHz
10.0
TPC 13. SINAD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
1.0
FREQUENCY – MHz
–55
–60
100
–65
90
–80
–85
–0.5dB
5V p-p
–80
–85
2V p-p
1.0
FREQUENCY – MHz
10.0
TPC 16. THD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
–100
0.4
70
SFDR
60
50
SNR
40
30
–95
–95
REV. E
–75
–90
–90
–100
0.1
SNR/SFDR – dB
–6.0dB
10.0
80
–70
–20.0dB
THD – dB
THD – dB
–75
1.0
FREQUENCY – MHz
TPC 15. SINAD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
–60
–70
–20.0dB
40
0.1
10.0
TPC 14. THD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
–50
–65
55
45
–95
–100
0.1
–6.0dB
60
50
–90
0.1
65
–85
50
–0.5dB
70
SINAD – dB
–0.5dB
N+1
80
–55
75
N
CODE
TPC 12. “Grounded-Input”
Histogram (Input Span = 2 V p-p)
–50
80
SINAD – dB
N–1
4095
CODE
20
0.6 0.8 1
2
3
SAMPLE RATE – MSPS
4
TPC 17. THD vs. Sample Rate
(AIN = –0.5 dB, fIN = 500 kHz,
VCM = 2.5 V)
–7–
5 6
10
–60
–50
–40
–30
–20
AIN – dBFS
–10
0
TPC 18. SNR/SFDR vs. AIN (Input
Amplitude) (fIN = 1.5 MHz, Input
Span = 2 V p-p, VCM = 2.5 V)
AD9221/AD9223/AD9220
1.0
1.0
0.8
0.8
0.6
0.6
0.4
0.4
INL – LSBs
0.2
0.0
–0.2
(AVDD = 5 V, DVDD = 5 V, fSAMPLE = 10 MSPS, TA = 25C)
8,123,672
0.2
HITS
DNL – LSBs
AD9220–Typical Performance Characteristics
0.0
–0.2
–0.4
–0.4
–0.6
–0.6
–0.8
–0.8
–1.0
–1.0
1
1
4095
CODE
TPC 19. Typical DNL
CODE
N–1
4095
TPC 20. Typical INL
–0.5dB
75
–60
70
70
55
–20dB
–70
THD – dB
60
SINAD – dB
–65
–6dB
65
–75
–6dB
–80
–20dB
N+1
80
–55
75
N
CODE
TPC 21. “Grounded-Input”
Histogram (Input Span = 2 V p-p)
–50
80
SINAD – dB
130,323
134,613
–85
–0.5dB
50
–0.5dB
–6.0dB
65
60
55
–20.0dB
50
–90
45
40
0.1
45
–95
1.0
FREQUENCY – MHz
–100
0.5
10.0
1.0
TPC 23. THD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
–50
–60
–55
–65
–60
–70
–6.0dB
TPC 24. SINAD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
80
SFDR
5V p-p
70
–75
SNR/SFDR – dB
THD – dB
THD – dB
–0.5dB
2V p-p
–80
60
SNR
50
–75
–85
–80
–90
30
–85
–95
20
–100
1.0
FREQUENCY – MHz
10.0
TPC 25. THD vs. Input Frequency
(Input Span = 5.0 V p-p, VCM = 2.5 V)
10.0
90
–20.0dB
–65
–90
0.1
1.0
FREQUENCY – MHz
FREQUENCY – MHz
TPC 22. SINAD vs. Input Frequency
(Input Span = 2.0 V p-p, VCM = 2.5 V)
–70
40
0.1
10.0
1
10
SAMPLE RATE – MSPS
15
TPC 26. THD vs. Clock Frequency
(AIN = –0.5 dB, fIN = 1.0 MHz,
VCM = 2.5 V)
–8–
40
10
–60
–50
–40
–30
–20
AIN – dBFS
–10
0
TPC 27. SNR/SFDR vs. AIN (Input
Amplitude) (fIN = 5.0 MHz, Input
Span = 2 V p-p, VCM = 2.5 V)
REV. E
AD9221/AD9223/AD9220
also similar. The data sheet is structured such that the designer
can make an informed decision in selecting the proper A/D and
optimizing its performance to fit the specific application.
INTRODUCTION
The AD9221/AD9223/AD9220 are members of a high performance, complete single-supply 12-bit ADC product family based
on the same CMOS pipelined architecture. The product family
allows the system designer an upward or downward component
selection path based on dynamic performance, sample rate, and
power. The analog input range of the AD9221/AD9223/AD9220
is highly flexible, allowing for both single-ended or differential inputs of varying amplitudes that can be ac or dc coupled.
Each device shares the same interface options, pinout, and
package offering.
0
AD9220
AMPLITUDE – dB
The AD9221/AD9223/AD9220 utilize a four-stage pipeline
architecture with a wideband input sample-and-hold amplifier
(SHA) implemented on a cost-effective CMOS process. Each
stage of the pipeline, excluding the last stage, consists of a low
resolution flash A/D connected to a switched capacitor DAC
and interstage residue amplifier (MDAC). The residue amplifier
amplifies the difference between the reconstructed DAC output
and the flash input for the next stage in the pipeline. One bit of
redundancy is used in each of the stages to facilitate digital
correction of flash errors. The last stage simply consists of a
flash A/D.
–6
AD9221
–9
–12
1
10
FREQUENCY – MHz
100
Figure 2. Full-Power Bandwidth
4000
AD9220
The pipeline architecture allows a greater throughput rate at the
expense of pipeline delay or latency. This means that while the
converter is capable of capturing a new input sample every clock
cycle, it actually takes three clock cycles for the conversion to be
fully processed and appear at the output. This latency is not a
concern in most applications. The digital output, together with
the out-of-range indicator (OTR), is latched into an output buffer
to drive the output pins. The output drivers can be configured to
interface with 5 V or 3.3 V logic families.
AD9223
AD9221
CODE
3000
2000
1000
The AD9221/AD9223/AD9220 use both edges of the clock in
their internal timing circuitry (see Figure 1 and Specifications
for exact timing requirements). The A/D samples the analog
input on the rising edge of the clock input. During the clock low
time (between the falling edge and rising edge of the clock), the
input SHA is in the sample mode; during the clock high time, it
is in hold. System disturbances just prior to the rising edge of
the clock and/or excessive clock jitter may cause the input SHA
to acquire the wrong value, and should be minimized.
0
0
10
20
30
40
SETTLING TIME – ns
50
60
Figure 3. Settling Time
ANALOG INPUT AND REFERENCE OVERVIEW
Figure 4, a simplified model of the AD9221/AD9223/AD9220,
highlights the relationship between the analog inputs, VINA,
VINB, and the reference voltage, VREF. Like the voltage
applied to the top of the resistor ladder in a flash A/D converter,
the value VREF defines the maximum input voltage to the A/D
core. The minimum input voltage to the A/D core is automatically defined to be –VREF.
The internal circuitry of both the input SHA and individual
pipeline stages of each member of the product family are optimized for both power dissipation and performance. An inherent
trade-off exists between the input SHA’s dynamic performance
and its power dissipation. Figures 2 and 3 show this trade-off by
comparing the full-power bandwidth and settling time of the
AD9221/AD9223/AD9220. Both figures reveal that higher fullpower bandwidths and faster settling times are achieved at the
expense of an increase in power dissipation. Similarly, a tradeoff exists between the sampling rate and the power dissipated
in each stage.
AD9221/AD9223/AD9220
+VREF
VINA
VCORE
As previously stated, the AD9221, AD9223, and AD9220 are
similar in most aspects except for the specified sampling rate,
power consumption, and dynamic performance. The product
family is highly flexible, providing several different input ranges
and interface options. As a result, many of the application issues
and trade-offs associated with these resulting configurations are
REV. E
AD9223
–3
VINB
A/D
CORE
12
–VREF
Figure 4. AD9221/AD9223/AD9220 Equivalent
Functional Input Circuit
–9–
AD9221/AD9223/AD9220
The addition of a differential input structure gives the user an
additional level of flexibility that is not possible with traditional
flash converters. The input stage allows the user to easily configure the inputs for either single-ended operation or differential
operation. The A/D’s input structure allows the dc offset of the
input signal to be varied independently of the input span of the
converter. Specifically, the input to the A/D core is the difference of the voltages applied at the VINA and VINB input
pins. Therefore, the equation,
VCORE = VINA – VINB
(1)
defines the output of the differential input stage and provides
the input to the A/D core.
The voltage, VCORE, must satisfy the condition,
–VREF ≤ VCORE ≤ VREF
(2)
where VREF is the voltage at the VREF pin.
While an infinite combination of VINA and VINB inputs exist
that satisfy Equation 2, there is an additional limitation placed
on the inputs by the power supply voltages of the AD9221/
AD9223/AD9220. The power supplies bound the valid operating range for VINA and VINB. The condition,
AVSS – 0.3V < VINA < AVDD + 0.3V
(3)
AVSS – 0.3V < VINB < AVDD + 0.3V
The SHA’s optimum distortion performance for a differential or
single-ended input is achieved under the following two conditions:
(1) the common-mode voltage is centered around midsupply
(i.e., AVDD/2 or approximately 2.5 V) and (2) the input signal
voltage span of the SHA is set at its lowest (i.e., 2 V input span).
This is due to the sampling switches, QS1, being CMOS switches
whose RON resistance is very low but has some signal dependency that causes frequency dependent ac distortion while the
SHA is in the track mode. The RON resistance of a CMOS
switch is typically lowest at its midsupply but increases symmetrically as the input signal approaches either AVDD or AVSS. A
lower input signal voltage span centered at midsupply reduces
the degree of RON modulation.
Figure 6 compares the AD9221/AD9223/AD9220’s THD vs.
frequency performance for a 2 V input span with a commonmode voltage of 1 V and 2.5 V. Note how each A/D with a
common-mode voltage of 1 V exhibits a similar degradation in
THD performance at higher frequencies (i.e., beyond 750 kHz).
Similarly, note how the THD performance at lower frequencies
becomes less sensitive to the common-mode voltage. As the
input frequency approaches dc, the distortion will be dominated
by static nonlinearities such as INL and DNL. It is important to
note that these dc static nonlinearities are independent of any
RON modulation.
–50
where AVSS is nominally 0 V and AVDD is nominally 5 V,
defines this requirement. Thus, the range of valid inputs for
VINA and VINB is any combination that satisfies both
Equations 2 and 3.
AD9220
1VCM
–60
THD – dB
For additional information showing the relationship between
VINA, VINB, VREF and the digital output of the AD9221/
AD9223/AD9220, see Table IV.
Refer to Table I and Table II at the end of this section for a
summary of both the various analog input and reference configurations.
AD9223
1VCM
AD9223
2.5VCM
–80
AD9221
2.5VCM
ANALOG INPUT OPERATION
Figure 5 shows the equivalent analog input of the AD9221/
AD9223/AD9220, which consists of a differential sample-andhold amplifier (SHA). The differential input structure of the
SHA is highly flexible, allowing the devices to be easily configured for either a differential or single-ended input. The dc
offset, or common-mode voltage, of the input(s) can be set to
accommodate either single-supply or dual-supply systems. Also,
note that the analog inputs, VINA and VINB, are interchangeable with the exception that reversing the inputs to the VINA
and VINB pins results in a polarity inversion.
–90
0.1
QS2
VINA
QS1
VINB
QH1
The optimum noise and dc linearity performance for either
differential or single-ended inputs is achieved with the largest
input signal voltage span (i.e., 5 V input span) and matched
input impedance for VINA and VINB. Note that only a slight
degradation in dc linearity performance exists between the 2 V
and 5 V input span as specified in the AD9221/AD9223/
AD9220 DC Specifications.
CS
–
CPIN
CPAR
10
Due to the high degree of symmetry within the SHA topology, a
significant improvement in distortion performance for differential input signals with frequencies up to and beyond Nyquist can
be realized. This inherent symmetry provides excellent cancellation of both common-mode distortion and noise. Also, the
required input signal voltage span is reduced by a half, which
further reduces the degree of RON modulation and its effects
on distortion.
CS
QS1
1
FREQUENCY – MHz
AD9220
2.5VCM
Figure 6. AD9221/AD9223/AD9220 THD vs. Frequency for
VCM = 2.5 V and 1.0 V (AIN = –0.5 dB, Input Span = 2.0 V p-p)
CH
CPIN+
CPAR
AD9221
1VCM
–70
QS2
CH
Figure 5. AD9221/AD9223/AD9220 Simplified Input Circuit
–10–
REV. E
AD9221/AD9223/AD9220
Referring to Figure 5, the differential SHA is implemented using a
switched-capacitor topology. Therefore, its input impedance
and its subsequent effects on the input drive source should be
understood to maximize the converter’s performance. The combination of the pin capacitance, CPIN, parasitic capacitance, CPAR,
and sampling capacitance, CS, is typically less than 16 pF.
When the SHA goes into track mode, the input source must
charge or discharge the voltage stored on CS to the new input
voltage. This action of charging and discharging CS, averaged
over a period of time and for a given sampling frequency, fS,
makes the input impedance appear to have a benign resistive
component. However, if this action is analyzed within a sampling
period (i.e., T = 1/fS), the input impedance is dynamic and therefore certain precautions on the input drive source should be
observed.
The resistive component to the input impedance can be computed by calculating the average charge that gets drawn by CH
from the input drive source. It can be shown that if CS is allowed
to fully charge up to the input voltage before switches QS1 are
opened, then the average current into the input is the same as if
there were a resistor of 1/(CS fS) ohms connected between the
inputs. This means that the input impedance is inversely proportional to the converter’s sample rate. Since CS is only 4 pF,
this resistive component is typically much larger than that of the
drive source (i.e., 25 kΩ at fS = 10 MSPS).
applications may require a larger resistor value to reduce the noise
bandwidth or possibly limit the fault current in an overvoltage
condition. Other applications may require a larger resistor value
as part of an antialiasing filter. In any case, since the THD
performance is dependent on the series resistance and the above
mentioned factors, optimizing this resistor value for a given
application is encouraged.
A slight improvement in SNR performance and dc offset
performance is achieved by matching the input resistance of VINA
and VINB. The degree of improvement is dependent on the
resistor value and the sampling rate. For series resistor values
greater than 100 Ω, the use of a matching resistor is encouraged.
Figure 8 shows a plot for THD performance versus RSERIES for
the AD9221/AD9223/AD9220 at their respective sampling rate
and Nyquist frequency. The Nyquist frequency typically represents the worst case scenario for an ADC. In this case, a high
speed, high performance amplifier (AD8047) was used as the
buffer op amp. Although not shown, the AD9221/AD9223/AD9220
exhibits a slight increase in SNR (i.e. 1 dB to 1.5 dB) as the
resistance is increased from 0 kΩ to 2.56 kΩ due to its bandlimiting
effect on wideband noise. Conversely, it exhibits slight decrease
in SNR (i.e., 0.5 dB to 2 dB) if VINA and VINB do not have a
matched input resistance.
–45
If one considers the SHA’s input impedance over a sampling
period, it appears as a dynamic input impedance to the input
drive source. When the SHA goes into the track mode, the input
source should ideally provide the charging current through RON
of switch QS1 in an exponential manner. The requirement of
exponential charging means that the most common input source,
an op amp, must exhibit a source impedance that is both low
and resistive up to and beyond the sampling frequency.
–55
THD – dB
AD9223
AD9220
The output impedance of an op amp can be modeled with a
series inductor and resistor. When a capacitive load is switched
onto the output of the op amp, the output will momentarily
drop due to its effective output impedance. As the output recovers, ringing may occur. To remedy the situation, a series resistor
can be inserted between the op amp and the SHA input as shown
in Figure 7. The series resistance helps isolate the op amp from
the switched-capacitor load.
VCC
VINA
RS
VINB
VEE
VREF
10F
0.1F
SENSE
REFCOM
Figure 7. Series Resistor Isolates Switched-Capacitor SHA
Input from Op Amp. Matching Resistors Improve SNR
Performance
The optimum size of this resistor is dependent on several factors,
which include the AD9221/AD9223/AD9220 sampling rate, the
selected op amp, and the particular application. In most applications, a 30 Ω to 50 Ω resistor is sufficient. However, some
REV. E
–75
AD9221
–85
1
10
100
RSERIES – 1k
10k
Figure 8. THD vs. RSERIES (fIN = fS / 2, AIN = –0.5 dB, Input
Span = 2 V p-p, VCM = 2.5 V)
Figure 8 shows that a small RSERIES between 30 Ω and 50 Ω
provides the optimum THD performance for the AD9220.
Lower values of RSERIES are acceptable for the AD9223 and
AD9221 as their lower sampling rates provide a longer transient
recovery period for the AD8047. Note that op amps with lower
bandwidths will typically have a longer transient recovery period
and therefore require a slightly higher value of RSERIES and/or
lower sampling rate to achieve the optimum THD performance.
AD9221/AD9223/
AD9220
RS
–65
As the value of RSERIES increases, a corresponding increase in
distortion is noted. This is due to its interaction with the SHA’s
parasitic capacitor, CPAR, which has a signal dependency. Thus,
the resulting R-C time constant is signal dependent and consequently a source of distortion.
The noise or small-signal bandwidth of the AD9221/AD9223/
AD9220 is the same as their full-power bandwidth as shown in
Figure 2. For noise sensitive applications, the excessive bandwidth
may be detrimental and the addition of a series resistor and/or
–11–
AD9221/AD9223/AD9220
shunt capacitor can help limit the wideband noise at the A/D’s
input by forming a low-pass filter. Note, however, that the
combination of this series resistance with the equivalent input
capacitance of the AD9221/AD9223/AD9220 should be evaluated for those time-domain applications that are sensitive to the
input signal’s absolute settling time. In applications where harmonic distortion is not a primary concern, the series resistance
may be selected in combination with the SHA’s nominal 16 pF of
input capacitance to set the filter’s 3 dB cutoff frequency.
disable the reference amplifier if the SENSE pin is tied to AVDD.
Disabling the reference amplifier allows the VREF pin to be
driven by an external voltage reference.
AD9221/AD9223/AD9220
TO
A/D
5k
CAPT
5k
A2
5k
A better method of reducing the noise bandwidth, while possibly establishing a real pole for an antialiasing filter, is to add
some additional shunt capacitance between the input (i.e., VINA
and/or VINB) and analog ground. Since this additional shunt
capacitance combines with the equivalent input capacitance of
the AD9221/AD9223/AD9220, a lower series resistance can
be selected to establish the filter’s cutoff frequency while not
degrading the distortion performance of the device. The shunt
capacitance also acts like a charge reservoir, sinking or sourcing
the additional charge required by the hold capacitor, CH, further
reducing current transients seen at the op amp’s output.
The effect of this increased capacitive load on the op amp driving the AD9221/AD9223/AD9220 should be evaluated. To
optimize performance when noise is the primary consideration,
increase the shunt capacitance as much as the transient response
of the input signal will allow. Increasing the capacitance too
much may adversely affect the op amp’s settling time, frequency
response, and distortion performance.
REFERENCE OPERATION
The AD9221/AD9223/AD9220 contain an on-board band gap
reference that provides a pin-strappable option to generate
either a 1 V or 2.5 V output. With the addition of two external
resistors, the user can generate reference voltages other than 1 V
and 2.5 V. Another alternative is to use an external reference for
designs requiring enhanced accuracy and/or drift performance.
See Table II for a summary of the pin-strapping options for the
AD9221/AD9223/AD9220 reference configurations.
CAPB
5k
DISABLE
A2
LOGIC
VREF
A1
1V
7.5k
SENSE
DISABLE
A1
LOGIC
5k
REFCOM
Figure 9. Equivalent Reference Circuit
The actual reference voltages used by the internal circuitry of
the AD9221/AD9223/AD9220 appear on the CAPT and CAPB
pins. For proper operation when using the internal or an external
reference, it is necessary to add a capacitor network to decouple
these pins. Figure 10 shows the recommended decoupling network. This capacitive network performs the following three
functions: (1) along with the reference amplifier, A2, it provides
a low source impedance over a large frequency range to drive
the A/D internal circuitry, (2) it provides the necessary compensation for A2, and (3) it band-limits the noise contribution from
the reference. The turn-on time of the reference voltage appearing between CAPT and CAPB is approximately 15 ms and
should be evaluated in any power-down mode of operation.
0.1F
CAPT
Figure 9 shows a simplified model of the internal voltage reference
of the AD9221/AD9223/AD9220. A pin-strappable reference
amplifier buffers a 1 V fixed reference. The output from the
reference amplifier, A1, appears on the VREF pin. The voltage
on the VREF pin determines the full-scale input span of the
A/D. This input span equals,
AD9221/
AD9223/
AD9220
0.1F
10F
CAPB
0.1F
Figure 10. Recommended CAPT/CAPB Decoupling
Network
Full-Scale Input Span = 2 ⫻ VREF
The voltage appearing at the VREF pin as well as the state of
the internal reference amplifier, A1, are determined by the voltage appearing at the SENSE pin. The logic circuitry contains
two comparators that monitor the voltage at the SENSE pin.
The comparator with the lowest set point (approximately 0.3 V)
controls the position of the switch within the feedback path
of A1. If the SENSE pin is tied to REFCOM, the switch is
connected to the internal resistor network, thus providing a
VREF of 2.5 V. If the SENSE pin is tied to the VREF pin via a
short or resistor, the switch is connected to the SENSE pin. A
short will provide a VREF of 1.0 V while an external resistor
network will provide an alternative VREF between 1.0 V and
2.5 V. The other comparator controls internal circuitry that will
The A/D’s input span may be varied dynamically by changing
the differential reference voltage appearing across CAPT and
CAPB symmetrically around 2.5 V (i.e., midsupply). To change
the reference at speeds beyond the capabilities of A2, it will be
necessary to drive CAPT and CAPB with two high speed, low
noise amplifiers. In this case, both internal amplifiers (i.e., A1
and A2) must be disabled by connecting SENSE to AVDD and
VREF to REFCOM, and the capacitive decoupling network
removed. The external voltages applied to CAPT and CAPB
must be 2.5 V + Input Span/4 and 2.5 V – Input Span/4, respectively, in which the input span can be varied between 2 V and
5 V. Note that those samples within the pipeline A/D during
any reference transition will be corrupted and should be
discarded.
–12–
REV. E
AD9221/AD9223/AD9220
Table I. Analog Input Configuration Summary
Input
Connection
Input Range (V)
Input
Coupling Span (V) VINA*
VINB*
Figure
No.
Single-Ended
DC
1
13, 14
Best for stepped input response applications, suboptimum THD, and noise
performance. Requires ± 5 V op amp.
2 × VREF 0 to
2 × VREF
VREF
13, 14
Same as above but with improved noise
performance due to increase in dynamic
range. Headroom/settling time requirements of ± 5 V op amp should be evaluated.
5
2.5
13, 14
Optimum noise performance, excellent
THD performance. Requires op amp with
VCC > 5 V due to headroom issue.
2.5
24
Optimum THD performance with VREF =
1. Noise to performance improves while
THD performance degrades as VREF
increases to 2.5 V. Single-supply operation
(i.e., 5 V) for many op amps.
2 or
0 to 1 or
1 or VREF
2 × VREF 0 to 2 × VREF
15
Suboptimum ac performance due to input
common-mode level not biased at optimum
midsupply level (i.e., 2.5 V).
5
2.5
15
Optimum noise performance, excellent THD
performance, ability to use ± 5 V op amp.
2 × VREF 2.5 – VREF
to
2.5 + VREF
2.5
16
Flexible input range, optimum THD
performance with VREF = 1. Noise
performance improves while THD performance degrades as VREF increases to 2.5 V.
Ability to use +5 V or ± 5 V op amp.
2
3 to 2
19
Optimum full-scale THD and SFDR
performance well beyond the A/D’s Nyquist
frequency. Preferred mode for undersampling applications.
2
0 to 2
0 to 5
2 × VREF 2.5 – VREF
to
2.5 + VREF
Single-Ended
AC
Differential
AC
(via Transformer)
0 to 5
2 to 3
2 × VREF 2.5 – VREF/2
to
2.5 + VREF/2
2.5 + VREF/2 19
to
2.5 – VREF/2
Same as 2 V to 3 V input range with the
exception that full-scale THD and SFDR
performance can be traded off for better
noise performance. Refer to discussion in AC
Coupling and Interface Issue section and
Simple AC Interface section.
5
3.25 to 1.75
Optimum Noise performance. Also, the
optimum THD and SFDR performance for
“less than” full-scale signals (i.e., –6 dBFS).
Refer to discussion in AC Coupling and
Interface Issue section and Simple AC
Interface section.
1.75 to 3.25
*VINA and VINB can be interchanged if signal inversion is required.
REV. E
Comments
–13–
19
AD9221/AD9223/AD9220
Table II. Reference Configuration Summary
Reference
Operating Mode
Input Span (VINA–VINB)
(V p-p)
Required VREF (V)
Connect
To
INTERNAL
INTERNAL
INTERNAL
2
5
2 ≤ SPAN ≤ 5 and
SPAN = 2 × VREF
1
2.5
1 ≤ VREF ≤ 2.5 and
VREF = (1 + R1/R2)
SENSE
SENSE
R1
R2
VREF
REFCOM
VREF and SENSE
SENSE and REFCOM
EXTERNAL
(Nondynamic)
2 ≤ SPAN ≤ 5
1 ≤ VREF ≤ 2.5
SENSE
VREF
AVDD
EXT. REF.
EXTERNAL
(Dynamic)
2 ≤ SPAN ≤ 5
CAPT and CAPB
Externally Driven
SENSE
VREF
EXT. REF.
EXT. REF.
AVDD
REFCOM
CAPT
CAPB
AD9221/AD9223/AD9220 offers the following benefits: (1)
Signal swings are smaller and therefore linearity requirements
placed on the input signal source may be easier to achieve, (2)
Signal swings are smaller and therefore may allow the use of op
amps that may otherwise have been constrained by headroom
DRIVING THE ANALOG INPUTS
Introduction
The AD9221/AD9223/AD9220 has a highly flexible input
structure, allowing it to interface with single-ended or differential input interface circuitry. The applications shown in sections
Driving the Analog Inputs and Reference Configurations, along
with the information presented in Input and Reference Overview of this data sheet, give examples of both single-ended and
differential operation. Refer to Tables I and II for a list of the
different possible input and reference configurations and their
associated figures in the data sheet.
20
30
40
CMR – dB
AD9223
The optimum mode of operation, analog input range, and associated interface circuitry will be determined by the particular
application’s performance requirements as well as power supply
options. For example, a dc coupled single-ended input would be
appropriate for most data acquisition and imaging applications.
Also, many communication applications that require a dc coupled
input for proper demodulation can take advantage of the excellent single-ended distortion performance of the AD9221/AD9223/
AD9220. The input span should be configured such that the
system’s performance objectives and the headroom requirements
of the driving op amp are simultaneously met.
Alternatively, the differential mode of operation with a transformer
coupled input provides the best THD and SFDR performance
over a wide frequency range. This mode of operation should be
considered for the most demanding spectral based applications
that allow ac coupling (e.g., Direct IF to Digital Conversion).
AD9221
50
60
AD9220
70
80
90
0.1
1
10
FREQUENCY– MHz
100
Figure 11. AD9221/AD9223/AD9220 Input CMR vs.
Input Frequency
limitations, (3) Differential operation minimizes even-order
harmonic products, and (4) Differential operation offers noise
immunity based on the device’s common-mode rejection.
Figure 11 depicts the common-mode rejection of the three devices.
Single-ended operation requires that VINA be ac- or dc-coupled
to the input signal source while VINB of the AD9221/AD9223/
AD9220 can be biased to the appropriate voltage corresponding
to a midscale code transition. Note that signal inversion may be
easily accomplished by transposing VINA and VINB. The rated
specifications for the AD9221/AD9223/AD9220 are characterized using single-ended circuitry with input spans of 5 V and
2 V as well as VINB = 2.5 V.
Differential operation requires that VINA and VINB be simultaneously driven with two equal signals that are in and out of
phase versions of the input signal. Differential operation of the
As is typical of most CMOS devices, exceeding the supply limits
will turn on internal parasitic diodes, resulting in transient currents within the device. Figure 12 shows a simple means of
clamping an ac- or dc-coupled single-ended input with the
addition of two series resistors and two diodes. An optional capacitor is shown for ac-coupled applications. Note that a larger
series resistor could be used to limit the fault current through
D1 and D2 but should be evaluated since it can cause a degradation in overall performance. A similar clamping circuit could also
be used for each input if a differential input signal is being applied.
–14–
REV. E
AD9221/AD9223/AD9220
VCC
OPTIONAL
AC COUPLING
CAPACITOR
network can be inserted between the op amp’s output and the
AD9221/AD9223/AD9220 input to provide a real pole.
AVDD
RS1
30
D2
1N4148
RS2
20
D1
1N4148
Simple Op Amp Buffer
AD9221/
AD9223/
AD9220
VEE
Figure 12. Simple Clamping Circuit
SINGLE-ENDED MODE OF OPERATION
The AD9221/AD9223/AD9220 can be configured for singleended operation using dc or ac coupling. In either case, the
input of the A/D must be driven from an operational amplifier
that will not degrade the A/D’s performance. Because the A/D
operates from a single-supply, it will be necessary to level-shift
ground-based bipolar signals to comply with its input requirements. Both dc and ac coupling provide this necessary function,
but each method results in different interface issues that may
influence the system design and performance.
In the simplest case, the input signal to the AD9221/AD9223/
AD9220 will already be biased at levels in accordance with the
selected input range. It is simply necessary to provide an
adequately low source impedance for the VINA and VINB
analog input pins of the A/D. Figure 13 shows the recommended
configuration for a single-ended drive using an op amp. In this
case, the op amp is shown in a noninverting unity gain configuration driving the VINA pin. The internal reference drives the
VINB pin. Note that the addition of a small series resistor of
30 Ω to 50 Ω connected to VINA and VINB will be beneficial
in nearly all cases. Refer to the Analog Input Operation section
for a discussion on resistor selection. Figure 13 shows the
proper connection for a 0 V to 5 V input range. Alternative
single-ended input ranges of 0 V to 2 × VREF can also be realized with the proper configuration of VREF (refer to the Using
the Internal Reference section).
DC COUPLING AND INTERFACE ISSUES
+V
Many applications require the analog input signal to be dccoupled to the AD9221/AD9223/AD9220. An operational
amplifier can be configured to rescale and level shift the input
signal so that it is compatible with the selected input range of
the A/D. The input range to the A/D should be selected on the
basis of system performance objectives as well as the analog
power supply availability since this will place certain constraints
on the op amp selection.
5V
0V
VINA
RS
–V
2.5V
AD9221/
AD9223/
AD9220
VINB
VREF
10F
0.1F
SENSE
Many of the new high performance op amps are specified for
only ± 5 V operation and have limited input/output swing capabilities. Therefore, the selected input range of the AD9221/
AD9223/AD9220 should be sensitive to the headroom requirements of the particular op amp to prevent clipping of the signal.
Also, since the output of a dual supply amplifier can swing
below –0.3 V, clamping its output should be considered in some
applications.
In some applications, it may be advantageous to use an op
amp specified for single-supply 5 V operation since it will
inherently limit its output swing to within the power supply
rails. An amplifier like the AD8041, AD8011, and AD817 are
useful for this purpose. Rail-to-rail output amplifiers such as
the AD8041 allow the AD9221/AD9223/AD9220 to be configured for larger input spans, which improves the noise
performance.
RS
U1
Figure 13. Single-Ended AD9221/AD9223/AD9220
Op Amp Drive Circuit
Op Amp with DC Level Shifting
Figure 14 shows a dc-coupled level shifting circuit employing an
op amp, A1, to sum the input signal with the desired dc offset.
Configuring the op amp in the inverting mode with the given
resistor values results in an ac signal gain of –1. If the signal
inversion is undesirable, interchange the VINA and VINB connections to re-establish the original signal polarity. The dc voltage
at VREF sets the common-mode voltage of the AD9221/AD9223/
AD9220. For example, when VREF = 2.5 V, the output level
from the op amp will also be centered around 2.5 V. The use of
ratio matched, thin-film resistor networks will minimize gain
and offset errors. Also, an optional pull-up resistor, RP, may be
used to reduce the output load on VREF to ± 1 mA.
5001
If the application requires the largest input span (i.e., 0 V to
5 V) of the AD9221/AD9223/AD9220, the op amp will require
larger supplies to drive it. Various high speed amplifiers in the
Op Amp Selection Guide of this data sheet can be selected to
accommodate a wide range of supply options. Once again,
clamping the output of the amplifier should be considered for
these applications.
+VCC
0.1F
+VREF
0VDC
–VREF
AVDD
RP2
NC
2
7
1
6
A1
5001
0.1F
Two dc-coupled op amp circuits using a noninverting and
inverting topology are discussed below. Although not shown,
the noninverting and inverting topologies can be easily configured as part of an antialiasing filter by using a Sallen-Key or
Multiple-Feedback topology, respectively. An additional R-C
5001
3
5001
RS
VINA
5
AD9221/
AD9223/
AD9220
4
NC
RS
VREF
VINB
NOTES
1OPTIONAL RESISTOR NETWORK-OHMTEK ORNA500D
2OPTIONAL PULL-UP RESISTOR WHEN USING INTERNAL REFERENCE
Figure 14. Single-Ended Input with DC-Coupled
Level Shift
REV. E
–15–
AD9221/AD9223/AD9220
AC COUPLING AND INTERFACE ISSUES
For applications where ac coupling is appropriate, the op amp’s
output can be easily level shifted to the common-mode voltage,
VCM, of the AD9221/AD9223/AD9220 via a coupling capacitor.
This has the advantage of allowing the op amp’s common-mode
level to be symmetrically biased to its midsupply level (i.e.,
(VCC + VEE)/2). Op amps that operate symmetrically with respect
to their power supplies typically provide the best ac performance
as well as the greatest input/output span. Thus, various high
speed/performance amplifiers that are restricted to +5 V/–5 V
operation and/or specified for 5 V single-supply operation can be
easily configured for the 5 V or 2 V input span of the AD9221/
AD9223/AD9220. The best ac distortion performance is achieved
when the A/D is configured for a 2 V input span and commonmode voltage of 2.5 V. Note that differential transformer coupling,
which is another form of ac coupling, should be considered for
optimum ac performance.
in parallel to achieve a low cutoff frequency while maintaining a
low impedance over a wide frequency range. RS isolates the
buffer amplifier from the A/D input. The optimum performance
is achieved when VINA and VINB are driven via «Immetrical
networks. The f–3 dB point can be approximated by the equation,
(
f –3 dB = 1 / 2 × π × R / 2 × (C1 + C 2)
)
+5V
+5V
VIN
C1
R
C2
R
RS
AD9221/
AD9223/
AD9220
–5V
RS
R
+5V
R
C2
VINA
VINB
C1
Simple AC Interface
Figure 16. AC-Coupled Input-Flexible Input Span,
VCM = 2 V
Figure 15 shows a typical example of an ac-coupled, single-ended
configuration. The bias voltage shifts the bipolar, ground-referenced input signal to approximately VREF. The value for C1
and C2 will depend on the size of the resistor, R. The capacitors,
C1 and C2, are typically a 0.1 µF ceramic and 10 µF tantalum capacitor in parallel to achieve a low cutoff frequency
while maintaining a low impedance over a wide frequency
range. The combination of the capacitor and the resistor form a
high-pass filter with a high-pass –3 dB frequency determined
by the equation,
(
f3 dB = 1 / 2 × π × R × (C1 + C 2)
Op Amp Selection Guide
Op amp selection for the AD9221/AD9223/AD9220 is highly
dependent on a particular application. In general, the performance
requirements of any given application can be characterized by
either time domain or frequency domain parameters. In either
case, one should carefully select an op amp that preserves the
performance of the A/D. This task becomes challenging when
one considers the AD9221/AD9223/AD9220’s high performance capabilities coupled with other extraneous system level
requirements such as power consumption and cost.
)
The low impedance VREF voltage source both biases the VINB
input and provides the bias voltage for the VINA input. Figure 15
shows the VREF configured for 2.5 V; thus the input range
The ability to select the optimal op amp may be further complicated by either limited power supply availability and/or limited
acceptable supplies for a desired op amp. Newer, high performance op amps typically have input and output range limitations
in accordance with their lower supply voltages. As a result, some
op amps will be more appropriate in systems where ac-coupling
is allowable. When dc-coupling is required, op amps without
headroom constraints, such as rail-to-rail op amps or ones
where larger supplies can be used, should be considered. The
following section describes some op amps currently available
from Analog Devices. The system designer is always encouraged
to contact the factory or local sales office to be updated on Analog
Devices’ latest amplifier product offerings. Highlights of the
areas where the op amps excel and where they may limit the
performance of the AD9221/AD9223/AD9220 is also included.
C1
+5V
+VREF
0V
–VREF
C2
VIN
AD9221/
AD9223/
RS
VINA AD9220
R
RS
–5V
VINB
VREF
C2
C1
SENSE
Figure 15. AC-Coupled Input
of the A/D is 0 V to 5 V. Other input ranges could be selected
by changing VREF, but the A/D’s distortion performance will
degrade slightly as the input common-mode voltage deviates
from its optimum level of 2.5 V.
AD817:
50 MHz Unity GBW, 70 ns Settling to 0.01%, +5 V
to ± 15 V Supplies
Best Applications: Sample Rates < 7 MSPS, Low
Noise, 5 V p-p Input Range
Limits: THD above 100 kHz
AD826:
Dual Version of AD817
Best Applications: Differential and/or Low Impedance Input
Drivers, Low Noise
Limits: THD above 100 kHz
AD818:
130 MHz @ G = +2 BW, 80 ns Settling to 0.01%,
+5 V to ± 15 V Supplies
Best Applications: Sample Rates < 7 MSPS, Low
Noise, 5 V p-p Input Range, Gains ≥ +2
Limits: THD above 100 kHz
Alternative AC Interface
Figure 16 shows a flexible ac-coupled circuit that can be configured for different input spans. Since the common-mode voltage
of VINA and VINB are biased to midsupply independent of
VREF, VREF can be pin-strapped or reconfigured to achieve
input spans between 2 V and 5 V p-p. The AD9221/AD9223/
AD9220’s CMRR along with the symmetrical coupling R-C
networks will reject both power supply variations and noise. The
resistors, R, establish the common-mode voltage. They may
have a high value (e.g., 5 kΩ) to minimize power consumption
and establish a low cutoff frequency. The capacitors, C1 and
C2, are typically 0.1 µF ceramic and 10 µF tantalum capacitors
–16–
REV. E
AD9221/AD9223/AD9220
AD828:
Dual Version of AD818
Best Applications: Differential and/or Low Impedance Input
Drivers, Low Noise, Gains ≥ +2
Limits: THD above 100 kHz
AD812:
Dual, 145 MHz Unity GBW, Single-Supply Current Feedback, +5 V to ± 15 V Supplies
Best Applications: Differential and/or Low Impedance Input Drivers, Sample Rates < 7 MSPS
Limits: THD above 1 MHz
Note that although a single-ended-to-differential op amp topology would allow dc coupling of the input signal, no significant
improvement in THD performance was realized when compared
to the dc single-ended mode of operation up to the AD9221/
AD9223/AD9220’s Nyquist frequency (i.e., fIN < fS/2). Also,
the additional op amp required in the topology tends to increase
the total system noise, power consumption, and cost. Thus, a
single-ended mode of operation is recommended for most applications requiring dc coupling.
AD8011:
f–3 dB = 300 MHz, +5 V or ± 5 V Supplies, Current
Feedback
Best Applications: Single-Supply, AC-DC-Coupled,
Good AC Specs, Low Noise, Low Power (5 mW)
Limits: THD above 5 MHz, Usable Input/Output
Range
AD8013:
Triple, f–3 dB = 230 MHz, +5 V or ± 5 V Supplies,
Current Feedback, Disable Function
Best Applications: 3:1 Multiplexer, Good AC Specs
Limits: THD above 5 MHz, Input Range
AD9631:
220 MHz Unity GBW, 16 ns Settling to 0.01%,
± 5 V Supplies
Best Applications: Best AC Specs, Low Noise,
AC-Coupled
Limits: Usable Input/Output Range, Power
Consumption
AD8047:
130 MHz Unity GBW, 30 ns Settling to 0.01%,
± 5 V Supplies
Best Applications: Good AC Specs, Low Noise,
AC-Coupled
Limits: THD > 5 MHz, Usable Input Range
AD8041:
Rail-to-Rail, 160 MHz Unity GBW, 55 ns Settling
to 0.01%, 5 V Supply, 26 mW
Best Applications: Low Power, Single-Supply
Systems, DC-Coupled, Large Input Range
Limits: Noise with 2 V Input Range
The figures are also helpful in determining the appropriate A/D
for Direct IF down conversion or undersampling applications.
Refer to Analog Devices application notes AN-301 and AN-302
for an informative discussion on undersampling. One should
select the A/D that meets or exceeds the distortion performance
requirements measured over the required frequency passband.
For example, the AD9220 achieves the best distortion performance over an extended frequency range as a result of its greater
full-power bandwidth and thus would represent the best selection for an IF undersampling application at 21.4 MHz. Refer to
the Applications section of this data sheet for more detailed
information and characterization of this particular application.
–50
–60
Dual AD8041
Best Applications: Differential and/or Low Impedance Input Drivers
Limits: Noise with 2 V Input Range
THD – dB
AD8042:
A dramatic improvement in THD and SFDR performance can
be realized by operating the AD9221/AD9223/AD9220 in the
differential mode using a transformer. Figure 17 shows a plot of
THD versus Input Frequency for the differential transformer
coupled circuit for each A/D while Figure 18 shows a plot of
SFDR versus Input Frequency. Both figures demonstrate the
enhancement in spectral performance for the differential-mode
of operation. The performance enhancement between the differential and single-ended mode is most noteworthy as the input
frequency approaches and goes beyond the Nyquist frequency
(i.e., fIN > fS/2) corresponding to the particular A/D.
–70
AD9221
AD9223
–80
DIFFERENTIAL MODE OF OPERATION
AD9220
Since not all applications have a signal preconditioned for
differential operation, there is often a need to perform a
single-ended-to-differential conversion. In systems that do not
need to be dc-coupled, an RF transformer with a center tap is
the best method to generate differential inputs for the AD9221/
AD9223/AD9220. It provides all the benefits of operating the
A/D in the differential mode without contributing additional
noise or distortion. An RF transformer also has the added benefit of providing electrical isolation between the signal source
and the A/D.
REV. E
–90
1
10
FREQUENCY – MHz
100
Figure 17. AD9221/AD9223/AD9220 THD vs. Input
Frequency (VCM = 2.5 V, 2 V p-p Input Span,
AIN = –0.5 dB)
–17–
AD9221/AD9223/AD9220
Figure 23 of this data sheet. Figure 20 demonstrates how both
spans of the AD9220 achieve the high degree of linearity and
SFDR over a wide range of amplitudes required by the most
demanding communication applications. Similar performance is
achievable with the AD9221 and AD9223 at their corresponding Nyquist frequency.
–55
–65
SFDR – dB
AD9221
AD9223
–75
90
SFDR – 5.0V p-p
80
AD9220
–85
SFDR – 2.0V p-p
SNR/SFDR – dB
70
–95
1
10
FREQUENCY – MHz
100
Figure 18. AD9221/AD9223/AD9220 SFDR vs. Input
Frequency (VCM = 2.5 V, 2 V p-p Input Span,
AIN = –0.5 dB)
SNR – 5.0V p-p
20
–50
MINI-CIRCUITS
T4-1
RS
33
CS
15pF
AD9221/
AD9223/
AD9220
0
REFERENCE CONFIGURATIONS
VINB
The figures associated with this section on internal and external
reference operation do not show recommended matching series resistors
for VINA and VINB for the purpose of simplicity. Please refer to the
Driving the Analog Inputs, Introduction section for a discussion of
this topic. Also, the figures do not show the decoupling network associated with the CAPT and CAPB pins. Please refer to the Reference
Operation section for a discussion of the internal reference circuitry
and the recommended decoupling network shown in Figure 10.
Figure 19. Transformer Coupled Input
Transformers with other turns ratios may also be selected to
optimize the performance of a given application. For example, a
given input signal source or amplifier may realize an improvement in distortion performance at reduced output power levels
and signal swings. Therefore, selecting a transformer with a
higher impedance ratio (e.g., Mini-Circuits T16-6T with a 1:16
impedance ratio) effectively “steps up” the signal level, thus
further reducing the driving requirements of the signal source.
USING THE INTERNAL REFERENCE
Single-Ended Input with 0 to 2 VREF Range
Referring to Figure 19, a series resistor, RS, and shunt capacitor,
CS, were inserted between the AD9221/AD9223/AD9220 and
the secondary of the transformer. The values of 33 Ω and 15 pF
were selected to specifically optimize both the THD and SNR
performance of the A/D. RS and CS help provide some isolation from transients at the A/D inputs reflected back through the
primary of the transformer.
The AD9221/AD9223/AD9220 can be easily configured for
either a 2 V p-p input span or 5.0 V p-p input span by setting
the internal reference (see Table II). Other input spans can be
realized with two external gain setting resistors as shown in
–30
–20
–10
INPUT AMPLITUDE – dBFS
Figure 20 also reveals a noteworthy difference in the SFDR and
SNR performance of the AD9220 between the 2 V p-p and 5 V p-p
input span options. First, the SNR performance improves by 2 dB
with a 5.0 V p-p input span due to the increase in dynamic
range. Second, the SFDR performance of the AD9220 will
improve for input signals below approximately –6.0 dBFS. A 3 dB
to 5 dB improvement was typically realized for input signal levels
between –6.0 dBFS and –36 dBFS. This improvement in SNR
and SFDR for a 5.0 V p-p span may be advantageous for communication systems that have additional margin or headroom
to minimize clipping of the ADC.
CML
0.1F
–40
Figure 20. AD9220 SFDR, SNR vs. Input Amplitude
(fIN = 5 MHz, fCLK = 10 MSPS, VCM = 2.5 V, Differential)
VINA
200
SNR – 2.0V p-p
30
CS
15pF
49.9
50
40
Figure 19 shows the schematic of the suggested transformer
circuit. The circuit uses a Mini-Circuits RF transformer, model
#T4-6T, which has an impedance ratio of 4 (turns ratio of 2).
The schematic assumes that the signal source has a 50 Ω source
impedance. The 1:4 impedance ratio requires the 200 Ω secondary termination for optimum power transfer and VSWR.
The center tap of the transformer provides a convenient means
of level shifting the input signal to a desired common-mode
voltage. Optimum performance can be realized when the center
tap is tied to CML of the AD9221/AD9223/AD9220, which is
the common-mode bias level of the internal SHA.
RS
33
60
Figure 21 shows how to connect the AD9221/AD9223/AD9220
for a 0 V to 2 V or 0 V to 5 V input range via pin strapping the
SENSE pin. An intermediate input range of 0 to 2 × VREF can
be established using the resistor programmable configuration in
Figure 23 and connecting VREF to VINB.
In either case, both the common-mode voltage and input span
are directly dependent on the value of VREF. More specifically,
the common-mode voltage is equal to VREF while the input
span is equal to 2 × VREF. Thus, the valid input range extends
from 0 to 2 × VREF. When VINA is ≤ 0 V, the digital output
will be 000 Hex; when VINA is ≥ 2 × VREF, the digital output
will be FFF Hex.
–18–
REV. E
AD9221/AD9223/AD9220
Shorting the VREF pin directly to the SENSE pin places the
internal reference amplifier in unity-gain mode and the resultant
VREF output is 1 V. Therefore, the valid input range is 0 V to 2 V.
However, shorting the SENSE pin directly to the REFCOM pin
configures the internal reference amplifier for a gain of 2.5 and
the resultant VREF output is 2.5 V. Thus, the valid input range
becomes 0 V to 5 V. The VREF pin should be bypassed to the
REFCOM pin with a 10 µF tantalum capacitor in parallel with a
low inductance 0.1 µF ceramic capacitor.
2VREF
VINA
0V
VINB
10F
AD9221/
AD9223/
AD9220
Resistor Programmable Reference
Figure 23 shows an example of how to generate a reference
voltage other than 1 V or 2.5 V with the addition of two external
resistors and a bypass capacitor. Use the equation,
VREF = 1V × (1 + R1 / R2)
to determine appropriate values for R1 and R2. These resistors
should be in the 2 kΩ to 100 kΩ range. For the example shown,
R1 equals 2.5 kΩ and R2 equals 5 kΩ. From the equation above,
the resultant reference voltage on the VREF pin is 1.5 V. This
sets the input span to be 3 V p-p. To assure stability, place a
0.1 µF ceramic capacitor in parallel with R1.
The common-mode voltage can be set to VREF by connecting
VINB to VREF to provide an input span of 0 to 2 × VREF.
Alternatively, the common-mode voltage can be set to VREF by
connecting VINB to a low impedance 2.5 V source. For the
example shown, the valid input single range for VINA is 1 V to
4 V since VINB is set to an external, low impedance 2.5 V source.
The VREF pin should be bypassed to the REFCOM pin with a
10 µF tantalum capacitor in parallel with a low inductance
0.1 µF ceramic capacitor.
0.1F
VREF
SHORT FOR 0V TO 2V
INPUT SPAN
SENSE
SHORT FOR 0V TO 5V
INPUT SPAN
REFCOM
Figure 21. Internal Reference—2 V p-p Input Span,
VCM = 1 V, or 5 V p-p Input Span, VCM = 2.5 V
4V
VINA
Single-Ended or Differential Input, V CM = 2.5 V
1V
Figure 22 shows the single-ended configuration that gives the best
dynamic performance (SINAD, SFDR). To optimize dynamic
specifications, center the common-mode voltage of the analog
input at approximately by 2.5 V by connecting VINB to a low
impedance 2.5 V source. As described above, shorting the VREF
pin directly to the SENSE pin results in a 1 V reference voltage
and a 2 V p-p input span. The valid range for input signals is 1.5 V
to 3.5 V. The VREF pin should be bypassed to the REFCOM
pin with a 10 µF tantalum capacitor in parallel with a low inductance 0.1 µF ceramic capacitor.
This reference configuration could also be used for a differential
input in which VINA and VINB are driven via a transformer as
shown in Figure 19. In this case, the common-mode voltage,
VCM, is set at midsupply by connecting the transformer’s center
tap to CML of the AD9221/AD9223/AD9220. VREF can be
configured for 1 V or 2.5 V by connecting SENSE to either VREF
or REFCOM respectively. Note that the valid input range for
each of the differential inputs is one-half of single-ended input
and thus becomes VCM – VREF/2 to VCM + VREF/2.
2.5V
VINB
AD9220
1.5V
VREF
10F
0.1F
R1
2.5k
C1
0.1F
SENSE
R2
5k
REFCOM
Figure 23. Resistor Programmable Reference—3 V p-p
Input Span, VCM = 2.5 V
USING AN EXTERNAL REFERENCE
Using an external reference may enhance the dc performance of
the AD9221/AD9223/AD9220 by improving drift and accuracy.
Figures 24 through 26 show examples of how to use an external
reference with the A/D. Table III is a list of suitable voltage
references from Analog Devices. To use an external reference,
the user must disable the internal reference amplifier and drive
the VREF pin. Connecting the SENSE pin to AVDD disables
the internal reference amplifier.
3.5V
VINA
1.5V
VINB
2.5V
1V
10F
0.1F
AD9221/
AD9223/
AD9220
Table III. Suitable Voltage References
VREF
SENSE
REFCOM
Figure 22. Internal Reference—2 V p-p Input Span,
VCM = 2.5 V
REV. E
Internal
AD589
AD1580
REF191
Internal
REF192
REF43
AD780
–19–
Output
Voltage
Drift
(ppm/C)
Initial
Accuracy
% (max)
Operating
Current
(A)
1.00
1.235
1.225
2.048
2.50
2.50
2.50
2.50
26
10–100
50–100
5–25
26
5–25
10–25
3–7
1.4
1.2–2.8
0.08–0.8
0.1–0.5
1.4
0.08–0.4
0.06–0.1
0.04–0.2
N/A
50
50
45
N/A
45
600
1000
AD9221/AD9223/AD9220
The AD9221/AD9223/AD9220 contains an internal reference
buffer, A2 (see Figure 9), that simplifies the drive requirements
of an external reference. The external reference must be able to
drive a ≈5 kΩ (± 20%) load. Note that the bandwidth of the
reference buffer is deliberately left small to minimize the reference noise contribution. As a result, it is not possible to change
the reference voltage rapidly in this mode without the removal
of the CAPT/CAPB Decoupling Network.
conjunction with 1/2 of an OP282 to provide a very low impedance drive for VINB. The selected op amp need not be a high
speed op amp and may be selected based on cost, power, and
accuracy.
3.75V
VINA
1.25V
Figure 24 shows an example of the AD9221/AD9223/AD9220
configured for an input span of 2 × VREF centered at 2.5 V. An
external 2.5 V reference drives the VINB pin, thus setting the
common-mode voltage at 2.5 V. The input span can be independently set by a voltage divider consisting of R1 and R2,
which generates the VREF signal. A1 buffers this resistor network and drives VREF. Choose this op amp based on accuracy
requirements. It is essential that a minimum of a 10 µF capacitor in parallel with a 0.1 µF low inductance ceramic capacitor
decouple the reference output to ground.
2.5V+VREF
2.5V
2.5V–VREF
VINA
2.5V
REF
+5V
0.1F
AD9221/
AD9223/
AD9220
A1
0.1F
VINB
10F
2N2222
1k
7.5k
1/2
OP282
0.1F
316
1.225V
5V
AD1580
VREF
0.1F
10F
5V
SENSE
Figure 26. External Reference Using the AD1580
and Low Impedance Buffer
The AD9221/AD9223/AD9220 output data is presented in
positive true straight binary for all input ranges. Table IV indicates the output data formats for various input ranges regardless
of the selected input range. A twos complement output data
format can be created by inverting the MSB.
0.1F
R1
0.1F
1k
DIGITAL INPUTS AND OUTPUTS
Digital Outputs
VINB
22F
AD9221/
AD9223/
AD9220
820
5V
1k
Variable Input Span with V CM = 2.5 V
VREF
R2
SENSE
+5V
Table IV. Output Data Format
Figure 24. External Reference—VCM = 2.5 V (2.5 V
on VINB, Resistor Divider to Make VREF)
Single-Ended Input with 0 to 2 VREF Range
Figure 25 shows an example of an external reference driving
both VINB and VREF. In this case, both the common-mode
voltage and input span are directly dependent on the value of
VREF. More specifically, the common-mode voltage is equal to
VREF while the input span is equal to 2 × VREF. Thus, the
valid input range extends from 0 to 2 × VREF. For example, if
the REF-191, a 2.048 external reference was selected, the valid
input range extends from 0 to 4.096 V. In this case, 1 LSB of
the AD9221/AD9223/AD9220 corresponds to 1 mV. It is essential that a minimum of a 10 µF capacitor in parallel with a 0.1 µF
low inductance ceramic capacitor decouple the reference output
to ground.
Input (V)
Condition (V)
Digital Output
OTR
VINA –VINB
VINA –VINB
VINA –VINB
VINA –VINB
VINA –VINB
< – VREF
= – VREF
=0
= + VREF – 1 LSB
≥ + VREF
0000 0000 0000
0000 0000 0000
1000 0000 0000
1111 1111 1111
1111 1111 1111
1
0
0
0
1
+FS –1 1/2 LSB
OTR DATA OUTPUTS
1
0
0
1111 1111 1111
1111 1111 1111
1111 1111 1110
OTR
–FS+1/2 LSB
0
0
1
0000 0000 0001
0000 0000 0000
0000 0000 0000
–FS
–FS –1/2 LSB
2REF
VINA
0V
+5V
VINB
VREF
0.1F
AD9221/
AD9223/
AD9220
10F
0.1F
VREF
0.1F
+5V
SENSE
Figure 25. Input Range = 0 V to 2 × VREF
Low Cost/Power Reference
The external reference circuit shown in Figure 26 uses a low
cost 1.225 V external reference (e.g., AD580 or AD1580) along
with an op amp and transistor. The 2N2222 transistor acts in
+FS
+FS –1/2 LSB
Figure 27. Output Data Format
Out Of Range (OTR)
An out-of-range condition exists when the analog input voltage
is beyond the input range of the converter. OTR is a digital
output that is updated along with the data output corresponding
to the particular sampled analog input voltage. Thus, OTR has
the same pipeline delay (latency) as the digital data. It is LOW
when the analog input voltage is within the analog input range.
It is HIGH when the analog input voltage exceeds the input
range as shown in Figure 27. OTR will remain HIGH until the
analog input returns within the input range and another conversion is completed. By logical ANDing OTR with the MSB and
its complement, overrange high or underrange low conditions
–20–
REV. E
AD9221/AD9223/AD9220
can be detected. Table V is a truth table for the over/underrange
circuit in Figure 28, which uses NAND gates. Systems requiring
programmable gain conditioning of the AD9221/AD9223/
AD9220 input signal can immediately detect an out-of-range
condition, thus eliminating gain selection iterations. Also, OTR
can be used for digital offset and gain calibration.
In the equation, the rms aperture jitter, tA, represents the rootsum square of all the jitter sources, which include the clock
input, analog input signal, and A/D aperture jitter specification.
For example, if a 5 MHz full-scale sine wave is sampled by an
A/D with a total rms jitter of 15 ps, the SNR performance of the
A/D will be limited to 66.5 dB. Undersampling applications are
particularly sensitive to jitter.
Table V. Out-of-Range Truth Table
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the AD9221/
AD9223/AD9220. As such, supplies for clock drivers should be
separated from the A/D output driver supplies to avoid modulating
the clock signal with digital noise. Low jitter crystal controlled
oscillators make the best clock sources. If the clock is generated
from another type of source (by gating, dividing, or other method),
it should be retimed by the original clock at the last step.
OTR
MSB
Analog Input Is
0
0
1
1
0
1
0
1
In Range
In Range
Underrange
Overrange
MSB
Most of the power dissipated by the AD9221/AD9223/AD9220
is from the analog power supplies. However, lower clock speeds
will reduce digital current slightly. Figure 29 shows the relationship between power and clock rate for each A/D.
OVER = “1”
OTR
UNDER = “1”
MSB
66
Figure 28. Overrange or Underrange Logic
64
Digital Output Driver Considerations (DVDD)
62
POWER – mW
The AD9221, AD9223 and AD9220 output drivers can be
configured to interface with 5 V or 3.3 V logic families by setting
DVDD to 5 V or 3.3 V respectively. The AD9221/AD9223/
AD9220 output drivers are sized to provide sufficient output
current to drive a wide variety of logic families. However, large
drive currents tend to cause glitches on the supplies and may
affect SINAD performance. Applications requiring the AD9221/
AD9223/AD9220 to drive large capacitive loads or large fanout
may require additional decoupling capacitors on DVDD. In
extreme cases, external buffers or latches may be required.
60
5V p-p
58
2V p-p
56
54
52
50
48
0.5
Clock Input and Considerations
1.0
1.5
2.0
2.5
3.0
CLOCK FREQUENCY – MHz
The AD9221/AD9223/AD9220 internal timing uses the two
edges of the clock input to generate a variety of internal timing
signals. The clock input must meet or exceed the minimum
specified pulsewidth high and low (tCH and tCL) specifications
for the given A/D as defined in the Switching Specifications to
meet the rated performance specifications. For example, the
clock input to the AD9220 operating at 10 MSPS may have a
duty cycle between 45% to 55% to meet this timing requirement
since the minimum specified tCH and tCL is 45 ns. For clock
rates below 10 MSPS, the duty cycle may deviate from this
range to the extent that both tCH and tCL are satisfied.
Figure 29a. AD9221 Power Consumption vs. Clock
Frequency
125
120
POWER – mW
115
All high speed high resolution A/Ds are sensitive to the quality
of the clock input. The degradation in SNR at a given full-scale
input frequency (fIN) due to only aperture jitter (tA) can be
calculated with the following equation:
110
5V p-p
105
2V p-p
100
95
SNR = 20 log10 [1 / 2π fIN t A ]
90
0
1
2
3
4
5
6
CLOCK FREQUENCY – MHz
Figure 29b. AD9223 Power Consumption vs. Clock
Frequency
REV. E
–21–
AD9221/AD9223/AD9220
impedance over a wide frequency range. Note that the
AVDD and AVSS pins are co-located on the AD9221/
AD9223/AD9220 to simplify the layout of the decoupling
capacitors and provide the shortest possible PCB trace
lengths. The AD9221/AD9223/AD9220/EB power plane
layout, shown in Figure 40 depicts a typical arrangement
using a multilayer PCB.
300
280
POWER – mW
INPUT = 5V p-p
260
INPUT = 2V p-p
240
26 AVDD
0.1F
220
25 AVSS
200
0
2
4
6
8
10
12
AD9221/
AD9223/
AD9220
15 AVDD
14
0.1F
CLOCK FREQUENCY – MHz
16 AVSS
Figure 29c. AD9220 Power Consumption vs. Clock
Frequency
Figure 30. Analog Supply Decoupling
The CML is an internal analog bias point used internally by the
AD9221/AD9223/AD9220. This pin must be decoupled with at
least a 0.1 µF capacitor as shown in Figure 31. The dc level of
CML is approximately AVDD/2. This voltage should be buffered if it is to be used for any external biasing.
GROUNDING AND DECOUPLING
Analog and Digital Grounding
Proper grounding is essential in any high speed, high resolution
system. Multilayer printed circuit boards (PCBs) are recommended to provide optimal grounding and power schemes. The
use of ground and power planes offers distinct advantages:
1. The minimization of the loop area encompassed by a signal
and its return path.
2. The minimization of the impedance associated with ground
and power paths.
3. The inherent distributed capacitor formed by the power
plane, PCB insulation, and ground plane.
22
AD9221/
AD9223/
AD9220
Figure 31. CML Decoupling
These characteristics result in both a reduction of electromagnetic interference (EMI) and an overall improvement in
performance.
It is important to design a layout that prevents noise from coupling onto the input signal. Digital signals should not be run in
parallel with input signal traces and should be routed away from
the input circuitry. While the AD9221/AD9223/AD9220 features
separate analog and digital ground pins, it should be treated as
an analog component. The AVSS and DVSS pins must be joined
together directly under the AD9221/AD9223/AD9220. A solid
ground plane under the A/D is acceptable if the power and
ground return currents are managed carefully. Alternatively,
the ground plane under the A/D may contain serrations to steer
currents in predictable directions where cross-coupling between
analog and digital would otherwise be unavoidable. The AD9221/
AD9223/AD9220/EB ground layout, shown in Figure 39, depicts
the serrated type of arrangement. The analog and digital grounds
are connected by a jumper below the A/D.
The digital activity on the AD9221/AD9223/AD9220 chip falls
into two general categories: correction logic and output drivers.
The internal correction logic draws relatively small surges of
current, mainly during the clock transitions. The output drivers
draw large current impulses while the output bits are changing.
The size and duration of these currents are a function of the
load on the output bits: large capacitive loads are to be avoided.
Note, the internal correction logic of the AD9221, AD9223,
and AD9220 is referenced to AVDD while the output drivers
are referenced to DVDD.
The decoupling shown in Figure 32, a 0.1 µF ceramic chip
capacitor, is appropriate for a reasonable capacitive load on
the digital outputs (typically 20 pF on each pin). Applications
involving greater digital loads should consider increasing the
digital decoupling proportionally, and/or using external buffers/latches.
28 DVDD
0.1F
27 DVSS
Analog and Digital Supply Decoupling
The AD9221/AD9223/AD9220 features separate analog and
digital supply and ground pins, helping to minimize digital
corruption of sensitive analog signals. In general, AVDD, the
analog supply, should be decoupled to AVSS, the analog
common, as close to the chip as physically possible. Figure 30
shows the recommended decoupling for the analog supplies;
0.1 µF ceramic chip capacitors should provide adequately low
CML
0.1F
AD9221/
AD9223/
AD9220
Figure 32. Digital Supply Decoupling
A complete decoupling scheme will also include large tantalum
or electrolytic capacitors on the PCB to reduce low frequency
ripple to negligible levels. Refer to the AD9221/AD9223/
AD9220/EB schematic and layouts in Figures 36 to 42 for more
information regarding the placement of decoupling capacitors.
–22–
REV. E
AD9221/AD9223/AD9220
90
APPLICATIONS
Direct IF Down Conversion Using the AD9220
80
As previously noted, the AD9220’s performance in the differential mode of operation extends well beyond its baseband region
and into several Nyquist zone regions. Thus, the AD9220 may
be well suited as a mix down converter in both narrow and
wideband applications. Various IF frequencies exist over the
frequency range in which the AD9220 maintains excellent
dynamic performance (e.g., refer to Figure 17 and 18). The IF
signal will be aliased to the ADC’s baseband region due to the
sampling process in a similar manner that a mixer will downconvert an IF signal. For signals in various Nyquist zones, the
following equation may be used to determine the final frequency
after aliasing.
f1 NYQUIST = fSIGNAL
f2 NYQUIST = fSAMPLE – fSIGNAL
f3 NYQUIST = abs (fSAMPLE – fSIGNAL)
f4 NYQUIST = 2 × fSAMPLE – fSIGNAL
f5 NYQUIST = abs (2 × fSAMPLE – fSIGNAL)
1
ENCODE = 10MSPS
AIN = 21.4MHz
AMPLITUDE – dB
–40
–60
–80
8
7
2
6
5
3
4
9
–100
–120
1
FREQUENCY – MHz
5
Figure 33. IF Sampling a 21.4 MHz Input Using
the AD9220 (VCM = 2.5 V, Input Span = 2 V p-p)
REV. E
SNR/SFDR – dB
50
SNR
40
30
20
10
0
–50
–40
–30
–20
AIN – dB
–10
0
Multichannel Data Acquisition with Autocalibration
One common example is the digitization of a 21.4 MHz IF
using a low jitter 10 MHz sample clock. Using the equation
above for the fifth Nyquist zone, the resultant frequency after
sampling is 1.4 MHz. Figure 33 shows the typical performance
of the AD9220 operating under these conditions. Figure 34
demonstrates how the AD9220 is still able to maintain a high
degree of linearity and SFDR over a wide amplitude.
–20
SFDR
60
Figure 34. AD9220 Differential Input SNR/SFDR
vs. Input Amplitude (AIN) @ 21.4 MHz
There are several potential benefits in using the ADC to alias
(i.e., or mix) down a narrow-band or wideband IF signal. First
and foremost is the elimination of a complete mixer stage with
its associated amplifiers and filters, reducing cost and power
dissipation. Second is the ability to apply various DSP techniques to perform such functions as filtering, channel selection,
quadrature demodulation, data reduction, and detection.
0
70
The AD9221/AD9223/AD9220 is well suited for high performance, low power data acquisition systems. Aside from its
exceptional ac performance, it exhibits true 12-bit linearity and
temperature drift performance (i.e., excluding internal reference). Furthermore, the A/D product family provides the system
designer with an upward or downward component selection
path based on power consumption and sampling rate.
A typical multichannel data acquisition system is shown in
Figure 35. Also shown is some additional inexpensive gain and
offset autocalibration circuitry that is often required in high
accuracy data acquisition systems. These additional peripheral
components were selected based on their performance, power
consumption, and cost.
Referring to Figure 35, the AD9221/AD9223/AD9220 is configured for single-ended operation with a 2.5 V p-p input span and
a 2.5 V common-mode voltage using an external, precision 2.5
voltage reference, U1. This configuration and input span allows
the buffer amplifier, U4, to be single supply. Also, it simplifies
the design of the low temperature drift autocalibration circuitry
that uses thin-film resistors for temperature stability and ratiometric accuracy. The input of the AD9221/AD9223/AD9220
can be easily configured for a wider span but it should remain
within the input/output swing capabilities of a high speed, railto-rail, single-supply amplifier, U4 (e.g., AD8041).
The gain and offset calibration circuitry is based on two 8-bit,
current-output DAC08s, U3 and U5. The gain calibration
circuitry consisting of U3, and an op amp, U2A, is configured
to provide a low drift nominal 1.25 V reference to the AD9221/
AD9223/AD9220. The resistor values that set the gain calibration range were selected to provide a nominal adjustment span
of ±128 LSBs with 1 LSB resolution with respect to the A/D. Note
that the bandwidth of the reference is low and, as a result, it is
not possible to change the reference voltage rapidly in this mode.
–23–
AD9221/AD9223/AD9220
The offset calibration circuitry consists of a DAC, U5 and
the buffer amplifier, U4. The DAC is configured for a bipolar
adjustment span of ± 64 LSB with a 1/2 LSB resolution span
with respect to the AD9221/AD9223/AD9220. Note that both
current outputs of U5 were configured to provide a bipolar
adjustment span. Also, RC is used to decouple the output of
both DACs, U3 and U5, from their respective op amps.
input that corresponds to –FS of the A/D. Before the value is
read, U4 should be preloaded with a code of 00 (Hex). Several
readings can also be taken and averaged to determine the digital
gain adjustment code to U2A. In this case, 1 LSB of the A/D
corresponds to 1 LSB of U4.
Due to the AD9221/AD9223/AD9220’s excellent INL performance, a two-point calibration procedure (i.e., –FS to midscale)
instead of an endpoint calibration procedure was chosen. Also,
since the bipolar offset is insensitive to any gain adjustment (due
to the differential SHA of the A/D), an iterative calibration
process is not required. The temperature stability of the circuit
is enhanced by selecting a dual precision op amp for U2 (e.g.,
OP293) and low temperature drift, thin film resistors. Note that
this application circuit was not built at the release of this data
sheet. Please consult Analog Devices for application assistance
or comments.
The calibration procedure consists of a two step process. First,
the bipolar offset is calibrated by selecting CH2, the 2.5 V system reference, of the analog multiplexer and preloading the DAC,
U5, with a midscale code of 1000 0000. If possible, several
readings of the A/D should be taken and averaged to determine
the required digital offset adjustment code, U5. This averaged
offset code requires an extra bit of resolution since 1 LSB of U5
equates to 1/2 LSB of the AD9221/AD9223/AD9220. The
required offset correction code to U5 can then be determined.
Second, the system gain is calibrated by selecting CH2, a 1.25 V
0.1F
1.25k
1.25V
U2B
2.5k
2.5k
10F
162
2.5k
0.1F
0.1F
CH1
2.5k
1.1k
U1
REF43
2 39
+5V
CH2
2.5k
CH3
CH4
U6
CH5
ADG608
1.25V
39mV
U2A
0.1F
IOUT
VREF(+)
OUT
VREF(–)
CH6
VREF
U3
DAC08
AD9221/
AD9223/
AD9220
IOUT
2.5k
CH7
SENSE
RC
100
CH8
BIT 1 – BIT 12
OTR
39
U4
VINA
39
2.50V
VINB
RC
100
2.5k
VREF(+)
VREF(–)
2.5k
U5
DAC08
RC
100
IOUT
IOUT
Figure 35. Typical Multichannel Data Acquisition System
–24–
REV. E
AD9221/AD9223/AD9220
+5A
TPA
+5A
JP19
D2
1N5711
VINA
D3
1N5711
C19
0.1F
C13
15pF
A
A
A
NOT
INSTALLED
C26
0.1F
C24
10F
16V
A
C25
0.1F
C23
0.1F
TP1
C28
0.1F
+5A
A
TPB
D4
1N5711
VINB
C15
15pF
D5
1N5711
JP10
A
C14
0.1F
A
15
26
23
20
21
22
24
18
17
19
AGND
27
DGND
28
25
+5D
16
1
13
14
12
11
10
9
8
7
6
5
4
3
2
1
1
19
BIT 7 9
BIT 6 8
BIT 5 7
BIT 4 6
BIT 3 5
BIT 2 4
MSB 3
2
10
U8
U8
5
3
CLK 74HC04N
JP15
6
74HC04N
TPE
JP11
J7
+5A
C33
0.1F
JP12
R12
10k
CLK IN
C17
0.1F
JP13
R17 TP14
Y4A 22
20 +5D2
G1
Y7
G2
Y6
A7
Y5
A6
Y4
A5
U7 Y3
A4
Y2
A3
Y1
A2
Y0
A1
A0
GND +5VD
11
12
13
14
15
16
17
18
Y3A
R18 TP13
22
Y2A
R19 TP12
22
Y1A
R20 TP11
22
Y0A
R21 TP10
22
TP9
Y7B
R22
22
TP8
Y6B
R23
22
TP7
Y5B
R24
22
TP6
Y4B
R25
22
Y3B
R26 TP5
22
Y2B
Y3B
Y4B
Y5B
Y6B
Y7B
+5D2
20
C22
0.1F
R14
50
C16
10F
16V
JP14
R13
10k
R16 TP15
Y5A 22
Y0A
Y1A
Y2A
Y3A
Y4A
Y5A
13
14
15
16
17
18
11
12
C21
0.1F
1
19
9
8
OTR 7
LSB 6
BIT 11 5
BIT 10 4
BIT 9 3
BIT 8 2
10
R15 TP16
22
1 J8
G1
Y5
G2
Y4
A7
Y3
A6
Y2
A5
U6
Y1
A4
Y0
A3
Y7
A2
Y6
A1
A0
GND +5VD
74HC541N
4
REFOUT
JP20
74HC541N
MSB
OTR
BIT 2
BIT 3
BIT 4
BIT 5
BIT 6
BIT 7
BIT 8
BIT 9
BIT 10
BIT 11
LSB
CLK
JP16
JP21
2 MSB
74HC04N
BIT 1
AVDD
OTR
AVDD
BIT 2
VINA
BIT 3
CAPB
BIT 4
CAPT
CML U5 BIT 5
BIT 6
VINB
BIT 7
VREF
BIT 8
SENSE
REFCOM BIT 9
BIT 10
DVSS
BIT 11
DVDD
BIT 12
AVSS
CLK
AVSS
A
C20
0.1F
A
NOT
INSTALLED REMOVE
FOR DIFF.
MODE
TPC
TPD
AD9221/
AD9223/
AD9220
C18
0.1F
U8
5 J8
7 J8
9 J8
11 J8
13 J8
15 J8
17 J8
19 J8
21 J8
23 J8
R27
Y2B 22
TP4
39 J8
JP18
+5REFBUF
A
3 J8
JP17
R10
820
R28
22
TP3
33 J8
REFOUT
+5REFBUF
C12
0.1F
U4
U3
REF43
2
+5REFBUF
9
U8
8
U8
R7
15k R9
50
JP9
4
EXTERNAL REFERENCE
AND REFERENCE BUFFER
AD817
C8
10F
16V
A
R8
10k
Q1
2N2222
U8
12 J8
A
A
R6
10k
U2
3
JP5
JP1
AIN
+SUPPLY
C2
0.1F
U1
A
A
2
4
–SUPPLY
R2
261
R3
261
A
TPH
27 J8
A
JP7
C5
0.1F
J6
GJ1
AGND
A
(GJ1-WIRE
JUMPER CKT SIDE)
–25–
31 J8
+5D
32 J8
34 J8
L4
POWER
SUPPLY
29 J8
30 J8
L3
TPK TPL
Figure 36. Evaluation Board Schematic
REV. E
C4
0.1F
A
DGND
TPJ
28 J8
–SUPPLY
C31
22F
25V
J5
VINA
JP3
L2
A
TPI
A
25 J8
26 J8
+5 DIG
JP4
22 J8
24 J8
2
C30
22F
25V
J4
R4
33
6
C1
0.1F
JP2
C29
22F
25V
A
–VEE
A
7
3
AD8047
R1
50
TPG
J3
IN OUT 1
GND
18 J8
20 J8
C32
0.1F
A
78L05P
+VCC
A
J1
16 J8
L6
JP6
J2
C3
0.1F
A
14 J8
+5REFBUF
74HC04N
R5
10k
10 J8
+5A
TPF
SPARE GATES
8 J8
R29
316
L1
L5
2 J8
4 J8
6 J8
+SUPPLY
VINB
12
A
A
74HC04N
13
C11
0.1F
6
C34
0.1F
4
–SUPPLY
A
C27
0.1F
A
7
3
2
U8
DECOUPLING
+5D2
10
R11
1k
F.S./GAIN ADJ
GND
A
74HC04N
11
VOUT 6
VIN
C7
0.1F
A
C9
0.1F
C10
10F
16V
NC 35 J8
C6
0.1F
+5D2
36 J8
NC 37 J8
38 J8
40 J8
A
AD9221/AD9223/AD9220
Figure 37. Evaluation Board Component Side Layout (Not to Scale)
Figure 38. Evaluation Board Solder Side Layout (Not to Scale)
–26–
REV. E
AD9221/AD9223/AD9220
Figure 39. Evaluation Board Ground Plane Layout (Not to Scale)
Figure 40. Evaluation Board Power Plane Layout
REV. E
–27–
AD9221/AD9223/AD9220
Figure 41. Evaluation Board Component Side Silkscreen (Not to Scale)
Figure 42. Evaluation Board Component Side Silkscreen (Not to Scale)
–28–
REV. E
AD9221/AD9223/AD9220
OUTLINE DIMENSIONS
28-Lead Standard SmWall Outline Package [SOIC]
Wide Body
(R-28)
Dimensions shown in millimeters and (inches)
18.10 (0.7126)
17.70 (0.6969)
28
15
7.60 (0.2992)
7.40 (0.2913)
1
10.65 (0.4193)
10.00 (0.3937)
14
2.65 (0.1043)
2.35 (0.0925)
0.75 (0.0295)
45
0.25 (0.0098)
0.30 (0.0118)
0.10 (0.0039)
COPLANARITY
0.10
8
0
1.27 (0.0500) 0.51 (0.0201) SEATING
0.32 (0.0126)
PLANE
BSC
0.33 (0.0130)
0.23 (0.0091)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-013AE
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
28-Lead Shrink Small Outline Package [SSOP]
(RS-28)
Dimensions shown in millimeters
10.50
10.20
9.90
28
15
5.60
5.30
5.00
8.20
7.80
7.40
14
1
1.85
1.75
1.65
2.00 MAX
0.10
COPLANARITY
0.25
0.09
0.05
MIN
0.65
BSC
0.38
0.22
SEATING
PLANE
8
4
0
COMPLIANT TO JEDEC STANDARDS MO-150AH
REV. E
–29–
0.95
0.75
0.55
AD9221/AD9223/AD9220
Revision History
Location
Page
2/03—Data Sheet changed from REV. D to REV. E.
Updated graphic captions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Global
Changes to DC SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Changes to INTRODUCTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Changes to Digital Output Driver Considerations (DVDD) section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
–30–
REV. E
–31–
–32–
PRINTED IN U.S.A.
C00576–0–2/03(E)