AD AD549K Ultralow input bias current operational amplifier Datasheet

a
FEATURES
Ultralow Bias Current: 60 fA max (AD549L)
250 fA max (AD549J)
Input Bias Current Guaranteed Over Common-Mode
Voltage Range
Low Offset Voltage: 0.25 mV max (AD549K)
1.00 mV max (AD549J)
Low Offset Drift: 5 mV/8C max (AD549K)
20 mV/8C max (AD549J)
Low Power: 700 mA max Supply Current
Low Input Voltage Noise: 4 mV p-p 0.1 Hz to 10 Hz
MIL-STD-883B Parts Available
APPLICATIONS
Electrometer Amplifiers
Photodiode Preamp
pH Electrode Buffer
Vacuum lon Gage Measurement
PRODUCT DESCRIPTION
The AD549 is a monolithic electrometer operational amplifier
with very low input bias current. Input offset voltage and input
offset voltage drift are laser trimmed for precision performance.
The AD549’s ultralow input current is achieved with “Topgate”
JFET technology, a process development exclusive to Analog
Devices. This technology allows the fabrication of extremely low
input current JFETs compatible with a standard junctionisolated bipolar process. The 1015 Ω common-mode impedance,
a result of the bootstrapped input stage, insures that the input
current is essentially independent of common-mode voltage.
The AD549 is suited for applications requiring very low input
current and low input offset voltage. It excels as a preamp for a
wide variety of current output transducers such as photodiodes,
photomultiplier tubes, or oxygen sensors. The AD549 can also
be used as a precision integrator or low droop sample and hold.
The AD549 is pin compatible with standard FET and electrometer op amps, allowing designers to upgrade the performance of
present systems at little additional cost.
The AD549 is available in a TO-99 hermetic package. The case
is connected to Pin 8 so that the metal case can be independently
connected to a point at the same potential as the input terminals, minimizing stray leakage to the case.
*Protected by Patent No. 4,639,683.
Ultralow Input Bias Current
Operational Amplifier
AD549*
CONNECTION DIAGRAM
GUARD PIN, CONNECTED TO CASE
NC
OFFSET NULL
8
1
AD549
V+
7
INVERTING 2
INPUT
6 OUTPUT
3
NONINVERTING
INPUT
5
4
OFFSET
NULL
V–
10kΩ
5
1
4 –15V
VOS TRIM
NC = NO CONNECTION
The AD549 is available in four performance grades. The J, K,
and L versions are rated over the commercial temperature range
0°C to +70°C. The S grade is specified over the military temperature range of –55°C to +125°C and is available processed to
MIL-STD-883B, Rev C. Extended reliability PLUS screening is
also available. Plus screening includes 168-hour burn-in, as
well as other environmental and physical tests derived from
MIL-STD-883B, Rev C.
PRODUCT HIGHLIGHTS
1. The AD549’s input currents are specified, 100% tested and
guaranteed after the device is warmed up. Input current is
guaranteed over the entire common-mode input voltage
range.
2. The AD549’s input offset voltage and drift are laser trimmed
to 0.25 mV and 5 µV/°C (AD549K), 1 mV and 20 µV/°C
(AD549J).
3. A maximum quiescent supply current of 700 µA minimizes
heating effects on input current and offset voltage.
4. AC specifications include 1 MHz unity gain bandwidth and
3 V/µs slew rate. Settling time for a 10 V input step is 5 µs to
0.01%.
5. The AD549 is an improved replacement for the AD515,
OPA104, and 3528.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD549–SPECIFICATIONS (@ +258C and V = +15 V dc, unless otherwise noted)
S
Model
Min
INPUT BIAS CURRENT1
Either Input, VCM = 0 V
Either Input, VCM = ± 10 V
Either Input at TMAX,
VCM = 0 V
Offset Current
Offset Current at TMAX
AD549J
Typ
Max
150
150
250
250
Min
11
50
2.2
AD549K
Typ
Max
75
75
100
100
10
32
32
15
INPUT VOLTAGE NOISE
f = 0.1 Hz to 10 Hz
f = 10 Hz
f = 100 Hz
f = 1 kHz
f = 10 kHz
4
90
60
35
35
4
90
60
35
35
INPUT CURRENT NOISE
f = 0.1 Hz to 10 Hz
f = 1 kHz
0.7
0.22
INPUT IMPEDANCE
Differential
VDIFF = ± 1
Common Mode
VCM = ± 10
OPEN-LOOP GAIN
VO @ ± 10 V, RL = 10 k
VO @ ± 10 V, RL = 10 k,
TMIN to TMAX
VO = ± 10 V, RL = 2 k
VO = ± 10 V, RL = 2 k,
TMIN to TMAX
INPUT VOLTAGE RANGE
Differential3
Common-Mode Voltage
Common-Mode Rejection Ratio
V = +10 V, –10 V
TMIN to TMAX
OUTPUT CHARACTERISTICS
Voltage @ RL = 10 k,
TMIN to TMAX
Voltage @ RL = 2 k,
TMIN to TMAX
Short Circuit Current
TMIN to TMAX
Load Capacitance Stability
G = +1
FREQUENCY RESPONSE
Unity Gain, Small Signal
Full Power Response
Slew Rate
Settling Time, 0.1%
0.01%
Overload Recovery,
50% Overdrive, G = –1
0.15
1.0
1.9
20
100
100
2
10
10
15
AD549L
Typ
40
40
4.2
30
1.3
INPUT OFFSET VOLTAGE2
Initial Offset
Offset at TMAX
vs. Temperature
vs. Supply
vs. Supply, TMIN to TMAX
Long-Term Offset Stability
0.5
Min
Max
Min
60
60
2.8
20
0.85
0.3
0.25
0.4
5
32
32
5
10
10
15
AD549S
Typ
Max
Units
75
75
100
100
fA
fA
420
30
125
0.3
0.5
0.9
10
32
32
10
10
32
15
pA
fA
pA
0.5
2.0
15
32
50
mV
mV
µV/°C
µV/V
µV/V
µV/Month
4
90
60
35
35
4
90
60
35
35
µV p-p
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
0.5
0.16
0.36
0.11
0.5
0.16
fA rms
fA/√Hz
1013i1
1013i1
1013i1
1013i1
ΩipF
1015i0.8
1015i0.8
1015i0.8
1015i0.8
ΩipF
6
300
1000
300
1000
300
1000
300
1000
V/mV
300
100
800
250
300
100
800
250
300
100
800
250
300
100
800
250
V/mV
V/mV
80
200
80
200
80
200
25
150
V/mV
± 20
+10
–10
80
76
90
80
–12
–10
15
9
20
90
80
+12
–12
+10
35
–10
15
9
4000
0.7
2
1.0
50
3
4.5
5
2
± 20
+10
–10
100
90
20
90
80
+12
–12
+10
35
–10
15
9
4000
0.7
2
1.0
50
3
4.5
5
2
± 20
+10
–10
100
90
20
2
1.0
50
3
4.5
5
2
–2–
90
80
+12
–12
+10
35
–10
15
6
4000
0.7
± 20
+10
–10
0.7
2
100
90
20
V
V
dB
dB
+12
V
+10
35
V
mA
mA
4000
pF
1.0
50
3
4.5
5
MHz
kHz
V/µs
µs
µs
2
µs
REV. A
AD549
Model
Min
POWER SUPPLY
Rated Performance
Operating
Quiescent Current
TEMPERATURE RANGE
Operating, Rated Performance
Storage
PACKAGE OPTION
TO-99 (H-08A)
Chips
AD549J
Typ
Max
Min
618
0.70
65
+70
+150
0
–65
± 15
65
0.60
0
–65
AD549K
Typ
Max
Min
618
0.70
65
+70
+150
0
–65
± 15
AD549JH
AD549JChips
AD549L
Typ
Max
Min
618
0.70
65
+70
+150
–55
–65
± 15
0.60
AD549KH
0.60
AD549S
Typ
Max
Units
618
0.70
V
V
mA
+125
+150
°C
°C
± 15
AD549LH
0.60
AD549SH, AD549SH/883B
NOTES
1
Bias current specifications are guaranteed after 5 minutes of operation at T A = +25°C. Bias current increases by a factor of 2.3 for every 10°C rise in temperature.
2
Input offset voltage specifications are guaranteed after 5 minutes of operation at T A = +25°C.
3
Defined as max continuous voltage between the inputs such that neither input exceeds ± 10 V from ground.
Specifications subject to change without notice.
All min and max specifications are guaranteed. Specifications in boldface are tested on all production units at final electrical test. Results from those tests are used to
calculate outgoing quality levels.
ABSOLUTE MAXIMUM RATINGS 1
METALIZATION PHOTOGRAPH
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Internal Power Dissipation . . . . . . . . . . . . . . . . . . . . . . 500 mW
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V2
Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and –VS
Storage Temperature Range (H) . . . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD549J (K, L) . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
AD549S . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C
Dimensions shown in inches and (mm).
Contact factory for latest dimensions.
NOTES
1
Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in the
operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
For supply voltages less than ± 18 V, the absolute maximum input voltage is equal
to the supply voltage.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD549 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. A
–3–
WARNING!
ESD SENSITIVE DEVICE
AD549–Typical Characteristics
+VIN
10
–VIN
5
0
+25°C
RL = 10k
5
10
15
SUPPLY VOLTAGE ± V
–VOUT
10
5
5
10
15
SUPPLY VOLTAGE ± V
500
110
100
90
20
3000
1000
∆ |VOS| – µV
–25
5
35
65
TEMPERATURE – °C
95
Figure 7. Open-Loop Gain vs.
Temperature
125
300
0
30
50
25
45
15
5
10
15
SUPPLY VOLTAGE ± V
20
40
35
30
25
5
–55
100k
Figure 6. Open-Loop Gain vs.
Supply Voltage
10
100
100
1k
10k
LOAD RESISTANCE – Ω
1000
–10
0
+10
+15
INPUT COMMON-MODE VOLTAGE – V
20
300
10
100
–15
Figure 5. CMRR vs. Input
Common-Mode Voltage
Figure 4. Quiescent Current
vs. Supply Voltage
5
80
INPUT CURRENT – fA
5
10
15
SUPPLY VOLTAGE ± V
10
3000
70
0
15
Figure 3. Output Voltage
Swing vs. Load Resistance
OPEN-LOOP GAIN – V/mV
600
VS = ±15 VOLTS
20
20
120
COMMON-MODE REJECTION RATIO – dB
AMPLIFIER QUIESCENT CURRENT – µA
0
Figure 2. Output Voltage
Swing vs. Supply Voltage
700
25
0
0
20
800
OPEN-LOOP GAIN – V/mV
+VOUT
15
Figure 1. Input Voltage Range
vs. Supply Voltage
400
OUTPUT VOLTAGE SWING – Volts p-p
15
0
30
20
OUTPUT VOLTAGE SWING ± V
INPUT VOLTAGE ± V
20
20
0
0
1
2
3
4
5
WARM-UP TIME – Minutes
Figure 8. Change in Offset
Voltage vs. Warm-Up Time
–4–
6
7
–10
–5
0
5
10
COMMON-MODE VOLTAGE ± V
Figure 9. Input Bias Current
vs. Common-Mode Voltage
REV. A
AD549
50
40
35
30
25
120
100
80
60
40
0
5
10
15
POWER SUPPLY VOLTAGE ± V
10
20
80
80
35
60
60
40
40
20
20
0
PHASE MARGIN – °
40
OUTPUT VOLTAGE SWING – V
100
–20
–40
10
100
1k
10k
100k
FREQUENCY – Hz
1M
100
1k
10
10Hz
BANDWIDTH
1
AMPLIFIER GENERATED NOISE
100k
1M
10M
100M
1G
10G 100G
SOURCE RESISTANCE – Ω
Figure 12. Noise vs. Source
Resistance
–40
10M
100
80
30
60
25
20
15
40
20
10
0
5
–20
0
10
Figure 13. Open-Loop
Frequency Response
100
1k
10k
FREQUENCY – Hz
100k
1M
Figure 14. Large Signal
Frequency Response
10
100
1k
10k
100k
FREQUENCY – Hz
OUTPUT VOLTAGE SWING – V
100
80
+ SUPPLY
60
40
– SUPPLY
20
10mV
5
5mV
1mV
0
10mV
5mV
–5
1mV
0
–20
–10
10
100
1k
10k
100k
FREQUENCY – Hz
1M
10M
Figure 16. PSRR vs. Frequency
REV. A
0
1
2
3
SETTLING TIME – µs
4
Figure 17. Output Voltage
Swing and Error vs.
Settling Time
–5–
1M
10M
Figure 15. CMRR vs. Frequency
10
120
PSRR – dB
100
10k
Figure 11. Input Voltage Noise
Spectral Density
100
–20
1kHz BANDWIDTH
RESISTOR
JOHNSON NOISE
1k
FREQUENCY – Hz
Figure 10. Input Bias Current
vs. Supply Voltage
0
10k
WHENEVER JOHNSON NOISE IS GREATER THAN
AMPLIFIER NOISE, AMPLIFIER NOISE CAN BE
CONSIDERED NEGLIGIBLE FOR THE APPLICATION
0.1
20
20
OPEN LOOP GAIN – dB
140
CMRR – dB
INPUT CURRENT – fA
45
100k
INPUT NOISE VOLTAGE – µV p-p
NOISE SPECTRAL DENSITY – nV/√Hz
160
5
AD549
Figure 18. Unity Gain
Follower
Figure 19. Unity Gain Follower
Large Signal Pulse Response
Figure 20. Unity Gain Follower
Small Signal Pulse Response
Figure 21. Unity Gain Inverter
Figure 22. Unity Gain Inverter
Large Signal Pulse Response
Figure 23. Unity Gain Inverter
Small Signal Pulse Response
The AD549 has been optimized for low input current and offset
voltage. Careful attention to how the amplifier is used will reduce
input currents in actual applications.
The amplifier operating temperature should be kept as low as possible to minimize input current. Like other JFET input amplifiers,
the AD549’s input current is sensitive to chip temperature, rising
by a factor of 2.3 for every 10°C rise. This is illustrated in Figure
24, a plot of AD549 input current versus ambient temperature.
1nA
100pA
10pA
1pA
However, heavy output loads can cause a significant increase
in chip temperature and a corresponding increase in input
current. Maintaining a minimum load resistance of 10 Ω is recommended. Input current versus additional power dissipation
due to output drive current is plotted in Figure 25.
6.0
NORMALIZED INPUT BIAS CURRENT
MINIMIZING INPUT CURRENT
5.0
BASED ON
TYPICAL IB = 40fA
4.0
3.0
2.0
1.0
100fA
0
25
50
75 100 125 150 175 200
ADDITIONAL INTERNAL POWER DISSIPATION – mW
10fA
Figure 25. AD549 Input Bias Current vs.
Additional Power Dissipation
1fA
–55
–25
5
35
65
TEMPERATURE – °C
95
125
CIRCUIT BOARD NOTES
There are a number of physical phenomena that generate
spurious currents that degrade the accuracy of low current
measurements. Figure 26 is a schematic of an I-to-V converter
with these parasitic currents modeled.
Finite resistance from input lines to voltages on the board,
modeled by resistor RP, results in parasitic leakage. Insulation
resistance of over 1015 Ω must be maintained between the
amplifier’s signal and supply lines in order to capitalize on the
AD549’s low input currents. Standard PC board material
Figure 24. AD549 Input Bias Current vs.
Ambient Temperature
On-chip power dissipation will raise chip operating temperature
causing an increase in input bias current. Due to the AD549’s
low quiescent supply current, chip temperature when the (unloaded) amplifier is operated with 15 V supplies, is less than 3°C
higher than ambient. The difference in input current is negligible.
–6–
REV. A
AD549
does not have high enough insulation resistance. Therefore, the
AD549’s input leads should be connected to standoffs made of
insulating material with adequate volume resistivity (e.g.,
Teflon*). The surface of the insulator’s surface must be kept
clean in order to preserve surface resistivity. For Teflon, an effective cleaning procedure consists of swabbing the surface with
high-grade isopropyl alcohol, rinsing with deionized water, and
baking the board at 80°C for 10 minutes.
mized. Input capacitance can substantially degrade signal bandwidth and the stability of the I-to-V converter. The case of the
AD549 is connected to Pin 8 so that it can be bootstrapped
near the input potential. This minimizes pin leakage and input
common-mode capacitance due to the case. Guard schemes for
inverting and noninverting amplifier topologies are illustrated in
Figures 28 and 29.
Figure 28. Inverting Amplifier with Guard
Figure 26. Sources of Parasitic Leakage Currents
In addition to high volume and surface resistivity, other properties are desirable in the insulating material chosen. Resistance to
water absorption is important since surface water films drastically reduce surface resistivity. The insulator chosen should also
exhibit minimal piezoelectric effects (charge emission due to
mechanical stress) and triboelectric effects (charge generated by
friction). Charge imbalances generated by these mechanisms can
appear as parasitic leakage currents. These effects are modeled
by variable capacitor CP in Figure 26. The table in Figure 27
lists various insulators and their properties.1
Material
Volume
Resistivity
(V–CM)
Teflon*
Kel-F**
Sapphire
Polyethylene
Polystyrene
Ceramic
Glass Epoxy
PVC
Phenolic
1017–1018
1017–1018
1016–1018
1014–1018
1012–1018
1012–1014
1010–1017
1010–1015
105–1012
Minimal
Triboelectric
Effects
W
W
M
M
W
W
W
G
W
Minimal
Resistance
Piezoelectric to Water
Effects
Absorption
W
M
G
G
M
M
M
M
G
G
G
G
M
M
W
W
G
W
Figure 29. Noninverting Amplifier with Guard
Other guidelines include keeping the circuit layout as compact
as possible and input lines short. Keeping the assembly rigid
and minimizing sources of vibration will reduce triboelectric and
piezoelectric effects. All precision high impedance circuitry requires shielding against interference noise. Low noise coax or
triax cables should be used for remote connections to the input
signal lines.
OFFSET NULLING
The AD549’s input offset voltage can be nulled by using balance
Pins 1 and 5, as shown in Figure 30. Nulling the input offset
voltage in this fashion will introduce an added input offset voltage drift component of 2.4 µV/°C per millivolt of nulled offset
(a maximum additional drift of 0.6 µV/°C for the AD549K,
1.2 µV/°C for the AD549L, 2.4 µV/°C for the AD549J).
G–Good with Regard to Property
M–Moderate with Regard to Property
W–Weak with Regard to Property
Figure 27. Insulating Materials and Characteristics
Guarding the input lines by completely surrounding them with a
metal conductor biased near the input lines’ potential has two
major benefits. First, parasitic leakage from the signal line is
reduced since the voltage between the input line and the guard
is very low. Second, stray capacitance at the input node is miniFigure 30. Standard Offset Null Circuit
1
Electronic Measurements, pp. 15–17, Keithley Instruments, Inc., Cleveland,
Ohio, 1977.
*Teflon is a registered trademark of E.I. DuPont Co.
**Kel-F is a registered trademark of 3-M Company.
REV. A
The approach in Figure 31 can be used when the amplifier is
used as an inverter. This method introduces a small voltage
referenced to the power supplies in series with the amplifier’s
–7–
AD549
positive input terminal. The amplifier’s input offset voltage drift
with temperature is not affected. However, variation of the
power supply voltages will cause offset shifts.
In an inverting configuration, the differential input capacitance
forms a pole in the circuit’s loop transmission. This can create
peaking in the ac response and possible instability. A feedback
capacitance can be used to stabilize the circuit. The inverter
pulse response with RF and RS equal to 1 MΩ appears in Figure
34. Figure 35 shows the response of the same circuit with a I pF
feedback capacitance. Typical differential input capacitance for
the AD549 is 1 pF.
COMMON-MODE INPUT VOLTAGE OVERLOAD
Figure 31. Alternate Offset Null Circuit for Inverter
AC RESPONSE WITH HIGH VALUE SOURCE AND
FEEDBACK RESISTANCE
Source and feedback resistances greater than 100 kΩ will magnify the effect of input capacitances (stray and inherent to the
AD549) on the ac behavior of the circuit. The effects of
common-mode and differential input capacitances should be
taken into account since the circuit’s bandwidth and stability
can be adversely affected.
The rated common-mode input voltage range of the AD549 is
from 3 V less than the positive supply voltage to 5 V greater
than the negative supply voltage. Exceeding this range will degrade the amplifier’s CMRR. Driving the common-mode voltage
above the positive supply will cause the amplifier’s output to
saturate at the upper limit of output voltage. Recovery time is
typically 2 µs after the input has been returned to within the normal operating range. Driving the input common-mode voltage
within 1 V of the negative supply causes phase reversal of the
output signal. In this case, normal operation is typically resumed
within 0.5 µs of the input voltage returning within range.
Figure 34. Inverter Pulse Response with 1 MΩ Source and
Feedback Resistance
Figure 32. Follower Pulse Response from 1 MΩ Source
Resistance, Case Not Bootstrapped
Figure 35. Inverter Pulse Response with 1 MΩ Source and
Feedback Resistance, 1 pF Feedback Capacitance
DIFFERENTIAL INPUT VOLTAGE OVERLOAD
Figure 33. Follower Pulse Response from 1 MΩ Source
Resistance, Case Bootstrapped
In a follower, the source resistance and input common-mode
capacitance form a pole that limits the bandwidth to 1/2 π RSCS.
Bootstrapping the metal case by connecting Pin 8 to the output
minimizes capacitance due to the package. Figures 32 and 33
show the follower pulse response from a 1 MΩ source resistance
with and without the package connected to the output. Typical
common-mode input capacitance for the AD549 is 0.8 pF.
A plot of the AD549’s input currents versus differential input
voltage (defined as VIN+ –VIN–) appears in Figure 36. The input
current at either terminal stays below a few hundred femtoamps
until one input terminal is forced higher than 1 V to 1.5 V above
the other terminal. Under these conditions, the input current
limits at 30 µA.
–8–
REV. A
AD549
100µ
IIN–
10µ
IIN+
INPUT CURRENT – Amps
1µ
100n
10n
1n
Figure 39. Input Voltage Clamp with Diodes
100p
10p
SAMPLE AND DIFFERENCE CIRCUIT TO MEASURE
ELECTROMETER LEAKAGE CURRENTS
1p
100f
10f
–5 –4
–3
–2
–1
0
1
2
3
4
5
DIFFERENTIAL INPUT VOLTAGE – V (VIN– – VIN–)
Figure 36. Input Current vs. Differential Input Voltage
INPUT PROTECTION
The AD549 safely handles any input voltage within the supply
voltage range. Subjecting the input terminals to voltages beyond
the power supply can destroy the device or cause shifts in input
current or offset voltage if the amplifier is not protected.
A protection scheme for the amplifier as an inverter is shown in
Figure 37. RP is chosen to limit the current through the inverting input to 1 mA for expected transient (less than 1 second)
overvoltage conditions, or to 100 µA for a continuous overload.
Since RP is inside the feedback loop, and is much lower in value
than the amplifier’s input resistance, it does not affect the
inverter’s dc gain. However, the Johnson noise of the resistor
will add root sum of squares to the amplifier’s input noise.
There are a number of methods used to test electrometer leakage currents, including current integration and direct current to
voltage conversion. Regardless of the method used, board and
interconnect cleanliness, proper choice of insulating materials
(such as Teflon or Kel-F), correct guarding and shielding techniques and care in physi-cal layout are essential to making accurate leakage measurements.
Figure 40 is a schematic of the sample and difference circuit. It
uses two AD549 electrometer amplifiers (A and B) as current-to
voltage converters with high value (1010 Ω) sense resistors (RSa
and RSb). R1 and R2 provide for an overall circuit sensitivity of
10 fA/mV (10 pA full scale). CC and CF provide noise suppression and loop compensation. CC should be a low leakage polystyrene capacitor. An ultralow leakage Kel-F test socket is used
for contacting the device under test. Rigid Teflon coaxial cable
is used to make connections to all high impedance nodes. The
Figure 37. Inverter with Input Current Limit
In the corresponding version of this scheme for a follower,
shown in Figure 38, RP and the capacitance at the positive input
terminal will produce a pole in the signal frequency response at
a f = 1/2 π RC. Again, the Johnson noise RP will add to the
amplifier’s input voltage noise.
Figure 38. Follower with Input Current Limit
Figure 39 is a schematic of the AD549 as an inverter with an
input voltage clamp. Bootstrapping the clamp diodes at the inverting input minimizes the voltage across the clamps and keeps
the leakage due to the diodes low. Low leakage diodes, such as
the FD333’s should be used, and should be shielded from light
to keep photocurrents from being generated. Even with these
precautions, the diodes will measurably increase the input current and capacitance.
REV. A
Figure 40. Sample and Difference Circuit for Measuring
Electrometer Leakage Currents
–9–
AD549
use of rigid coax affords immunity to error induced by mechanical vibration and provides an outer conductor for shielding. The
entire circuit is enclosed in a grounded metal box.
The test apparatus is calibrated without a device under test
present. A five minute stabilization period after the power is
turned on is required. First, VERR1 and VERR2 are measured.
These voltages are the errors caused by offset voltages and leakage currents of the current to voltage converters.
VERR1 = 10 (VOSA – IBA × RSa)
VERR2 = 10 (VOSB – IBB × RSb)
Figure 42. Photodiode Preamp
DC Error Sources
Once measured, these errors are subtracted from the readings
taken with a device under test present. Amplifier B closes the
feedback loop to the device under test, in addition to providing
current to voltage conversion. The offset error of the device under test appears as a common-mode signal and does not affect
the test measurement. As a result, only the leakage current of
the device under test is measured.
Input current, IB, will contribute an output voltage error, VE1,
proportional to the feedback resistance:
VE1 = IB × RF
The op amp’s input voltage offset will cause an error current
through the photodiode’s shunt resistance, RS:
VA – VERR1 = 10[RSa × IB(+)]
I = VOS/RS
VX – VERR2 = 10[RSb × IB(–)]
Although a series of devices can be tested after only one calibration measurement, calibration should be updated periodically to
compensate for any thermal drift of the current to voltage converters or changes in the ambient environment. Laboratory results have shown that repeatable measurements within 10 fA can
be realized when this apparatus is properly implemented. These
results are achieved in part by the design of the circuit, which
eliminates relays and other parasitic leakage paths in the high
impedance signal lines, and in part by the inherent cancellation
of errors through the calibration and measurement procedure.
PHOTODIODE INTERFACE
The AD549’s low input current and low input offset voltage
make it an excellent choice for very sensitive photodiode
preamps (Figure 41). The photodiode develops a signal current,
IS equal to:
IS = R × P
where P is light power incident on the diode’s surface in Watts
and R is the photodiode responsivity in Amps/Watt. RF converts
the signal current to an output voltage:
VOUT = RF × IS
Figure 41. Photodiode Preamp
DC error sources and an equivalent circuit for a small area
(0.2 mm square) photodiode are indicated in Figure 42.
The error current will result in an error voltage (VE2) at the
amplifier’s output equal to:
VE2 = ( I + RF/RS) VOS
Given typical values of photodiode shunt resistance (on the
order of 109 Ω), RF/RS can easily be greater than one, especially
if a large feedback resistance is used. Also, RF/RS will increase
with temperature, as photodiode shunt resistance typically drops
by a factor of two for every 10°C rise in temperature. An op
amp with low offset voltage and low drift must be used in order to
maintain accuracy. The AD549K offers guaranteed maximum 0.25
mV offset voltage, and 5 mV/°C drift for very sensitive applications.
Photodiode Preamp Noise
Noise limits the signal resolution obtainable with the preamp.
The output voltage noise divided by the feedback resistance is
the minimum current signal that can be detected. This minimum detectable current divided by the responsivity of the photodiode represents the lowest light power that can be detected
by the preamp.
Noise sources associated with the photodiode, amplifier, and
feedback resistance are shown in Figure 43; Figure 44 is the
spectral density versus frequency plot of each of the noise
source’s contribution to the output voltage noise (circuit parameters in Figure 42 are assumed). Each noise source’s rms contribution to the total output voltage noise is obtained by integrating
the square of its spectral density function over frequency. The rms
value of the output voltage noise is the square root of the sum of all
contributions. Minimizing the total area under these curves will optimize the preamplifier’s resolution for a given bandwidth.
The photodiode preamp in Figure 41 can detect a signal current
of 26 fA rms at a bandwidth of 16 Hz, which assuming a photodiode responsivity of 0.5 A/W, translates to a 52 fW rms minimum detectable power. The photodiode used has a high source
resistance and low junction capacitance. CF sets the signal bandwidth with RF and also limits the “peak” in the noise gain that
multiplies the op amp’s input voltage noise contribution. A
single pole filter at the amplifier’s output limits the op amp’s output voltage noise bandwidth to 26 Hz, a frequency comparable to
the signal bandwidth. This greatly improves the preamplifier’s
signal to noise ratio (in this case, by a factor of three).
–10–
REV. A
AD549
tracter section’s gain for positive and negative inputs matched
over temperature.
Frequency compensation is provided by R11, R12, and C1 and
C2. The bandwidth of the circuit is 300 kHz at input signals
greater than 50 µA, and decreases smoothly with decreasing
signal levels.
To trim the circuit, set the input currents to 10 µA and trim
A3’s offset using the amplifier’s trim potentiometer so the output equals 0. Then set I1 to 1 µA and adjust the output to equal
1 V by trimming R10. Additional offset trims on the amplifiers
A1 and A2 can be used to increase the voltage input accuracy
and dynamic range.
Figure 43. Photodiode Preamp Noise Sources
The very low input current of the AD549 makes this circuit useful over a very wide range of signal currents. The total input
current (which determines the low level accuracy of the circuit)
is the sum of the amplifier input current, the leakage across the
compensating capacitor (negligible if polystyrene or Teflon capacitor is used), and the collector to collector, and collector to
base leakages of one side of the dual log transistors. The magnitude of these last two leakages depend on the amplifier’s input
offset voltage and are typically less than 10 fA with 1 mV offsets.
The low level accuracy is limited primarily by the amplifier’s input current, only 60 fA maximum when the AD549L is used.
Figure 44. Photodiode Preamp Noise Sources’ Spectral
Density vs. Frequency
Log Ratio Amplifier
Logarithmic ratio circuits are useful for processing signals with
wide dynamic range. The AD549L’s 60 fA maximum input current makes it possible to build a log ratio amplifier with 1% log
conformance for input current ranging from 10 pA to 1 mA, a
dynamic range of 160 dB.
The log ratio amplifier in Figure 45 provides an output voltage
proportional to the log base 10 of the ratio of the input currents
I1 and I2. Resistors R1 and R2 are provided for voltage inputs.
Since NPN devices are used in the feedback loop of the frontend amplifiers that provide the log transfer function, the output
is valid only for positive input voltages and input currents. The
input currents set the collector currents IC1 and IC2 of a
matched pair of log transistors Q1 and Q2 to develop voltages
VA and VB:
VA, B = – (kT/q) ln IC/IES
where IES is the transistors’ saturation current.
The difference of VA and VB is taken by the subtractor section
to obtain:
Figure 45. Log Ratio Amplifier
VC = (kT/q) ln (IC2/IC1)
VC is scaled up by the ratio of (R9 + R10)/R8, which is equal to
approximately 16 at room temperature, resulting in the output
voltage:
VOUT = 1 × log (IC2/IC1) V.
R8 is a resistor with a positive 3500 ppm/°C temperature coefficient to provide the necessary temperature compensation. The
parallel combination of R15 and R7 is provided to keep the sub
REV. A
The effects of the emitter resistance of Q1 and Q2 can degrade
the circuit’s accuracy at input currents above 100 µA. The networks composed of R13, D1, R16, and R14, D2, R17 compensate for these errors, so that this circuit has less than 1% log
conformance error at 1 mA input currents. The correct value
for R13 and R14 depends on the type of log transistors used.
49.9 kΩ resistors were chosen for use with LM394 transistors.
Smaller resistance values will be needed for smaller log
transistors.
–11–
The pH probe output is ideally zero volts at a pH of 7 independent of temperature. The slope of the probe’s transfer function,
though predictable, is temperature dependent (–54.2 mV/pH at
0 and –74.04 mV/pH at 100°C). By using an AD590 temperature sensor and an AD535 analog divider, an accurate temperature compensation network can be added to the basic pH probe
amplifier. The table in Figure 47 shows voltages at various points
and illustrates the compensation. The AD549 is set for a noninverting gain of 13.51. The output of the AD590 circuitry (point
C) will be equal to 10 V at 100°C and decrease by 26.8 mV/°C.
The output of the AD535 analog divider (point D) will be a
temperature compensated output voltage centered at zero volts
for a pH of 7, and having a transfer function of –1.00 V/pH
unit. The output range spans from –7.00 V (pH = 14) to +7.00 V
(pH = 0).
TEMPERATURE COMPENSATED pH PROBE
AMPLIFIER
A pH probe can be modeled as a mV-level voltage source with a
series source resistance dependent upon the electrode’s composition and configuration. The glass bulb resistance of a typical
pH electrode pair falls between 106 and 109 Ω. It is therefore
important to select an amplifier with low enough input currents
such that the voltage drop produced by the amplifier’s input
bias current and the electrode resistance does not become an
appreciable percentage of a pH unit.
The circuit in Figure 46 illustrates the use of the AD549 as a
pH probe amplifier. As with other electrometer applications, the
use of guarding, shielding, Teflon standoffs, etc., is a must in
order to capitalize on the AD549’s low input current. If an
AD549L (60 fA max input current) is used, the error contributed by input current will be held below 60 µV for pH electrode
source impedances up to 109 Ω. Input offset voltage (which can
be trimmed) will be below 0.5 mV.
PROBE
TEMP
A
(PROBE OUTPUT)
B
(A 3 13.51)
C
(590 OUTPUT)
D
(10 B/C)
0
258C
378C
608C
1008C
54.20 mV
59.16 mV
61.54 mV
66.10 mV
74.04 mV
0.732 V
0.799 V
0.831 V
0.893 V
1.000 V
7.32 V
7.99 V
8.31 V
8.93 V
10.00 V
1.00 V
1.00 V
1.00 V
1.00 V
1.00 V
C1073a–10–10/87
AD549
Figure 47. Table Illustrating Temperature Compensation
Figure 46. Temperature Compensated pH Probe Amplifier
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
0.370 (9.40)
0.335 (8.50)
0.335 (8.50)
0.305 (7.75)
0.040
(1.0)
MAX
SEATING
PLANE
INSULATION
0.05 (1.27) MAX
8 LEADS
0.2
(5.1)
TYP
3
0.185 (4.70)
0.165 (4.19)
REFFERENCE
PLANE
2
4
8
6
1
PRINTED IN U.S.A.
TO-99 (H) Package
45° EQUALLY
SPACED
5
7
0.500
(12.70)
MIN
0.019 (0.48)
DIA
0.016 (0.41)
0.034 (0.86)
0.028 (0.41)
0.045 (1.1)
0.020 (0.51)
BOTTOM VIEW
–12–
REV. A