Burr-Brown OPA860 Wide bandwidth operational transconductance amplifier (ota) and buffer Datasheet

 OPA860
SBOS331 – JUNE 2005
Wide Bandwidth
OPERATIONAL TRANSCONDUCTANCE
AMPLIFIER (OTA) and BUFFER
OTA FEATURES
DESCRIPTION
•
•
•
•
The OPA860 is a versatile monolithic component
designed for wide-bandwidth systems, including high
performance video, RF and IF circuitry. It includes a
wideband, bipolar operational transconductance
amplifier (OTA), and voltage buffer amplifier.
Wide Bandwidth (80MHz, Open-Loop, G = +5)
High Slew Rate (900V/µs)
High Transconductance (95mA/V)
External IQ-Control
BUFFER FEATURES
•
•
•
•
Closed-Loop Buffer
Wide Bandwidth (1600MHz, VO = 1VPP)
High Slew Rate (4000V/µs)
60mA Output Current
OPA860 FEATURES
•
•
Low Quiescent Current (11.2mA)
Versatile Circuit Function
APPLICATIONS
•
•
•
•
•
•
•
•
•
Baseline Restore Circuits
Video/Broadcast Equipment
Communications Equipment
High-Speed Data Acquisition
Wideband LED Driver
AGC-Multiplier
ns-Pulse Integrator
Control Loop Amplifier
OPA660 Upgrade
The OTA or voltage-controlled current source can be
viewed as an ideal transistor. Like a transistor, it has
three terminals—a high impedance input (base), a
low-impedance input/output (emitter), and the current
output (collector). The OTA, however, is self-biased
and bipolar. The output collector current is zero for a
zero base-emitter voltage. AC inputs centered about
zero produce an output current, which is bipolar and
centered about zero. The transconductance of the
OTA can be adjusted with an external resistor,
allowing bandwidth, quiescent current, and gain
trade-offs to be optimized.
Also included in the OPA860 is an uncommited
closed-loop, unity-gain buffer. This provides
1600MHz bandwidth and 4000V/µs slew rate.
Used as a basic building block, the OPA860 simplifies the design of AGC amplifiers, LED driver
circuits for fiber optic transmission, integrators for fast
pulses, fast control loop amplifiers and control amplifiers for capacitive sensors and active filters. The
OPA860 is available in an SO-8 surface-mount package.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005, Texas Instruments Incorporated
OPA860
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SBOS331 – JUNE 2005
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated
circuits be handled with appropriate precautions. Failure to observe proper handling and installation
procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision
integrated circuits may be more susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
(1)
PRODUCT
PACKAGE
PACKAGE
DESIGNATOR
OPA860
SO-8
D
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
–45°C to +85°C
OPA860
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
OPA860ID
Rails, 75
OPA860IDR
Tape and Reel, 2500
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
±6.5VDC
Power Supply
Internal Power Dissipation
See Thermal Information
±1.2V
Differential Input Voltage
±VS
Input Common-Mode Voltage Range
Storage Temperature Range: D
–40°C to +125°C
Lead Temperature (soldering, 10s)
+300°C
Junction Temperature (TJ)
+150°C
ESD Rating:
(1)
(2)
Human Body Model (HBM) (2)
1500V
Charge Device Model (CDM)
1000V
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress Ratings only, and functional operations of the device at these and any other conditions
beyond those specified is not supported.
Pin 2 > 500V HBM.
PIN CONFIGURATION
Top View
2
SO
IQ Adjust
1
8
C
E
2
7
V+ = +5V
B
3
6
Out
V− = −5V
4
5
In
+1
OPA860
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SBOS331 – JUNE 2005
ELECTRICAL CHARACTERISTICS: VS = ±5V
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA860ID
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
VO = 200mVPP
470
380
375
370
MHz
min
B
VO = 1VPP
470
MHz
typ
C
CONDITIONS
TEST
LEVEL (1)
Closed Loop OTA + BUFFER (see Figure 53)
AC PERFORMANCE
Bandwidth
G = +2, See Figure 53
VO = 5VPP
350
MHz
typ
C
Bandwidth for 0.1dB Gain Flatness
VO = 200mVPP
42
MHz
typ
C
Slew Rate
VO = 5V Step
3500
V/µs
typ
C
Rise Time and Fall Time
VO = 1V Step
0.7
ns
typ
C
RL = 100Ω
–54
dBc
typ
C
RL = 500Ω
–77
dBc
typ
C
RL = 100Ω
–66
dBc
typ
C
RL = 500Ω
–79
dBc
typ
C
G = +5, VO = 200mVPP,
RL = 500Ω
80
MHz
min
B
G = +5, VO = 1VPP
80
MHz
typ
C
G = +5, VO = 5VPP
80
MHz
typ
C
G = +5, VO = 5V Step
900
V/µs
min
B
VO = 1V Step
4.4
ns
typ
C
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
3000
2800
2700
G = +2, VO = 2VPP, 5MHz
OTA - Open-Loop (see Figure 48)
AC PERFORMANCE
Bandwidth
Slew Rate
Rise Time and Fall Time
Harmonic Distortion
77
860
75
850
74
840
G = +5, VO = 2VPP, 5MHz
2nd-Harmonic
RL = 500Ω
–68
–55
–54
–53
dB
max
B
3rd-Harmonic
RL = 500Ω
–57
–52
–51
–49
dB
max
B
Base Input Voltage Noise
f > 100kHz
2.4
3.0
3.3
3.4
nV/√Hz
max
B
Base Input Current Noise
f > 100kHz
1.65
2.4
2.45
2.5
pA/√Hz
max
B
Emitter Input Current Noise
f > 100kHz
5.2
15.3
16.6
17.5
pA/√Hz
max
B
Min OTA Transconductance
VO = ±10mV, RC = 0Ω, RE = 0Ω
95
80
77
75
mA/V
min
A
Max OTA Transconductance
VO = ±10mV, RC = 0Ω, RE = 0Ω
95
150
155
160
mA/V
min
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±12
±15
±20
mV
max
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±67
±120
µV/°C
max
B
VB = 0V, RC = 0Ω, RE = 100Ω
±1
±6
±6.6
µA
max
A
±20
±25
nA/°C
max
B
±125
±140
µA
max
A
±500
±600
nA/°C
max
B
±30
±38
µA
max
A
±250
±300
nA/°C
max
B
±3.6
±3.6
V
min
B
kΩ || pF
typ
C
OTA DC PERFORMANCE (4) (see Figure 48)
B-Input Offset Voltage
Average B-Input Offset Voltage Drift
B-Input Bias Current
Average B-Input Bias Current Drift
E-Input Bias Current
Average E-Input Bias Current Drift
C-Output Bias Current
Average C-Output Bias Current Drift
±5
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
±30
±100
VB = 0V, VC = 0V
VB = 0V, VC = 0V
±5
±18
VB = 0V, VC = 0V
OTA INPUT (see Figure 48)
B-Input Voltage Range
B-Input Impedance
±4.2
±3.7
455 || 2.1
Min E-Input Input Resistance
10.5
12.5
13.0
13.3
Ω
min
B
Max E-Input Input Resistance
10.5
6.7
6.5
6.3
Ω
max
B
(1)
(2)
(3)
(4)
Test levels: (A) 100% tested at 25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for 25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient + 8°C at high temperature limit for over
temperature specifications.
Current is considered positive out of node. VCM is the input common-mode voltage.
3
OPA860
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SBOS331 – JUNE 2005
ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA860ID
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
IE = ±1mA
±4.2
±3.7
±3.6
VE = 0
±15
±10
±9
±3.6
V
min
A
±9
mA
min
IC = ±1mA
±4.7
±4.0
A
±3.9
±3.9
V
min
VC = 0
±15
±10
A
±9
±9
mA
min
A
kΩ || pF
typ
C
MHz
min
B
MHz
typ
C
MHz
typ
C
V/µs
min
B
OTA OUTPUT
E-Output Voltage Compliance
E-Output Current, Sinking/Sourcing
C-Output Voltage Compliance
C-Output Current, Sinking/Sourcing
C-Output Impedance
54 || 2
BUFFER (see Figure Figure 45)
AC PERFORMANCE
Bandwidth
VO = 200mVPP
1200
VO = 1VPP
1600
750
720
700
VO = 5VPP
1000
Slew Rate
VO = 5V Step
4000
Rise Time and Fall Time
VO = 1V Step
0.4
ns
typ
C
VO = 1V Step
6
ns
typ
C
Settling Time to 0.05%
Harmonic Distortion
2nd-Harmonic
3500
3200
3000
VO = 2VPP, 5MHz
RL = 100Ω
–52
–47
–46
–44
dBc
max
B
RL≥ 500Ω
–72
–65
–63
–61
dBc
max
B
RL = 100Ω
–67
–63
–63
–62
dBc
max
B
RL≥ 500Ω
–96
–86
–85
–83
dBc
max
B
Input Voltage Noise
f > 100kHz
4.8
5.1
5.6
6.0
nV/√Hz
max
B
Input Current Noise
f > 100kHz
2.1
2.6
2.7
2.8
pA/√Hz
max
B
Differential Gain
NTSC, PAL
0.06
%
typ
C
Differential Phase
NTSC, PAL
0.02
Degrees
typ
C
RL = 500Ω
1
0.98
0.98
0.98
V/V
min
A
RL = 500Ω
1
1
1
1
V/V
max
A
±16
±30
±36
±38
mV
max
A
±125
±125
µV/°C
max
B
±8
±8.5
µA
max
A
±20
±20
nA/°C
max
B
MΩ || pF
typ
C
A
3rd-Harmonic
BUFFER DC PERFORMANCE
Gain
Input Offset Voltage
Average Input Offset Voltage Drift
±3
Input Bias Current
±7
Average Input Bias Current Drift
BUFFER INPUT
Input Impedance
1.0 || 2.1
BUFFER OUTPUT
Output Voltage Swing
Output Current
Closed-Loop Output Impedance
RL = 500Ω
±4.0
±3.8
±3.8
±3.8
V
min
VO = 0
±60
±50
±49
±48
mA
min
A
f ≤ 100kHz
1.4
Ω
typ
C
POWER SUPPLY (OTA + BUFFER)
±5
Specified Operating Voltage
V
typ
C
Maximum Operating Voltage
±6.5
±6.5
±6.5
V
max
A
Minimum Operating Voltage
±2.5
±2.5
±2.5
V
min
B
Maximum Quiescent Current
RADJ = 250Ω
11.2
12
13.5
14.5
mA
max
A
Minimum Quiescent Current
RADJ = 250Ω
11.2
10.5
9.5
7.9
mA
min
A
OTA Power-Supply Rejection Ratio
(+PSRR)
∆IC/∆VS
±20
±50
±60
±65
µA/V
max
A
Buffer Power-Supply Rejection Ratio
(–PSRR)
∆VO/∆VS
54
48
46
45
dB
min
A
–40 to +85
°C
typ
C
125
°C/W
typ
C
THERMAL CHARACTERISTICS
Specification: ID
Thermal Resistance θJA
D
4
SO-8
Junction-to-Ambient
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted. (See Figure 53.)
OTA + BUF Performance
SMALL-SIGNAL FREQUENCY RESPONSE
9
LARGE-SIGNAL FREQUENCY RESPONSE
9
G = +2V/V
RL = 500Ω
6
6
3
0
Gain (dB)
3
Gain (dB)
G = +2V/V
RL = 500Ω
VOUT = 0.5VPP
−3
0
VOUT = 1VPP
−3
VOUT = 0.2VPP
VOUT = 2VPP
−6
−6
−9
−9
VOUT = 5VPP
1M
10M
100M
1G
2G
1M
1G
Figure 1.
Figure 2.
SMALL-SIGNAL FREQUENCY RESPONSE
vs QUIESCENT CURRENT
GAIN FLATNESS vs QUIESCENT CURRENT
2G
6.5
G = +2V/V
R L = 500Ω
VO = 0.2VPP
6.4
6
6.3
IQ = 12mA
6.2
Gain (dB)
3
Gain (dB)
100M
Frequency (Hz)
9
0
IQ = 8mA
−3
6.1
IQ = 11.2mA
6.0
5.9
IQ = 8mA
5.8
G = +2V/V
RL = 500Ω
VO = 0.2VPP
−6
−9
IQ = 9mA
5.7
IQ = 11.2mA
5.6
I Q = 12mA
IQ = 9mA
5.5
1M
10M
100M
1G
2G
1
Frequency (MHz)
Figure 3.
Figure 4.
100
1.0
Output Voltage (V)
1.5
50
0
−50
−150
G = +2V/V
VIN = 0.125VPP
fIN = 20MHz
0.5
0
−0.5
−1.0
−1.5
Time (5ns/div)
Figure 5.
100
LARGE-SIGNAL PULSE RESPONSE
150
−100
10
Frequency (Hz)
SMALL-SIGNAL PULSE RESPONSE
Output Voltage (mV)
10M
Frequency (Hz)
G = +2V/V
VIN = 1.25VPP
fIN = 20MHz
Time (5ns/div)
Figure 6.
5
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted. (See Figure 53.)
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs OUTPUT RESISTANCE
−55
−50
G = +2V/V
R L = 500Ω
VO = 2VPP
See Figure 53
−65
−55
−70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
2nd−Harmonic
−75
−80
3rd−Harmonic
−85
−65
3rd−Harmonic
−70
−75
G = +2V/V
VO = 2VPP
f = 5MHz
See Figure 53
−80
−90
0.1
1
10
−85
100
20
1k
Frequency (MHz)
Output Resistance (Ω)
Figure 7.
Figure 8.
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs SUPPLY VOLTAGE
−65
−60
−75
Harmonic Distortion (dBc)
G = +2V/V
RL = 500Ω
f = 5MHz
See Figure 53
−70
Harmonic Distortion (dBc)
2nd−Harmonic
−60
−80
2nd−Harmonic
−85
3rd−Harmonic
−90
−95
−100
G = +2V/V
RL = 500Ω
VO = 2VPP
f = 5MHz
See Figure 53
−65
−70
3rd−Harmonic
−75
−80
2nd−Harmonic
−85
−90
0.1
1
10
2.0
2.5
3.0
3.5
Figure 9.
−55
12
−60
2nd−Harmonic
−70
−80
−85
−90
G = +2V/V
RL = 500Ω
VO = 2VPP
f = 5MHz
See Figure 53
5.5
6.0
3rd−Harmonic
11
10
9
8
+IQ
7
6
−IQ
5
8.0
8.5
9.0
9.5
10.0
10.5
IQ (mA)
Figure 11.
6
5.0
QUIESCENT CURRENT vs RADJ
13
Quiescent Current (mA)
Harmonic Distortion (dBc)
HARMONIC DISTORTION vs QUIESCENT CURRENT
−75
4.5
Figure 10.
−50
−65
4.0
±Supply Voltage (±VS)
Output Voltage (VPP)
11.0
11.5
12.0
0.1
1
10
100
RADJ (Ω)
Figure 12.
1k
10k
100k
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
OTA Performance
OTA TRANSCONDUCTANCE vs FREQUENCY
1000
OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT
150
IQ = 12.5mA (117mA/V)
VIN = 100mVPP
Transconductance (mA/V)
Transconductance (mA/V)
IQ = 11.2mA (102mA/V)
IQ = 9mA (79mA/V)
100
IQ = 7.5mA (51mA/V)
IO U T
VIN
50Ω
120
90
IOUT
60
VIN
50Ω
RL = 50Ω
VIN = 10mVPP
50Ω
10
0
1M
10M
100M
1G
6
7
Frequency (Hz)
140
6
IQ = 12mA
100
IQ = 9mA
80
60
IQ = 7mA
40
Small signal around input voltage.
−40
−30
−20
−10
0
20
30
IQ = 11.2mA
2
IQ = 9mA
0
I Q = 7mA
−2
IOUT
VIN
−4
50Ω
50Ω
−70 −60 −50 −40 −30 −20 −10
40
10
20
30
OTA Input Voltage (mV)
Figure 15.
Figure 16.
40
50
60
70
OTA LARGE-SIGNAL PULSE RESPONSE
3
0.2
0
G = +5V/V
RL = 500Ω
VIN = 0.25VPP
fIN = 20MHz
See Figure 48
Time (10ns/div)
Figure 17.
Output Voltage (V)
2
0.4
−0.8
0
Input Voltage (mV)
0.6
−0.6
13
4
OTA SMALL-SIGNAL PULSE RESPONSE
−0.4
12
−8
10
0.8
−0.2
11
I Q = 12mA
−6
20
Output Voltage (V)
10
OTA TRANSFER CHARACTERISTICS
8
OTA Output Current (mA)
Transconductance (mA/V)
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
IQ = 11.2mA
9
Figure 14.
160
120
8
Quiescent Current (mA)
Figure 13.
0
50Ω
30
1
0
−1
−2
−3
G = +5V/V
RL = 500Ω
VIN = 1VPP
fIN = 20MHz
See Figure 48
Time (10ns/div)
Figure 18.
7
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
490
110
OTA C−Output Resistance (kΩ )
OTA B−Input Resistance (kΩ )
B-INPUT RESISTANCE vs QUIESCENT CURRENT
500
480
470
460
450
440
100
430
80
70
60
50
40
7
8
9
10
11
12
13
7
8
9
11
Quiescent Current (mA)
Figure 19.
Figure 20.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
12
13
INPUT VOLTAGE AND CURRENT NOISE DENSITY
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
100
50
40
30
20
10
0
E−Input Current Noise (5.2pA/√Hz)
10
B−Input Voltage Noise (2.4nV/√Hz)
B−Input Current Noise (1.65pA/√Hz)
1
7
8
9
10
11
12
13
100
1k
Quiescent Current (mA)
100k
Figure 22.
1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY
vs QUIESCENT CURRENT ADJUST RESISTOR
16
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
10k
Frequency (Hz)
Figure 21.
E−Input Current Noise (pA/√Hz)
14
12
10
8
B−Input Voltage Noise (nV/√Hz)
6
B−Input Current Noise (pA/√Hz)
4
2
0
0
200 400 600 800 1000 1200 1400 1600 1800 2000
Quiescent Current Adjust Resistor (Ω )
Figure 23.
8
10
Quiescent Current (mA)
60
OTA E−Output Resistance (Ω )
90
1M
10M
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
BUF Performance
BUFFER BANDWIDTH vs OUTPUT VOLTAGE
BUFFER BANDWIDTH vs LOAD RESISTANCE
6
6
RL = 500Ω
3
Gain (dB)
3
Gain (dB)
RL = 1kΩ
VO = 0.2VPP
VO = 0.6VPP
0
VO = 5VPP
−3
VO = 2.8VPP
−3
RL = 500Ω
VO = 1.4VPP
−6
0
−6
VO = 0.2VPP
RL = 100Ω
−9
−9
1M
10M
100M
1G
2G
1M
100M
Frequency (Hz)
Figure 24.
Figure 25.
BUFFER GAIN FLATNESS
1G
2G
BUFFER SMALL-SIGNAL PULSE RESPONSE
0.5
0.20
0.4
0.15
Output Voltage (V)
0.3
0.2
Gain (dB)
10M
Frequency (Hz)
0.1
0
−0.1
−0.2
−0.3
RL = 500Ω
VIN = 0.2VPP
fIN = 20MHz
0.10
Output
Voltage
Input
Voltage
0.05
0
−0.05
−0.10
−0.15
−0.4
−0.20
−0.5
1
10
100
400
Time (10ns/div)
Frequency (MHz)
Figure 26.
Figure 27.
BUFFER LARGE-SIGNAL PULSE RESPONSE
Output Voltage (V)
2
1
0
HARMONIC DISTORTION vs FREQUENCY
−40
RL = 500Ω
VIN = 3VPP
fIN = 20MHz
Input
Voltage
Output
Voltage
−1
−2
Harmonic Distortion (dBc)
3
RL = 500Ω
VO = 2VPP
−50
2nd−Harmonic
−60
−70
−80
3rd−Harmonic
−90
−100
−3
Time (10ns/div)
1
10
100
Frequency (MHz)
Figure 28.
Figure 29.
9
OPA860
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
5MHz HARMONIC DISTORTION vs LOAD RESISTANCE
Harmonic Distortion (dBc)
−50
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−60
R L = 500Ω
VO = 2VPP
−60
Harmonic Distortion (dBc)
−40
2nd−Harmonic
−70
−80
3rd−Harmonic
−90
−100
100
RL = 500Ω
f = 5MHz
2nd−Harmonic
−70
−80
−90
3rd−Harmonic
−100
−110
1k
0.5
1.0
1.5
2.0
Figure 30.
5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
4.0
4.5
5.0
BUFFER TRANSFER FUNCTION
RL = 500Ω
VO = 2VPP
4
2nd−Harmonic
−70
3
Output Voltage (V)
Harmonic Distortion (dBc)
3.5
5
−65
−75
−80
−85
3rd−Harmonic
−90
2
1
0
−1
−2
−3
−95
−4
−100
−5
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
−5
−4
−3
±Supply Voltage (±VS)
−2
−1
0
1
2
3
4
5
Input Voltage (V)
Figure 32.
Figure 33.
INPUT VOLTAGE AND CURRENT NOISE DENSITY
BUFFER OUTPUT IMPEDANCE
100
Output Impedance (Ω )
100
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
3.0
Figure 31.
−60
10
Input Current Noise (2.1pA/√Hz)
10
Input Voltage Noise (4.8nV/√Hz)
1
1
100
1k
10k
100k
Frequency (Hz)
Figure 34.
10
2.5
Output Voltage (VPP)
Load Resistance (Ω )
1M
10M
10k
100k
1M
10M
Frequency (Hz)
Figure 35.
100M
1G
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
BUFFER GROUP DELAY TIME vs FREQUENCY
BUFFER OUTPUT VOLTAGE AND CURRENT LIMITATIONS
1.2
5
3
0.8
Output Voltage (V)
0.6
0.4
0.2
100Ω
Load Line
2
25Ω Load Line
1
50Ω Load Line
0
−1
−2
−3
250
100
50
0
−50
Frequency (MHz)
−100
700 800 900 1000
−150
400 500 600
−250
100 200 300
−300
−5
0
200
−0.2
300
1W Internal
Power Limit
−4
150
0
−200
Group Delay Time (ns)
1W Internal
Power Limit
4
1.0
Output Current (mA)
Figure 36.
Figure 37.
POWER-SUPPLY REJECTION RATIO vs FREQUENCY
50
15
45
14
40
13
35
PSRR (dB)
Quiescent Current (mA)
QUIESCENT CURRENT vs TEMPERATURE
16
12
11
10
30
25
20
9
15
8
10
7
5
6
−40
−20
+PSRR
−PSRR
0
0
20
40
60
80
100
120
10k
100k
Ambient Temperature ( C)
1M
10M
100M
Frequency (Hz)
Figure 38.
Figure 39.
VOLTAGE RANGE vs TEMPERATURE
OUTPUT CURRENT vs TEMPERATURE
4.10
56.0
55.6
4.05
Output Current (mA)
±Output Voltage Swing (V)
55.8
+VO
4.00
−VO
3.95
55.4
55.2
Output Current Sinking, Sourcing
55.0
54.8
54.6
54.4
54.2
3.90
−40
−20
0
20
40
60
80
100
120
54.0
−40
−20
0
20
40
60
Ambient Temperature ( C)
Temperature ( C)
Figure 40.
Figure 41.
80
100
120
11
OPA860
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SBOS331 – JUNE 2005
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 11.2mA, and RL = 500Ω, unless otherwise noted.
DC DRIFT vs TEMPERATURE
C-OUTPUT BIAS CURRENT vs TEMPERATURE
6
5
20
4
Buffer Input Offset Voltage (VOS)
15
3
Buffer Input Bias Current (IB)
10
2
5
1
0
−40
−20
0
0
20
40
60
80
Ambient Temperature ( C)
Figure 42.
12
100
120
OTA C−Output Bias Current (µA)
25
40
Input Bias Current (µA)
Input Offset Voltage (mV)
30
Five Representative Units
30
20
10
0
−10
−20
−30
−40
−40
−20
0
20
40
60
80
Ambient Temperature ( C)
Figure 43.
100
120
OPA860
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SBOS331 – JUNE 2005
APPLICATION INFORMATION
The OPA860 combines a high-performance buffer
with
a
transconductance
section.
This
transconductance section is discussed in the OTA
(Operational Transconductance Amplifier) section of
this data sheet. Over the years and depending on the
writer, the OTA section of an op amp has been
referred to as a Diamond Transistor, Voltage-Controlled Current source, Transconductor,
Macro Transistor, or positive second-generation current conveyor (CCII+). Corresponding symbols for
these terms are shown in Figure 44.
3
1
C
VIN1
IOUT
E
VIN2
Diamond Transistor
Voltage−Controlled Current Source
Transconductor
Macro Transistor
Current Conveyor II+
C
VIN1
VIN2
Z
CCII+
I OUT
The buffer section of the OPA860 is an 1600MHz,
4000V/µs closed-loop buffer that can be used as a
building block for AGC amplifiers, LED driver circuit,
integrator for fast pulse, fast control loop amplifiers,
and control amplifiers for capacitive sensors and
active filters. The Buffer section does not share the
bias circuit of the OTA section; thus, it is not affected
by changes in the IQ adjust resistor (RADJ).
TRANSCONDUCTANCE (OTA) SECTION—AN
OVERVIEW
B
2
BUFFER SECTION—AN OVERVIEW
B
E
Figure 44. Symbols and Terms
Regardless of its depiction, the OTA section has a
high-input impedance (B input), a low-input/output
impedance (E input), and a high impedance current
source output (C output).
The symbol for the OTA section is similar to a
transistor (see Figure 44). Applications circuits for the
OTA look and operate much like transistor circuits—the transistor is also a voltage-controlled current source. Not only does this characteristic simplify
the understanding of application circuits, it aids the
circuit optimization process as well. Many of the
same intuitive techniques used with transistor designs
apply to OTA circuits. The three terminals of the OTA
are labeled B, E, and C. This labeling calls attention
to its similarity to a transistor, yet draws distinction for
clarity. While the OTA is similar to a transistor, one
essential difference is the sense of the C-output
current: it flows out the C terminal for positive B-to-E
input voltage and in the C terminal for negative B-to-E
input voltage. The OTA offers many advantages over
a discrete transistor. The OTA is self-biased, simplifying the design process and reducing component
count. In addition, the OTA is far more linear than a
transistor. Transconductance of the OTA is constant
over a wide range of collector currents—this feature
implies a fundamental improvement of linearity.
13
OPA860
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SBOS331 – JUNE 2005
BASIC CONNECTIONS
Figure 46 shows basic connections required for
operation. These connections are not shown in subsequent circuit diagrams. Power-supply bypass capacitors should be located as close as possible to the
device pins. Solid tantalum capacitors are generally
best.
It is also possible to vary the quiescent current with a
control signal. The control loop in Figure 45 shows
1/2 of a REF200 current source used to develop
100mV on R1. The loop forces 125mV to appear on
R2. Total quiescent current of the OPA860 is approximately 37 × I1, where I1 is the current made to flow
out of pin 1.
QUIESCENT CURRENT CONTROL PIN
V+
The quiescent current of the transconductance
portion of the OPA860 is set with a resistor, RADJ,
connected from pin 1 to –VS. It affects only the
operating currents of OTA sections. The bias circuitry
of the Buffer section is independent of the bias
circuitry for the OTA section; therefore, the quiescent
current cannot go below 5.8mA. The maximum
quiescent current is 12.7mA. RADJ should be set
between 50Ω and 1kΩ for optimal performance of the
OTA section. This range corresponds to the 12.5mA
quiescent current for RADJ = 50Ω, and 9mA for RADJ =
1kΩ. If the IQ adjust pin is connected to the negative
supply, the quiescent current will be set by the 250Ω
internal resistor.
Reducing or increasing the quiescent current for the
OTA section controls the bandwidth and AC behavior
as well as the transconductance. With RADJ = 250Ω,
this sets approximately 11.2mA total quiescent current at 25°C. It may be appropriate in some applications to trim this resistor to achieve the desired
quiescent current or AC performance.
Applications circuits generally do not show the
resistor RQ, but it is required for proper operation.
With a fixed RADJ resistor, quiescent current increases with temperature (see Figure 43 in the
Typical Characteristics section). This variation of
current with temperature holds the transconductance,
gm, of the OTA relatively constant with temperature
(another advantage over a transistor).
14
OPA860
1/2 REF200
100µA
R1
1.25kΩ
IQ Adjust
1 I1
R2
425Ω
TLV2262
Figure 45. Optional Control Loop for Setting
Quiescent Current
OPA860
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SBOS331 – JUNE 2005
RQ = 250Ω, roughly sets IQ = 11.2mA.
RS
(25Ωto 200Ω)
RADJ
250Ω
+5V(1)
1
8
2
7
+
+1
3
−VS
−5V(1)
0.1µF
+VS
2.2µF
6
Solid Tantalum
4
5
49.9Ω
0.1µF
+
VO
RS
(25Ωto 200Ω)
2.2µF
VI
Solid
Tantalum
49.9Ω
NOTE: (1) VS = ±6.5V absolute maximum.
Figure 46. Basic Connections
With this control loop, quiescent current will be nearly
constant with temperature. Since this differs from the
temperature-dependent behavior of the internal current source, other temperature-dependent behavior
may differ from that shown in the Typical Characteristics. The circuit of Figure 45 will control the IQ of the
OTA section of the OPA860 somewhat more accurately than with a fixed external resistor, RQ.
Otherwise, there is no fundamental advantage to
using this more complex biasing circuitry. It does,
however, demonstrate the possibility of signal-controlled quiescent current. This capability may
suggest other possibilities such as AGC, dynamic
control of AC behavior, or VCO.
BASIC APPLICATIONS CIRCUITS
Most applications circuits for the OTA section consist
of a few basic types, which are best understood by
analogy to a transistor. Used in voltage-mode, the
OTA section can operate in three basic operating
states—common emitter, common base, and common collector. In the current-mode, the OTA can be
useful for analog computation such as current amplifier, current differentiator, current integrator, and current summer.
Common-E Amplifier or Forward Amplifier
Figure 47 compares the common-emitter configuration for a BJT with the common-E amplifier for the
OTA section. There are several advantages in using
the OTA section in place of a BJT in this configuration. Notably, the OTA does not require any biasing,
and the transconductance gain remains constant over
temperature. The output offset voltage is close to 0,
compared with several volts for the common-emitter
amplifier.
The gain is set in a similar manner as for the BJT
equivalent with Equation 1:
R
G 1 L
gm R E
(1)
Just as transistor circuits often use emitter degeneration, OTA circuits may also use degeneration. This
option can be used to reduce the effects that offset
voltage and offset current might otherwise have on
the DC operating point of the OTA. The
E-degeneration resistor may be bypassed with a
large capacitor to maintain high AC gain. Other
circumstances may suggest a smaller value capacitor
used to extend or optimize high-frequency performance.
15
OPA860
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SBOS331 – JUNE 2005
The forward amplifier shown in Figure 48 and Figure 49 corresponds to one of the basic circuits used
to
characterize
the
OPA860.
Extended
characterization of this topology appears in the Typical Characteristics section of this datasheet.
V+
RS
RL
VO
VO
VI
R1
160Ω
Inverting Gain
VOS = Several Volts
RS
VI
8
C
3 B
RC
500Ω
OTA
RE
E
2
G = 5V/V
IQ = 11.2mA
RE
78Ω
V−
(a) Common−Emitter Amplifier
Transconductance varies over temperature.
100Ω
VI
8
C
3 B
Figure 48. Forward Amplifier Configuration and
Test Circuit
VO
OTA
RL
E
2
RL1
RE
VO
Noninverting Gain
VOS = 0V
3
(b) Common−E Amplifier
Transconductance remains constant over temperature.
RIN
50Ω
OTA
R1
100Ω
RL2
rE
2
VI
RL = RL1 + RL2 || RIN
RE
Figure 47. Common-Emitter vs Common-E
Amplifier
The transconductance of the OTA with degeneration
can be calculated by Equation 2:
g m_deg 1 1
gm R E
(2)
A positive voltage at the B-input, pin 3, causes a
positive current to flow out of the C-input, pin 8.
Figure 47b shows an amplifier connection of the
OTA, the equivalent of a common-emitter transistor
amplifier. Input and output can be ground-referenced
without any biasing. The amplifier is non-inverting
because of the sense of the output current.
16
Network
Analyzer
8
G
RL
r E g1
m
RE rE
At I Q 11.2mA
G
rE 1
8
125mAV
RL
at I Q 11.2mA
R E 8
Figure 49. Forward Amplifier Design Equations
OPA860
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SBOS331 – JUNE 2005
Common-C Amplifier
Figure 50b shows the OTA connected as an
E-follower—a voltage buffer. It is interesting to notice
that the larger the RE resistor, the closer to unity gain
the buffer will be. If the OTA section is to be used as
a buffer, use RE ≥ 500Ω for best results. For the OTA
section used as a buffer, the gain is given by
Equation 3:
1
G
1
1
1 g R
m
(3)
E
V+
G=1
VOS = 0.7V
G
RL
R
L
1
RE
R E gm
This low impedance can be converted to a high
impedance by inserting the buffer amplifier in series.
Current-Mode Analog Computations
As mentioned earlier, the OTA section of the OPA860
can be used advantageously for analog computation.
Among the application possibilities are functionality
as a current amplifier, current differentiator, current
integrator, current summer, and weighted current
summer. Table 1 lists these different uses with the
associated transfer functions.
These functions can easily be combined to form
active filters. Some examples using these current-mode functions are shown later in this document.
VI
VO
RE
OPA860 APPLICATIONS
V−
(a) Common−Collector Amplifier
(Emitter Follower)
G
1
1g
1
1
mR E
The OPA860 is comprised of both the OTA section
and the Buffer section. This applications information
focuses more on using both sections together to form
various useful amplifiers. A more thorough description
of the OTA section in filter applications can be found
in the OPA861 datasheet, available for download at
www.ti.com.
R O g1
m
100Ω
VI
V+
8
C
3 B
RE
(4)
OTA
VO
G=1
VO = 0V
RL
E
2
Noninverting Gain
VOS = Several Volts
VO
RE
(b) Common−C Amplifier
(Buffer)
V−
(a) Common−Base Amplifier
Figure 50. Common-Collector vs Common-C
Amplifier
A low value resistor in series with the B OTA and
buffer inputs is recommended. This resistor helps
isolate trace parasitic from the inputs, reduces any
tendency to oscillate, and controls frequency response peaking. Typical resistor values are from 25Ω
to 200Ω.
G
100Ω
8
C
3 B
VO
OTA
E
2
Inverting Gain
VOS = 0V
RL
RE
Common-B Amplifier
Figure 51 shows the Common-B amplifier. This configuration produces an inverting gain and a low
impedance input. Equation 4 shows the gain for this
configuration.
RL
R
L
RE
R E g1m
V−
(b) Common−B Amplifier
Figure 51. Common-Base Transistor vs Common-B OTA
17
OPA860
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SBOS331 – JUNE 2005
Direct Feedback Amplifier
The gain for this topology is given by Equation 5:
R3
The direct feedback amplifier (shown in Figure 53)
topology has been used to characterize the OPA860.
Extended characterization of this topology appears in
the Typical Characteristics section of this data sheet.
This topology is obtained by closing the loop between
the C-output and the E-input of the common-E
topology, and then buffered.
G
2
R5
R5 1
2g m
1
R3
2R5
(5)
Table 1. Current-Mode Analog Computation Using the OTA Section
FUNCTIONAL ELEMENT
TRANSFER FUNCTION
Current Amplifier
R
I OUT 1 I IN
R2
IMPLEMENTATION WITH THE OTA SECTION
IOUT
IIN
R1
R2
IOUT
1
I OUT Current Integrator
CR
IIN
I dt
C
IN
R
IOUT
n
I OUT I
j
j1
Current Summer
I1
I2
In
I OUT
n
I OUT I RR
j
j
j1
Weighted Current Summer
R
R1
I1
18
R
Rn
In
OPA860
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SBOS331 – JUNE 2005
Current-Feedback Amplifier
portional) behavior versus frequency. The control
loop amplifiers show an integrator behavior from DC
to the frequency, represented by the RC time constant of the network from the C-output to GND. Above
this frequency, they operate as an amp with constant
gain. The series connection increases the overall gain
to about 110dB and thus minimizes the control loop
deviation. The differential configuration at the inputs
enables one to apply the measured output signal and
the reference voltage to two identical high-impedance
inputs. The output buffer decouples the C-output of
the second OTA in order to insure the AC performance and to drive subsequent output stages.
Building a current-feedback amplifier with the
OPA860 is extremely simple. One advantage of
building a current-feedback amplifier with the
OPA860 instead of getting an off-the-shelf current-feedback amplifier is the control gained on the
bandwidth though the use of external capacitors.
Figure 54 shows a typical circuit for the OPA860 in a
noninverting current-feedback amplifier configuration.
Input and output parasitic capacitances are shown.
R1 is the output impedance of the C-output of the
OTA section. C1 is the output parasitic capacitance
on the C-output pin of the OTA-section. C2 is the
input parasitic capacitance for the input of the Buffer
section. As shown in Equation 6, the poles formed by
R1, C1, R2, and C2 control the frequency response.
The frequency response in this configuration is shown
in Figure 52. Setting an external capacitor on the
C-output to ground allows adjusting the bandwidth.
1R
RF
1 1R
G
1
g mR 1
6
3
Gain (dB)
V OUT
V IN
RF
9
G
[1 s(R 1C1 R1C 2 R 2C 2) s R1C 1C 2]
2
0
−3
(6)
−6
Note that both peaking and bandwidth can be adjusted by changing the feedback resistance, RF.
−9
−12
Control-Loop Amplifier
1M
R1
100Ω
VI
7
3 B
50Ω
8
C
OTA
1 4
RQ
250Ω
100M
1G
Figure 52. Current-Feedback Architecture
Frequency Response
R2
80.6Ω
+5V
10M
Frequency (Hz)
A new type of control loop amplifier for fast and
precise control circuits can be designed with the
OPA860. The circuit of Figure 55 shows a series
connection of two voltage control current sources that
have an integral (and at higher frequencies, a pro-
50Ω
Source
G = +2V/V
RL = 500Ω
VO = 2VPP
R3
301Ω
E
2
5
+1
6
VO
R4
453Ω
Network
Analyzer
RIN
50Ω
G = +2V/V
I Q = 20mA
R5
133Ω
−5V
Figure 53. Direct Feedback Amplifier Specification and Test Circuit
19
OPA860
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SBOS331 – JUNE 2005
OPA860
R2
50Ω
VOUT
+1
200Ω
VIN
C1
R1
C2
500Ω
50Ω
rE
RF
259Ω
RG
249Ω
Figure 54. OPA860 Used in a Noninverting Current-Feedback Architecture
8
5
6
+1
3
8
180Ω
2
10pF
VREF
10pF
3
2
10Ω
180Ω
VIN
5
33Ω
10Ω
33Ω
6
+1
Figure 55. Control-Loop Amplifier Using Two OPA860s
20
VOUT
OPA860
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SBOS331 – JUNE 2005
DC-Restore Circuit
Comparator
The OPA860 can be used advantageously with an
operational amplifier, here the OPA820, as a
DC-restore circuit. Figure 56 illustrates this design.
Depending on the collector current of the
transconductance amplifier (OTA) of the OPA860, a
switching function is realized with the diodes D1 and
D2.
An interesting and also cost-effective circuit solution
using the OPA860 as a low-jitter comparator is shown
in Figure 57. At the same time, this circuit uses a
positive and negative feedback. The input is connected to the inverting E-input. The output signal is
applied in a direct feedback over the two antiparallel,
connected gallium-arsenide diodes back to the emitter. A second feedback path over the RC combination
to the base, which is a positive feedback, accelerates
the output voltage change when the input voltage
crosses the threshold voltage. The output voltage is
limited to the threshold voltage of the back-to-back
diodes.
When the C-output is sourcing current, the capacitor
C1 is being charged. When the C-output is sinking
current, D1 is turned off and D2 is turned on, letting
the voltage across C1 be discharged through R2.
The condition to charge C1 is set by the voltage
difference between VREF and VOUT. For the OTA
C-output to source current, VREF has to be greater
than VOUT. The rate of charge of C1 is set by both R1
and C1. The discharge rate is given by R2 and C1.
150Ω
VIN
5
+1
6
C1
100pF
20Ω
D1
D 1, D2 = 1N4148
RQ = 1kΩ
OPA656
R2
100kΩ
VOUT
20Ω
D2
CCII
8 C
The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1.
R1 and C1 set the DC restoration time constant.
D1 adds a propogation delay to the DC restoration.
R2 and C1 set the decay time constant.
E 2
R1
40.2Ω
B
3
R2
100Ω
VREF
Figure 56. DC Restorer Circuit
21
OPA860
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SBOS331 – JUNE 2005
C3
2.2pF
Offset
Trim
0.5pF …2.5pF
+5V
R5
47kΩ
R2
10kΩ
R8
27kΩ
+5V
−5V
R1
100kΩ
RC5
150Ω
VIN
5
+1
R2
100kΩ
6
RC5
150Ω
3
2
+5V
C3
2.2µF
RC5
150Ω
7
OTA
8
RS
47Ω
1
4
1
VOUT
BUF602
5
4
RQ
250Ω
C3
2.2µF
C3
2.2µF
C3
2.2µF
−5V
−5V
D1
D2
DMF3068A
Figure 57. Comparator (Low Jitter)
T
Integrator for ns Pulse
One very interesting application using the OPA860 in
physical measurement technology is an open-loop
ns-integrator (shown in Figure 58) which can process
pulses with an amplitude of ±2.5V, have a rise/fall
time of as little as 2ns, and also have a pulse width of
more than 8ns. The voltage-controlled current source
charges the integration capacitor linearly according to
Equation 7:
V C VBE gm t
C
(7)
Where:
• VC = Voltage At Pin 8
• VBE = Base-Emitter Voltage
• gm = Transconductance
• t = Time
• C = Integration Capacitance
gm
VO C
V
BE
dt
(8)
O
Where:
• VO = Output Voltage
• T = Integration Time
• C = Integration Capacitance
200Ω
780Ω
VI
8
C
3 B
The output voltage is the time integral of the input
voltage. It can be calculated from Equation 8:
27pF
620Ω
820Ω
1µF
50Ω
+5V
+1
OTA
E
2
50Ω
5
−5V
Figure 58. Integrator for ns-Pulses
22
6
VO
OPA860
www.ti.com
SBOS331 – JUNE 2005
Video Luminance Matrix
The inverting amplifier in Figure 59 amplifies the
three input voltages that correspond to the luminance
section of the RGB color signal. Different feedback
resistances weight the voltages differently, resulting
in an output voltage consisting of 30% of the red,
59% of the green, and 11% of the blue section of the
input voltage. The way in which the signal is weighted
corresponds to the transformation equation for converting RGB pictures into B/W pictures. The output
signal is the black/white replay. It might drive a
monochrome control monitor or an analog printer
(hardcopy output).
VIN
VOUT
Figure 60. State Variable Filter Block Diagram
C
200Ω
150Ω
5
+1
6
E
C
VLUMINANCE
8
C
3 B
OTA
VBLUE
R2
x1
R3
665Ω(1)
200Ω
RV
1820Ω(1)
C1
E
x1
VRED
VGREEN
C1
B
E
2
340Ω(1)
B
RQ = 250Ω
(IQ = 11.2mA)
NOTE: (1) Resistors shown are 1% values
that produce 30%/59%/11% R/G/B mix.
R1
VIN
VOUT
Figure 61. State Variable Filter Using the OPA860
The transfer function is then:
Figure 59. Video Luminance Matrix
H(s) a0
R
1
s 2 C 1s C 0
R V 1 sC
R3
1
s 2C 1C 2R 1R 2
(9)
State-Variable Filters
The ability of the OPA860 to easily drive a capacitor
can be put to good use in implementing state-variable
filters. A state-variable filter, or KHN filter, can be
represented with integrators and coefficients. For
example, the filter represented in the block diagram
of Figure 60 can easily be implemented with two
OPA860s, as shown in Figure 61.
R 1R 2
2
0 Q
1
C1C 2R 1R2
C1
R3
C 2 R 1R2
(10)
(11)
23
OPA860
www.ti.com
SBOS331 – JUNE 2005
DESIGN-IN TOOLS
DEMONSTRATION BOARDS
A printed circuit board (PCB) is available to assist in
the initial evaluation of circuit performance using the
OPA860. This module is available free, as an
unpopulated PCB delivered with descriptive documentation. The summary information for the board is
shown below:
PRODUCT
PACKAGE
BOARD PART
NUMBER
OPA860ID
SO-8
DEM-OPA86xD
LITERATURE
REQUEST
NUMBER
SBOU035
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 12 shows the
general form for the output noise voltage using the
terms shown in Figure 62.
eO 2
e RSi bn 4kTR S
2
n
RL
RG g1m
2
2
R
R Gibi 4kTR G 1L
gm
(12)
For the buffer, the noise model is shown in Figure 63.
Equation 13 shows the general form for the output
noise voltage using the terms shown in Figure 63.
The board can be requested on Texas Instruments
web site (www.ti.com).
en
VO
MACROMODELS AND APPLICATIONS
SUPPORT
RS
Computer simulation of circuit performance using
SPICE is often useful when analyzing the performance of analog circuits and systems. This principle is
particularly true for Video and RF amplifier circuits
where parasitic capacitance and inductance can have
a major effect on circuit performance. A SPICE model
for the OPA860 is available through the Texas
Instruments web page (www.ti.com). These models
do a good job of predicting small-signal AC and
transient performance under a wide variety of
operating conditions. They do not do as well in
predicting the harmonic distortion. These models do
not attempt to distinguish between the package types
in their small-signal AC performance.
NOISE PERFORMANCE
The OTA noise model consists of three elements: a
voltage noise on the B-input; a current noise on the
B-input; and a current noise on the E-input. Figure 62
shows the OTA noise analysis model with all the
noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms
in either nV/√Hz or pA/√Hz.
en
VO
RL
RS
√4kTRS
ibn
RG
ibi
√4kTRS
Figure 62. OTA Noise Analysis Model
24
in
√4kTRS
Figure 63. Buffer Noise Analysis Model
eO e
2
n
2
inR S 4kTR S
(13)
THERMAL ANALYSIS
Due to the high output power capability of the
OPA860, heatsinking or forced airflow may be required under extreme operating conditions. Maximum
desired junction temperature will set the maximum
allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by
TA + PD ×θJA. The total internal power dissipation (PD)
is the sum of quiescent power (PDQ) and additional
power dissipated in the output stage (PDL) to deliver
load power. Quiescent power is simply the specified
no-load supply current times the total supply voltage
across the part. PDL will depend on the required
output signal and load but would, for a grounded
resistive load, be at a maximum when the output is
fixed at a voltage equal to 1/2 of either supply voltage
(for equal bipolar supplies). Under this condition,
PDL = VS2/(4 × RL) where RL includes feedback
network loading.
Note that it is the power in the output stage and not
into the load that determines internal power dissipation.
OPA860
www.ti.com
SBOS331 – JUNE 2005
As a worst-case example, compute the maximum TJ
using an OPA860ID in the circuit of Figure 53
operating at the maximum specified ambient temperature of +85°C and driving a grounded 20Ω load.
PD = 10V × 11.2mA + 52/(4 × 20Ω) = 424mW
Maximum TJ = +85°C + (0.43W × 125°C/W) = 139°C.
Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower tested junction temperatures. The highest possible internal dissipation
will occur if the load requires current to be forced into
the output for positive output voltages or sourced
from the output for negative output voltages. This
puts a high current through a large internal voltage
drop in the output transistors. The output V-I plot
shown in the Typical Characteristics include a boundary for 1W maximum internal power dissipation under
these conditions.
BOARD LAYOUT GUIDELINES
Achieving
optimum
performance
with
a
high-frequency amplifier like the OPA860 requires
careful attention to board layout parasitics and external component types. Recommendations that will
optimize performance include:
a) Minimize parasitic capacitance to any AC ground
for all of the signal I/O pins. Parasitic capacitance on
the output and inverting input pins can cause instability: on the noninverting input, it can react with the
source
impedance
to
cause
unintentional
bandlimiting. To reduce unwanted capacitance, a
window around the signal I/O pins should be opened
in all of the ground and power planes around those
pins. Otherwise, ground and power planes should be
unbroken elsewhere on the board.
b) Minimize the distance (< 0.25") from the
power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and
power-plane layout should not be in close proximity to
the signal I/O pins. Avoid narrow power and ground
traces to minimize inductance between the pins and
the decoupling capacitors. The power-supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.1µF)
across the two power supplies (for bipolar operation)
will improve 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the
main supply pins. These may be placed somewhat
farther from the device and may be shared among
several devices in the same area of the PC board.
c) Careful selection and placement of external
components will preserve the high-frequency performance of the OPA860. Resistors should be a
very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal film
or carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
Again, keep their leads and PC board traces as short
as possible. Never use wirewound type resistors in a
high-frequency application.
d) Connections to other wideband devices on the
board may be made with short, direct traces or
through onboard transmission lines. For short connections, consider the trace and the input to the next
device as a lumped capacitive load. Relatively wide
traces (50mils to 100mils) should be used, preferably
with ground and power planes opened up around
them. Estimate the total capacitive load and set RS
from the plot of Recommended RS vs Capacitive
Load. Low parasitic capacitive loads (< 5pF) may not
need an RS since the OPA860 is nominally compensated to operate with a 2pF parasitic load. Higher
parasitic capacitive loads without an RS are allowed
as the signal gain increases (increasing the unloaded
phase margin). If a long trace is required, and the
6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched
impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook
for microstrip and stripline layout techniques). A 50Ω
environment is normally not necessary on board, and
in fact, a higher impedance environment will improve
distortion as shown in the distortion versus load plots.
e) Socketing a high-speed part like the OPA860 is
not recommended. The additional lead length and
pin-to-pin capacitance introduced by the socket can
create an extremely troublesome parasitic network
that makes it almost impossible to achieve a smooth,
stable frequency response. Best results are obtained
by soldering the OPA860 onto the board.
25
OPA860
www.ti.com
SBOS331 – JUNE 2005
INPUT AND ESD PROTECTION
The OPA860 is built using a very high-speed complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD protection
diodes to the power supplies as shown in Figure 64.
+VCC
External
Pin
Internal
Circuitry
−VCC
Figure 64. Internal ESD Protection
26
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA continuous current. Where higher currents are possible (for
example, in systems with ±15V supply parts driving
into the OPA860), current-limiting series resistors
should be added into the two inputs. Keep these
resistor values as low as possible since high values
degrade both noise performance and frequency response.
PACKAGE OPTION ADDENDUM
www.ti.com
29-Jun-2005
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
OPA860ID
ACTIVE
SOIC
D
8
75
TBD
Call TI
Call TI
OPA860IDR
ACTIVE
SOIC
D
8
2500
TBD
Call TI
Call TI
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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