ON NCP4303BDR2G Secondary side synchronous rectification driver for high efficiency smps topology Datasheet

NCP4303A/B
Secondary Side
Synchronous Rectification
Driver for High Efficiency
SMPS Topologies
The NCP4303A/B is a full featured controller and driver tailored to
control synchronous rectification circuitry in switch mode power
supplies. Thanks to its versatility, it can be used in various topologies
such as flyback, forward and Half Bridge Resonant LLC.
The combination of externally adjustable minimum on and off times
helps to fight the ringing induced by the PCB layout and other
parasitic elements. Therefore, a reliable and noise less operation of the
SR system is insured.
The extremely low turn off delay time, high sink current capability
of the driver and automatic package parasitic inductance
compensation system allow to maximize synchronous rectification
MOSFET conduction time that enables further increase of SMPS
efficiency.
Finally, a wide operating VCC range combined with two versions of
driver voltage clamp eases implementation of the SR system in 24 V
output applications.
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MARKING
DIAGRAM
8
8
1
SOIC−8
D SUFFIX
CASE 751
xx
A
WL, L
YY, Y
WW, W
G or G
Features
• Self−Contained Control of Synchronous Rectifier in CCM, DCM,
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
and QR Flyback Applications
Precise True Secondary Zero Current Detection with Adjustable
Threshold
Automatic Parasitic Inductance Compensation Input
Typically 40 ns Turn off Delay from Current Sense Input to Driver
Zero Current Detection Pin Capability up to 200 V
Ultrafast Trigger Interface to External Signal for CCM operation
Disable Input to Enter Standby or Low Consumption Mode
Adjustable Minimum On Time Independent of VCC Level
Adjustable Minimum Off Time Independent of VCC Level
5 A / 2.5 A Peak Current Sink / Source Drive Capability
Operating Voltage Range up to 30 V
Gate Drive Clamp of Either 12 V (NCP4303A) or 6 V (NCP4303B)
Low Startup and Standby Current Consumption
Maximum Frequency of Operation up to 500 kHz
SOIC−8 Package
These are Pb−Free Devices
1
XXXXX
ALYWX
G
= Specific Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
PINOUT INFORMATION
VCC
Min_Toff
Min_Ton
Trig/Disable
1
2
3
4
8
7
6
5
DRV
GND
COMP
CS
ORDERING INFORMATION
Package
Shipping†
NCP4303ADR2G
SOIC−8
(Pb−Free)
2500 /
Tape & Reel
NCP4303BDR2G
SOIC−8
(Pb−Free)
2500 /
Tape & Reel
Device
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
Typical Applications
•
•
•
•
Notebook Adapters
High Power Density AC/DC Power Supplies
Gaming Consoles
All SMPS with High Efficiency Requirements
© Semiconductor Components Industries, LLC, 2010
April, 2010 − Rev. 1
1
Publication Order Number:
NCP4303/D
NCP4303A/B
Figure 1. Typical Application Example – LLC Converter
Figure 2. Typical Application Example − DCM or QR Flyback Converter
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2
NCP4303A/B
PIN FUNCTION DESCRIPTION
Pin No.
Pin Name
Function
1
VCC
Supplies the driver
Pin Description
2
Min_toff
Minimum off time adjust
Adjust the minimum off time period by connecting resistor to ground.
3
Min_ton
Minimum on time adjust
Adjust the minimum on time period by connecting resistor to ground.
4
TRIG/Disable
Forced reset input
This ultrafast input turns off the SR MOSFET in CCM applications. Activates
sleep mode if pulled up for more than 100 ms.
5
CS
Current sense of the SR
MOSFET
This pin detects if the current flows through the SR MOSFET and/or its body
diode. Basic turn off detection threshold is 0 mV. A resistor in series with this
pin can modify the turn off threshold if needed.
6
COMP
Compensation inductance
connection
7
GND
IC ground
8
DRV
Gate driver output
VCC supply terminal of the controller. Accepts up to 30 V continuously.
Use as a Kelvin connection to auxiliary compensation inductance. If SR
MOSFET package parasitic inductance compensation is not used (like for
SMT MOSFETs), connect this pin directly to GND pin.
Ground connection for the SR MOSFET driver and VCC decoupling capacitor.
Ground connection for minimum ton, toff adjust resistors and trigger input.
GND pin should be wired directly to the SR MOSFET source terminal/soldering point using Kelvin connection.
Driver output for the SR MOSFET.
Figure 3. Internal Circuit Architecture
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3
NCP4303A/B
MAXIMUM RATINGS
Symbol
Value
Unit
VCC
IC supply voltage
Rating
−0.3 to 30
V
VDRV
Driver output voltage
−0.3 to 17
V
VCS
Current sense input dc voltage
−4 to 200
V
VCsdyn
Current sense input dynamic voltage (tpw = 200 ns)
−10 to 200
V
VTRIG
Trigger input voltage
−0.3 to 10
V
VMin_ton, VMin_toff
Min_Ton and Min_Toff input voltage
−0.3 to 10
V
I_Min_Toff, I_Min_Toff
Min_Ton and Min_Toff current
−10 to +10
mA
Static voltage difference between GND and COMP pins (internally clamped)
−3 to 10
V
Dynamic voltage difference between GND and COMP pins (tpw = 200 ns)
−10 to 10
V
−5 to 5
mA
180
°C/W
VGND−COMP
VGND−COMP_dyn
ICOMP
Current into COMP pin
RqJA
Thermal Resistance Junction−to−Air, SOIC version, A/B version
TJmax
Maximum junction temperature
TSmax
Storage Temperature Range
TLmax
Lead temperature (Soldering, 10 s)
ESD Capability, Human Body Model except pin VCS – pin 5, HBM ESD
Capability on pin 5 is 650 V
ESD Capability, Machine Model
150
°C
−60 to +150
°C
300
°C
2
kV
200
V
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
1. This device series contains ESD protection and exceeds the following tests:
Pin 1*8: Human Body Model 2000 V per JEDEC Standard JESD22−A114E.
Machine Model Method 200 V pre JEDEC Standard JESD22−A115−A
2. This device meets latchup tests defined by JEDEC Standard JESD78.
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NCP4303A/B
ELECTRICAL CHARACTERISTICS (For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, Max TJ = 150°C, VCC
= 12 V, Cload = 0 nF, R_min_ton = R_min_toff = 10 kW, Vtrig = 0 V, f_CS = 100 kHz, DC_CS = 50%, VCS_high = 4 V, VCS_low=−1 V unless
otherwise noted)
Symbol
Rating
Pin
Min
Typ
Max
Unit
SUPPLY SECTION
VCC_on
Turn−on threshold level (VCC going up)
1
9.3
9.9
10.5
V
VCC_off
Minimum operating voltage after turn−on (VCC going down)
1
8.3
8.9
9.5
V
VCC hysteresis
1
0.8
1.0
1.3
V
ICC1_A
ICC1_B
Internal IC consumption (no output load on pin 8, Fsw = 500 kHz, RTon_min
= RToff_min = 5 kW)
1
−
−
4.7
4
−
−
mA
ICC2_A
ICC2_B
Internal IC consumption (Cload = 1 nF on pin 8, Fsw = 400 kHz, RTon_min =
RToff_min = 5 kW)
1
−
−
9.3
6.4
−
−
mA
ICC3_A
ICC3_B
Internal IC consumption (Cload = 10 nF on pin 8, Fsw = 400 kHz, RTon_min =
RToff_min = 5 kW)
1
−
−
54
34
−
−
mA
ICC_SDM
Startup current consumption (VCC = VCC_on − 0.1 V) and consumption
during light load (disable) mode, (Fsw = 500 kHz, Vtrig = 5 V)
1
−
390
550
mA
ICC_SDM NS
Startup current consumption (VCC = VCC_on − 0.1 V) and consumption
during light load (disable) mode, (Vcs = 0 V, Vtrig = 5 V)
1
−
280
450
mA
tr_A
Output voltage rise−time for A version (Cload = 10 nF), (Note 3)
8
−
120
−
ns
tr_B
Output voltage rise−time for B version (Cload = 10 nF), (Note 3)
8
−
80
−
ns
tf_A
Output voltage fall−time for A version (Cload = 10 nF), (Note 3)
8
−
50
−
ns
tf_B
Output voltage fall−time for B version (Cload = 10 nF), (Note 3)
8
−
35
−
ns
Roh
Driver source resistance (Note 3)
8
−
1.8
7
W
Rol
Driver sink resistance
8
−
1
2
W
Output source peak current (Note 3)
8
−
2.5
−
A
VCC_hyste
DRIVE OUTPUT
IDRV_pk(source)
IDRV_pk(sink)
Output sink peak current (Note 3)
8
−
5
−
A
VDRV(H)_A
Driver high level output voltage on A version (Cload = 1 nF)
8
10
−
−
V
VDRV(H)_A
Driver high level output voltage on A version (Cload = 10 nF)
8
11.8
−
−
V
VDRV(H)_B
Driver high level output voltage on B version (Cload = 1 nF)
8
5
−
−
V
VDRV(H)_B
Driver high level output voltage on B version (Cload = 10 nF)
8
6
−
−
V
VDRV(min_A)
Minimum drive output voltage for A version (VCC = VCC_off + 200 mV)
8
8.3
−
−
V
VDRV(min_B)
Minimum drive output voltage for B version (VCC = VCC_off + 200 mV)
8
4.5
−
−
V
VDRV(CLMP_A)
Driver clamp voltage for A version, (12 V < VCC < 28 V, minimum Cload =
1 nF)
8
−
12
16
V
VDRV(CLMP_B)
Driver clamp voltage for B version, (12 V < VCC < 28 V, minimum Cload =
1 nF)
8
−
7
8.3
V
CS INPUT
Tpd_on
The total propagation delay from CS input to DRV output turn on (VCS goes
down from 4 V to −1 V, tf_CS = 5 ns, COMP pin connected to GND)
5, 8
−
60
90
ns
Tpd_off
The total propagation delay from CS input to DRV output turn off (VCS goes
up from −1 V to 4 V, tr_CS = 5 ns, COMP pin connected to GND), (Note 3)
5, 8
−
40
55
ns
Ishift_CS
Current sense input current source (VCS = 0 V)
5
95
100
105
mA
Vth_cs_on
Turn on current sense input threshold voltage
5, 8
−120
−85
−50
mV
Vth_cs_off
Current sense pin turn off threshold voltage, COMP pin connected to GND
(Note 3)
5, 8
−1
−
0
mV
Compensation inverter gain (Note 3)
5,6,8
−
−1
−
−
Gcomp
3. Guaranteed by design.
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NCP4303A/B
ELECTRICAL CHARACTERISTICS (For typical values TJ = 25°C, for min/max values TJ = −40°C to +125°C, Max TJ = 150°C, VCC
= 12 V, Cload = 0 nF, R_min_ton = R_min_toff = 10 kW, Vtrig = 0 V, f_CS = 100 kHz, DC_CS = 50%, VCS_high = 4 V, VCS_low=−1 V unless
otherwise noted)
Symbol
Rating
Pin
Min
Typ
Max
Unit
5
−
−
1
mA
Minimum trigger pulse duration
4
30
−
−
ns
Vtrig
Trigger input threshold voltage (Vtrig goes up)
4
1.5
−
2.5
V
tp_trig
Propagation delay from trigger input to the DRV output (Vtrig goes up from 0
to 5 V tr_trig = 5 ns)
4
−
−
30
ns
Light load turn off filter duration
4
−
100
−
ms
4
−
−
550
ns
4
−
10
−
uA
CS INPUT
ICS_Leakage
CS input leakage current, VCS = 200 Vdc
TRIGGER/DISABLE INPUT
Ttrig_pw
ttrig_light_load
ttrig_light_load_rec IC operation recovery time when leaving light load disable mode (Vtrig goes
down from 5 to 0 V tf_trig = 5 ns)
Itrig
Trigger input pull down current (Vtrig = 5 V)
MINIMUM Ton AND Toff ADJUST
Ton_min
Minimum Ton period (RT_on_min = 0 W)
3
−
300
−
ns
Toff_min
Minimum Toff period (RT_off_min = 0 W)
2
−
620
−
ns
Ton_min
Minimum Ton period (RT_on_min = 10 kW)
3
0.9
1.0
1.1
ms
Toff_min
Minimum Toff period (RT_off_min = 10 kW)
2
0.9
1.0
1.1
ms
Ton_min
Minimum Ton period (RT_on_min = 50 kW)
3
−
4.8
−
ms
Toff_min
Minimum Toff period (RT_off_min = 50 kW)
2
−
4.8
−
ms
3. Guaranteed by design.
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NCP4303A/B
9.96
8.9
9.94
8.88
9.92
8.86
9.9
8.84
VCCoff (V)
VCCon (V)
TYPICAL CHARACTERISTICS
9.88
9.86
8.82
8.8
9.84
8.78
9.82
8.76
9.8
−40 −25 −10 5
20
35
50
65
80
95
110 125
8.74
−40 −25 −10 5
50
65
80
95
TEMPERATURE (°C)
Figure 4. VCC Startup Voltage
Figure 5. VCC Turn−off Voltage
420
1.065
410
ICC_SDM (mA)
1.06
VCC_Hyste (V)
35
TEMPERATURE (°C)
1.07
1.055
1.05
1.045
110 125
400
390
380
370
1.04
1.035
−40 −25 −10 5
20
35
50
65
80
95
360
−40 −25 −10 5
110 125
20
35
50
65
80
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 6. VCC Hysteresis
Figure 7. Startup Current
11.8
95
110 125
95
110 125
12.06
12.04
11.6
12.02
VDRV(H)_A (V)
11.4
VDRV(H)_A (V)
20
11.2
11
10.8
12
11.98
11.96
11.94
11.92
10.6
11.9
10.4
−40 −25 −10 5
20 35 50 65 80
TEMPERATURE (°C)
95
11.88
−40 −25 −10 5
110 125
Figure 8. Driver High Level – A Version,
VCC = 12 V and Cload = 1 nF
20 35 50 65 80
TEMPERATURE (°C)
Figure 9. Driver High Level− A Version, VCC =
12 V and Cload = 10 nF
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NCP4303A/B
5.95
7.02
5.9
7
5.85
6.98
5.8
6.96
VDRV(H)_B (V)
VDRV(H)_B (V)
TYPICAL CHARACTERISTICS
5.75
5.7
5.65
6.92
6.9
5.6
6.88
5.55
6.86
5.5
−40 −25 −10 5
20
35
50
65
80
95
6.84
−40 −25 −10 5
110 125
35
50
65
80
95
110 125
TEMPERATURE (°C)
Figure 10. Driver High Level – B Version, VCC
= 12 V and Cload = 1 nF
Figure 11. Driver High Level – B Version, VCC =
12 V and Cload = 10 nF
9.96
5.9
9.94
5.8
9.92
5.7
9.9
9.88
9.86
9.84
5.6
5.5
5.4
5.3
9.82
5.2
9.8
9.78
−40 −25 −10 5
20
35
50
65
80
95
110 125
5.1
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 12. Minimal Driver High Level – A
Version, VCC = VCC_OFF + 0.2 V and Cload = 0 nF
Figure 13. Minimal Driver High Level – B
Version, VCC = VCC_OFF + 0.2 V and Cload = 0 nF
12.8
14.2
14
12.6
13.8
VDRV(CLMP_A) (V)
12.4
VDRV(CLMP_A) (V)
20
TEMPERATURE (°C)
VDRV(min_B) (V)
VDRV(min_A) (V)
6.94
12.2
12
11.8
11.6
13.6
13.4
13.2
13
12.8
12.6
11.4
12.4
11.2
−40 −25 −10 5
20
35
50
65
80
95
110 125
12.2
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 14. Driver Clamp Level – A Version,
VCC = 28 V and Cload = 1 nF
Figure 15. Driver Clamp Level – A Version,
VCC = 28 V and Cload = 10 nF
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NCP4303A/B
TYPICAL CHARACTERISTICS
7.35
6.15
7.3
6.1
7.25
6.05
VDRV(CLMP_B) (V)
VDRV(CLMP_B) (V)
6.2
6
5.95
5.9
5.85
5.8
7
6.95
6.9
5.7
6.85
20
35
50
65
80
95
6.8
−40 −25 −10 5
110 125
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 16. Driver Clamp Level – B Version,
VCC = 28 V and Cload = 1 nF
Figure 17. Driver Clamp Level – B Version,
VCC = 28 V and Cload = 10 nF
70
45
40
60
35
50
30
TPD_off (ns)
TPD_on (ns)
7.1
7.05
5.75
5.65
−40 −25 −10 5
40
30
20
25
20
15
10
10
5
0
−40 −25 −10 5
20
35
50
65
80
95
110 125
0
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 18. CS to DRV Turn−on Propagation
Delay
Figure 19. CS to DRV Turn−off Propagation
Delay
100.5
−40
100
−50
−60
Vth_CS_on (mV)
99.5
Ishift_CS (mA)
7.2
7.15
99
98.5
98
−70
−80
−90
−100
97.5
−110
97
−40 −25 −10 5
20 35 50 65 80
TEMPERATURE (°C)
95
110 125
−120
−40 −25 −10 5
20 35 50 65 80
TEMPERATURE (°C)
95
Figure 21. CS Turn−on Threshold
Figure 20. CS Pin Shift Current
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110 125
NCP4303A/B
TYPICAL CHARACTERISTICS
18
2.15
16
14
12
2.05
Tp_trig (ns)
Vtrig (V)
2.1
2
10
8
6
4
1.95
2
1.9
−40 −25 −10 5
20
35
50
65
80
95
110 125
0
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 22. Trigger Input Threshold Voltage
Figure 23. Propagation Delay from Trigger
Input to DRV Turn−off
117
485
116.5
480
Ttrig−light_load_rec (ns)
Ttrig−light_load (ms)
116
115.5
115
114.5
114
475
470
465
113.5
113
−40 −25 −10 5
20 35 50 65 80
TEMPERATURE (°C)
95
110 125
460
−40 −25 −10 5
Figure 24. Light Load Transition Timer
Duration
95
110 125
Figure 25. Light Load to Normal Operation
Recovery Time
12
290
10
285
Ton_min (ns)
8
Itrig (mA)
20 35 50 65 80
TEMPERATURE (°C)
6
4
280
275
270
2
0
−40 −25 −10
5
20
35
50
65
80
95
110 125
265
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 26. Trigger Input Pulldown Current
Figure 27. Minimum on Time @ Rt_on_min = 0 W
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NCP4303A/B
TYPICAL CHARACTERISTICS
1047
976
1046
975
974
1044
Toff_min (ns)
Ton_min (ns)
1045
1043
1042
973
972
971
1041
970
1040
1039
−40 −25 −10 5
20
35
50
65
80
95
110 125
969
−40 −25 −10 5
20
35
50
65
80
95
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 28. Minimum on Time @ Rt_on_min =
10 kW
Figure 29. Minimum Off Time @
Rt_off_min = 10 kW
5340
110 125
5050
5320
5000
Toff_min (ns)
5280
5260
5240
5220
4950
4900
4850
5200
5180
−40 −25 −10 5
20
35
50
65
80
95
110 125
4800
−40 −25 −10 5
20
35
50
65
80
95
TEMPERATURE (°C)
TEMPERATURE (°C)
Figure 30. Minimum on Time @
Rt_on_min = 53 kW
Figure 31. Minimum Off Time @
Rt_off_min = 53 kW
640
635
630
Toff_min (ns)
Ton_min (ns)
5300
625
620
615
610
605
−40 −25 −10 5
20
35
50
65
80
95
110 125
TEMPERATURE (°C)
Figure 32. Minimum Off Time @ Rt_off_min = 0 W
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110 125
NCP4303A/B
APPLICATION INFORMATION
General Description
To overcome issues after turn on and off events, the
NCP4303 provides adjustable minimum on time and off
time blanking periods. Blanking times can be adjusted
independently of IC VCC using resistors connected to GND.
If needed, blanking periods can be modulated using
additional components.
An ultrafast trigger input helps to implement synchronous
rectification systems in CCM applications (like CCM
flyback or forward). The time delay from trigger input to
driver turn off event is 12 ns (typically). Additionally, the
trigger input can be used to disable the IC and activate a low
consumption standby mode. This feature can be used to
decrease standby consumption of an SMPS.
Finally, the NCP4303 features a special input that can be
used to automatically compensate for SR MOSFET
parasitic inductance effect. This technique achieves the
maximum available on−time and thus optimizes efficiency
when a MOSFET in standard package (like TO220 or
TO247) is used. If a SR MOSFET in SMT package with
negligible inductance is used, the compensation input is
connected to GND pin.
The NCP4303 is designed to operate either as a standalone
IC or as a companion IC to a primary side controller to help
achieve efficient synchronous rectification in switch mode
power supplies. This controller features a high current gate
driver along with high−speed logic circuitry to provide
appropriately timed drive signals to a synchronous
rectification MOSFET. With its novel architecture, the
NCP4303 has enough versatility to keep the synchronous
rectification system efficient under any operating mode.
The NCP4303 works from an available bias supply with
voltage range from 10.4 V to 28 V (typical). The wide VCC
range allows direct connection to the SMPS output voltage
of most adapters such as notebook and LCD TV adapters. As
a result, the NCP4303 simplifies circuit operation compared
to other devices that require specific bias power supply (e.g.
5 V). The high voltage capability of the VCC pin is also a
unique feature designed to allow operation for a broader
range of applications.
Precise turn off threshold of the current sense comparator
together with accurate offset current source allows the user
to adjust for any required turn off current threshold of the SR
MOSFET switch using a single resistor. Compared to other
SR controllers that provide turn off thresholds in the range
of −10 mV to −5 mV, the NCP4303 offers a turn off
threshold of 0 mV that in combination with a low RDS(on) SR
MOSFET significantly reduces the turn off current
threshold and improves efficiency.
Zero Current Detection and parasitic inductance
compensation
Figure 33 shows the internal connection of the ZCD
circuitry on the current sense input. The synchronous
rectification MOSFET is depicted with it’s parasitic
inductances to demonstrate operation of the compensation
system.
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NCP4303A/B
Figure 33. ZCD Sensing Circuitry Functionality
When the voltage on the secondary winding of the SMPS
reverses, the body diode of M1 starts to conduct current and
the voltage of M1’s drain drops approximately to −1 V. The
CS pin sources current of 100 mA that creates a voltage drop
on the Rshift_cs resistor. Once the voltage on the CS pin is
lower than Vth_cs_on threshold, M1 is turned on. Because of
parasitic impedances, significant ringing can occur in the
application. To overcome sudden turn−off due to mentioned
ringing, the minimum conduction time of the SR MOSFET
is activated. Minimum conduction time can be adjusted
using R_Min_Ton resistor.
The SR MOSFET is turned−off as soon as the voltage on
the CS pin is higher than Vth_cs_off. For the same ringing
reason, a minimum off time timer is asserted once the
turn−off is detected. The minimum off time can be
externally adjusted using R_Min_Toff resistor. MOSFET M1
channel conducts when the secondary current decreases,
therefore the turn−off time depends on its RDS(on). The 0 mV
threshold provides an optimum switching period usage
while keeping enough time margin for the gate turn off. The
Rshift_cs resistor provides the designer with the possibility to
modify (increase) the actual turn−off current threshold.
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NCP4303A/B
Figure 34. ZCD Comparators Thresholds and Blanking Periods Timing
If no Rshift_cs resistor is used, the turn−on and turn−off
thresholds are fully given by the CS input specification
(please refer to parametric table). Once non−zero Rshift_cs
resistor is used, both thresholds move down (i.e. higher
MOSFET turn off current) as the CS pin offset current
causes a voltage drop that is equal to:
V_Rshift_cs + Rshift_cs * Ishift_cs
Note that Rshift_cs impact on turn−on threshold is less critical
compare to turn−off threshold.
If using a SR MOSFET in TO220 package (or other
package which features leads), the parasitic inductance of
the package leads causes a turn−off current threshold
increase. This is because current that flows through the SR
MOSFET has quite high di(t)/dt that induces error voltage
on the SR MOSFET leads inductance. This error voltage,
that is proportional to the secondary current derivative,
shifts the CS input voltage to zero when significant current
still flows through the channel. Zero current threshold is thus
detected when current still flows through the SR MOSFET
channel – please refer to Figure 35 for better understanding.
As a result, the SR MOSFET is turned−off prematurely and
the efficiency of the SMPS is not optimized.
(eq. 1)
Final turn−on and turn off thresholds can be then calculated
as:
VCS_turn_on + Vth_CS_on * (Rshift_cs * Ishift_cs)
(eq. 2)
VCS_turn_off + Vth_CS_off * (Rshift_cs * Ishift_cs)
(eq. 3)
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NCP4303A/B
Figure 35. Waveforms from SR System Using MOSFET in TO220 Package without Parasitic Inductance
Compensation – SR MOSFET Channel Conduction Time is Reduced
Note that the efficiency impact of the error caused by
parasitic inductance increases with lower RDS(on)
MOSFETs and/or higher operating frequency.
The NCP4303 offers a way to compensate for MOSFET
parasitic inductances effect − refer to Figure 36.
Figure 36. Package Parasitic Inductances Compensation Principle
Dedicated input (COMP) offers the possibility to use an
external compensation inductance (wire strap or PCB). If
the value of this compensation inductance is Lcomp = Ldrain
+ Lsource, the compensation voltage created on this
inductance is exactly the same as the sum of error voltages
created on drain and source parasitic inductances i.e. VLdrain
+ VLsource. The internal analog inverter (Figure 33) inverts
compensation voltage Vl_comp and offsets the current
sense comparator turn−off threshold. The current sense
comparator thus “sees” between its terminals a voltage that
would be seen on the SR MOSFET channel resistance in
case the lead inductances wouldn’t exist. The current sense
comparator of the NCP4303 is thus able to detect the
secondary current zero crossing very precisely. More over,
the secondary current turn−off threshold is then di(t)/t
independent thus the NCP4303 allows to increase operating
frequency of the SR system. One should note that the
parasitic resistance of compensation inductance should be as
low as possible compared to the SR MOSFET channel and
leads resistance otherwise compensation is not efficient.
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NCP4303A/B
Typical value of compensation inductance for a TO220
package is 7 nH. The parasitic inductance can differ depends
on how much are the leads shortened during the assembly
process. The compensation inductance design has to be done
with enough margin to overcome situation that the system
will become overcompensated due to packaging and
assembly process variations. Waveforms from the
application with compensated SR system can be seen in
Figure 37. One can see the conduction time has been
significantly increased and turn−off current reduced.
Figure 37. Waveforms from SR System Using MOSFET in TO220 Package with Parasitic Inductance
Compensation – SR MOSFET Channel Conduction Time Optimized
comparator. Ideally the CS turn–off comparator should
detect voltage that is caused by secondary current directly on
the SR MOSFET channel resistance. Practically this is not
possible because of the bonding wires, leads and soldering.
To assure the best efficiency results, a Kelvin connection of
the SR controller to the power circuitry should be
implemented (i.e. GND pin should be connected to the SR
MOSFET source soldering point and current sense pin
should be connected to the SR MOSFET drain soldering
point). Any impact of PCB parasitic elements on the SR
controller functionality is then avoided. Figures 38 and 39
show examples of SR system layouts using parasitic
inductance compensation (i.e. for low RDS(on) MOSFET in
TO220 package ) and not using compensation (i.e. for higher
RDS(on) MOSFET in TO220 package or SMT package
MOSFETs ).
Note that using the compensation system is only
beneficial in applications that are using a low RDS(on)
MOSFET in non−SMT package. Using the compensation
method allows for optimized efficiency with a standard
TO220 package that in turn results in reduced costs, as the
SMT MOSFETs usually require reflow soldering process
and more expensive PCB.
From the above paragraphs and parameter tables it is
evident that turn−off threshold precision is quite critical. If
we consider a SR MOSFET with RDS(on) of 1 mW, the 1 mV
error voltage on the CS pin results in a 1 A turn−off current
threshold difference. Thus the PCB layout is very critical
when implementing the SR system. Note that the CS
turn−off comparator as well as compensation inputs are
referred to the GND pin. Any parasitic impedance (resistive
or inductive − talking about mW and nH values) can cause
a high error voltage that is then evaluated by the CS
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NCP4303A/B
Figure 38. Recommended Layout When Parasitic
Inductance Compensation is Used
Figure 39. Recommended Layout When Parasitic
Inductance Compensation is Not Used
Trigger/Disable input
disabled from the end of the minimum off time period to the
end of the minimum on time period. This technique is used
to:
a) Overcome false turn−off of the gate driver in case the
synchronization pulse is too wide and comes twice per
switching period (in HB and HB LLC applications).
b) Increase trigger input noise immunity against the parasitic
ringing that is present in the SMPS layout during the turn on
process.
The NCP4303 features an ultrafast trigger input that
exhibits a typically of 12 ns delay from its activation to the
turn−off of the SR MOSFET. The main purpose of this input
is to turn−off the SR MOSFET in applications operating in
CCM mode via a signal coming from the primary side. The
primary trigger signal rising edge should come to the trigger
input before the secondary voltage reverses. Thus the driver
signal for primary switch should be delayed – refer to figure
46 for one possible method of delaying the primary driving
signal in CCM flyback topology. The trigger signal is
Figure 40. Trigger Input Internal Connection
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NCP4303A/B
Figure 41. Trigger Input Functionality Waveforms
standby mode. The IC consumption is reduced to 390 mA
during the standby mode. When trigger input voltage is
decreased again the device recovers operation in 500 ns. If
the IC is enabled in the time the current sense input voltage
is negative (secondary current flows through the Shottky or
body diode) the IC waits for another switching cycle to
turn−on the SR MOSFET – refer to Figures 42, 43 and 44.
The NCP4303 operation can be disabled using the
trigger/disable input. If the trigger/disable input is pulled up
(above 1.5 V) the driver is disabled immediately. In some
cases, the driver is activated one more time by the current
sense because the trigger signal is still blanked. This final
drive pulse lasts only for the minimum on time period. If the
trigger signal is high for more than 100 ms, the driver enters
Figure 42. Operating Waveforms for the Trig/Disable Input – Device Sleep Mode Transition – Case 1
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NCP4303A/B
Figure 43. Operating Waveforms for the Trig/Disable Input – Device Sleep Mode Transition – Case 2
Figure 44. Operating Waveforms for the Trig/Disable Input –Wake−up from Sleep Mode
If the trigger signal comes periodically and the trigger
pulse overlaps the SR MOSFET drain positive voltage (i.e.
overlaps the whole SR MOSFET body diode off time
period), the driver is disabled for the next cycle – refer to
Figure 45.
Figure 45. Operating Waveforms for the Trig/Disable Input with a Trigger Signal that is Periodical and Overlaps
CS (SR MOSFET Vds) High Level
Note that the trigger input is an ultrafast input that doesn’t
feature any internal filtering and reacts even on very narrow
voltage pulses. Thus it is wise to keep this input on a low
impedance path and provide it with a clean triggering signal.
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NCP4303A/B
A typical application schematic of a CCM flyback
converter with the NCP4303 driver can be seen in Figure 46.
In this application the trigger signal is taken directly from the
flyback controller driver output and transmitted to the
secondary side by pulse transformer TR2. Because the
trigger input is rising edge sensitive, it is not necessary to
transmit the entire primary driver pulse to the secondary.
The coupling capacitor C5 is used to allow pulse transformer
core reset and also to prepare a needle pulse (a pulse with
width lower than 100 ns) to be transmitted to the NCP4303
trigger input. The advantage of needle trigger pulse usage is
that the required volt−second product of the pulse
transformer is very low and that allows the designer to use
very small and cheap magnetics. The trigger transformer can
be for instance prepared on a small toroidal ferrite core with
diameter of 8 mm. Proper safety insulation between primary
and secondary sides can be easily assured by using triple
insulated wire for one or even both windings.
The primary MOSFET gate voltage rising edge is delayed
by external circuitry consisting of transistors Q1, Q2 and
surrounding components. The primary MOSFET is thus
turned−on with a slight delay so that the secondary
controller turns−off the SR MOSFET by trigger signal prior
to the primary switching. This method reduces the
commutation losses and the SR MOSFET drain voltage
spike, which results in improved efficiency.
It is also possible to use capacitive coupling (use
additional capacitor with safety insulation) between the
primary and secondary to transmit the trigger signal. We do
not recommend this technique as the parasitic capacitive
currents between primary and secondary may affect the
trigger signal and thus overall system functionality.
Figure 46. Typical Application Schematic When NCP4303 is Used in CCM Flyback Converter
Minimum Ton and Toff Adjustment
timers avoid false triggering on the CS input after the
MOSFET is turned on or off. The adjustment is based on an
internal timing capacitance and external resistors connected
to the GND pin – refer to Figure 47 for better understanding.
The NCP4303 offers adjustable minimum ON and OFF
time periods that ease the implementation of the
synchronous rectification system in a power supply. These
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NCP4303A/B
Figure 47. Internal Connection of the Min_Ton Generator (the Min_Toff Works in the Same Way)
Current through the Min_Ton adjust resistor can be
calculated as:
I R_Ton_min +
V ref
internal capacitor size would be too high if we would use
directly IR_Ton_min current thus this current is decreased by
the internal current mirror ratio. One can then calculate the
minimum Ton and Toff blanking periods using below
equations:
(eq. 4)
R Ton_min
As the same current is used for the internal timing
capacitor (Ct) charging, one can calculate the minimum
on−time duration using this equation.
T on_min + C t @
V ref
I R_Ton_min
+ Ct +
V
ref
V
ref
T on_min + 9.82 * 10 −11 * R T_on_min ) 4.66 * 10 −8 [ms]
(eq. 6)
T off_min + 9.56 * 10 −11 * R T_off_min ) 5.397 * 10 −8 [ms]
(eq. 7)
R Ton_min
Note that the internal timing comparator delay affects the
accuracy of Equations 6 and 7 when Ton/Toff times are
selected near to their minimum possible values. Please refer
to Figure 48 and 49 for measured minimum on and off time
charts.
(eq. 5)
+ C t @ R Ton_min
6
6
5
5
4
4
Toff_MIN (ms)
Ton_MIN (ms)
As can be seen from Equation 5, the minimum ON and
OFF times are independent of the Vref or VCC level. The
3
2
1
0
3
2
1
0
10
20
30
40
50
0
60
0
10
20
30
40
50
Rmin_Ton (kW)
Rmin_Toff (kW)
Figure 48. Min Ton Adjust Characteristic
Figure 49. Min Toff Adjust Characteristic
The absolute minimum Ton duration is internally clamped
to 300 ns and minimum Toff duration to 600 ns in order to
prevent any potential issues with the minimum Ton and/or
Toff input being shorted to GND.
60
Some applications may require adaptive minimum on and
off time blanking periods. With NCP4303 it is possible to
modulate blanking periods by using an external NPN
transistor – refer to Figure 50. The modulation signal can be
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NCP4303A/B
derived based on the load current or feedback regulator
voltage.
Figure 50. Possible Connection for Min Ton and Toff Modulation
In LLC applications with a very wide operating frequency
range it is necessary to have very short minimum on time and
off time periods in order to reach the required maximum
operating frequency. However, when a LLC converter
operates under low frequency, the minimum off time period
may then be too short. To overcome possible issues with the
LLC operating under low line and light load conditions, one
can prolong the minimum off time blanking period by using
resistors Rdrain1 and Rdrain2 connected from the opposite SR
MOSFET drain – refer to Figure 51.
Figure 51. Possible Connection for Min Toff Prolongation in LLC Applications with Wide Operating Frequency
Range
Note that Rdrain1 and Rdrain2 should be designed in such
a way that the maximum pulse current into the Min_Toff
adjust pin is below 10 mA. Voltage on the min Toff and Ton
pins is clamped by internal zener protection to 10 V.
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NCP4303A/B
Power Dissipation Calculation
always operates under Zero Voltage Switching (ZVS)
conditions when implemented in a synchronous
rectification system.
The following steps show how to approximately calculate
the power dissipation and DIE temperature of the
NCP4303A/B controller. Note that real results can vary due
to the effects of the PCB layout on the thermal resistance.
It is important to consider the power dissipation in the
MOSFET driver of a SR system. If no external gate resistor
is used and the internal gate resistance of the MOSFET is
very low, nearly all energy losses related to gate charge are
dissipated in the driver. Thus it is necessary to check the SR
driver power losses in the target application to avoid over
temperature and to optimize efficiency.
In SR systems the body diode of the SR MOSFET starts
conducting before turn on because the Vth_cs_on threshold
level is below 0 V. On the other hand, the SR MOSFET turn
off process always starts before the drain to source voltage
rises up significantly. Therefore, the MOSFET switch
Step 1 – MOSFET gate to source capacitance:
During ZVS operation the gate to drain capacitance does
not have a Miller effect like in hard switching systems
because the drain to source voltage is close to zero and its
change is negligible.
C iss + C gs ) C gd
C rss + C gd
C oss + C ds ) C gd
Figure 52. Typical MOSFET Capacitances Dependency on Vds and Vgs Voltages
to source voltage − in this case the NCP4303A should be
used.
The total driving loss can be calculated using the selected
gate driver clamp voltage and the input capacitance of the
MOSFET:
Therefore, the input capacitance of a MOSFET operating
in ZVS mode is given by the parallel combination of the gate
to source and gate to drain capacitances (i.e. Ciss capacitance
for given gate to source voltage). The total gate charge,
Qg_total, of most MOSFETs on the market is defined for hard
switching conditions. In order to accurately calculate the
driving losses in a SR system, it is necessary to determine the
gate charge of the MOSFET for operation specifically in a
ZVS system. Some manufacturers define this parameter as
Qg_ZVS. Unfortunately, most datasheets do not provide this
data. If the Ciss (or Qg_ZVS) parameter is not available then
it will need to be measured. Please note that the input
capacitance is not linear (as shown figure 52) and it needs to
be characterized for a given gate voltage clamp level.
P DRV_total + V CC @ V clamp @ C g_ZVS @ f SW
(eq. 8)
Where:
Vcc is the NCP4303x supply voltage
Vclamp is the driver clamp voltage
Cg_ZVS is the gate to source capacitance of the MOSFET in
ZVS mode
fsw is the switching frequency of the target application
The total driving power loss won’t only be dissipated in
the IC, but also in external resistances like the external gate
resistor (if used) and the MOSFET internal gate resistance
(Figure 53). Because NCP4303A/B features a clamped
driver, it’s high side portion can be modeled as a regular
driver switch with equivalent resistance and a series voltage
source. The low side driver switch resistance does not drop
immediately at turn−off, thus it is necessary to use an
equivalent value (Rdrv_low_eq) for calculations. This method
simplifies power losses calculations and still provides
acceptable accuracy. Internal driver power dissipation can
then be calculated using Equation 9:
Step 2 – Gate drive losses calculation:
Gate drive losses are affected by the gate driver clamp
voltage. Gate driver clamp voltage selection depends on the
type of MOSFET used (threshold voltage versus channel
resistance). The total power losses (driving loses and
conduction losses) should be considered when selecting the
gate driver clamp voltage. Most of today’s MOSFETs for SR
systems feature low RDS(on) for 5 V Vgs voltage and thus it
is beneficial to use NCP4303B. However, there is still a big
group of MOSFETs on the market that require higher gate
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NCP4303A/B
Figure 53. Equivalent Schematic of Gate Drive Circuitry
P DRV_IC +
ǒ
Ǔ
R drv_low_eq
1
@ C g_ZVS @ V clamp 2 @ f SW @
) C g_ZVS @ V clamp @ f SW @ ǒV CC * V clampǓ
2
R drv_low_eq ) R g_ext ) R g_int
)
ǒ
R drv_high_eq
Ǔ
(eq. 9)
1
@ C g_ZVS @ V clamp 2 @ f SW @
2
R drv_high_eq ) R g_ext ) R g_int
Step 4 – IC DIE Temperature Arise Calculation:
Where:
Rdrv_low_eq is the NCP4303x driver low side switch
equivalent resistance (1.55 W)
Rdrv_high_eq is the NCP4303x driver high side switch
equivalent resistance (7 W)
Rg_ext is the external gate resistor (if used)
Rg_int is the internal gate resistance of the MOSFET
The DIE temperature can be calculated now that the total
internal power losses have been determined (driver losses
plus internal IC consumption losses). The SO−8 package
thermal resistance is specified in the maximum ratings table
for a 35 mm thin copper layer with no extra copper plates on
any pin (i.e. just 0.5 mm trace to each pin with standard
soldering points are used).
The DIE temperature is calculated as:
Step 3 – IC Consumption Calculation:
In this step, power dissipation related to the internal IC
consumption is calculated. This power loss is given by the
ICC current and the IC supply voltage. The ICC current
depends on switching frequency and also on the selected
min Ton and Toff periods because there is current flowing out
from the min Ton and Toff pins. The most accurate method
for calculating these losses is to measure the Icc current when
Cload = 0 nF and the IC is switching at the target frequency
with given Min_Ton and Min_Toff adjust resistors. Refer
also to Figure 54 for typical IC consumption charts when the
driver is not loaded. IC consumption losses can be calculated
as:
P ICC + V CC @ I CC
T DIE + ǒP DRV_IC ) P ICCǓ @ R qJ*A ) T A (eq. 11)
Where:
PDRV_IC is the IC driver internal power dissipation
PIcc is the IC control internal power dissipation
RqJA is the thermal resistance from junction to ambient
TA is the ambient temperature
(eq. 10)
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NCP4303A/B
400
160
POWER CONSUMTION (mW)
NCP4303B,
VCC = 30 V
140
120
NCP4303A,
VCC = 30 V
100
80
NCP4303A,
VCC = 12 V
60
40
NCP4303B,
VCC = 12 V
20
0
50
100
150 200 250 300 350 400
OPERATING FREQUENCY (kHz)
450
350
300
NCP4303A,
VCC = 30 V
250
200
NCP4303B,
VCC = 30 V
NCP4303A,
VCC = 12 V
150
100
NCP4303B,
VCC = 12 V
50
0
50
500
100
150
200
250
300
350
400
450
OPERATING FREQUENCY (kHz)
Figure 54. IC Power Consumption as a
Function of Frequency for Cload = 0 nF,
Rton_min = Rtoff_min = 5 kW
Figure 55. IC Power Consumption as a
Function of Frequency for Cload = 1 nF,
Rton_min = Rtoff_min = 5 kW
800
POWER CONSUMTION (mW)
POWER CONSUMTION (mW)
180
NCP4303A,
VCC = 30 V
700
NCP4303B,
VCC = 30 V
600
500
400
300
200
NCP4303B,
VCC = 12 V
100
0
50
NCP4303A,
VCC = 12 V
100
150
200
250
300
350
400
450
500
OPERATING FREQUENCY (kHz)
Figure 56. IC Power Consumption as a Function of Frequency for Cload = 10 nF, Rton_min = Rtoff_min = 5 kW
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500
NCP4303A/B
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AJ
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
SOLDERING FOOTPRINT*
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
The products described herein (NCP4303) has patents pendings.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent
rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur.
Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries,
affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury
or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an
Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
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NCP4303/D