LTC3728L-1 Dual, 550kHz, 2-Phase Synchronous Regulator DESCRIPTION FEATURES n n n n n n n n n n n n n n n Dual, 180° Phased Controllers Reduce Required Input Capacitance and Power Supply Induced Noise OPTI-LOOP® Compensation Minimizes COUT ±1.5% Output Voltage Accuracy Power Good Output Voltage Indicator Phase-Lockable Fixed Frequency 250kHz to 550kHz Over Current Latchoff Disabled Wide VIN Range: 4.5V to 28V (35V for LTC3728LI-1) Operation Very Low Dropout Operation: 99% Duty Cycle Adjustable Soft-Start Current Ramping Foldback Output Current Limiting Output Overvoltage Protection Low Shutdown IQ: 20μA 5V and 3.3V Standby Regulators 3 Selectable Operating Modes: Constant Frequency, Burst Mode® Operation and PWM 5mm × 5mm QFN and 28-Lead Narrow SSOP Packages The LTC®3728L-1 is a dual high performance step-down switching regulator controller that drives all N-channel synchronous power MOSFET stages. A constant frequency current mode architecture allows phase-lockable frequency of up to 550kHz. Power loss and noise due to the ESR of the input capacitors are minimized by operating the two controller output stages out of phase. The LTC3728L-1 is identical to the LTC3728L except that the LTC3728L-1 lacks the over current latchoff feature. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The precision 0.8V reference and power good output indicator are compatible with future microprocessor generations, and a wide 4.5V to 28V (30V maximum/ 35V for LTC3728LI-1) input supply range encompasses all battery chemistries. A RUN/SS pin for each controller provides soft-start. Current foldback limits MOSFET dissipation during shortcircuit conditions. The FCB mode pin can select among Burst Mode operation, constant frequency mode and continuous inductor current mode or regulate a secondary winding. The LTC3728L-1 includes a power good output pin that indicates when both outputs are within 7.5% of their designed set point. APPLICATIONS n n n n n Notebook and Palmtop Computers Telecom Systems Portable Instruments Battery-Operated Digital Devices DC Power Distribution Systems L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst Mode and OPTI-LOOP are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5929620, 6177787, 6144199, 5471178, 5994885, 6100678. TYPICAL APPLICATION + 4.7μF D3 TG1 L1 3.2μH CB1, 0.1μF RSENSE1 0.01Ω + R2 105k COUT1 1% 47μF 6V SP BOOST2 SW1 SW2 LTC3728L-1 PLLIN R1 20k 1% L2 3.2μH CB2, 0.1μF BG2 PGND SENSE1+ SENSE2+ SENSE1– VOSENSE1 ITH1 SENSE2– VOSENSE2 ITH2 1000pF CC1 220pF RC1 15k M2 TG2 BOOST1 BG1 fIN 500kHz VOUT1 5V 5A D4 VIN PGOOD INTVCC M1 RSENSE2 0.01Ω 1000pF RUN/SS1 SGND RUN/SS2 CSS1 0.1μF VIN 5.2V TO 28V CIN 22μF 50V CERAMIC 1μF CERAMIC CSS2 0.1μF CC2 220pF RC2 15k R3 20k 1% R4 63.4k 1% M1, M2: FDS6982S COUT 56μF 6V SP + VOUT2 3.3V 5A 3728l1 F01 Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter 3728l1fc 1 LTC3728L-1 ABSOLUTE MAXIMUM RATINGS (Note 1) Input Supply Voltage (VIN) .........................30V to – 0.3V Input Supply Voltage (VIN) LTC3728LI-1 ....35V to – 0.3V Top Side Driver Voltages (BOOST1, BOOST2) ............................... 36V to –0.3V (BOOST1, BOOST2) LTC3728LI-1 .......... 38V to –0.3V Switch Voltage (SW1, SW2) ........................... 30V to –5V Switch Voltage (SW1, SW2) LTC3728LI-1... 35V to –0.3V INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1), (BOOST2-SW2), PGOOD ......................... 7V to –0.3V INTVCC, EXTVCC LTC3728LI-1 ....................... 8V to –0.3V SENSE1+, SENSE2+, SENSE1–, SENSE2– Voltages .....................(1.1)INTVCC to –0.3V PLLIN, PLLFLTR, FCB, Voltage .............. INTVCC to –0.3V ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages .... 2.7V to –0.3V Peak Output Current <10μs (TG1, TG2, BG1, BG2) .....3A INTVCC Peak Output Current ..................................40mA Operating Temperature Range (Note 7).... –40°C to 85°C Junction Temperature (Note 2) ............................. 125°C Storage Temperature Range................... –65°C to 125°C Reflow Peak Body Temperature (UH Package) ...... 260°C Lead Temperature (Soldering, 10 sec) GN Package ...................................................... 300°C PIN CONFIGURATION 6 FCB 7 22 EXTVCC ITH1 8 21 INTVCC SGND 9 20 PGND ITH2 11 23 BG1 17 SW2 16 TG2 SENSE2+ 14 15 RUN/SS2 GN PACKAGE 28-LEAD NARROW PLASTIC SSOP TJMAX = 125°C, θJA = 95°C/W SW1 TG1 PGOOD 20 INTVCC SGND 6 19 PGND 3.3VOUT 7 18 BG2 ITH2 8 18 BOOST2 SENSE2– 13 21 EXTVCC 33 ITH1 5 19 BG2 VOSENSE2 12 22 BG1 FCB 4 17 BOOST2 9 10 11 12 13 14 15 16 VOSENSE2 3.3VOUT 10 23 VIN PLLIN 3 NC PLLIN 24 BOOST1 PLLFLTR 2 SW2 24 VIN RUN/SS1 5 32 31 30 29 28 27 26 25 VOSENSE1 1 TG2 PLLFLTR NC 25 BOOST1 RUN/SS2 4 SENSE2+ 26 SW1 VOSENSE1 SENSE1– 27 TG1 3 SENSE2– 2 SENSE– NC SENSE1+ NC RUN/SS1 28 PGOOD SENSE1+ TOP VIEW TOP VIEW 1 UH PACKAGE 32-LEAD (5mm s 5mm) PLASTIC QFN TJMAX = 125°C, θJA = 34°C/W EXPOSED PAD (PIN 33) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING LTC3728LEGN-1#PBF LTC3728LEGN-1#TRPBF LTC3728LEGN-1 LTC3728LIGN-1#PBF LTC3728LIGN-1#TRPBF LTC3728LIGN-1 LTC3728LEUH-1#PBF LTC3728LEUH-1#TRPBF 728LE1 LTC3728LIUH-1#PBF LTC3728LIUH-1#TRPBF 3728L1 LEAD BASED FINISH TAPE AND REEL PART MARKING LTC3728LEGN-1 LTC3728LEGN-1#TR LTC3728LEGN-1 LTC3728LIGN-1 LTC3728LIGN-1#TR LTC3728LIGN-1 LTC3728LEUH-1 LTC3728LEUH-1#TR 728LE1 LTC3728LIUH-1 LTC3728LIUH-1#TR 3728L1 Consult LTC Marketing for parts specified with wider operating temperature ranges. PACKAGE DESCRIPTION 28-Lead Narrow Plastic SSOP 28-Lead Narrow Plastic SSOP 32-Lead (5mm × 5mm) Plastic DFN 32-Lead (5mm × 5mm) Plastic DFN PACKAGE DESCRIPTION 28-Lead Narrow Plastic SSOP 28-Lead Narrow Plastic SSOP 32-Lead (5mm × 5mm) Plastic DFN 32-Lead (5mm × 5mm) Plastic DFN TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C –40°C to 85°C –40°C to 85°C TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C –40°C to 85°C –40°C to 85°C For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3728l1fc 2 LTC3728L-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 0.788 0.800 0.812 V –5 –50 nA 0.002 0.02 %/V 0.1 – 0.1 0.5 –0.5 % % Main Control Loops VOSENSE1, 2 Regulated Feedback Voltage (Note 3); ITH1, 2 Voltage = 1.2V IVOSENSE1, 2 Feedback Current (Note 3) VREFLNREG Reference Voltage Line Regulation VIN = 3.6V to 30V (Note 3) VLOADREG Output Voltage Load Regulation (Note 3) Measured in Servo Loop; ΔITH Voltage = 1.2V to 0.7V Measured in Servo Loop; ΔITH Voltage = 1.2V to 2.0V gm1, 2 Transconductance Amplifier gm gmGBW1, 2 Transconductance Amplifier GBW ITH1, 2 = 1.2V; (Note 3) IQ Input DC Supply Current Normal Mode Shutdown (Note 4) VIN = 15V; EXTVCC Tied to VOUT1; VOUT1 = 5V VRUN/SS1, 2 = 0V ● ● ● ITH1, 2 = 1.2V; Sink/Source 5μA; (Note 3) 1.3 mmho 3 ● MHz 450 20 35 μA μA 0.76 0.800 0.84 V – 0.50 –0.18 –0.1 μA 4.3 4.8 V 3.5 4 V 0.86 0.88 V VFCB Forced Continuous Threshold IFCB Forced Continuous Pin Current VFCB = 0.85V VBINHIBIT Burst Inhibit (Constant Frequency) Threshold Measured at FCB pin UVLO Undervoltage Lockout VIN Ramping Down ● VOVL Feedback Overvoltage Lockout Measured at VOSENSE1, 2 ● ISENSE Sense Pins Total Source Current (Each Channel); VSENSE1–, 2– = VSENSE1+, 2+ = 0V –90 –60 μA DFMAX Maximum Duty Factor In Dropout 98 99.4 % IRUN/SS1, 2 Soft-Start Charge Current VRUN/SS1, 2 = 1.9V 0.5 1.2 μA VRUN/SS1, 2 ON RUN/SS Pin ON Threshold 0.84 VRUN/SS1, VRUN/SS2 Rising 1.0 1.5 2.0 V VSENSE(MAX) Maximum Current Sense Threshold VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V 65 62 75 75 85 88 mV mV TG1, 2 tr TG1, 2 tf TG Transition Time: Rise Time Fall Time (Note 5) CLOAD = 3300pF CLOAD = 3300pF 55 55 100 100 ns ns BG1, 2 tr BG1, 2 tf BG Transition Time: Rise Time Fall Time (Note 5) CLOAD = 3300pF CLOAD = 3300pF 45 45 100 90 ns ns TG/BG t1D Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time CLOAD = 3300pF Each Driver 80 ns BG/TG t2D Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time CLOAD = 3300pF Each Driver 80 ns tON(MIN) Minimum On-Time Tested with a Square Wave (Note 6) 100 ns ● INTVCC Linear Regulator VINTVCC Internal VCC Voltage 6V < VIN < 30V, VEXTVCC = 4V 5.0 5.2 V VLDO INT INTVCC Load Regulation ICC = 0 to 20mA, VEXTVCC = 4V 4.8 0.2 2.0 % VLDO EXT EXTVCC Voltage Drop ICC = 20mA, VEXTVCC = 5V 100 200 mV VEXTVCC EXTVCC Switchover Voltage ICC = 20mA, EXTVCC Ramping Positive VLDOHYS EXTVCC Hysteresis ● 4.5 4.7 V 0.2 V Oscillator and Phase-Locked Loop fNOM Nominal Frequency VPLLFLTR = 1.2V 360 400 440 kHz fLOW Lowest Frequency VPLLFLTR = 0V 230 260 290 kHz fHIGH Highest Frequency VPLLFLTR ≥ 2.4V 480 550 590 kHz RPLLIN PLLIN Input Resistance 50 kΩ 3728l1fc 3 LTC3728L-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted. SYMBOL PARAMETER CONDITIONS I PLLFLTR Phase Detector Output Current Sinking Capability Sourcing Capability fPLLIN < fOSC fPLLIN > fOSC MIN TYP MAX UNITS –15 15 μA μA 3.3V Linear Regulator ● V3.3OUT 3.3V Regulator Output Voltage No Load 3.35 3.45 V V3.3IL 3.3V Regulator Load Regulation I3.3 = 0 to 10mA 3.2 0.5 2 % V3.3VL 3.3V Regulator Line Regulation 6V < VIN < 30V 0.05 0.2 % I3.3LEAK Leakage Current in Shutdown VRUN/SS1 = 0V; VRUN/SS2 = 0V, VIN = 3V 10 50 μA VPGL PGOOD Voltage Low IPGOOD = 2mA 0.1 0.3 V IPGOOD PGOOD Leakage Current VPGOOD = 5V ±1 μA VPG PGOOD Trip Level, Either Controller VOSENSE with Respect to Set Output Voltage VOSENSE Ramping Negative VOSENSE Ramping Positive – 9.5 9.5 % % ● PGOOD Output –6 6 –7.5 7.5 Note 5: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 6: The minimum on-time condition is specified for an inductor peak-to-peak ripple current ≥ 40% of IMAX (see minimum on-time considerations in the Applications Information section). Note 7: The LTC3728LE-1 are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3728LI-1 is guaranteed to meet performance specifications over the full –40°C to 85°C operating temperature range Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC3728LUH-1: TJ = TA + (PD • 34°C/W) LTC3728LGN-1: TJ = TA + (PD • 95°C/W) Note 3: The IC is tested in a feedback loop that servos VITH1, 2 to a specified voltage and measures the resultant VOSENSE1, 2. Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. TYPICAL PERFORMANCCE CHARACTERISTICS Efficiency vs Output Current and Mode (Figure 13) Efficiency vs Output Current (Figure 13) 100 VIN = 7V 70 50 40 30 FORCED CONTINUOUS MODE (PWM) CONSTANT FREQUENCY (BURST DISABLE) 20 VIN = 15V VOUT = 5V f = 250kHz 10 0 0.001 90 90 EFFICIENCY (%) EFFICIENCY (%) 80 60 100 100 Burst Mode OPERATION 0.1 0.01 1 OUTPUT CURRENT (A) 10 3728L1 G01 80 VIN = 10V VIN = 15V EFFICIENCY (%) 90 Efficiency vs Input Voltage (Figure 13) VIN = 20V 70 80 70 60 60 VOUT = 5V IOUT = 3A f = 250kHz VOUT = 5V f = 250kHz 50 0.001 50 0.1 0.01 1 OUTPUT CURRENT (A) 10 3728L1 G02 5 25 15 INPUT VOLTAGE (V) 35 3728L1 G03 3728l1fc 4 LTC3728L-1 TYPICAL PERFORMANCE CHARACTERISTICS Supply Current vs Input Voltage and Mode (Figure 13) 200 800 600 BOTH CONTROLLERS ON 400 200 SHUTDOWN 0 0 5 150 100 50 0 20 15 10 INPUT VOLTAGE (V) 25 5.05 INTVCC AND EXTVCC SWITCH VOLTAGE (V) EXTVCC VOLTAGE DROP (mV) 1000 SUPPLY CURRENT (μA) INTVCC and EXTVCC Switch Voltage vs Temperature EXTVCC Voltage Drop 30 0 10 20 CURRENT (mA) 30 4.85 4.80 EXTVCC SWITCHOVER THRESHOLD 4.75 50 25 75 0 TEMPERATURE (°C) 100 3728L1 G06 Maximum Current Sense Threshold vs Duty Factor 75 ILOAD = 1mA 125 Maximum Current Sense Threshold vs Percent of Nominal Output Voltage (Foldback) 80 70 5.0 60 4.8 4.7 50 VSENSE (mV) 4.9 VSENSE (mV) INTVCC VOLTAGE (V) 4.90 3728L1 G05 Internal 5V LDO Line Regulation 25 4.6 50 40 30 20 4.5 10 4.4 0 5 20 15 10 INPUT VOLTAGE (V) 25 0 30 0 20 40 60 DUTY FACTOR (%) 80 3728L1 G07 80 0 100 50 0 100 25 75 PERCENT ON NOMINAL OUTPUT VOLTAGE (%) 3728L1 G08 Maximum Current Sense Threshold vs VRUN/SS (Soft-Start) 3728L1 G09 Current Sense Threshold vs ITH Voltage Maximum Current Sense Threshold vs Sense Common Mode Voltage 90 80 VSENSE(CM) = 1.6V 80 70 76 40 60 VSENSE (mV) VSENSE (mV) 60 VSENSE (mV) 4.95 4.70 –50 –25 40 3728L1 G04 5.1 INTVCC VOLTAGE 5.00 72 68 50 40 30 20 10 20 0 64 –10 –20 0 0 1 2 3 4 5 6 VRUN/SS (V) 3728L1 G10 60 –30 0 1 3 4 2 COMMON MODE VOLTAGE (V) 5 3728L1 G11 0 0.5 1 1.5 VITH (V) 2 2.5 3728L1 G12 3728l1fc 5 LTC3728L-1 TYPICAL PERFORMANCE CHARACTERISTICS Load Regulation VITH vs VRUN/SS 2.5 FCB = 0V VIN = 15V FIGURE 13 SENSE Pins Total Source Current 100 VOSENSE = 0.7V 2.0 –0.2 50 ISENSE (μA) –0.1 VITH (V) NORMALIZED VOUT (%) 0.0 1.5 1.0 –0.3 0 –50 0.5 –0.4 0 1 3 2 LOAD CURRENT (A) 4 0 5 0 2 1 3 4 5 6 –100 3728L1 G13 2 0 VRUN/SS (V) 4 3728L1 G14 Maximum Current Sense Threshold vs Temperature 3728L1 G15 Dropout Voltage vs Output Current (Figure 14) 80 4 6 VSENSE COMMON MODE VOLTAGE (V) RUN/SS Current vs Temperature 1.8 VOUT = 5V 1.6 76 74 3 RUN/SS CURRENT (μA) DROPOUT VOLTAGE (V) VSENSE (mV) 78 2 RSENSE = 0.015Ω 1 1.4 1.2 1.0 0.8 0.6 0.4 72 RSENSE = 0.010Ω 0.2 70 –50 0 –25 50 25 0 75 TEMPERATURE (°C) 100 0 125 0.5 1.0 1.5 2.0 2.5 3.0 OUTPUT CURRENT (A) 3.5 4.0 0 –50 0 25 50 75 TEMPERATURE (°C) 3728L1 G18 3728L1 G17 Soft-Start Up (Figure 13) 100 125 3728L1 G25 Load Step (Figure 13) VOUT 5V/DIV –25 Load Step (Figure 13) VOUT 200mV/DIV VOUT 200mV/DIV IL 2A/DIV IL 2A/DIV VRUN/SS 5V/DIV IL 2A/DIV VIN = 15V VOUT = 5V 5ms/DIV 3728L1 G19 VIN = 15V 20μs/DIV VOUT = 5V VPLLFLTR = 50V LOAD STEP = 0A TO 3A Burst Mode OPERATION 3728L1 G20 VIN = 15V 20μs/DIV VOUT = 5V VPLLFLTR = 50V LOAD STEP = 0A TO 3A CONTINUOUS MODE 3728L1 G21 3728l1fc 6 LTC3728L-1 TYPICAL PERFORMANCE CHARACTERISTICS Input Source/Capacitor Instantaneous Current (Figure 13) IIN 2A/DIV VIN 200mV/DIV Constant Frequency (Burst Inhibit) Operation (Figure 13) Burst Mode Operation (Figure 13) VOUT 20mV/DIV VOUT 20mV/DIV IL 0.5A/DIV IL 0.5A/DIV VSW1 10V/DIV VSW2 10V/DIV VIN = 15V 1μs/DIV VOUT1 = 5V, VOUT2 = 3.3V VPLLFLTR = 0V IOUT5 = IOUT3.3 = 2A 3728L1 G22 VIN = 15V VOUT = 5V VPLLFLTR = 0V VFCB = 5V IOUT = 20mA 3728L1 G23 10μs/DIV EXTVCC SWITCH RESISTANCE (Ω) CURRENT SENSE INPUT CURRENT (μA) 33 31 29 27 –25 50 25 0 75 TEMPERATURE (°C) 100 8 6 4 2 0 –50 125 –25 50 25 0 75 TEMPERATURE (°C) 3728L1 G26 3.50 UNDERVOLTAGE LOCKOUT (V) FREQUENCY (kHz) VPLLFLTR = 2.4V 500 300 VPLLFLTR = 1.2V VPLLFLTR = 0V 200 100 0 –50 –25 125 Undervoltage Lockout vs Temperature 700 400 100 3728L1 G27 Oscillator Frequency vs Temperature 600 3728L1 G24 10 VOUT = 5V 25 –50 2μs/DIV EXTVCC Switch Resistance vs Temperature Current Sense Pin Input Current vs Temperature 35 VIN = 15V VOUT = 5V VPLLFLTR = 0V VFCB = 5V IOUT = 20mA 50 25 75 0 TEMPERATURE (°C) 100 125 3728L1 G28 3.45 3.40 3.35 3.30 3.25 3.20 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 3728L1 G29 3728l1fc 7 LTC3728L-1 PIN FUNCTIONS VOSENSE1, VOSENSE2: Error Amplifier Feedback Input. Receives the remotely-sensed feedback voltage for each controller from an external resistive divider across the output. TG2, TG1: High Current Gate Drives for Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V superimposed on the switch node voltage SW. PLLFLTR: Filter Connection for Phase-Locked Loop. Alternatively, this pin can be driven with an AC or DC voltage source to vary the frequency of the internal oscillator. SW2, SW1: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external) voltage drop below ground to VIN. PLLIN: External Synchronization Input to Phase Detector. This pin is internally terminated to SGND with 50kΩ. The phase-locked loop will force the rising top gate signal of controller 1 to be synchronized with the rising edge of the PLLIN signal. BOOST2, BOOST1: Bootstrapped Supplies to the Top Side Floating Drivers. Capacitors are connected between the boost and switch pins and Schottky diodes are tied between the boost and INTVCC pins. Voltage swing at the boost pins is from INTVCC to (VIN + INTVCC). FCB: Forced Continuous Control Input. This input acts on both controllers and is normally used to regulate a secondary winding. Pulling this pin below 0.8V will force continuous synchronous operation. BG2, BG1: High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is from ground to INTVCC. ITH1, ITH2: Error Amplifier Output and Switching Regulator Compensation Point. Each associated channels’ current comparator trip point increases with this control voltage. SGND: Small Signal Ground. Common to both controllers, this pin must be routed separately from high current grounds to the common (–) terminals of the COUT capacitors. 3.3VOUT: Linear Regulator Output. Capable of supplying 10mA DC with peak currents as high as 50mA. NC: No Connect. SENSE2 –, SENSE1–: The (–) Input to the Differential Current Comparators. SENSE2+, SENSE1+: The (+) Input to the Differential Current Comparators. The ITH pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold. RUN/SS2, RUN/SS1: Combination of soft-start and run control inputs. A capacitor to ground at each of these pins sets the ramp time to full output current. Forcing either of these pins back below 1.0V causes the IC to shut down the circuitry required for that particular controller. PGND: Driver Power Ground. Connects to the sources of bottom (synchronous) N-channel MOSFETs, anodes of the Schottky rectifiers and the (–) terminal(s) of CIN. INTVCC: Output of the Internal 5V Linear Low Dropout Regulator and the EXTVCC Switch. The driver and control circuits are powered from this voltage source. Must be decoupled to power ground with a minimum of 4.7μF tantalum or other low ESR capacitor. EXTVCC: External Power Input to an Internal Switch Connected to INTVCC. This switch closes and supplies VCC power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection in Applications section. Do not exceed 7V on this pin. VIN: Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. PGOOD: Open-Drain Logic Output. PGOOD is pulled to ground when the voltage on either VOSENSE pin is not within ±7.5% of its set point. Exposed Pad (UH Package Only): Signal Ground. Must be soldered to the PCB, providing a local ground for the control components of the IC, and be tied to the PGND pin under the IC. 3728l1fc 8 LTC3728L-1 FUNCTIONAL DIAGRAM PLLIN FIN INTVCC PHASE DET DUPLICATE FOR SECOND CONTROLLER CHANNEL 50k BOOST VIN DB PLLFLTR DROP OUT DET CLK1 RLP OSCILLATOR CLK2 CLP – 0.86V S Q R Q BOT VOSENSE1 – 0.86V SWITCH LOGIC + 0.55V INTVCC BG COUT PGND B – + + SHDN VOSENSE2 CIN SW TOP ON BOT 0.74V + D1 – + CB FCB + PGOOD TG TOP VOUT RSENSE – INTVCC + 3V 4.5V 0.18μA R6 0.74V I1 – + – BINH FCB + R5 – 3.3VOUT + 0.8V + – ++ SLOPE COMP + 30k SENSE + – 30k SENSE 45k 45k 2.4V VREF – EA + – OV VIN VIN EXTVCC + – 5V LDO REG VOSENSE 0.80V SHDN RST 4(VFB) + R2 R1 0.86V ITH 1.2μA 6V INTVCC VFB + – 4.8V 5V INTVCC I2 – 3mV 0.86V 4(VFB) FCB – RUN SOFT START CC CC2 RC RUN/SS SGND (UH PACKAGE PAD) INTERNAL SUPPLY CSS 3728L1 F02 Figure 2 3728l1fc 9 LTC3728L-1 OPERATION (Refer to Functional Diagram) Main Control Loop The LTC3728L-1 is a constant frequency, current mode step-down controller with two channels operating 180 degrees out of phase. During normal operation, each top MOSFET is turned on when the clock for that channel sets the RS latch, and turned off when the main current comparator, I1, resets the RS latch. The peak inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin, which is the output of each error amplifier EA. The VOSENSE pin receives the voltage feedback signal, which is compared to the internal reference voltage by the EA. When the load current increases, it causes a slight decrease in VOSENSE relative to the 0.8V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. After the top MOSFET has turned off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during each off cycle through an external diode when the top MOSFET turns off. As VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about 400ns every tenth cycle to allow CB to recharge. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 1.2μA current source to charge soft-start capacitor CSS. When CSS reaches 1.5V, the main control loop is enabled with the ITH voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, the ITH pin voltage is gradually released allowing normal, full-current operation. When both RUN/SS1 and RUN/SS2 are low, all controller functions are shut down, including the 5V regulator. Low Current Operation The FCB pin is a multifunction pin providing two functions: 1) to provide regulation for a secondary winding by temporarily forcing continuous PWM operation on both controllers; and 2) to select between two modes of low current operation. When the FCB pin voltage is below 0.8V, the controller forces continuous PWM current mode operation. In this mode, the top and bottom MOSFETs are alternately turned on to maintain the output voltage independent of direction of inductor current. When the FCB pin is below VINTVCC – 2V but greater than 0.8V, the controller enters Burst Mode operation. Burst Mode operation sets a minimum output current level before inhibiting the top switch and turns off the synchronous MOSFET(s) when the inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a voltage threshold that will temporarily inhibit turn-on of both output MOSFETs until the output voltage drops. There is 60mV of hysteresis in the burst comparator B tied to the ITH pin. This hysteresis produces output signals to the MOSFETs that turn them on for several cycles, followed by a variable “sleep” interval depending upon the load current. The resultant output voltage ripple is held to a very small value by having the hysteretic comparator after the error amplifier gain block. Frequency Synchronization The phase-locked loop allows the internal oscillator to be synchronized to an external source via the PLLIN pin. The output of the phase detector at the PLLFLTR pin is also the DC frequency control input of the oscillator that operates over a 260kHz to 550kHz range corresponding to a DC voltage input from 0V to 2.4V. When locked, the PLL aligns the turn on of the top MOSFET to the rising edge of the synchronizing signal. When PLLIN is left open, the PLLFLTR pin goes low, forcing the oscillator to minimum frequency. Constant Frequency Operation When the FCB pin is tied to INTVCC, Burst Mode operation is disabled and the forced minimum output current requirement is removed. This provides constant frequency, discontinuous current (preventing reverse inductor current) operation over the widest possible output current range. This constant frequency operation is not as efficient 3728l1fc 10 LTC3728L-1 OPERATION (Refer to Functional Diagram) as Burst Mode operation, but does provide a lower noise, constant frequency operating mode down to approximately 1% of the designed maximum output current. turned off within 10μs and the pin is allowed to be pulled up by an external resistor to a source of up to 7V. Foldback Current Continuous Current (PWM) Operation Tying the FCB pin to ground will force continuous current operation. This is the least efficient operating mode, but may be desirable in certain applications. The output can source or sink current in this mode. When sinking current while in forced continuous operation, the controller will cause current to flow back into the input filter capacitor. If large enough, this element will prevent the input supply from boosting to unacceptably high levels; see COUT Selection in the Applications Information Section. INTVCC /EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open, an internal 5V low dropout linear regulator supplies INTVCC power. If EXTVCC is taken above 4.7V, the 5V regulator is turned off and an internal switch is turned on connecting EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information section. Output Overvoltage Protection An overvoltage comparator, OV, guards against transient overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. Power Good (PGOOD) Pin The PGOOD pin is connected to an open drain of an internal MOSFET. The MOSFET turns on and pulls the pin low when either output is not within ± 7.5% of the nominal output level as determined by the resistive feedback divider. When both outputs meet the ±7.5% requirement, the MOSFET is The RUN/SS capacitors are used initially to limit the inrush current of each switching regulator. Foldback current limiting is activated when the output voltage falls below 70% of its nominal level. If a short is present, a safe, low output current is provided due to the internal current foldback and actual power wasted is low due to the efficient nature of the current mode switching regulator. THEORY AND BENEFITS OF 2-PHASE OPERATION The LTC1628 and the LTC3728L-1 family of dual high efficiency DC/DC controllers brings the considerable benefits of 2-phase operation to portable applications for the first time. Notebook computers, PDAs, handheld terminals and automotive electronics will all benefit from the lower input filtering requirement, reduced electromagnetic interference (EMI) and increased efficiency associated with 2-phase operation. Why the need for 2-phase operation? Up until the 2-phase family, constant-frequency dual switching regulators operated both channels in phase (i.e., single-phase operation). This means that both switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor and battery. These large amplitude current pulses increased the total RMS current flowing from the input capacitor, requiring the use of more expensive input capacitors and increasing both EMI and losses in the input capacitor and battery. With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency. 3728l1fc 11 LTC3728L-1 OPERATION (Refer to Functional Diagram) 5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV 3728L1 F03a IIN(MEAS) = 2.53ARMS IIN(MEAS) = 1.55ARMS (a) 3728L1 F03b (b) Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the LTC1628 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the LTC1628 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2phase operation dropped the input current from 2.53ARMS to 1.55ARMS. While this is an impressive reduction in itself, remember that the power losses are proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative duty cycles which, in turn, are dependent upon the input voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how the RMS input current varies for single-phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but in fact extend over a wide region. A good rule of thumb for most applications is that 2-phase operation will reduce the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle. 3.0 SINGLE PHASE DUAL CONTROLLER INPUT RMS CURRENT (A) 2.5 2.0 1.5 2-PHASE DUAL CONTROLLER 1.0 0.5 0 VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40 3728L1 F04 Figure 4. RMS Input Current Comparison 3728l1fc 12 LTC3728L-1 APPLICATIONS INFORMATION 2.5 PLLFLTR PIN VOLTAGE (V) Figure 1 on the first page is a basic LTC3728L-1 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). RSENSE Selection For Output Current RSENSE is chosen based on the required output current. The current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of SGND to 1.1(INTVCC). The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current, ΔIL. Allowing a margin for variations in the IC and external component values yields: 50mV RSENSE = IMAX When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability criteria for buck regulators operating at greater than 50% duty factor. A curve is provided to estimate this reduction in peak output current level depending upon the operating duty factor. Operating Frequency The IC uses a constant frequency phase-lockable architecture with the frequency determined by an internal capacitor. This capacitor is charged by a fixed current plus an additional current which is proportional to the voltage applied to the PLLFLTR pin. Refer to Phase-Locked Loop and Frequency Synchronization in the Applications Information section for additional information. A graph for the voltage applied to the PLLFLTR pin vs frequency is given in Figure 5. As the operating frequency is increased the gate charge losses will be higher, reducing 2.0 1.5 1.0 0.5 0 200 300 400 500 OPERATING FREQUENCY (kHz) 600 3728L1 F05 Figure 5. PLLFLTR Pin Voltage vs Frequency efficiency (see Efficiency Considerations). The maximum switching frequency is approximately 550kHz. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current ΔIL decreases with higher inductance or frequency and increases with higher VIN: IL = V 1 VOUT 1– OUT (f)(L) VIN Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ΔIL=0.3(IMAX). The maximum ΔIL occurs at the maximum input voltage. The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average inductor current required results in a peak current below 3728l1fc 13 LTC3728L-1 APPLICATIONS INFORMATION 25% of the current limit determined by RSENSE. Lower inductor values (higher ΔIL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. Inductor Core Selection Usually, high inductance is preferred for small current ripple and low core loss. Unfortunately, increased inductance requires more turns of wire or a smaller air gap in the inductor core, resulting in high copper loss or low saturation current. Once the value of L is known, the actual inductor must be selected. There are two popular types of core material of commercial available inductors. Ferrite core inductors usually have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. However, ferrite core saturates “hard”, which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. One advantage of the LTC3728L-1 is its current mode control that detects and limits cycle-by-cycle peak inductor current. Therefore, accurate and fast protection is achieved if the inductor is saturated in steady state or during transient mode. Powdered iron core inductors usually saturate “soft”, which means the inductance drops in a linear fashion when the current increases. However, the core loss of the powder iron inductor is usually higher than the ferrite inductor. So designs with high switching frequency should also address inductor core loss. Inductor manufacturers usually provide inductance, DCR, (peak) saturation current and (DC) heating current ratings in the inductor data sheet. A good supply design should not exceed the saturation and heating current rating of the inductor. Power MOSFET and D1 Selection Two external power MOSFETs must be selected for each controller in the LTC3728L-1: One N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. The only exception is if low input voltage is expected (VIN < 5V); then, sub-logic level threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the “ON” resistance RDS(ON), Miller capacitance CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in VDS. This result is then multiplied by the ratio of the application applied VDS to the Gate charge curve specified VDS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT VIN Synchronous Switch Duty Cycle = VIN – VOUT VIN The MOSFET power dissipations at maximum output current are given by: PMAIN = VOUT 2 IMAX ) (1+ )RDS(ON) + ( VIN ( VIN )2 IMAX (RDR )(CMILLER ) • 2 1 1 + VINTVCC – VTHMIN VTHMIN PSYNC = ( f) VIN – VOUT 2 IMAX ) (1+ )RDS(ON) ( VIN 3728l1fc 14 LTC3728L-1 APPLICATIONS INFORMATION where δ is the temperature dependency of RDS(ON) and RDR (approximately 4Ω) is the effective driver resistance at the MOSFET’s Miller threshold voltage. VTHMIN is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1+δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead-time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. Schottky diodes should be placed in parallel with the synchronous MOSFETs when operating in pulse-skip mode or in Burst Mode operation. CIN and COUT Selection The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst case RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used in the formula below to determine the maximum RMS current requirement. Increasing the output current, drawn from the other out-of-phase controller, will actually decrease the input RMS ripple current from this maximum value (see Figure 4). The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor of 30% to 70% when compared to a single phase power supply solution. The type of input capacitor, value and ESR rating have efficiency effects that need to be considered in the selection process. The capacitance value chosen should be sufficient to store adequate charge to keep high peak battery currents down. 20μF to 40μF is usually sufficient for a 25W output supply operating at 200kHz. The ESR of the capacitor is important for capacitor power dissipation as well as overall battery efficiency. All of the power (RMS ripple current • ESR) not only heats up the capacitor but wastes power from the battery. Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors, but each has drawbacks: ceramics have very high voltage coefficients and may have audible piezoelectric effects; tantalums need to be surge-rated; OS-CONs suffer from higher inductance, larger case size and limited surface-mount applicability; electrolytics’ higher ESR and dryout possibility require several to be used. Multiphase systems allow the lowest amount of capacitance overall. As little as one 22μF or two to three 10μF ceramic capacitors are an ideal choice in a 20W to 35W power supply due to their extremely low ESR. Even though the capacitance at 20V is substantially below their rating at zero-bias, very low ESR loss makes ceramics an ideal candidate for highest efficiency battery operated systems. Also consider parallel ceramic and high quality electrolytic capacitors as an effective means of achieving ESR and bulk capacitance goals. In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: CIN Required IRMS IMAX VOUT ( VIN VOUT ) VIN 1/2 3728l1fc 15 LTC3728L-1 APPLICATIONS INFORMATION This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The benefit of the LTC3728L-1 multiphase clocking can be calculated by using the equation above for the higher power controller and then calculating the loss that would have resulted if both controller channels switched on at the same time. The total RMS power lost is lower when both controllers are operating due to the interleaving of current pulses through the input capacitor’s ESR. This is why the input capacitor’s requirement calculated above for the worst-case controller is adequate for the dual controller design. Remember that input protection fuse resistance, battery resistance and PC board trace resistance losses are also reduced due to the reduced peak currents in a multiphase system. The overall benefit of a multiphase design will only be fully realized when the source impedance of the power supply/battery is included in the efficiency testing. The drains of the two top MOSFETS should be placed within 1cm of each other and share a common CIN(s). Separating the drains and CIN may produce undesirable voltage and current resonances at VIN. The selection of COUT is driven by the required effective series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (ΔVOUT) is determined by: 1 VOUT IL ESR + 8fCOUT Where f = operating frequency, COUT = output capacitance, and ΔIL= ripple current in the inductor. The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. With ΔIL = 0.3IOUT(MAX) the output ripple will typically be less than 50mV at the maximum VIN assuming: COUT Recommended ESR < 2 RSENSE and COUT > 1/(8fRSENSE) The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. Manufacturers such as Nichicon, Nippon Chemi-Con and Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest (ESR)(size) product of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the inductance effects. In surface mount applications multiple capacitors may need to be used in parallel to meet ESR, RMS current handling and load step requirements. Aluminum electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR but have lower storage capacity per unit volume than other capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when combined with a controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for switching regulators having controlled soft-start. Several excellent surge-tested choices are the AVX TPS, AVX TPSV or the KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors can be used 3728l1fc 16 LTC3728L-1 APPLICATIONS INFORMATION in cost-driven applications providing that consideration is given to ripple current ratings, temperature and long term reliability. A typical application will require several to many aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor types include Nichicon PL series, Panasonic SP, NEC Neocap, Cornell Dubilier ESRE and Sprague 595D series. Consult manufacturers for other specific recommendations. INTVCC Regulator An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the IC. The INTVCC pin regulator can supply a peak current of 50mA and must be bypassed to ground with a minimum of 4.7μF tantalum, 10μF special polymer, or low ESR type electrolytic capacitor. A 1μF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers and to prevent interaction between channels. Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the IC to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC and 3.3V linear regulators also needs to be taken into account for the power dissipation calculations. The total INTVCC current can be supplied by either the 5V internal linear regulator or by the EXTVCC input pin. When the voltage applied to the EXTVCC pin is less than 4.7V, all of the INTVCC current is supplied by the internal 5V linear regulator. Power dissipation for the IC in this case is highest: (VIN)(IINTVCC), and overall efficiency is lowered. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, the IC VIN current is thermally limited to less than 67mA from a 24V supply when not using the EXTVCC pin as follows: TJ = 70°C + (67mA)(24V)(34°C/W) = 125°C Use of the EXTVCC input pin reduces the junction temperature to: TJ = 70°C + (67mA)(5V)(34°C/W) = 81°C The absolute maximum rating for the INTVCC Pin is 40mA. Dissipation should be calculated to also include any added current drawn from the internal 3.3V linear regulator. To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. EXTVCC Connection The IC contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. When the voltage applied to EXTVCC rises above 4.7V, the internal regulator is turned off and the switch closes, connecting the EXTVCC pin to the INTVCC pin thereby supplying internal power. The switch remains closed as long as the voltage applied to EXTVCC remains above 4.5V. This allows the MOSFET driver and control power to be derived from the output during normal operation (4.7V < VOUT < 7V) and from the internal regulator when the output is out of regulation (start-up, short-circuit). If more current is required through the EXTVCC switch than is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. Do not apply greater than 7V to the EXTVCC pin and ensure that EXTVCC < VIN. Significant efficiency gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Efficiency). For 5V regulators this supply means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC Left Open (or Grounded). This will cause INTVCC to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 3728l1fc 17 LTC3728L-1 APPLICATIONS INFORMATION 2. EXTVCC Connected directly to VOUT. This is the normal connection for a 5V regulator and provides the highest efficiency. 3. EXTVCC Connected to an External supply. If an external supply is available in the 5V to 7V range, it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. 4. EXTVCC Connected to an Output-Derived Boost Network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an outputderived voltage that has been boosted to greater than 4.7V. This can be done with either the inductive boost winding as shown in Figure 6a or the capacitive charge pump shown in Figure 6b. The charge pump has the advantage of simple magnetics. Topside MOSFET Driver Supply (CB, DB) External bootstrap capacitors CB connected to the BOOST pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the functional diagram is charged through external diode DB from INTVCC when the SW pin is low. When one of the topside MOSFETs is to be turned on, the driver places the CB voltage across the gate-source of the desired MOSFET. This enhances the MOSFET and turns on the topside switch. The switch node voltage, SW, rises to VIN and the BOOST pin follows. With the topside MOSFET on, the boost voltage is above the input supply: VBOOST = VIN + VINTVCC. The value of the boost capacitor CB needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than VIN(MAX). When adjusting the gate drive level, the final arbiter is the total input current for the regulator. If a change is made and the input current decreases, then the efficiency has improved. If there is no change in input current, then there is no change in efficiency. Output Voltage The output voltages are each set by an external feedback resistive divider carefully placed across the output capacitor. The resultant feedback signal is compared with the internal precision 0.800V voltage reference by the error amplifier. The output voltage is given by the equation: R2 VOUT = 0.8V 1+ R1 where R1 and R2 are defined in Figure 2. SENSE+/SENSE– Pins The common mode input range of the current comparator sense pins is from 0V to (1.1)INTVCC. Continuous linear operation is guaranteed throughout this range allowing output voltage setting from 0.8V to 7.7V, depending upon the voltage applied to EXTVCC. A differential NPN input + CIN CIN BAT 85 VIN + VIN VIN OPTIONAL EXTVCC CONNECTION 5V < VSEC < 7V VSEC + BAT85 VIN 1μF RSENSE N-CH SW FCB BG1 EXTVCC + COUT COUT BG1 N-CH N-CH SGND VOUT L1 SW + R5 BAT85 RSENSE N-CH VOUT T1 1:N R6 BAT85 VN2222LL TG1 TG1 EXTVCC 0.22μF LTC3728L-1 + LTC3728L-1 1μF PGND PGND 3728L1 F06a Figure 6a. Secondary Output Loop & EXTVCC Connection 3728L1 F06b Figure 6b. Capacitive Charge Pump for EXTVCC 3728l1fc 18 LTC3728L-1 APPLICATIONS INFORMATION stage is biased with internal resistors from an internal 2.4V source as shown in the Functional Diagram. This requires that current either be sourced or sunk from the SENSE pins depending on the output voltage. If the output voltage is below 2.4V current will flow out of both SENSE pins to the main output. The output can be easily preloaded by the VOUT resistive divider to compensate for the current comparator’s negative input bias current. The maximum current flowing out of each pair of SENSE pins is: ISENSE+ + ISENSE– = (2.4V – VOUT)/24k Since VOSENSE is servoed to the 0.8V reference voltage, we can choose R1 in Figure 2 to have a maximum value to absorb this current. 0.8V R1(MAX) = 24k 2.4V – VOUT for VOUT < 2.4V Regulating an output voltage of 1.8V, the maximum value of R1 should be 32k. Note that for an output voltage above 2.4V, R1 has no maximum value necessary to absorb the sense currents; however, R1 is still bounded by the VOSENSE feedback current. Soft-Start/Run Function The RUN/SS1 and RUN/SS2 pins are multipurpose pins that provide a soft-start function and a means to shut down the LTC3728L-1. Soft-start reduces the input power source’s surge currents by gradually increasing the controller’s current limit (proportional to VITH). This pin can also be used for power supply sequencing. An internal 1.2μA current source charges up the CSS capacitor. When the voltage on RUN/SS1 (RUN/SS2) reaches 1.5V, the particular controller is permitted to start operating. As the voltage on RUN/SS increases from 1.5V to 3.0V, the internal current limit is increased from 25mV/RSENSE to 75mV/RSENSE. The output current limit ramps up slowly, taking an additional 1.25s/μF to reach full current. The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If RUN/SS has been pulled all the way to ground there is a delay before starting of approximately: tDELAY = 1.5V C = (1.25s / μF ) CSS 1.2μA SS tIRAMP = 3V 1.5V C = (1.25s / μF ) CSS 1.2μA SS By pulling both RUN/SS pins below 1V, the IC is put into low current shutdown (IQ = 20μA). The RUN/SS pins can be driven directly from logic as shown in Figure 7. Diode D1 in Figure 7 reduces the start delay but allows CSS to ramp up slowly providing the soft-start function. Each RUN/SS pin has an internal 6V zener clamp (See Functional Diagram). Because the LTC3728L-1 is designed for applications not requiring over current latchoff, no pull-up resistor is required on the RUN/SS pin to defeat latchoff. Refer to the LTC3728L/LTC3728LX datasheet if this feature is required. Fault Conditions: Current Limit and Current Foldback The current comparators have a maximum sense voltage of 75mV resulting in a maximum MOSFET current of 75mV/RSENSE. The maximum value of current limit generally occurs with the largest VIN at the highest ambi3.3V OR 5V RUN/SS RUN/SS D1 CSS CSS (a) (b) 3728L1 F07 Figure 7. RUN/SS Pin Interfacing 3728l1fc 19 LTC3728L-1 APPLICATIONS INFORMATION ent temperature, conditions that cause the highest power dissipation in the top MOSFET. Each controller includes current foldback to help further limit load current when the output is shorted to ground. The foldback circuit is active even when the overload shutdown latch described above is overridden. If the output falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered from 75mV to 17mV. Under short-circuit conditions with very low duty cycles, the controller will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short-circuit ripple current is determined by the minimum on-time tON(MIN) of each controller (typically 100ns), the input voltage and inductor value: ΔIL(SC) = tON(MIN) (VIN /L) The resulting short-circuit current is: ISC = 25mV 1 – I RSENSE 2 L(SC) Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the controller is operating. A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults greater than 7.5% above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The output of this comparator is only latched by the overvoltage condition itself and will therefore allow a switching regulator system having a poor PC layout to function while the design is being debugged. The bottom MOSFET remains on continuously for as long as the OV condition persists; if VOUT returns to a safe level, normal operation automatically resumes. A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. Phase-Locked Loop and Frequency Synchronization The IC has a phase-locked loop comprised of an internal voltage controlled oscillator and phase detector. This allows the top MOSFET turn-on to be locked to the rising edge of an external source. The frequency range of the voltage controlled oscillator is ± 50% around the center frequency fO. A voltage of 1.2V applied to the PLLFLTR pin corresponds to a frequency of approximately 400kHz. The nominal operating frequency range of the IC is 260kHz to 550kHz. The phase detector used is an edge sensitive digital type which provides zero degrees phase shift between the external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the harmonics of the VCO center frequency. The PLL hold-in range, ΔfH, is equal to the capture range, ΔfC: ΔfH = ΔfC = ±0.5 fO (260kHz-550kHz) The output of the phase detector is a complementary pair of current sources charging or discharging the external filter network on the PLLFLTR pin. If the external frequency (fPLLIN) is greater than the oscillator frequency f0SC, current is sourced continuously, 3728l1fc 20 LTC3728L-1 APPLICATIONS INFORMATION pulling up the PLLFLTR pin. When the external frequency is less than f0SC, current is sunk continuously, pulling down the PLLFLTR pin. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the PLLFLTR pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable operating point the phase comparator output is open and the filter capacitor CLP holds the voltage. The IC’s PLLIN pin must be driven from a low impedance source such as a logic gate located close to the pin. When using multiple ICs for a phase-locked system, the PLLFLTR pin of the master oscillator should be biased at a voltage that will guarantee the slave oscillator(s) ability to lock onto the master’s frequency. A DC voltage of 0.7V to 1.7V applied to the master oscillator’s PLLFLTR pin is recommended in order to meet this requirement. The resultant operating frequency can range from 300kHz to 500kHz. The loop filter components (CLP, RLP) smooth out the current pulses from the phase detector and provide a stable input to the voltage controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires lock. Typically RLP =10kΩ and CLP is 0.01μF to 0.1μF. Minimum On-Time Considerations Minimum on-time tON(MIN) is the smallest time duration that each controller is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that tON(MIN) < VOUT VIN(f) If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. The minimum on-time for each controller is approximately 100ns. However, as the peak sense voltage decreases the minimum on-time gradually increases up to about 150ns. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. FCB Pin Operation The FCB pin can be used to regulate a secondary winding or as a logic level input. Continuous operation is forced on both controllers when the FCB pin drops below 0.8V. During continuous mode, current flows continuously in the transformer primary. The secondary winding(s) draw current only when the bottom, synchronous switch is on. When primary load currents are low and/or the VIN/VOUT ratio is low, the synchronous switch may not be on for a sufficient amount of time to transfer power from the output capacitor to the secondary load. Forced continuous operation will support secondary windings providing there is sufficient synchronous switch duty factor. Thus, the FCB input pin removes the requirement that power must be drawn from the inductor primary in order to extract power from the auxiliary windings. With the loop in continuous mode, the auxiliary outputs may nominally be loaded without regard to the primary output load. The secondary output voltage VSEC is normally set as shown in Figure 6a by the turns ratio N of the transformer: VSEC ≅ (N + 1) VOUT However, if the controller goes into Burst Mode operation and halts switching due to a light primary load current, then VSEC will droop. An external resistive divider from VSEC to the FCB pin sets a minimum voltage VSEC(MIN): R6 VSEC(MIN) 0.8V 1+ R5 where R5 and R6 are shown in Figure 2. 3728l1fc 21 LTC3728L-1 APPLICATIONS INFORMATION If VSEC drops below this level, the FCB voltage forces temporary continuous switching operation until VSEC is again above its minimum. capacitance can be reduced for a particular application. A complete explanation is included in Design Solutions 10. (See www.linear.com) In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18μA internal current source pulling the pin high. Include this current when choosing resistor values R5 and R6. Efficiency Considerations The following table summarizes the possible states available on the FCB pin: Table 1 FCB PIN CONDITION 0V to 0.75V Forced Continuous Both Controllers (Current Reversal Allowed— Burst Inhibited) 0.85V < VFCB < 4.3V Minimum Peak Current Induces Burst Mode Operation No Current Reversal Allowed Feedback Resistors Regulating a Secondary Winding >4.8V Burst Mode Operation Disabled Constant Frequency Mode Enabled No Current Reversal Allowed No Minimum Peak Current Voltage Positioning Voltage positioning can be used to minimize peak-to-peak output voltage excursions under worst-case transient loading conditions. The open-loop DC gain of the control loop is reduced depending upon the maximum load step specifications. Voltage positioning can easily be added to either or both controllers by loading the ITH pin with a resistive divider having a Thevenin equivalent voltage source equal to the midpoint operating voltage range of the error amplifier, or 1.2V (see Figure 8). The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier. The maximum output voltage deviation can theoretically be reduced to half or alternatively the amount of output The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentINTVCC RT2 ITH RT1 RC LTC3728L-1 CC 3728L1 F08 Figure 8. Active Voltage Positioning Applied to the LTC3728L-1 age of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3728L-1 circuits: 1) IC VIN current (including loading on the 3.3V internal regulator), 2) INTVCC regulator current, 3) I2R losses, 4) Topside MOSFET transition losses. 1. The VIN current has two components: the first is the DC supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents; the second is the current drawn from the 3.3V linear regulator output. VIN current typically results in a small (<0.1%) loss. 3728l1fc 22 LTC3728L-1 APPLICATIONS INFORMATION 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the control circuit current. In continuous mode, IGATECHG =f(QT+QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. Supplying INTVCC power through the EXTVCC switch input from an output-derived source will scale the VIN current required for the driver and control circuits by a factor of (Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 2.5mA of VIN current. This reduces the mid-current loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous mode the average output current flows through L and RSENSE, but is “chopped” between the topside MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L, RSENSE and ESR to obtain I2R losses. For example, if each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and RESR = 40mΩ (sum of both input and output capacitance losses), then the total resistance is 130mΩ. This results in losses ranging from 3% to 13% as the output current increases from 1A to 5A for a 5V output, or a 4% to 20% loss for a 3.3V output. Efficiency varies as the inverse square of VOUT for the same external components and output power level. The combined effects of increasingly lower output voltages and higher currents required by high performance digital systems is not doubling but quadrupling the importance of loss terms in the switching regulator system! 4. Transition losses apply only to the topside MOSFET(s), and become significant only when operating at high input voltages (typically 15V or greater). Transition losses can be estimated from: 2 I Transition Loss = ( VIN ) • MAX (RDR ) • 2 (CMILLER )( f ) 5V –1V TH + 1 VTH Other “hidden” losses such as copper trace and internal battery resistances can account for an additional 5% to 10% efficiency degradation in portable systems. It is very important to include these “system” level losses during the design phase. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a minimum of 20μF to 40μF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. The LTC3728L-1 2-phase architecture typically halves this input capacitance requirement over competing solutions. Other losses including Schottky conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load current transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating the feedback error signal that forces the regulator to adapt to the current change and return VOUT to its steady-state value. During this recovery time VOUT can be monitored for excessive overshoot or ringing, which would indicate a stability problem. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The availability of the ITH pin not only allows optimization of control loop behavior but also provides a DC coupled and AC filtered closed loop response test point. The DC step, rise time and settling 3728l1fc 23 LTC3728L-1 APPLICATIONS INFORMATION at this test point truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at the pin. The ITH external components shown in the Figure 1 circuit will provide an adequate starting point for most applications. CLOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10μF capacitor would require a 250μs rise time, limiting the charging current to about 200mA. The ITH series RC-CC filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop gain and phase. An output current pulse of 20% to 80% of full-load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly across the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC is decreased, the zero frequency will be kept the same, thereby keeping the phase shift the same in the most critical frequency range of the feedback loop. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main power line in an automobile is the source of a number of nasty potential transients, including load-dump, reverse-battery, and double-battery. A second, more severe transient is caused by switching in loads with large (>1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of Automotive Considerations: Plugging into the Cigarette Lighter Load-dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse-battery is just what it says, while double-battery is a consequence of tow-truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 9 is the most straightforward approach to protect a DC/DC converter from the ravages of an automotive power line. The series diode prevents current from flowing during reverse-battery, while the transient suppressor clamps the input voltage during load-dump. Note that the transient suppressor should not conduct during double-battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LTC3728L-1 has a maximum input voltage of 30V, most applications will also be limited to 30V by the MOSFET BVDSS. 50A IPK RATING 12V VIN LTC3728L-1 TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A 3728L1 F09 Figure 9. Automotive Application Protection 3728l1fc 24 LTC3728L-1 APPLICATIONS INFORMATION Design Example As a design example for one channel, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A, and f = 300kHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the PLLFLTR pin to a resistive divider from the INTVCC pin, generating 0.7V for 300kHz operation. The minimum inductance for 30% ripple current is: V –V V L IN OUT • OUT (f)(IRIPPLE ) VIN or 3.7μH. Using standard inductor values: IL = VOUT VOUT 1– (f)(L) VIN A 4.7μH inductor will produce 23% ripple current and a 3.3μH will result in 33%. The peak inductor current will be the maximum DC value plus one half the ripple current, or 5.84A, for the 3.3μH value. Increasing the ripple current will also help ensure that the minimum on-time of 100ns is not violated. The minimum on-time occurs at maximum VIN: tON(MIN) = VOUT VIN(MAX)f = 1.8V = 273ns 22V(300kHz) The RSENSE resistor value can be calculated by using the maximum current sense voltage specification with some accommodation for tolerances: RSENSE 60mV 0.01 5.84A Since the output voltage is below 2.4V the output resistive divider will need to be sized to not only set the output voltage but also to absorb the SENSE pin’s specified input current. 0.8V R1(MAX) = 24k 2.4V – VOUT 0.8V = 24k = 32k 2.4V – 1.8V Choosing 1% resistors: R1 = 25.5k and R2 = 32.4k yields an output voltage of 1.816V. The power dissipation on the top side MOSFET can be easily estimated. Choosing a Fairchild FDS6982S dual MOSFET results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At maximum input voltage with T(estimated) = 50°C: 1.8V 2 PMAIN = (5) [1+ (0.005)(50°C – 25°C)] • 22V (0.035) + (22V )2 5A ( 4)(215pF ) • 2 1 1 5 – 2.3 + 2.3 ( 300kHz ) = 332mW A short-circuit to ground will result in a folded back current of: ISC = 25mV 1 120ns(22V) – = 2.1A 0.01 2 3.3μH with a typical value of RDS(ON) and δ = (0.005/°C)(20) = 0.1. The resulting power dissipated in the bottom MOSFET is: 22V – 1.8V 2 2.1A ) (1.125) ( 0.022 ) ( 22V = 100mW PSYNC = which is less than under full-load conditions. CIN is chosen for an RMS current rating of at least 3A at temperature assuming only this channel is on. COUT is chosen with an ESR of 0.02Ω for low output ripple. The output ripple in continuous mode will be highest at the 3728l1fc 25 LTC3728L-1 APPLICATIONS INFORMATION maximum input voltage. The output voltage ripple due to ESR is approximately: at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large resonant loop. VORIPPLE = RESR (ΔIL) = 0.02Ω(1.67A) = 33mVP–P When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. These items are also illustrated graphically in the layout diagram of Figure 10. The Figure 11 illustrates the current waveforms present in the various branches of the 2-phase synchronous regulators operating in the continuous mode. Check the following in your layout: 2. Are the signal and power grounds kept separate? The combined IC signal ground pin and the ground return of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET, Schottky diode and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. 1. Are the top N-channel MOSFETs M1 and M3 located within 1cm of each other with a common drain connection 3. Do the LTC3728L-1 VOSENSE pins’ resistive dividers connect to the (+) terminals of COUT? The resistive divider must PC Board Layout Checklist RPU RUN/SS1 PGOOD SENSE1+ TG1 SENSE1– SW1 VOSENSE1 BOOST1 PLLFLTR VIN VPULL-UP (<7V) PGOOD L1 VOUT1 CB1 R1 INTVCC RIN 1μF CERAMIC D1 COUT1 FCB 3.3VOUT ITH2 PGND CVIN CINTVCC VIN CIN COUT2 1μF CERAMIC BG2 M3 BOOST2 GND + EXTVCC LTC3728L-1 ITH1 INTVCC SGND 3.3V BG1 M2 + PLLIN + fIN M1 + R2 RSENSE M4 D2 CB2 R3 R4 VOSENSE2 SW2 SENSE2– TG2 RSENSE VOUT2 L2 SENSE2+ RUN/SS2 3728L1 F10 Figure 10. LTC3728L-1 Recommended Printed Circuit Layout Diagram 3728l1fc 26 LTC3728L-1 APPLICATIONS INFORMATION SW1 L1 D1 RSENSE1 COUT1 VOUT1 + RL1 CERAMIC VIN RIN CIN + SW2 BOLD LINES INDICATE HIGH SWITCHING CURRENT. KEEP LINES TO A MINIMUM LENGTH. D2 L2 RSENSE2 COUT2 VOUT2 + RL2 CERAMIC 3728L1 F11 Figure 11. Branch Current Waveforms be connected between the (+) terminal of COUT and signal ground. The R2 and R4 connections should not be along the high current input feeds from the input capacitor(s). 4. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The filter capacitor between SENSE+ and SENSE– should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. 5. Is the INTVCC decoupling capacitor connected close to the IC, between the INTVCC and the power ground pins? This capacitor carries the MOSFET drivers current peaks. An additional 1μF ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 6. Keep the switching nodes (SW1, SW2), top gate nodes (TG1, TG2), and boost nodes (BOOST1, BOOST2) away from sensitive small-signal nodes, especially from the opposites channel’s voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the “output side” of the LTC3728L-1 and occupy minimum PC trace area. 3728l1fc 27 LTC3728L-1 APPLICATIONS INFORMATION 7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTVCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the SGND pin of the IC. PC Board Layout Debugging Start with one controller on at a time. It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be maintained over the input voltage range down to dropout and until the output load drops below the low current operation threshold—typically 10% to 20% of the maximum designed current level in Burst Mode operation. The duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after each controller is checked for its individual performance should both controllers be turned on at the same time. A particularly difficult region of operation is when one controller channel is nearing its current comparator trip point when the other channel is turning on its top MOSFET. This occurs around 50% duty cycle on either channel due to the phasing of the internal clocks and may cause minor duty cycle jitter. Reduce VIN from its nominal level to verify operation of the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while monitoring the outputs to verify operation. Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems coincide with high input voltages and low output currents, look for capacitive coupling between the BOOST, SW, TG, and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between CIN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC. An embarrassing problem, which can be missed in an otherwise properly working switching regulator, results when the current sensing leads are hooked up backwards. The output voltage under this improper hookup will still be maintained but the advantages of current mode control will not be realized. Compensation of the voltage loop will be much more sensitive to component selection. This behavior can be investigated by temporarily shorting out the current sensing resistor—don’t worry, the regulator will still maintain control of the output voltage. 3728l1fc 28 LTC3728L-1 TYPICAL APPLICATIONS 59k 1M 100k 0.1μF 180pF TG1 SENSE1– SW1 0.015Ω VOSENSE1 BOOST1 PLLFLTR VIN PLLIN D1 LTC3728L-1 INTVCC SGND PGND 3.3VOUT ITH2 1000pF 180pF 1 CMDSH-3TR 22μF 50V 150μF, 6.3V PANASONIC SP VOUT3 12V 120mA + 100k 0.1μF 1μF 10V 1μF 25V GND 4.7μF 180μF, 4V PANASONIC SP VIN 7V TO 28V M2 BG2 Q3 Q4 D2 BOOST2 0.1μF 20k 1% 63.4k 1% ON/OFF 2 220k CMDSH-3TR 3.3V 3 + ITH1 LT1121 + EXTVCC 5 Q2 BG1 FCB VOUT1 5V 3A; 4A PEAK 8 Q1 10Ω 1000pF + M1 0.1μF 33pF 15k SENSE1+ 1000pF 105k, 1% 33pF 15k PGOOD 33μF 25V T1, 1:1.8 10μH + 20k 1% RUN/SS1 MBRS1100T3 VPULL-UP (<7V) PGOOD VOSENSE2 SW2 SENSE2– TG2 1000pF SENSE2+ RUN/SS2 L1 6.3μH VOUT2 3.3V 5A; 6A PEAK 0.01Ω 0.1μF 3728L1 F12 VIN: 7V TO 28V VOUT: 5V, 3A/3.3V, 6A/12V, 150mA SWITCHING FREQUENCY = 250kHz MI, M2: FDS6982S OR VISHAY Si4810DY L1: SUMIDA CEP123-6R3MC T1: 10μH 1:1.8 — DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID Figure 12. LTC3728L-1 High Efficiency Low Noise 5V/3A, 3.3V/5A, 12V/120mA Regulator 3728l1fc 29 LTC3728L-1 TYPICAL APPLICATIONS VPULL-UP (<7V) 0.1μF 180pF 20k 1% 10k RUN/SS1 PGOOD SENSE1+ TG1 SENSE1– SW1 VOSENSE1 BOOST1 PLLFLTR VIN 0.1μF 0.01μF 1000pF PLLIN 1μF 50V BG1 + PGND 180μF, 4V CMDSH-3TR 3.3V 3.3VOUT ITH2 PIN 4 BOOST2 Q3 Q4 0.1μF VOSENSE2 SW2 SENSE2– TG2 1000pF SENSE2+ RUN/SS2 M2 L2 4.3μH VOUT2 3.3V/5A 0.008Ω 0.1μF VIN: 7V TO 28V VOUT: 5V, 4A/3.3V, 5A VIN 7V TO 28V BG2 1000pF 20k 1% 180pF 1μF 50V 4.7μF, 10V GND + SGND 150μF, 6.3V 0.1μF 1μF 1500pF 22μF 50V CMDSH-3TR EXTVCC LTC3728L-1 ITH1 INTVCC 100pF Q2 M1 10Ω 63.4k 1% VOUT1 5V/4A + fSYNC Q1 PIN 4 FCB 4.75k 0.008Ω 1000pF 105k 1% 100pF 8.06k L1 4.3μH PGOOD 3728L1 F13 SWITCHING FREQUENCY = 250kHz TO 550kHz M1, M2: FDS6982S OR VISHAY Si4810DY L1, L2: SUMIDA CDEP105-4R3MC-88 OUTPUT CAPACITORS: PANASONIC SP SERIES Figure 13. LTC3728L-1 5V/4A, 3.3V/5A Regulator with External Frequency Synchronization 3728l1fc 30 LTC3728L-1 PACKAGE DESCRIPTION GN Package 28-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .386 – .393* (9.804 – 9.982) .045 ±.005 .033 (0.838) REF 28 27 26 25 24 23 22 21 20 19 18 17 1615 .254 MIN .150 – .165 .229 – .244 (5.817 – 6.198) .0165 ± .0015 .150 – .157** (3.810 – 3.988) .0250 BSC 1 RECOMMENDED SOLDER PAD LAYOUT .015 ± .004 × 45° (0.38 ± 0.10) .0075 – .0098 (0.19 – 0.25) 2 3 4 5 6 7 8 9 10 11 12 13 14 .0532 – .0688 (1.35 – 1.75) .004 – .0098 (0.102 – 0.249) 0° – 8° TYP .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) .0250 (0.635) BSC .008 – .012 (0.203 – 0.305) TYP GN28 (SSOP) 0204 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE UH Package 32-Lead Plastic QFN (5mm × 5mm) (Reference LTC DWG # 05-08-1693 Rev D) 5.00 ± 0.10 (4 SIDES) 0.70 ±0.05 5.50 ±0.05 4.10 ±0.05 R = 0.05 TYP 0.75 ± 0.05 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD PIN 1 NOTCH R = 0.30 TYP R = 0.115 OR 0.35 × 45° CHAMFER TYP 31 32 0.40 ± 0.10 PIN 1 TOP MARK (NOTE 6) 1 2 3.50 REF (4 SIDES) 3.45 ± 0.05 3.50 REF (4-SIDES) 3.45 ± 0.05 3.45 ± 0.10 3.45 ± 0.10 PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.25 ± 0.05 0.50 BSC (UH32) QFN 0406 REV D 3728l1fc Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 31 LTC3728L-1 TYPICAL APPLICATION IIN 12VIN CIN I1 PHASMD OPEN TG1 U1 TG2 LTC3729 90° 0° IIN* BUCK: 2.5V/15A 180° BUCK: 2.5V/15A 2.5VO/30A I2 CLKOUT I2 I3 I3 TG1 TG2 90° BUCK: 1.5V/15A 270° U2 LTC3728L-1 90° BUCK: 1.8V/15A 1.5VO/15A 1.8VO/15A I4 *INPUT RIPPLE CURRENT CANCELLATION INCREASES THE RIPPLE FREQUENCY AND REDUCES THE RMS INPUT RIPPLE CURRENT THUS, SAVING INPUT CAPACITORS I4 PLLIN I1 3728L1 F14 Figure 14. Multioutput PolyPhase Application RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1628/LTC1628-PG/ 2-Phase, Dual Output Synchronous Step-Down LTC1628-SYNC DC/DC Controller Reduces CIN and COUT, Power Good Output Signal, Synchronizable, 3.5V ≤ VIN ≤ 36V, IOUT up to 20A, 0.8V ≤ VOUT ≤ 5V LTC1629/ LTC1629-PG 20A to 200A PolyPhase™ Synchronous Controllers Expandable from 2-Phase to 12-Phase, Uses All Surface Mount Components, No Heat Sink, VIN up to 36V LTC1708-PG 2-Phase, Dual Synchronous Controller with Mobile VID 3.5V ≤ VIN ≤ 36V, VID Sets VOUT1, PGOOD LT1709/ LT1709-8 High Efficiency, 2-Phase Synchronous Step-Down Switching Regulators with 5-Bit VID 1.3V ≤ VOUT ≤ 3.5V, Current Mode Ensures Accurate Current Sharing, 3.5V ≤ VIN ≤ 36V LTC1735 High Efficiency Synchronous Step-Down Switching Regulator Output Fault Protection, 16-Pin SSOP LTC1736 High Efficiency Synchronous Controller with 5-Bit Mobile Output Fault Protection, 24-Pin SSOP, VID Control 3.5V ≤ VIN ≤ 36V Up to 97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN), No RSENSE Current Mode Synchronous Step-Down Controllers IOUT up to 20A LTC1778/LTC1778-1 LTC1929/ LTC1929-PG 2-Phase Synchronous Controllers Up to 42A, Uses All Surface Mount Components, No Heat Sinks, 3.5V ≤ VIN ≤ 36V LTC3708 Dual, 2-Phase, DC/DC Controller with Output Tracking Current Mode, No RSENSE, Up/Down Tracking, Synchronizable LTC3711 No RSENSE Current Mode Synchronous Step-Down Controller with Digital 5-Bit Interface Up to 97% Efficiency, Ideal for Pentium® III Processors, 0.925V ≤ VOUT ≤ 2V, 4V ≤ VIN ≤ 36V, IOUT up to 20A LTC3728 Dual, 550kHz, 2-Phase Synchronous Step-Down Controller Dual 180° Phased Controllers, VIN 3.5V to 35V, 99% Duty Cycle, 5x5QFN, SSOP-28 LTC3729 20A to 200A, 550kHz PolyPhase Synchronous Controller Expandable from 2-Phase to 12-Phase, Uses all Surface Mount Components, VIN up to 36V LTC3731 3- to 12-Phase Step-Down Synchronous Controller 60A to 240A Output Current, 0.6V ≤ VOUT ≤ 6V, 4.5V ≤ VIN ≤ 32V LTC3802 No RSENSE 2-Phase Dual Synchronous Step-Down Controller 330kHz to 750kHz, No Sense Resistor, Output Voltage Tracking, Synchronizable for 4-Phase Operation No RSENSE and PolyPhase are trademarks of Linear Technology Corporation. 3728l1fc 32 Linear Technology Corporation LT 0708 REV C • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005