LINER LTC1708-PG Dual, 550khz, 2-phase synchronous regulator Datasheet

LTC3728L-1
Dual, 550kHz, 2-Phase
Synchronous Regulator
DESCRIPTION
FEATURES
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Dual, 180° Phased Controllers Reduce Required
Input Capacitance and Power Supply Induced Noise
OPTI-LOOP® Compensation Minimizes COUT
±1.5% Output Voltage Accuracy
Power Good Output Voltage Indicator
Phase-Lockable Fixed Frequency 250kHz to 550kHz
Over Current Latchoff Disabled
Wide VIN Range: 4.5V to 28V (35V for LTC3728LI-1)
Operation
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Output Overvoltage Protection
Low Shutdown IQ: 20μA
5V and 3.3V Standby Regulators
3 Selectable Operating Modes: Constant Frequency,
Burst Mode® Operation and PWM
5mm × 5mm QFN and 28-Lead Narrow SSOP
Packages
The LTC®3728L-1 is a dual high performance step-down
switching regulator controller that drives all N-channel
synchronous power MOSFET stages. A constant frequency
current mode architecture allows phase-lockable frequency
of up to 550kHz. Power loss and noise due to the ESR of
the input capacitors are minimized by operating the two
controller output stages out of phase. The LTC3728L-1
is identical to the LTC3728L except that the LTC3728L-1
lacks the over current latchoff feature.
OPTI-LOOP compensation allows the transient response to
be optimized over a wide range of output capacitance and
ESR values. The precision 0.8V reference and power good
output indicator are compatible with future microprocessor generations, and a wide 4.5V to 28V (30V maximum/
35V for LTC3728LI-1) input supply range encompasses
all battery chemistries.
A RUN/SS pin for each controller provides soft-start.
Current foldback limits MOSFET dissipation during shortcircuit conditions. The FCB mode pin can select among
Burst Mode operation, constant frequency mode and
continuous inductor current mode or regulate a secondary
winding. The LTC3728L-1 includes a power good output
pin that indicates when both outputs are within 7.5% of
their designed set point.
APPLICATIONS
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Notebook and Palmtop Computers
Telecom Systems
Portable Instruments
Battery-Operated Digital Devices
DC Power Distribution Systems
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents including 5929620, 6177787, 6144199, 5471178, 5994885, 6100678.
TYPICAL APPLICATION
+
4.7μF
D3
TG1
L1
3.2μH
CB1, 0.1μF
RSENSE1
0.01Ω
+
R2
105k
COUT1
1%
47μF
6V
SP
BOOST2
SW1
SW2
LTC3728L-1
PLLIN
R1
20k
1%
L2
3.2μH
CB2, 0.1μF
BG2
PGND
SENSE1+
SENSE2+
SENSE1–
VOSENSE1
ITH1
SENSE2–
VOSENSE2
ITH2
1000pF
CC1
220pF
RC1
15k
M2
TG2
BOOST1
BG1
fIN
500kHz
VOUT1
5V
5A
D4
VIN PGOOD INTVCC
M1
RSENSE2
0.01Ω
1000pF
RUN/SS1 SGND RUN/SS2
CSS1
0.1μF
VIN
5.2V TO 28V
CIN
22μF
50V
CERAMIC
1μF
CERAMIC
CSS2
0.1μF
CC2
220pF
RC2
15k
R3
20k
1%
R4
63.4k
1%
M1, M2: FDS6982S
COUT
56μF
6V
SP
+
VOUT2
3.3V
5A
3728l1 F01
Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter
3728l1fc
1
LTC3728L-1
ABSOLUTE MAXIMUM RATINGS
(Note 1)
Input Supply Voltage (VIN) .........................30V to – 0.3V
Input Supply Voltage (VIN) LTC3728LI-1 ....35V to – 0.3V
Top Side Driver Voltages
(BOOST1, BOOST2) ............................... 36V to –0.3V
(BOOST1, BOOST2) LTC3728LI-1 .......... 38V to –0.3V
Switch Voltage (SW1, SW2) ........................... 30V to –5V
Switch Voltage (SW1, SW2) LTC3728LI-1... 35V to –0.3V
INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1),
(BOOST2-SW2), PGOOD ......................... 7V to –0.3V
INTVCC, EXTVCC LTC3728LI-1 ....................... 8V to –0.3V
SENSE1+, SENSE2+, SENSE1–,
SENSE2– Voltages .....................(1.1)INTVCC to –0.3V
PLLIN, PLLFLTR, FCB, Voltage .............. INTVCC to –0.3V
ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages .... 2.7V to –0.3V
Peak Output Current <10μs (TG1, TG2, BG1, BG2) .....3A
INTVCC Peak Output Current ..................................40mA
Operating Temperature Range (Note 7).... –40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range................... –65°C to 125°C
Reflow Peak Body Temperature (UH Package) ...... 260°C
Lead Temperature (Soldering, 10 sec)
GN Package ...................................................... 300°C
PIN CONFIGURATION
6
FCB
7
22 EXTVCC
ITH1
8
21 INTVCC
SGND
9
20 PGND
ITH2 11
23 BG1
17 SW2
16 TG2
SENSE2+ 14
15 RUN/SS2
GN PACKAGE
28-LEAD NARROW PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/W
SW1
TG1
PGOOD
20 INTVCC
SGND 6
19 PGND
3.3VOUT 7
18 BG2
ITH2 8
18 BOOST2
SENSE2– 13
21 EXTVCC
33
ITH1 5
19 BG2
VOSENSE2 12
22 BG1
FCB 4
17 BOOST2
9 10 11 12 13 14 15 16
VOSENSE2
3.3VOUT 10
23 VIN
PLLIN 3
NC
PLLIN
24 BOOST1
PLLFLTR 2
SW2
24 VIN
RUN/SS1
5
32 31 30 29 28 27 26 25
VOSENSE1 1
TG2
PLLFLTR
NC
25 BOOST1
RUN/SS2
4
SENSE2+
26 SW1
VOSENSE1
SENSE1–
27 TG1
3
SENSE2–
2
SENSE–
NC
SENSE1+
NC
RUN/SS1
28 PGOOD
SENSE1+
TOP VIEW
TOP VIEW
1
UH PACKAGE
32-LEAD (5mm s 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W
EXPOSED PAD (PIN 33) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
LTC3728LEGN-1#PBF
LTC3728LEGN-1#TRPBF LTC3728LEGN-1
LTC3728LIGN-1#PBF
LTC3728LIGN-1#TRPBF
LTC3728LIGN-1
LTC3728LEUH-1#PBF
LTC3728LEUH-1#TRPBF 728LE1
LTC3728LIUH-1#PBF
LTC3728LIUH-1#TRPBF
3728L1
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
LTC3728LEGN-1
LTC3728LEGN-1#TR
LTC3728LEGN-1
LTC3728LIGN-1
LTC3728LIGN-1#TR
LTC3728LIGN-1
LTC3728LEUH-1
LTC3728LEUH-1#TR
728LE1
LTC3728LIUH-1
LTC3728LIUH-1#TR
3728L1
Consult LTC Marketing for parts specified with wider operating temperature ranges.
PACKAGE DESCRIPTION
28-Lead Narrow Plastic SSOP
28-Lead Narrow Plastic SSOP
32-Lead (5mm × 5mm) Plastic DFN
32-Lead (5mm × 5mm) Plastic DFN
PACKAGE DESCRIPTION
28-Lead Narrow Plastic SSOP
28-Lead Narrow Plastic SSOP
32-Lead (5mm × 5mm) Plastic DFN
32-Lead (5mm × 5mm) Plastic DFN
TEMPERATURE RANGE
–40°C to 85°C
–40°C to 85°C
–40°C to 85°C
–40°C to 85°C
TEMPERATURE RANGE
–40°C to 85°C
–40°C to 85°C
–40°C to 85°C
–40°C to 85°C
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3728l1fc
2
LTC3728L-1
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.788
0.800
0.812
V
–5
–50
nA
0.002
0.02
%/V
0.1
– 0.1
0.5
–0.5
%
%
Main Control Loops
VOSENSE1, 2
Regulated Feedback Voltage
(Note 3); ITH1, 2 Voltage = 1.2V
IVOSENSE1, 2
Feedback Current
(Note 3)
VREFLNREG
Reference Voltage Line Regulation
VIN = 3.6V to 30V (Note 3)
VLOADREG
Output Voltage Load Regulation
(Note 3)
Measured in Servo Loop; ΔITH Voltage = 1.2V to 0.7V
Measured in Servo Loop; ΔITH Voltage = 1.2V to 2.0V
gm1, 2
Transconductance Amplifier gm
gmGBW1, 2
Transconductance Amplifier GBW
ITH1, 2 = 1.2V; (Note 3)
IQ
Input DC Supply Current
Normal Mode
Shutdown
(Note 4)
VIN = 15V; EXTVCC Tied to VOUT1; VOUT1 = 5V
VRUN/SS1, 2 = 0V
●
●
●
ITH1, 2 = 1.2V; Sink/Source 5μA; (Note 3)
1.3
mmho
3
●
MHz
450
20
35
μA
μA
0.76
0.800
0.84
V
– 0.50
–0.18
–0.1
μA
4.3
4.8
V
3.5
4
V
0.86
0.88
V
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.85V
VBINHIBIT
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB pin
UVLO
Undervoltage Lockout
VIN Ramping Down
●
VOVL
Feedback Overvoltage Lockout
Measured at VOSENSE1, 2
●
ISENSE
Sense Pins Total Source Current
(Each Channel); VSENSE1–, 2– = VSENSE1+, 2+ = 0V
–90
–60
μA
DFMAX
Maximum Duty Factor
In Dropout
98
99.4
%
IRUN/SS1, 2
Soft-Start Charge Current
VRUN/SS1, 2 = 1.9V
0.5
1.2
μA
VRUN/SS1, 2 ON RUN/SS Pin ON Threshold
0.84
VRUN/SS1, VRUN/SS2 Rising
1.0
1.5
2.0
V
VSENSE(MAX)
Maximum Current Sense Threshold
VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V
VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V
65
62
75
75
85
88
mV
mV
TG1, 2 tr
TG1, 2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 5)
CLOAD = 3300pF
CLOAD = 3300pF
55
55
100
100
ns
ns
BG1, 2 tr
BG1, 2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 5)
CLOAD = 3300pF
CLOAD = 3300pF
45
45
100
90
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
80
ns
BG/TG t2D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
80
ns
tON(MIN)
Minimum On-Time
Tested with a Square Wave (Note 6)
100
ns
●
INTVCC Linear Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
5.0
5.2
V
VLDO INT
INTVCC Load Regulation
ICC = 0 to 20mA, VEXTVCC = 4V
4.8
0.2
2.0
%
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V
100
200
mV
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
●
4.5
4.7
V
0.2
V
Oscillator and Phase-Locked Loop
fNOM
Nominal Frequency
VPLLFLTR = 1.2V
360
400
440
kHz
fLOW
Lowest Frequency
VPLLFLTR = 0V
230
260
290
kHz
fHIGH
Highest Frequency
VPLLFLTR ≥ 2.4V
480
550
590
kHz
RPLLIN
PLLIN Input Resistance
50
kΩ
3728l1fc
3
LTC3728L-1
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
I PLLFLTR
Phase Detector Output Current
Sinking Capability
Sourcing Capability
fPLLIN < fOSC
fPLLIN > fOSC
MIN
TYP
MAX
UNITS
–15
15
μA
μA
3.3V Linear Regulator
●
V3.3OUT
3.3V Regulator Output Voltage
No Load
3.35
3.45
V
V3.3IL
3.3V Regulator Load Regulation
I3.3 = 0 to 10mA
3.2
0.5
2
%
V3.3VL
3.3V Regulator Line Regulation
6V < VIN < 30V
0.05
0.2
%
I3.3LEAK
Leakage Current in Shutdown
VRUN/SS1 = 0V; VRUN/SS2 = 0V, VIN = 3V
10
50
μA
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
0.1
0.3
V
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
±1
μA
VPG
PGOOD Trip Level, Either Controller
VOSENSE with Respect to Set Output Voltage
VOSENSE Ramping Negative
VOSENSE Ramping Positive
– 9.5
9.5
%
%
●
PGOOD Output
–6
6
–7.5
7.5
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 6: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥ 40% of IMAX (see minimum on-time
considerations in the Applications Information section).
Note 7: The LTC3728LE-1 are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization
and correlation with statistical process controls. The LTC3728LI-1 is
guaranteed to meet performance specifications over the full –40°C to 85°C
operating temperature range
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC3728LUH-1: TJ = TA + (PD • 34°C/W)
LTC3728LGN-1: TJ = TA + (PD • 95°C/W)
Note 3: The IC is tested in a feedback loop that servos VITH1, 2 to a
specified voltage and measures the resultant VOSENSE1, 2.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
TYPICAL PERFORMANCCE CHARACTERISTICS
Efficiency vs Output Current
and Mode (Figure 13)
Efficiency vs Output Current
(Figure 13)
100
VIN = 7V
70
50
40
30
FORCED
CONTINUOUS
MODE (PWM)
CONSTANT
FREQUENCY
(BURST DISABLE)
20
VIN = 15V
VOUT = 5V
f = 250kHz
10
0
0.001
90
90
EFFICIENCY (%)
EFFICIENCY (%)
80
60
100
100
Burst Mode
OPERATION
0.1
0.01
1
OUTPUT CURRENT (A)
10
3728L1 G01
80
VIN = 10V
VIN = 15V
EFFICIENCY (%)
90
Efficiency vs Input Voltage
(Figure 13)
VIN = 20V
70
80
70
60
60
VOUT = 5V
IOUT = 3A
f = 250kHz
VOUT = 5V
f = 250kHz
50
0.001
50
0.1
0.01
1
OUTPUT CURRENT (A)
10
3728L1 G02
5
25
15
INPUT VOLTAGE (V)
35
3728L1 G03
3728l1fc
4
LTC3728L-1
TYPICAL PERFORMANCE CHARACTERISTICS
Supply Current vs Input Voltage
and Mode (Figure 13)
200
800
600
BOTH
CONTROLLERS ON
400
200
SHUTDOWN
0
0
5
150
100
50
0
20
15
10
INPUT VOLTAGE (V)
25
5.05
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
EXTVCC VOLTAGE DROP (mV)
1000
SUPPLY CURRENT (μA)
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
30
0
10
20
CURRENT (mA)
30
4.85
4.80
EXTVCC SWITCHOVER THRESHOLD
4.75
50
25
75
0
TEMPERATURE (°C)
100
3728L1 G06
Maximum Current Sense Threshold
vs Duty Factor
75
ILOAD = 1mA
125
Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)
80
70
5.0
60
4.8
4.7
50
VSENSE (mV)
4.9
VSENSE (mV)
INTVCC VOLTAGE (V)
4.90
3728L1 G05
Internal 5V LDO Line Regulation
25
4.6
50
40
30
20
4.5
10
4.4
0
5
20
15
10
INPUT VOLTAGE (V)
25
0
30
0
20
40
60
DUTY FACTOR (%)
80
3728L1 G07
80
0
100
50
0
100
25
75
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
3728L1 G08
Maximum Current Sense Threshold
vs VRUN/SS (Soft-Start)
3728L1 G09
Current Sense Threshold
vs ITH Voltage
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
90
80
VSENSE(CM) = 1.6V
80
70
76
40
60
VSENSE (mV)
VSENSE (mV)
60
VSENSE (mV)
4.95
4.70
–50 –25
40
3728L1 G04
5.1
INTVCC VOLTAGE
5.00
72
68
50
40
30
20
10
20
0
64
–10
–20
0
0
1
2
3
4
5
6
VRUN/SS (V)
3728L1 G10
60
–30
0
1
3
4
2
COMMON MODE VOLTAGE (V)
5
3728L1 G11
0
0.5
1
1.5
VITH (V)
2
2.5
3728L1 G12
3728l1fc
5
LTC3728L-1
TYPICAL PERFORMANCE CHARACTERISTICS
Load Regulation
VITH vs VRUN/SS
2.5
FCB = 0V
VIN = 15V
FIGURE 13
SENSE Pins Total Source Current
100
VOSENSE = 0.7V
2.0
–0.2
50
ISENSE (μA)
–0.1
VITH (V)
NORMALIZED VOUT (%)
0.0
1.5
1.0
–0.3
0
–50
0.5
–0.4
0
1
3
2
LOAD CURRENT (A)
4
0
5
0
2
1
3
4
5
6
–100
3728L1 G13
2
0
VRUN/SS (V)
4
3728L1 G14
Maximum Current Sense
Threshold vs Temperature
3728L1 G15
Dropout Voltage vs Output Current
(Figure 14)
80
4
6
VSENSE COMMON MODE VOLTAGE (V)
RUN/SS Current vs Temperature
1.8
VOUT = 5V
1.6
76
74
3
RUN/SS CURRENT (μA)
DROPOUT VOLTAGE (V)
VSENSE (mV)
78
2
RSENSE = 0.015Ω
1
1.4
1.2
1.0
0.8
0.6
0.4
72
RSENSE = 0.010Ω
0.2
70
–50
0
–25
50
25
0
75
TEMPERATURE (°C)
100
0
125
0.5
1.0 1.5 2.0 2.5 3.0
OUTPUT CURRENT (A)
3.5
4.0
0
–50
0
25
50
75
TEMPERATURE (°C)
3728L1 G18
3728L1 G17
Soft-Start Up (Figure 13)
100
125
3728L1 G25
Load Step (Figure 13)
VOUT
5V/DIV
–25
Load Step (Figure 13)
VOUT
200mV/DIV
VOUT
200mV/DIV
IL
2A/DIV
IL
2A/DIV
VRUN/SS
5V/DIV
IL
2A/DIV
VIN = 15V
VOUT = 5V
5ms/DIV
3728L1 G19
VIN = 15V
20μs/DIV
VOUT = 5V
VPLLFLTR = 50V
LOAD STEP = 0A TO 3A
Burst Mode OPERATION
3728L1 G20
VIN = 15V
20μs/DIV
VOUT = 5V
VPLLFLTR = 50V
LOAD STEP = 0A TO 3A
CONTINUOUS MODE
3728L1 G21
3728l1fc
6
LTC3728L-1
TYPICAL PERFORMANCE CHARACTERISTICS
Input Source/Capacitor
Instantaneous Current (Figure 13)
IIN
2A/DIV
VIN
200mV/DIV
Constant Frequency (Burst Inhibit)
Operation (Figure 13)
Burst Mode Operation (Figure 13)
VOUT
20mV/DIV
VOUT
20mV/DIV
IL
0.5A/DIV
IL
0.5A/DIV
VSW1
10V/DIV
VSW2
10V/DIV
VIN = 15V
1μs/DIV
VOUT1 = 5V, VOUT2 = 3.3V
VPLLFLTR = 0V
IOUT5 = IOUT3.3 = 2A
3728L1 G22
VIN = 15V
VOUT = 5V
VPLLFLTR = 0V
VFCB = 5V
IOUT = 20mA
3728L1 G23
10μs/DIV
EXTVCC SWITCH RESISTANCE (Ω)
CURRENT SENSE INPUT CURRENT (μA)
33
31
29
27
–25
50
25
0
75
TEMPERATURE (°C)
100
8
6
4
2
0
–50
125
–25
50
25
0
75
TEMPERATURE (°C)
3728L1 G26
3.50
UNDERVOLTAGE LOCKOUT (V)
FREQUENCY (kHz)
VPLLFLTR = 2.4V
500
300
VPLLFLTR = 1.2V
VPLLFLTR = 0V
200
100
0
–50 –25
125
Undervoltage Lockout
vs Temperature
700
400
100
3728L1 G27
Oscillator Frequency
vs Temperature
600
3728L1 G24
10
VOUT = 5V
25
–50
2μs/DIV
EXTVCC Switch Resistance
vs Temperature
Current Sense Pin Input Current
vs Temperature
35
VIN = 15V
VOUT = 5V
VPLLFLTR = 0V
VFCB = 5V
IOUT = 20mA
50
25
75
0
TEMPERATURE (°C)
100
125
3728L1 G28
3.45
3.40
3.35
3.30
3.25
3.20
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
3728L1 G29
3728l1fc
7
LTC3728L-1
PIN FUNCTIONS
VOSENSE1, VOSENSE2: Error Amplifier Feedback Input.
Receives the remotely-sensed feedback voltage for each
controller from an external resistive divider across the
output.
TG2, TG1: High Current Gate Drives for Top N-Channel
MOSFETs. These are the outputs of floating drivers with
a voltage swing equal to INTVCC – 0.5V superimposed on
the switch node voltage SW.
PLLFLTR: Filter Connection for Phase-Locked Loop. Alternatively, this pin can be driven with an AC or DC voltage
source to vary the frequency of the internal oscillator.
SW2, SW1: Switch Node Connections to Inductors. Voltage
swing at these pins is from a Schottky diode (external)
voltage drop below ground to VIN.
PLLIN: External Synchronization Input to Phase Detector.
This pin is internally terminated to SGND with 50kΩ. The
phase-locked loop will force the rising top gate signal of
controller 1 to be synchronized with the rising edge of
the PLLIN signal.
BOOST2, BOOST1: Bootstrapped Supplies to the Top
Side Floating Drivers. Capacitors are connected between
the boost and switch pins and Schottky diodes are tied
between the boost and INTVCC pins. Voltage swing at the
boost pins is from INTVCC to (VIN + INTVCC).
FCB: Forced Continuous Control Input. This input acts
on both controllers and is normally used to regulate a
secondary winding. Pulling this pin below 0.8V will force
continuous synchronous operation.
BG2, BG1: High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins
is from ground to INTVCC.
ITH1, ITH2: Error Amplifier Output and Switching Regulator
Compensation Point. Each associated channels’ current
comparator trip point increases with this control voltage.
SGND: Small Signal Ground. Common to both controllers,
this pin must be routed separately from high current
grounds to the common (–) terminals of the COUT
capacitors.
3.3VOUT: Linear Regulator Output. Capable of supplying
10mA DC with peak currents as high as 50mA.
NC: No Connect.
SENSE2 –, SENSE1–: The (–) Input to the Differential Current Comparators.
SENSE2+, SENSE1+: The (+) Input to the Differential Current
Comparators. The ITH pin voltage and controlled offsets
between the SENSE– and SENSE+ pins in conjunction with
RSENSE set the current trip threshold.
RUN/SS2, RUN/SS1: Combination of soft-start and run
control inputs. A capacitor to ground at each of these pins
sets the ramp time to full output current. Forcing either of
these pins back below 1.0V causes the IC to shut down
the circuitry required for that particular controller.
PGND: Driver Power Ground. Connects to the sources of
bottom (synchronous) N-channel MOSFETs, anodes of the
Schottky rectifiers and the (–) terminal(s) of CIN.
INTVCC: Output of the Internal 5V Linear Low Dropout
Regulator and the EXTVCC Switch. The driver and control
circuits are powered from this voltage source. Must be
decoupled to power ground with a minimum of 4.7μF
tantalum or other low ESR capacitor.
EXTVCC: External Power Input to an Internal Switch Connected to INTVCC. This switch closes and supplies VCC
power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications section. Do not exceed 7V on this pin.
VIN: Main Supply Pin. A bypass capacitor should be tied
between this pin and the signal ground pin.
PGOOD: Open-Drain Logic Output. PGOOD is pulled to
ground when the voltage on either VOSENSE pin is not
within ±7.5% of its set point.
Exposed Pad (UH Package Only): Signal Ground. Must
be soldered to the PCB, providing a local ground for the
control components of the IC, and be tied to the PGND
pin under the IC.
3728l1fc
8
LTC3728L-1
FUNCTIONAL DIAGRAM
PLLIN
FIN
INTVCC
PHASE DET
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
50k
BOOST
VIN
DB
PLLFLTR
DROP
OUT
DET
CLK1
RLP
OSCILLATOR
CLK2
CLP
–
0.86V
S
Q
R
Q
BOT
VOSENSE1
–
0.86V
SWITCH
LOGIC
+
0.55V
INTVCC
BG
COUT
PGND
B
–
+
+
SHDN
VOSENSE2
CIN
SW
TOP ON
BOT
0.74V
+
D1
–
+
CB
FCB
+
PGOOD
TG
TOP
VOUT
RSENSE
–
INTVCC
+
3V
4.5V
0.18μA
R6
0.74V
I1
–
+
–
BINH
FCB
+
R5
–
3.3VOUT
+
0.8V
+
–
++
SLOPE
COMP
+
30k SENSE
+
–
30k SENSE
45k
45k
2.4V
VREF
–
EA
+
–
OV
VIN
VIN
EXTVCC
+
–
5V
LDO
REG
VOSENSE
0.80V
SHDN
RST
4(VFB)
+
R2
R1
0.86V
ITH
1.2μA
6V
INTVCC
VFB
+
–
4.8V
5V
INTVCC
I2
–
3mV
0.86V
4(VFB)
FCB
–
RUN
SOFT
START
CC
CC2
RC
RUN/SS
SGND (UH PACKAGE PAD)
INTERNAL
SUPPLY
CSS
3728L1 F02
Figure 2
3728l1fc
9
LTC3728L-1
OPERATION
(Refer to Functional Diagram)
Main Control Loop
The LTC3728L-1 is a constant frequency, current mode
step-down controller with two channels operating 180
degrees out of phase. During normal operation, each top
MOSFET is turned on when the clock for that channel sets
the RS latch, and turned off when the main current comparator, I1, resets the RS latch. The peak inductor current
at which I1 resets the RS latch is controlled by the voltage
on the ITH pin, which is the output of each error amplifier
EA. The VOSENSE pin receives the voltage feedback signal,
which is compared to the internal reference voltage by the
EA. When the load current increases, it causes a slight
decrease in VOSENSE relative to the 0.8V reference, which
in turn causes the ITH voltage to increase until the average
inductor current matches the new load current. After the
top MOSFET has turned off, the bottom MOSFET is turned
on until either the inductor current starts to reverse, as
indicated by current comparator I2, or the beginning of
the next cycle.
The top MOSFET drivers are biased from floating bootstrap
capacitor CB, which normally is recharged during each off
cycle through an external diode when the top MOSFET
turns off. As VIN decreases to a voltage close to VOUT, the
loop may enter dropout and attempt to turn on the top
MOSFET continuously. The dropout detector detects this
and forces the top MOSFET off for about 400ns every tenth
cycle to allow CB to recharge.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 1.2μA current source to charge soft-start capacitor CSS. When CSS
reaches 1.5V, the main control loop is enabled with the ITH
voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, the ITH pin voltage is
gradually released allowing normal, full-current operation.
When both RUN/SS1 and RUN/SS2 are low, all controller
functions are shut down, including the 5V regulator.
Low Current Operation
The FCB pin is a multifunction pin providing two functions: 1) to provide regulation for a secondary winding
by temporarily forcing continuous PWM operation on
both controllers; and 2) to select between two modes
of low current operation. When the FCB pin voltage is
below 0.8V, the controller forces continuous PWM current mode operation. In this mode, the top and bottom
MOSFETs are alternately turned on to maintain the output
voltage independent of direction of inductor current.
When the FCB pin is below VINTVCC – 2V but greater
than 0.8V, the controller enters Burst Mode operation.
Burst Mode operation sets a minimum output current
level before inhibiting the top switch and turns off the
synchronous MOSFET(s) when the inductor current goes
negative. This combination of requirements will, at low
currents, force the ITH pin below a voltage threshold that
will temporarily inhibit turn-on of both output MOSFETs
until the output voltage drops. There is 60mV of hysteresis in the burst comparator B tied to the ITH pin. This
hysteresis produces output signals to the MOSFETs that
turn them on for several cycles, followed by a variable
“sleep” interval depending upon the load current. The
resultant output voltage ripple is held to a very small
value by having the hysteretic comparator after the error
amplifier gain block.
Frequency Synchronization
The phase-locked loop allows the internal oscillator to
be synchronized to an external source via the PLLIN pin.
The output of the phase detector at the PLLFLTR pin is
also the DC frequency control input of the oscillator that
operates over a 260kHz to 550kHz range corresponding
to a DC voltage input from 0V to 2.4V. When locked, the
PLL aligns the turn on of the top MOSFET to the rising
edge of the synchronizing signal. When PLLIN is left
open, the PLLFLTR pin goes low, forcing the oscillator to
minimum frequency.
Constant Frequency Operation
When the FCB pin is tied to INTVCC, Burst Mode operation is disabled and the forced minimum output current
requirement is removed. This provides constant frequency,
discontinuous current (preventing reverse inductor current) operation over the widest possible output current
range. This constant frequency operation is not as efficient
3728l1fc
10
LTC3728L-1
OPERATION
(Refer to Functional Diagram)
as Burst Mode operation, but does provide a lower noise,
constant frequency operating mode down to approximately
1% of the designed maximum output current.
turned off within 10μs and the pin is allowed to be pulled
up by an external resistor to a source of up to 7V.
Foldback Current
Continuous Current (PWM) Operation
Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode, but
may be desirable in certain applications. The output can
source or sink current in this mode. When sinking current
while in forced continuous operation, the controller will
cause current to flow back into the input filter capacitor.
If large enough, this element will prevent the input supply from boosting to unacceptably high levels; see COUT
Selection in the Applications Information Section.
INTVCC /EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. When
the EXTVCC pin is left open, an internal 5V low dropout
linear regulator supplies INTVCC power. If EXTVCC is taken
above 4.7V, the 5V regulator is turned off and an internal
switch is turned on connecting EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency
external source such as the output of the regulator itself
or a secondary winding, as described in the Applications
Information section.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
Power Good (PGOOD) Pin
The PGOOD pin is connected to an open drain of an internal
MOSFET. The MOSFET turns on and pulls the pin low when
either output is not within ± 7.5% of the nominal output
level as determined by the resistive feedback divider. When
both outputs meet the ±7.5% requirement, the MOSFET is
The RUN/SS capacitors are used initially to limit the inrush
current of each switching regulator. Foldback current limiting is activated when the output voltage falls below 70%
of its nominal level. If a short is present, a safe, low output
current is provided due to the internal current foldback
and actual power wasted is low due to the efficient nature
of the current mode switching regulator.
THEORY AND BENEFITS OF 2-PHASE OPERATION
The LTC1628 and the LTC3728L-1 family of dual high efficiency DC/DC controllers brings the considerable benefits
of 2-phase operation to portable applications for the first
time. Notebook computers, PDAs, handheld terminals and
automotive electronics will all benefit from the lower input
filtering requirement, reduced electromagnetic interference
(EMI) and increased efficiency associated with 2-phase
operation.
Why the need for 2-phase operation? Up until the 2-phase
family, constant-frequency dual switching regulators
operated both channels in phase (i.e., single-phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of phase.
This effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
3728l1fc
11
LTC3728L-1
OPERATION
(Refer to Functional Diagram)
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
3728L1 F03a
IIN(MEAS) = 2.53ARMS
IIN(MEAS) = 1.55ARMS
(a)
3728L1 F03b
(b)
Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators
Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the LTC1628 2-Phase Regulator Allows
Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the LTC1628
2-phase dual switching regulator. An actual measurement of
the RMS input current under these conditions shows that 2phase operation dropped the input current from 2.53ARMS
to 1.55ARMS. While this is an impressive reduction in itself,
remember that the power losses are proportional to IRMS2,
meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means
less power is lost in the input power path, which could
include batteries, switches, trace/connector resistances
and protection circuitry. Improvements in both conducted
and radiated EMI also directly accrue as a result of the
reduced RMS input current and voltage.
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how
the RMS input current varies for single-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but
in fact extend over a wide region. A good rule of thumb
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.
3.0
SINGLE PHASE
DUAL CONTROLLER
INPUT RMS CURRENT (A)
2.5
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
3728L1 F04
Figure 4. RMS Input Current Comparison
3728l1fc
12
LTC3728L-1
APPLICATIONS INFORMATION
2.5
PLLFLTR PIN VOLTAGE (V)
Figure 1 on the first page is a basic LTC3728L-1 application circuit. External component selection is driven by
the load requirement, and begins with the selection of
RSENSE and the inductor value. Next, the power MOSFETs
and D1 are selected. Finally, CIN and COUT are selected.
The circuit shown in Figure 1 can be configured for
operation up to an input voltage of 28V (limited by the
external MOSFETs).
RSENSE Selection For Output Current
RSENSE is chosen based on the required output current.
The current comparator has a maximum threshold of
75mV/RSENSE and an input common mode range of SGND
to 1.1(INTVCC). The current comparator threshold sets the
peak of the inductor current, yielding a maximum average
output current IMAX equal to the peak value less half the
peak-to-peak ripple current, ΔIL.
Allowing a margin for variations in the IC and external
component values yields:
50mV
RSENSE =
IMAX
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability criteria
for buck regulators operating at greater than 50% duty
factor. A curve is provided to estimate this reduction in
peak output current level depending upon the operating
duty factor.
Operating Frequency
The IC uses a constant frequency phase-lockable architecture with the frequency determined by an internal
capacitor. This capacitor is charged by a fixed current plus
an additional current which is proportional to the voltage
applied to the PLLFLTR pin. Refer to Phase-Locked Loop
and Frequency Synchronization in the Applications Information section for additional information.
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 5. As the operating frequency
is increased the gate charge losses will be higher, reducing
2.0
1.5
1.0
0.5
0
200
300
400
500
OPERATING FREQUENCY (kHz)
600
3728L1 F05
Figure 5. PLLFLTR Pin Voltage vs Frequency
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 550kHz.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current ΔIL decreases with higher
inductance or frequency and increases with higher VIN:
IL =
V 1
VOUT 1– OUT (f)(L)
VIN Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple and
greater core losses. A reasonable starting point for setting
ripple current is ΔIL=0.3(IMAX). The maximum ΔIL occurs
at the maximum input voltage.
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
3728l1fc
13
LTC3728L-1
APPLICATIONS INFORMATION
25% of the current limit determined by RSENSE. Lower
inductor values (higher ΔIL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Usually, high inductance is preferred for small current
ripple and low core loss. Unfortunately, increased inductance requires more turns of wire or a smaller air gap in
the inductor core, resulting in high copper loss or low
saturation current. Once the value of L is known, the actual
inductor must be selected. There are two popular types
of core material of commercial available inductors. Ferrite
core inductors usually have very low core loss and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation. However, ferrite core saturates “hard”, which means
that inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. One advantage of the LTC3728L-1 is its current
mode control that detects and limits cycle-by-cycle peak
inductor current. Therefore, accurate and fast protection
is achieved if the inductor is saturated in steady state or
during transient mode.
Powdered iron core inductors usually saturate “soft”, which
means the inductance drops in a linear fashion when the
current increases. However, the core loss of the powder
iron inductor is usually higher than the ferrite inductor.
So designs with high switching frequency should also
address inductor core loss.
Inductor manufacturers usually provide inductance, DCR,
(peak) saturation current and (DC) heating current ratings
in the inductor data sheet. A good supply design should
not exceed the saturation and heating current rating of
the inductor.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for each
controller in the LTC3728L-1: One N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up
(see EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected (VIN
< 5V); then, sub-logic level threshold MOSFETs (VGS(TH)
< 3V) should be used. Pay close attention to the BVDSS
specification for the MOSFETs as well; most of the logic
level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), Miller capacitance CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the Gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN – VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
PMAIN =
VOUT
2
IMAX ) (1+ )RDS(ON) +
(
VIN
( VIN )2 IMAX
(RDR )(CMILLER ) •
2
1
1
+
VINTVCC – VTHMIN VTHMIN
PSYNC =
( f)
VIN – VOUT
2
IMAX ) (1+ )RDS(ON)
(
VIN
3728l1fc
14
LTC3728L-1
APPLICATIONS INFORMATION
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 4Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTHMIN is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1+δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two
power MOSFETs. This prevents the body diode of the
bottom MOSFET from turning on, storing charge during
the dead-time and requiring a reverse recovery period
that could cost as much as 3% in efficiency at high VIN.
A 1A to 3A Schottky is generally a good compromise for
both regions of operation due to the relatively small average current. Larger diodes result in additional transition
losses due to their larger junction capacitance. Schottky
diodes should be placed in parallel with the synchronous
MOSFETs when operating in pulse-skip mode or in Burst
Mode operation.
CIN and COUT Selection
The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current
drawn through the input network (battery/fuse/capacitor).
It can be shown that the worst case RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS current
requirement. Increasing the output current, drawn from
the other out-of-phase controller, will actually decrease the
input RMS ripple current from this maximum value (see
Figure 4). The out-of-phase technique typically reduces
the input capacitor’s RMS ripple current by a factor of
30% to 70% when compared to a single phase power
supply solution.
The type of input capacitor, value and ESR rating have
efficiency effects that need to be considered in the selection process. The capacitance value chosen should be
sufficient to store adequate charge to keep high peak
battery currents down. 20μF to 40μF is usually sufficient
for a 25W output supply operating at 200kHz. The ESR of
the capacitor is important for capacitor power dissipation
as well as overall battery efficiency. All of the power (RMS
ripple current • ESR) not only heats up the capacitor but
wastes power from the battery.
Medium voltage (20V to 35V) ceramic, tantalum, OS-CON
and switcher-rated electrolytic capacitors can be used
as input capacitors, but each has drawbacks: ceramics
have very high voltage coefficients and may have audible
piezoelectric effects; tantalums need to be surge-rated;
OS-CONs suffer from higher inductance, larger case size
and limited surface-mount applicability; electrolytics’
higher ESR and dryout possibility require several to be
used. Multiphase systems allow the lowest amount of
capacitance overall. As little as one 22μF or two to three
10μF ceramic capacitors are an ideal choice in a 20W to
35W power supply due to their extremely low ESR. Even
though the capacitance at 20V is substantially below their
rating at zero-bias, very low ESR loss makes ceramics
an ideal candidate for highest efficiency battery operated
systems. Also consider parallel ceramic and high quality
electrolytic capacitors as an effective means of achieving
ESR and bulk capacitance goals.
In continuous mode, the source current of the top N-channel
MOSFET is a square wave of duty cycle VOUT/VIN. To prevent
large voltage transients, a low ESR input capacitor sized for
the maximum RMS current of one channel must be used.
The maximum RMS capacitor current is given by:
CIN Required IRMS IMAX
VOUT ( VIN VOUT ) VIN
1/2
3728l1fc
15
LTC3728L-1
APPLICATIONS INFORMATION
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the manufacturer if there is any question.
The benefit of the LTC3728L-1 multiphase clocking can
be calculated by using the equation above for the higher
power controller and then calculating the loss that would
have resulted if both controller channels switched on at
the same time. The total RMS power lost is lower when
both controllers are operating due to the interleaving of
current pulses through the input capacitor’s ESR. This is
why the input capacitor’s requirement calculated above for
the worst-case controller is adequate for the dual controller
design. Remember that input protection fuse resistance,
battery resistance and PC board trace resistance losses are
also reduced due to the reduced peak currents in a multiphase system. The overall benefit of a multiphase design
will only be fully realized when the source impedance of
the power supply/battery is included in the efficiency testing. The drains of the two top MOSFETS should be placed
within 1cm of each other and share a common CIN(s).
Separating the drains and CIN may produce undesirable
voltage and current resonances at VIN.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (ΔVOUT) is determined by:
1 VOUT IL ESR +
8fCOUT Where f = operating frequency, COUT = output capacitance,
and ΔIL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ΔIL increases
with input voltage. With ΔIL = 0.3IOUT(MAX) the output
ripple will typically be less than 50mV at the maximum
VIN assuming:
COUT Recommended ESR < 2 RSENSE
and COUT > 1/(8fRSENSE)
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees
that the output capacitance does not significantly discharge
during the operating frequency period due to ripple current.
The choice of using smaller output capacitance increases
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage at or below 50mV. The ITH pin
OPTI-LOOP compensation components can be optimized
to provide stable, high performance transient response
regardless of the output capacitors selected.
Manufacturers such as Nichicon, Nippon Chemi-Con and
Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications multiple capacitors may
need to be used in parallel to meet ESR, RMS current handling and load step requirements. Aluminum
electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special
polymer surface mount capacitors offer very low ESR
but have lower storage capacity per unit volume than
other capacitor types. These capacitors offer a very
cost-effective output capacitor solution and are an ideal
choice when combined with a controller having high
loop bandwidth. Tantalum capacitors offer the highest
capacitance density and are often used as output capacitors for switching regulators having controlled soft-start.
Several excellent surge-tested choices are the AVX TPS,
AVX TPSV or the KEMET T510 series of surface mount
tantalums, available in case heights ranging from 2mm
to 4mm. Aluminum electrolytic capacitors can be used
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
in cost-driven applications providing that consideration
is given to ripple current ratings, temperature and long
term reliability. A typical application will require several
to many aluminum electrolytic capacitors in parallel. A
combination of the above mentioned capacitors will often
result in maximizing performance and minimizing overall
cost. Other capacitor types include Nichicon PL series,
Panasonic SP, NEC Neocap, Cornell Dubilier ESRE and
Sprague 595D series. Consult manufacturers for other
specific recommendations.
INTVCC Regulator
An internal P-channel low dropout regulator produces
5V at the INTVCC pin from the VIN supply pin. INTVCC
powers the drivers and internal circuitry within the IC.
The INTVCC pin regulator can supply a peak current of
50mA and must be bypassed to ground with a minimum
of 4.7μF tantalum, 10μF special polymer, or low ESR type
electrolytic capacitor. A 1μF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly
recommended. Good bypassing is necessary to supply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between channels.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the IC to be exceeded.
The system supply current is normally dominated by the
gate charge current. Additional external loading of the
INTVCC and 3.3V linear regulators also needs to be taken
into account for the power dissipation calculations. The
total INTVCC current can be supplied by either the 5V internal linear regulator or by the EXTVCC input pin. When
the voltage applied to the EXTVCC pin is less than 4.7V, all
of the INTVCC current is supplied by the internal 5V linear
regulator. Power dissipation for the IC in this case is highest: (VIN)(IINTVCC), and overall efficiency is lowered. The
gate charge current is dependent on operating frequency
as discussed in the Efficiency Considerations section.
The junction temperature can be estimated by using the
equations given in Note 2 of the Electrical Characteristics.
For example, the IC VIN current is thermally limited to less
than 67mA from a 24V supply when not using the EXTVCC
pin as follows:
TJ = 70°C + (67mA)(24V)(34°C/W) = 125°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + (67mA)(5V)(34°C/W) = 81°C
The absolute maximum rating for the INTVCC Pin is 40mA.
Dissipation should be calculated to also include any added
current drawn from the internal 3.3V linear regulator.
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
EXTVCC Connection
The IC contains an internal P-channel MOSFET switch
connected between the EXTVCC and INTVCC pins. When
the voltage applied to EXTVCC rises above 4.7V, the internal
regulator is turned off and the switch closes, connecting
the EXTVCC pin to the INTVCC pin thereby supplying internal
power. The switch remains closed as long as the voltage
applied to EXTVCC remains above 4.5V. This allows the
MOSFET driver and control power to be derived from the
output during normal operation (4.7V < VOUT < 7V) and
from the internal regulator when the output is out of regulation (start-up, short-circuit). If more current is required
through the EXTVCC switch than is specified, an external
Schottky diode can be added between the EXTVCC and
INTVCC pins. Do not apply greater than 7V to the EXTVCC
pin and ensure that EXTVCC < VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC Left Open (or Grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in an
efficiency penalty of up to 10% at high input voltages.
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
2. EXTVCC Connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC Connected to an External supply. If an external
supply is available in the 5V to 7V range, it may be used to
power EXTVCC providing it is compatible with the MOSFET
gate drive requirements.
4. EXTVCC Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an outputderived voltage that has been boosted to greater than 4.7V.
This can be done with either the inductive boost winding
as shown in Figure 6a or the capacitive charge pump
shown in Figure 6b. The charge pump has the advantage
of simple magnetics.
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the functional diagram is charged through
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on,
the driver places the CB voltage across the gate-source
of the desired MOSFET. This enhances the MOSFET and
turns on the topside switch. The switch node voltage, SW,
rises to VIN and the BOOST pin follows. With the topside
MOSFET on, the boost voltage is above the input supply:
VBOOST = VIN + VINTVCC. The value of the boost capacitor
CB needs to be 100 times that of the total input capacitance
of the topside MOSFET(s). The reverse breakdown of the
external Schottky diode must be greater than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the efficiency has
improved. If there is no change in input current, then there
is no change in efficiency.
Output Voltage
The output voltages are each set by an external feedback
resistive divider carefully placed across the output capacitor.
The resultant feedback signal is compared with the internal
precision 0.800V voltage reference by the error amplifier.
The output voltage is given by the equation:
R2 VOUT = 0.8V 1+ R1
where R1 and R2 are defined in Figure 2.
SENSE+/SENSE– Pins
The common mode input range of the current comparator
sense pins is from 0V to (1.1)INTVCC. Continuous linear
operation is guaranteed throughout this range allowing
output voltage setting from 0.8V to 7.7V, depending upon
the voltage applied to EXTVCC. A differential NPN input
+
CIN
CIN
BAT 85
VIN
+
VIN
VIN
OPTIONAL EXTVCC
CONNECTION
5V < VSEC < 7V
VSEC
+
BAT85
VIN
1μF
RSENSE
N-CH
SW
FCB
BG1
EXTVCC
+
COUT
COUT
BG1
N-CH
N-CH
SGND
VOUT
L1
SW
+
R5
BAT85
RSENSE
N-CH
VOUT
T1
1:N
R6
BAT85
VN2222LL
TG1
TG1
EXTVCC
0.22μF
LTC3728L-1
+
LTC3728L-1
1μF
PGND
PGND
3728L1 F06a
Figure 6a. Secondary Output Loop & EXTVCC Connection
3728L1 F06b
Figure 6b. Capacitive Charge Pump for EXTVCC
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LTC3728L-1
APPLICATIONS INFORMATION
stage is biased with internal resistors from an internal 2.4V
source as shown in the Functional Diagram. This requires
that current either be sourced or sunk from the SENSE
pins depending on the output voltage. If the output voltage
is below 2.4V current will flow out of both SENSE pins to
the main output. The output can be easily preloaded by
the VOUT resistive divider to compensate for the current
comparator’s negative input bias current. The maximum
current flowing out of each pair of SENSE pins is:
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Since VOSENSE is servoed to the 0.8V reference voltage,
we can choose R1 in Figure 2 to have a maximum value
to absorb this current.
0.8V
R1(MAX) = 24k 2.4V – VOUT for VOUT < 2.4V
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32k. Note that for an output voltage above
2.4V, R1 has no maximum value necessary to absorb the
sense currents; however, R1 is still bounded by the VOSENSE
feedback current.
Soft-Start/Run Function
The RUN/SS1 and RUN/SS2 pins are multipurpose pins that
provide a soft-start function and a means to shut down the
LTC3728L-1. Soft-start reduces the input power source’s
surge currents by gradually increasing the controller’s
current limit (proportional to VITH). This pin can also be
used for power supply sequencing.
An internal 1.2μA current source charges up the CSS capacitor. When the voltage on RUN/SS1 (RUN/SS2) reaches 1.5V,
the particular controller is permitted to start operating. As
the voltage on RUN/SS increases from 1.5V to 3.0V, the
internal current limit is increased from 25mV/RSENSE to
75mV/RSENSE. The output current limit ramps up slowly,
taking an additional 1.25s/μF to reach full current. The
output current thus ramps up slowly, reducing the starting surge current required from the input power supply.
If RUN/SS has been pulled all the way to ground there is
a delay before starting of approximately:
tDELAY =
1.5V
C = (1.25s / μF ) CSS
1.2μA SS
tIRAMP =
3V 1.5V
C = (1.25s / μF ) CSS
1.2μA SS
By pulling both RUN/SS pins below 1V, the IC is put into
low current shutdown (IQ = 20μA). The RUN/SS pins
can be driven directly from logic as shown in Figure 7.
Diode D1 in Figure 7 reduces the start delay but allows
CSS to ramp up slowly providing the soft-start function.
Each RUN/SS pin has an internal 6V zener clamp (See
Functional Diagram). Because the LTC3728L-1 is designed
for applications not requiring over current latchoff, no
pull-up resistor is required on the RUN/SS pin to defeat
latchoff. Refer to the LTC3728L/LTC3728LX datasheet if
this feature is required.
Fault Conditions: Current Limit and Current Foldback
The current comparators have a maximum sense voltage of 75mV resulting in a maximum MOSFET current
of 75mV/RSENSE. The maximum value of current limit
generally occurs with the largest VIN at the highest ambi3.3V OR 5V
RUN/SS
RUN/SS
D1
CSS
CSS
(a)
(b)
3728L1 F07
Figure 7. RUN/SS Pin Interfacing
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LTC3728L-1
APPLICATIONS INFORMATION
ent temperature, conditions that cause the highest power
dissipation in the top MOSFET.
Each controller includes current foldback to help further
limit load current when the output is shorted to ground.
The foldback circuit is active even when the overload
shutdown latch described above is overridden. If the
output falls below 70% of its nominal output level, then
the maximum sense voltage is progressively lowered from
75mV to 17mV. Under short-circuit conditions with very
low duty cycles, the controller will begin cycle skipping
in order to limit the short-circuit current. In this situation
the bottom MOSFET will be dissipating most of the power
but less than in normal operation. The short-circuit ripple
current is determined by the minimum on-time tON(MIN)
of each controller (typically 100ns), the input voltage and
inductor value:
ΔIL(SC) = tON(MIN) (VIN /L)
The resulting short-circuit current is:
ISC =
25mV 1
– I
RSENSE 2 L(SC)
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes huge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the controller
is operating.
A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage. When
this condition is sensed, the top MOSFET is turned off and
the bottom MOSFET is turned on until the overvoltage
condition is cleared. The output of this comparator is
only latched by the overvoltage condition itself and will
therefore allow a switching regulator system having a poor
PC layout to function while the design is being debugged.
The bottom MOSFET remains on continuously for as long
as the OV condition persists; if VOUT returns to a safe level,
normal operation automatically resumes. A shorted top
MOSFET will result in a high current condition which will
open the system fuse. The switching regulator will regulate
properly with a leaky top MOSFET by altering the duty
cycle to accommodate the leakage.
Phase-Locked Loop and Frequency Synchronization
The IC has a phase-locked loop comprised of an internal
voltage controlled oscillator and phase detector. This allows the top MOSFET turn-on to be locked to the rising
edge of an external source. The frequency range of the
voltage controlled oscillator is ± 50% around the center
frequency fO. A voltage of 1.2V applied to the PLLFLTR
pin corresponds to a frequency of approximately 400kHz.
The nominal operating frequency range of the IC is 260kHz
to 550kHz.
The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the external and internal oscillators. This type of phase detector
will not lock up on input frequencies close to the harmonics
of the VCO center frequency. The PLL hold-in range, ΔfH,
is equal to the capture range, ΔfC:
ΔfH = ΔfC = ±0.5 fO (260kHz-550kHz)
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLFLTR pin.
If the external frequency (fPLLIN) is greater than the oscillator frequency f0SC, current is sourced continuously,
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
pulling up the PLLFLTR pin. When the external frequency is
less than f0SC, current is sunk continuously, pulling down
the PLLFLTR pin. If the external and internal frequencies
are the same but exhibit a phase difference, the current
sources turn on for an amount of time corresponding to
the phase difference. Thus the voltage on the PLLFLTR pin
is adjusted until the phase and frequency of the external
and internal oscillators are identical. At this stable operating point the phase comparator output is open and the
filter capacitor CLP holds the voltage. The IC’s PLLIN pin
must be driven from a low impedance source such as a
logic gate located close to the pin. When using multiple
ICs for a phase-locked system, the PLLFLTR pin of the
master oscillator should be biased at a voltage that will
guarantee the slave oscillator(s) ability to lock onto the
master’s frequency. A DC voltage of 0.7V to 1.7V applied
to the master oscillator’s PLLFLTR pin is recommended
in order to meet this requirement. The resultant operating
frequency can range from 300kHz to 500kHz.
The loop filter components (CLP, RLP) smooth out the current pulses from the phase detector and provide a stable
input to the voltage controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires
lock. Typically RLP =10kΩ and CLP is 0.01μF to 0.1μF.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration that
each controller is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that
tON(MIN) <
VOUT
VIN(f)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for each controller is approximately
100ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases up to about 150ns.
This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle
drops below the minimum on-time limit in this situation,
a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple.
FCB Pin Operation
The FCB pin can be used to regulate a secondary winding
or as a logic level input. Continuous operation is forced
on both controllers when the FCB pin drops below 0.8V.
During continuous mode, current flows continuously in
the transformer primary. The secondary winding(s) draw
current only when the bottom, synchronous switch is on.
When primary load currents are low and/or the VIN/VOUT
ratio is low, the synchronous switch may not be on for a
sufficient amount of time to transfer power from the output
capacitor to the secondary load. Forced continuous operation will support secondary windings providing there is
sufficient synchronous switch duty factor. Thus, the FCB
input pin removes the requirement that power must be
drawn from the inductor primary in order to extract power
from the auxiliary windings. With the loop in continuous
mode, the auxiliary outputs may nominally be loaded
without regard to the primary output load.
The secondary output voltage VSEC is normally set as shown
in Figure 6a by the turns ratio N of the transformer:
VSEC ≅ (N + 1) VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current, then
VSEC will droop. An external resistive divider from VSEC to
the FCB pin sets a minimum voltage VSEC(MIN):
R6 VSEC(MIN) 0.8V 1+ R5 where R5 and R6 are shown in Figure 2.
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
If VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
capacitance can be reduced for a particular application. A
complete explanation is included in Design Solutions 10.
(See www.linear.com)
In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18μA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
Efficiency Considerations
The following table summarizes the possible states available on the FCB pin:
Table 1
FCB PIN
CONDITION
0V to 0.75V
Forced Continuous Both Controllers
(Current Reversal Allowed—
Burst Inhibited)
0.85V < VFCB < 4.3V
Minimum Peak Current Induces
Burst Mode Operation
No Current Reversal Allowed
Feedback Resistors
Regulating a Secondary Winding
>4.8V
Burst Mode Operation Disabled
Constant Frequency Mode Enabled
No Current Reversal Allowed
No Minimum Peak Current
Voltage Positioning
Voltage positioning can be used to minimize peak-to-peak
output voltage excursions under worst-case transient
loading conditions. The open-loop DC gain of the control
loop is reduced depending upon the maximum load step
specifications. Voltage positioning can easily be added
to either or both controllers by loading the ITH pin with
a resistive divider having a Thevenin equivalent voltage
source equal to the midpoint operating voltage range of
the error amplifier, or 1.2V (see Figure 8).
The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier.
The maximum output voltage deviation can theoretically
be reduced to half or alternatively the amount of output
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentINTVCC
RT2
ITH
RT1
RC
LTC3728L-1
CC
3728L1 F08
Figure 8. Active Voltage Positioning
Applied to the LTC3728L-1
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3728L-1 circuits: 1) IC VIN current (including
loading on the 3.3V internal regulator), 2) INTVCC regulator current, 3) I2R losses, 4) Topside MOSFET transition
losses.
1. The VIN current has two components: the first is the DC
supply current given in the Electrical Characteristics table,
which excludes MOSFET driver and control currents; the
second is the current drawn from the 3.3V linear regulator
output. VIN current typically results in a small (<0.1%)
loss.
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high
to low again, a packet of charge dQ moves from INTVCC
to ground. The resulting dQ/dt is a current out of INTVCC
that is typically much larger than the control circuit current. In continuous mode, IGATECHG =f(QT+QB), where QT
and QB are the gate charges of the topside and bottom
side MOSFETs.
Supplying INTVCC power through the EXTVCC switch input
from an output-derived source will scale the VIN current
required for the driver and control circuits by a factor of
(Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately
2.5mA of VIN current. This reduces the mid-current loss
from 10% or more (if the driver was powered directly from
VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of
the fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance of
one MOSFET can simply be summed with the resistances
of L, RSENSE and ESR to obtain I2R losses. For example, if
each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and
RESR = 40mΩ (sum of both input and output capacitance
losses), then the total resistance is 130mΩ. This results
in losses ranging from 3% to 13% as the output current
increases from 1A to 5A for a 5V output, or a 4% to 20%
loss for a 3.3V output. Efficiency varies as the inverse
square of VOUT for the same external components and
output power level. The combined effects of increasingly
lower output voltages and higher currents required by
high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high input
voltages (typically 15V or greater). Transition losses can
be estimated from:
2 I
Transition Loss = ( VIN ) • MAX (RDR ) •
2 (CMILLER )( f ) 5V –1V
TH
+
1 VTH Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses during
the design phase. The internal battery and fuse resistance
losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching
frequency. A 25W supply will typically require a minimum
of 20μF to 40μF of capacitance having a maximum of 20mΩ
to 50mΩ of ESR. The LTC3728L-1 2-phase architecture
typically halves this input capacitance requirement over
competing solutions. Other losses including Schottky conduction losses during dead-time and inductor core losses
generally account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by
an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to
charge or discharge COUT generating the feedback error
signal that forces the regulator to adapt to the current
change and return VOUT to its steady-state value. During
this recovery time VOUT can be monitored for excessive
overshoot or ringing, which would indicate a stability
problem. OPTI-LOOP compensation allows the transient
response to be optimized over a wide range of output
capacitance and ESR values. The availability of the ITH pin
not only allows optimization of control loop behavior but
also provides a DC coupled and AC filtered closed loop
response test point. The DC step, rise time and settling
3728l1fc
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LTC3728L-1
APPLICATIONS INFORMATION
at this test point truly reflects the closed loop response.
Assuming a predominantly second order system, phase
margin and/or damping factor can be estimated using the
percentage of overshoot seen at this pin. The bandwidth
can also be estimated by examining the rise time at the
pin. The ITH external components shown in the Figure 1
circuit will provide an adequate starting point for most
applications.
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is in
the feedback loop and is the filtered and compensated
control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main power line in an
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery, and
double-battery.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
Automotive Considerations: Plugging into the
Cigarette Lighter
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 9 is the most straightforward
approach to protect a DC/DC converter from the ravages
of an automotive power line. The series diode prevents
current from flowing during reverse-battery, while the
transient suppressor clamps the input voltage during
load-dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamp the input voltage below breakdown of the converter.
Although the LTC3728L-1 has a maximum input voltage
of 30V, most applications will also be limited to 30V by
the MOSFET BVDSS.
50A IPK RATING
12V
VIN
LTC3728L-1
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
3728L1 F09
Figure 9. Automotive Application Protection
3728l1fc
24
LTC3728L-1
APPLICATIONS INFORMATION
Design Example
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A,
and f = 300kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the PLLFLTR
pin to a resistive divider from the INTVCC pin, generating
0.7V for 300kHz operation. The minimum inductance for
30% ripple current is:
V –V
V
L IN OUT • OUT
(f)(IRIPPLE ) VIN
or 3.7μH. Using standard inductor values:
IL =
VOUT VOUT 1–
(f)(L) VIN A 4.7μH inductor will produce 23% ripple current and a
3.3μH will result in 33%. The peak inductor current will be
the maximum DC value plus one half the ripple current, or
5.84A, for the 3.3μH value. Increasing the ripple current
will also help ensure that the minimum on-time of 100ns
is not violated. The minimum on-time occurs at maximum
VIN:
tON(MIN) =
VOUT
VIN(MAX)f
=
1.8V
= 273ns
22V(300kHz)
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances:
RSENSE 60mV
0.01
5.84A
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the SENSE pin’s specified
input current.
0.8V
R1(MAX) = 24k 2.4V – VOUT 0.8V = 24k = 32k
2.4V – 1.8V Choosing 1% resistors: R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
The power dissipation on the top side MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At
maximum input voltage with T(estimated) = 50°C:
1.8V 2
PMAIN =
(5) [1+ (0.005)(50°C – 25°C)] •
22V
(0.035) + (22V )2 5A
( 4)(215pF ) •
2
1
1
5 – 2.3 + 2.3 ( 300kHz ) = 332mW
A short-circuit to ground will result in a folded back
current of:
ISC =
25mV 1 120ns(22V) –
= 2.1A
0.01 2 3.3μH with a typical value of RDS(ON) and δ = (0.005/°C)(20) = 0.1.
The resulting power dissipated in the bottom MOSFET is:
22V – 1.8V
2
2.1A ) (1.125) ( 0.022 )
(
22V
= 100mW
PSYNC =
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
3728l1fc
25
LTC3728L-1
APPLICATIONS INFORMATION
maximum input voltage. The output voltage ripple due to
ESR is approximately:
at CIN? Do not attempt to split the input decoupling for
the two channels as it can cause a large resonant loop.
VORIPPLE = RESR (ΔIL) = 0.02Ω(1.67A) = 33mVP–P
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 10. The Figure 11 illustrates the
current waveforms present in the various branches of the
2-phase synchronous regulators operating in the continuous mode. Check the following in your layout:
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return of
CINTVCC must return to the combined COUT (–) terminals.
The path formed by the top N-channel MOSFET, Schottky
diode and the CIN capacitor should have short leads and
PC trace lengths. The output capacitor (–) terminals should
be connected as close as possible to the (–) terminals
of the input capacitor by placing the capacitors next to
each other and away from the Schottky loop described
above.
1. Are the top N-channel MOSFETs M1 and M3 located
within 1cm of each other with a common drain connection
3. Do the LTC3728L-1 VOSENSE pins’ resistive dividers connect to the (+) terminals of COUT? The resistive divider must
PC Board Layout Checklist
RPU
RUN/SS1
PGOOD
SENSE1+
TG1
SENSE1–
SW1
VOSENSE1
BOOST1
PLLFLTR
VIN
VPULL-UP
(<7V)
PGOOD
L1
VOUT1
CB1
R1
INTVCC
RIN
1μF
CERAMIC
D1
COUT1
FCB
3.3VOUT
ITH2
PGND
CVIN
CINTVCC
VIN
CIN
COUT2
1μF
CERAMIC
BG2
M3
BOOST2
GND
+
EXTVCC
LTC3728L-1
ITH1
INTVCC
SGND
3.3V
BG1
M2
+
PLLIN
+
fIN
M1
+
R2
RSENSE
M4
D2
CB2
R3
R4
VOSENSE2
SW2
SENSE2–
TG2
RSENSE
VOUT2
L2
SENSE2+
RUN/SS2
3728L1 F10
Figure 10. LTC3728L-1 Recommended Printed Circuit Layout Diagram
3728l1fc
26
LTC3728L-1
APPLICATIONS INFORMATION
SW1
L1
D1
RSENSE1
COUT1
VOUT1
+
RL1
CERAMIC
VIN
RIN
CIN
+
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
D2
L2
RSENSE2
COUT2
VOUT2
+
RL2
CERAMIC
3728L1 F11
Figure 11. Branch Current Waveforms
be connected between the (+) terminal of COUT and signal
ground. The R2 and R4 connections should not be along
the high current input feeds from the input capacitor(s).
4. Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible to
the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor.
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
This capacitor carries the MOSFET drivers current peaks.
An additional 1μF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve noise
performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from the
opposites channel’s voltage and current sensing feedback
pins. All of these nodes have very large and fast moving
signals and therefore should be kept on the “output side”
of the LTC3728L-1 and occupy minimum PC trace area.
3728l1fc
27
LTC3728L-1
APPLICATIONS INFORMATION
7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the same
side of the PC board as the input and output capacitors with
tie-ins for the bottom of the INTVCC decoupling capacitor,
the bottom of the voltage feedback resistive divider and
the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output voltage
as well. Check for proper performance over the operating
voltage and current range expected in the application. The
frequency of operation should be maintained over the input
voltage range down to dropout and until the output load
drops below the low current operation threshold—typically
10% to 20% of the maximum designed current level in
Burst Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
3728l1fc
28
LTC3728L-1
TYPICAL APPLICATIONS
59k
1M
100k
0.1μF
180pF
TG1
SENSE1–
SW1
0.015Ω
VOSENSE1
BOOST1
PLLFLTR
VIN
PLLIN
D1
LTC3728L-1
INTVCC
SGND
PGND
3.3VOUT
ITH2
1000pF
180pF
1
CMDSH-3TR
22μF
50V
150μF, 6.3V
PANASONIC SP
VOUT3
12V
120mA
+
100k
0.1μF
1μF
10V
1μF
25V
GND
4.7μF
180μF, 4V
PANASONIC SP
VIN
7V TO
28V
M2
BG2
Q3
Q4
D2
BOOST2
0.1μF
20k
1%
63.4k
1%
ON/OFF
2
220k
CMDSH-3TR
3.3V
3
+
ITH1
LT1121
+
EXTVCC
5
Q2
BG1
FCB
VOUT1
5V
3A; 4A PEAK
8
Q1
10Ω
1000pF
+
M1
0.1μF
33pF
15k
SENSE1+
1000pF
105k, 1%
33pF
15k
PGOOD
33μF
25V
T1, 1:1.8
10μH
+
20k
1%
RUN/SS1
MBRS1100T3
VPULL-UP
(<7V)
PGOOD
VOSENSE2
SW2
SENSE2–
TG2
1000pF
SENSE2+
RUN/SS2
L1
6.3μH
VOUT2
3.3V
5A; 6A PEAK
0.01Ω
0.1μF
3728L1 F12
VIN: 7V TO 28V
VOUT: 5V, 3A/3.3V, 6A/12V, 150mA
SWITCHING FREQUENCY = 250kHz
MI, M2: FDS6982S OR VISHAY Si4810DY
L1: SUMIDA CEP123-6R3MC
T1: 10μH 1:1.8 — DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
Figure 12. LTC3728L-1 High Efficiency Low Noise 5V/3A, 3.3V/5A, 12V/120mA Regulator
3728l1fc
29
LTC3728L-1
TYPICAL APPLICATIONS
VPULL-UP
(<7V)
0.1μF
180pF
20k
1%
10k
RUN/SS1
PGOOD
SENSE1+
TG1
SENSE1–
SW1
VOSENSE1
BOOST1
PLLFLTR
VIN
0.1μF
0.01μF
1000pF
PLLIN
1μF 50V
BG1
+
PGND
180μF, 4V
CMDSH-3TR
3.3V
3.3VOUT
ITH2
PIN 4
BOOST2
Q3
Q4
0.1μF
VOSENSE2
SW2
SENSE2–
TG2
1000pF
SENSE2+
RUN/SS2
M2
L2
4.3μH
VOUT2
3.3V/5A
0.008Ω
0.1μF
VIN: 7V TO 28V
VOUT: 5V, 4A/3.3V, 5A
VIN
7V TO
28V
BG2
1000pF
20k
1%
180pF
1μF 50V
4.7μF, 10V
GND
+
SGND
150μF, 6.3V
0.1μF
1μF
1500pF
22μF
50V
CMDSH-3TR
EXTVCC
LTC3728L-1
ITH1
INTVCC
100pF
Q2
M1
10Ω
63.4k
1%
VOUT1
5V/4A
+
fSYNC
Q1
PIN 4
FCB
4.75k
0.008Ω
1000pF
105k
1%
100pF
8.06k
L1
4.3μH
PGOOD
3728L1 F13
SWITCHING FREQUENCY = 250kHz TO 550kHz
M1, M2: FDS6982S OR VISHAY Si4810DY
L1, L2: SUMIDA CDEP105-4R3MC-88
OUTPUT CAPACITORS: PANASONIC SP SERIES
Figure 13. LTC3728L-1 5V/4A, 3.3V/5A Regulator with External Frequency Synchronization
3728l1fc
30
LTC3728L-1
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 ±.005
.033
(0.838)
REF
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 ± .004
× 45°
(0.38 ± 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
9 10 11 12 13 14
.0532 – .0688
(1.35 – 1.75)
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.0250
(0.635)
BSC
.008 – .012
(0.203 – 0.305)
TYP
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693 Rev D)
5.00 ± 0.10
(4 SIDES)
0.70 ±0.05
5.50 ±0.05
4.10 ±0.05
R = 0.05
TYP
0.75 ± 0.05
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.30 TYP
R = 0.115
OR 0.35 × 45° CHAMFER
TYP
31 32
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.50 REF
(4 SIDES)
3.45 ± 0.05
3.50 REF
(4-SIDES)
3.45 ± 0.05
3.45 ± 0.10
3.45 ± 0.10
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
0.200 REF
NOTE:
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
0.25 ± 0.05
0.50 BSC
(UH32) QFN 0406 REV D
3728l1fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3728L-1
TYPICAL APPLICATION
IIN
12VIN
CIN
I1
PHASMD
OPEN
TG1
U1 TG2
LTC3729
90°
0°
IIN*
BUCK: 2.5V/15A
180°
BUCK: 2.5V/15A
2.5VO/30A
I2
CLKOUT
I2
I3
I3
TG1
TG2
90°
BUCK: 1.5V/15A
270°
U2
LTC3728L-1
90°
BUCK: 1.8V/15A
1.5VO/15A
1.8VO/15A
I4
*INPUT RIPPLE CURRENT CANCELLATION
INCREASES THE RIPPLE FREQUENCY AND
REDUCES THE RMS INPUT RIPPLE CURRENT
THUS, SAVING INPUT CAPACITORS
I4
PLLIN
I1
3728L1 F14
Figure 14. Multioutput PolyPhase Application
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1628/LTC1628-PG/ 2-Phase, Dual Output Synchronous Step-Down
LTC1628-SYNC
DC/DC Controller
Reduces CIN and COUT, Power Good Output Signal, Synchronizable,
3.5V ≤ VIN ≤ 36V, IOUT up to 20A, 0.8V ≤ VOUT ≤ 5V
LTC1629/
LTC1629-PG
20A to 200A PolyPhase™ Synchronous Controllers
Expandable from 2-Phase to 12-Phase, Uses All
Surface Mount Components, No Heat Sink, VIN up to 36V
LTC1708-PG
2-Phase, Dual Synchronous Controller with Mobile VID
3.5V ≤ VIN ≤ 36V, VID Sets VOUT1, PGOOD
LT1709/
LT1709-8
High Efficiency, 2-Phase Synchronous Step-Down
Switching Regulators with 5-Bit VID
1.3V ≤ VOUT ≤ 3.5V, Current Mode Ensures
Accurate Current Sharing, 3.5V ≤ VIN ≤ 36V
LTC1735
High Efficiency Synchronous Step-Down
Switching Regulator
Output Fault Protection, 16-Pin SSOP
LTC1736
High Efficiency Synchronous Controller with 5-Bit Mobile Output Fault Protection, 24-Pin SSOP,
VID Control
3.5V ≤ VIN ≤ 36V
Up to 97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN),
No RSENSE Current Mode Synchronous Step-Down
Controllers
IOUT up to 20A
LTC1778/LTC1778-1
LTC1929/
LTC1929-PG
2-Phase Synchronous Controllers
Up to 42A, Uses All Surface Mount Components,
No Heat Sinks, 3.5V ≤ VIN ≤ 36V
LTC3708
Dual, 2-Phase, DC/DC Controller with Output Tracking
Current Mode, No RSENSE, Up/Down Tracking, Synchronizable
LTC3711
No RSENSE Current Mode Synchronous Step-Down
Controller with Digital 5-Bit Interface
Up to 97% Efficiency, Ideal for Pentium® III Processors,
0.925V ≤ VOUT ≤ 2V, 4V ≤ VIN ≤ 36V, IOUT up to 20A
LTC3728
Dual, 550kHz, 2-Phase Synchronous Step-Down
Controller
Dual 180° Phased Controllers, VIN 3.5V to 35V, 99% Duty Cycle,
5x5QFN, SSOP-28
LTC3729
20A to 200A, 550kHz PolyPhase Synchronous Controller
Expandable from 2-Phase to 12-Phase, Uses all Surface Mount
Components, VIN up to 36V
LTC3731
3- to 12-Phase Step-Down Synchronous Controller
60A to 240A Output Current, 0.6V ≤ VOUT ≤ 6V, 4.5V ≤ VIN ≤ 32V
LTC3802
No RSENSE 2-Phase Dual Synchronous Step-Down
Controller
330kHz to 750kHz, No Sense Resistor, Output Voltage Tracking,
Synchronizable for 4-Phase Operation
No RSENSE and PolyPhase are trademarks of Linear Technology Corporation.
3728l1fc
32 Linear Technology Corporation
LT 0708 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005
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