Burr-Brown OPA2652E3K Dual, 700mhz, voltage-feedback operational amplifier Datasheet

 OP
A2
OPA2652
652
SBOS125A – JUNE 2000 – REVISED MAY 2006
TM
Dual, 700MHz, Voltage-Feedback
OPERATIONAL AMPLIFIER
FEATURES
DESCRIPTION
•
•
•
•
•
•
•
•
The OPA2652 is a dual, low-cost, wideband voltage
feedback amplifier intended for price-sensitive
applications. It features a high gain bandwidth
product of 200MHz on only 5.5mA/channel quiescent
current. Intended for operation on ±5V supplies, it
also supports applications on a single supply from
+6V to +12V with 140mA output current. Its classical
differential input, voltage-feedback design allows
wide application in active filters, integrators,
transimpedance amplifiers, and differential receivers.
WIDEBAND BUFFER: 700MHz, G = +1
WIDEBAND LINE DRIVER: 200MHz, G = +2
HIGH OUTPUT CURRENT: 140mA
LOW SUPPLY CURRENT: 5.5mA/Ch
ULTRA-SMALL PACKAGE: SOT23-8
LOW dG/dφ: 0.05%/0.03°
HIGH SLEW RATE: 335V/µsec
SUPPLY VOLTAGE: ±3V to ±6V
The OPA2652 is internally compensated for unity
gain stability. It has exceptional bandwidth (700MHz)
as a unity gain buffer, with little peaking (0dB typ).
Excellent DC accuracy is achieved with a low 1.5mV
input offset voltage and 300nA input offset current.
APPLICATIONS
•
•
•
•
•
A/D DRIVERS
CONSUMER VIDEO
ACTIVE FILTERS
PULSE DELAY CIRCUITS
LOW COST UPGRADE TO THE AD8056
OR EL2210
200W
402W
RELATED PRODUCTS
24.9W
SINGLES
DUALS
TRIPLES
QUADS
OPA650
OPA680
OPA2650
—
OPA4650
±5V Spec
OPA2680
OPA3680
—
+5V Capable
OPA631
OPA2631
—
—
+3V Capable
OPA634
OPA2634
—
—
+3V Capable
0.1mF
NOTES
+5V
22pF
+5V
1.00kW
1/2
OPA2652
+In
0.1mF
VIN
ADS807
CM
12-Bit
133W
-In
53MHz
1.00kW
200W
402W
24.9W
0.1mF
+
22pF
1/2
OPA2652
133W
-5V
Differential ADC Driver
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2000–2006, Texas Instruments Incorporated
OPA2652
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SBOS125A – JUNE 2000 – REVISED MAY 2006
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
PACKAGE/ORDERING INFORMATION (1)
(1)
PRODUCT
PACKAGELEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
OPA2652U
SO-8
D
–40°C to +85°C
OPA2652U
OPA2652E
SOT23-8
DCN
–40°C to +85°C
C52
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
OPA2652U
Rails
OPA2652U/2K5
Tape and Reel, 2500
OPA2652E/250
Tape and Reel, 250
OPA2652E/3K
Tape and Reel, 3000
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
Supply voltage
Internal power dissipation
OPA2652
UNIT
±6.5
V
See Thermal Characteristics
Differential input voltage
±1.2
V
Input voltage range
±VS
V
Storage temperature range
–40 to +125
°C
Lead temperature (SO-8)
+260
°C
Junction temperature, TJ
+175
°C
Human body model
2000
V
Machine model
200
V
ESD rating:
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
PIN CONFIGURATION
Top View
SO-8
SOT23-8
SOT23-8 Marking / Pin Orientation
OPA2652
Out A
1
8
+VS
-In A
2
7
Out B
+In A
3
6
-In B
-VS
4
5
+In B
C52
Pin 1
2
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SBOS125A – JUNE 2000 – REVISED MAY 2006
ELECTRICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29 for AC
performance only.
OPA2652U, E
MIN/MAX OVER
TEMPERATURE
TYP
PARAMETER
AC PERFORMANCE
Small-Signal Bandwidth
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
(Figure 28 and Figure 29)
G = +1, RF = 25Ω, VO = 200mVPP
700
MHz
typ
C
G = +2,VO = 200mVPP
200
MHz
typ
C
G = +5,VO = 200mVPP
45
MHz
typ
C
G ≥ +10
200
MHz
typ
C
VO = 200mVPP
50
MHz
typ
C
G = +1, RF = 25Ω, VO = 200mVPP
0
dB
typ
C
4V step
335
V/µs
typ
C
200mV step
2.0
ns
typ
C
4V step
10
ns
typ
C
VO = 4VPP
50
MHz
typ
C
VO = 2VPP, 5MHz
66
dB
typ
C
Input Voltage Noise
f > 1MHz
8
nV/√Hz
typ
C
Input Current Noise
f > 1MHz
1.4
pA/√Hz
typ
C
Differential Gain Error
NTSC, RL = 150Ω
0.05
%
typ
C
Differential Phase Error
NTSC, RL = 150Ω
0.03
degrees
typ
C
Channel-to-Channel Crosstalk
f = 5MHz
–100
dBc
typ
C
DC PERFORMANCE (4)
VCM = 0V
Gain Bandwidth Product
Bandwidth for 0.1dB Flatness
Peaking at a Gain of +1
Slew Rate
Rise-and-Fall Time
Large-Signal Bandwidth
SFDR
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
63
56
±1.5
±7
Average Offset Drift
Input Bias Current
4
15
55
54
5
7
20
25
Input Bias Current Drift
±0.3
Input Offset Current
±1.0
±1.4
±2.0
Input Offset Current Drift
dB
min
A
mV
max
A
µV/°C
max
B
µA
max
A
µA/°C
max
B
µA
max
A
µA/°C
max
B
INPUT (4)
Common-Mode Input Range
Common-Mode Rejection Ratio
Input Impedance
(1)
(2)
(3)
(4)
±4.0
±3.0
95
75
±2.8
±2.7
V
min
A
dB
min
A
VCM = 0V
Differential
35 || 1
kΩ || pF
typ
C
Common-Mode
18 || 1
MΩ || pF
typ
C
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C tested specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over
temperature specifications.
Current is considered positive-out-of node. VCM is the input common-mode voltage.
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ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29 for AC
performance only.
OPA2652U, E
MIN/MAX OVER
TEMPERATURE
TYP
PARAMETER
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
OUTPUT
1kΩ load
±3.0
±2.4
V
min
A
100Ω load
±2.5
±2.2
V
min
A
Output Current, Sourcing
VO = 0V
140
100
85
75
mA
min
A
Output Current, Sinking
VO = 0V
140
100
85
75
mA
min
A
f < 100kHz
0.06
Ω
typ
C
Voltage Output Swing
Closed-Loop Output Impedance
POWER SUPPLY
±5
Specified Operating Voltage
Maximum Operating Voltage
V
typ
C
±6
±6
±6
V
max
A
Maximum Quiescent Current
Total both channels
11
13.2
14
15.5
mA
max
A
Minimum Quiescent Current
Total both channels
11
8.8
8
7.5
mA
min
A
Input-referred
58
54
dB
min
A
U, E Packages
–40 to
+85
°C
typ
C
Power-Supply Rejection Ratio
(–PSRR)
THERMAL CHARACTERISTICS
Specified Operating Temperature Range
Thermal Resistance, θJA
4
Junction-to-Ambient
U
SO-8
125
°C/W
typ
C
E
SOT23-8
150
°C/W
typ
C
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TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29.
NONINVERTING
SMALL-SIGNAL FREQUENCY RESPONSE
INVERTING
SMALL-SIGNAL FREQUENCY RESPONSE
6
6
0
-3
G = +2
-6
-9
G = +5
-12
-15
G = +10
-18
G = -1
0
-3
G = -2
-6
-9
-12
G = -5
-15
-18
-21
G = -10
-21
-24
-24
1M
10M
100M
1G
1M
10M
100M
Frequency (Hz)
Figure 1.
Figure 2.
NONINVERTING
LARGE-SIGNAL FREQUENCY RESPONSE
INVERTING
LARGE-SIGNAL FREQUENCY RESPONSE
6
G = +2
3
G = -1
3
Normalized Gain (dB)
VO < 1VPP
0
-3
-6
-9
VO = 2VPP
-12
-15
VO = 4VPP
VO = 0.5VPP
0
-3
-6
VO = 1.0VPP
-9
-12
-15
-18
-18
-21
-21
-24
VO = 2.0VPP
-24
10M
100M
1G
1M
10M
100M
1G
Frequency (Hz)
Frequency (Hz)
Figure 3.
Figure 4.
NONINVERTING PULSE RESPONSE
INVERTING PULSE RESPONSE
G = -1
200mVPP
Output Voltage (800mV/div)
4VPP
Output Voltage (50mV/div)
G = +2
Time (5ns/div)
4VPP
200mVPP
Output Voltage (50mV/div)
1M
Output Voltage (800mV/div)
1G
Frequency (Hz)
6
Normalized Gain (dB)
VO = 0.2VPP
3
Normalized Gain (dB)
Normalized Gain (dB)
G = +1
RF = 25W
VO = 0.2VPP
3
Time (5ns/div)
Figure 5.
Figure 6.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29.
HARMONIC DISTORTION vs NONINVERTING GAIN
VO = 2VPP
f = 5MHz
HARMONIC DISTORTION vs INVERTING GAIN
-50
VO = 2VPP
f = 5MHz
3rd Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
-50
-60
2nd Harmonic
-70
-80
-90
-60
2nd Harmonic
-70
-80
-90
1
10
1
Gain Magnitude (V/V)
Figure 7.
Figure 8.
f = 5MHz
VO = 2VPP
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
HARMONIC DISTORTION vs FREQUENCY
-50
-60
3rd Harmonic
-70
2nd Harmonic
-80
-60
3rd Harmonic
-70
-80
2nd Harmonic
-90
-90
0.1
1
4
0.1
1
10
Output Voltage (VPP)
Frequency (MHz)
Figure 9.
Figure 10.
HARMONIC DISTORTION vs LOAD RESISTANCE
-50
Harmonic Distortion (dBc)
VO = 2VPP
f = 5MHz
-60
3rd Harmonic
-70
2nd Harmonic
-80
-90
20
HARMONIC DISTORTION vs SUPPLY VOLTAGE
-50
Harmonic Distortion (dBc)
10
Gain Magnitude (V/V)
HARMONIC DISTORTION vs OUTPUT VOLTAGE
-50
VO = 2VPP
f = 5MHz
-60
3rd Harmonic
-70
2nd Harmonic
-80
-90
100
6
3rd Harmonic
1000
±3
±5
±4
RL (W)
Supply Voltage (V)
Figure 11.
Figure 12.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29.
TWO-TONE, 3rd-ORDER SPURIOUS LEVEL
COMPOSITE VIDEO dG/dφ
-50
0.30
3rd-Order Spurious Level (dBc)
df, Positive Video
0.25
-60
dG/df (%/°)
20MHz
10MHz
-70
5MHz
1MHz
-8
-6
-4
dG, Positive Video
0.05
Load Power at matched 50W load
-90
df, Negative Video
0.15
0.10
2MHz
-80
0.20
0
-2
2
dG, Negative Video
0.00
4
1
2
Single-Tone Load Power (dBm)
3
Figure 13.
Figure 14.
INPUT VOLTAGE AND CURRENT NOISE DENSITY
CHANNEL-TO-CHANNEL CROSSTALK
-30
Crosstalk, Input-Referred (dB)
Voltage Noise (nV/ÖHz)
Current Noise (pA/ÖHz)
100
Voltage Noise = 8.0nV/ÖHz
10
Current Noise = 1.4pA/ÖHz
1
-40
-50
-60
-70
-80
-90
100
1k
10k
100k
1M
10M
10
100
Frequency (Hz)
Figure 16.
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
70
60
50
40
30
20
10
0
1
10
1000
Frequency (MHz)
Figure 15.
RS (W)
4
Number of 150W Loads
100
1000
2
G = +2
1
CL = 10pF
0
CL = 22pF
-1
CL = 100pF
-2
-3
-4
1/2
OPA2652
-5
RS
VO
CL = 47pF
CL
-6
1kW
-7
-8
0
10M
100M
1G
Frequency (Hz)
Capacitive Load (pF)
Figure 17.
Figure 18.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29.
CMRR AND PSRR vs FREQUENCY
OPEN-LOOP GAIN AND PHASE
70
60
CMRR
80
+PSRR
70
0
60
-PSRR
50
40
30
20
-30
Open-Loop Phase
50
40
-90
30
-120
20
-150
Open-Loop Gain
10
-210
-10
1k
10k
100k
1M
10M
-240
10k
100M
100k
1M
10M
Figure 19.
CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
5
200W
10
1W Internal
Power Limit
4
1/2
OPA2652
Output Current Limited
3
ZO
2
VO (V)
402W
402W
0.1
1
100W
Load Line
0
-1
50W Load Line
-2
20W Load Line
-3
10W Load Line
Output Current Limit
1W Internal
Power Limit
-4
0.01
10k
100k
1M
10M
-5
-200
100M 400M
-50
0
50
Figure 21.
Figure 22.
G = +2
5
2.0
4
1.5
VOUT
2
1.0
1
0.5
0
0
Input and Output Voltage (V)
VIN
100
150
INVERTING OVERDRIVE RECOVERY
2.5
G = -1
VIN
3
2
1
0
-1
-1
-0.5
-2
-1.0
-3
-1.5
-4
-2.0
-4
-5
-2.5
-5
-2
VOUT
-3
Time (20ns/div)
Time (20ns/div)
Figure 23.
8
-100
IO (mA)
Input Voltage (V)
Output Voltage (V)
3
-150
Frequency (Hz)
NONINVERTING OVERDRIVE RECOVERY
4
1G
Figure 20.
100
1
100M
Frequency (Hz)
Frequency (Hz)
Output Impedance (W)
-180
0
10
0
5
-60
Figure 24.
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200
Open-Loop Phase (°)
90
Open-Loop Gain (dB)
Power Supply Rejection Ratio (dB)
Common-Mode Rejection Ratio (dB)
100
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted. See Figure 28 and Figure 29.
TYPICAL DC DRIFT OVER TEMPERATURE
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
250
25
5
Sourcing Output Current
200
Output Current (mA)
3
2
1
IOS
0
-1
VOS
-2
-3
Sinking Output Current
150
15
100
10
Quiescent Supply Current
(Both Channels)
50
IB
-4
20
Supply Current (mA)
4
5
-5
-6
-40
-20
0
20
40
60
80
100
0
-40
0
-20
Ambient Temperature (°C)
0
20
40
60
80
100
Ambient Temperature (°C)
Figure 25.
Figure 26.
COMMON-MODE INPUT VOLTAGE RANGE
AND OUTPUT SWING vs SUPPLY VOLTAGE
6
Positive Common-Mode Input Range
5
Negative Common-Mode Input Range
Voltage Range (V)
Input Offset Voltage (mV)
Input Bias and Offset Current (A)
6
4
3
2
Negative Output Voltage Range
1
Positive Output Voltage Range
0
±3
±5
±4
±6
Supply Voltage (V)
Figure 27.
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APPLICATIONS INFORMATION
Wideband Voltage Feedback Operation
The OPA2652 is a dual, low-power, wideband
voltage feedback operational amplifier. Each channel
is internally compensated to provide unity gain
stability. The OPA2652 voltage feedback architecture
features true differential and fully symmetrical inputs.
This architecture minimizes offset errors, making the
OPA2652 well-suited for implementing filter and
instrumentation designs. As a dual operational
amplifier, OPA2652 is an ideal choice for designs
that require multiple channels where reduction of
board space, power dissipation and cost are critical.
Its AC performance is optimized to provide a gain
bandwidth product of 200MHz and a fast rise time of
2.0ns, which is an important consideration in
high-speed data conversion applications. The low
DC input offset of ±1.5mV and drift of ±5µV/°C
support high accuracy requirements. In applications
requiring a higher slew rate and wider bandwidth,
such as video and high bit rate digital
communications, consider the dual current feedback
OPA2694, or the OPA2691.
Figure 28 shows the DC-coupled, gain of +2, dual
power-supply circuit configuration used as the basis
of the ±5V specifications and typical characteristics.
This configuration is for one channel. The other
channel is connected similarly. For test purposes, the
input impedance is set to 50Ω with a resistor to
ground and the output impedance is set to 50Ω with
a series output resistor.
Voltage swings reported in the specifications are
taken directly at the input and output pins, while
output powers (dBm) are at the matched 50Ω load.
For the circuit of Figure 28, the total effective load
will be 100Ω || 804Ω. Two optional components are
included in Figure 28.
An additional resistor (174Ω) is included in series
with the noninverting input. Combined with the 25Ω
DC source resistance looking back towards the
signal generator, this additional resistor gives an
input bias current cancelling resistance that matches
the 201Ω source resistance seen at the inverting
input (see the DC Accuracy and Offset Control
section). In addition to the usual power-supply
decoupling capacitors to ground, a 0.1µF capacitor is
included between the two power-supply pins. In
practical printed circuit board (PCB) layouts, this
optional-added capacitor typically improves the
2nd-harmonic distortion performance by 3dB to 6dB.
Figure 29 shows the DC-coupled, gain of –1, bipolar
supply circuit configuration that is the basis of the
specifications and typical characteristics at G = –1.
The input impedance matching resistor (57.6Ω) used
for testing gives a 50Ω input load. A resistor (205Ω)
connects the noninverting input to ground. This
configuration provides the DC source resistance
matching to cancel outputs errors arising from input
bias current.
+5V
+5V
+
0.1mF
0.1mF
6.8mF
+
6.8mF
0.1mF
RO
VO 49.9W
50W Source
174W
VI
49.9W
VO
1/2
OPA2652
49.9W
RB
205W
50W Load
50W
0.1mF
RF
402W
Source
1/2
OPA2652
50W Load
RG
402W
VO
= -1
VI
RF
402W
VI
RM
57.6W
RG
402W
+
6.8mF
0.1mF
+
6.8mF
0.1mF
-5V
-5V
Figure 28. DC-Coupled, G = +2, Bipolar Supply,
Specification and Test Circuit
10
Figure 29. DC-Coupled, G = –1, Bipolar Supply,
Specification and Test Circuit
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Differential ADC Driver
0
The circuit on the front page shows an OPA2652
driving the ADS807 analog-to-digital converter (ADC)
differentially, at a gain of +2V/V. The outputs are
AC-coupled to the converter to adjust for the
difference in supply voltages. The 133Ω resistors at
the noninverting inputs minimize DC offset errors.
The differential topology minimizes even-order
distortion products, such as second-harmonic
distortion.
-5
Gain (dB)
-10
-15
-20
-25
-30
-35
Bandpass Filter
-40
10k
Figure 31 shows a single OPA2652 implementing a
sixth-order bandpass filter. This filter cascades two
second-order Sallen-Key sections with transmission
zeros, and a double real pole section. It has 0.3dB of
ripple, –3dB frequencies of 450kHz and 11MHz, and
–23dB frequencies of 315kHz and 16MHz. The
20.0Ω resistor isolates the first OPA2652 output from
capacitive loading. This configuration improves
stability with minimal impact on the filter response.
Figure 30 shows the nominal response simulated by
SPICE.
100k
1M
10M
100M
Frequency (Hz)
Figure 30. Nominal Filter Response
2.2nF
1% Resistors
5% Capacitors
+5V
140W
2.10kW
VIN
1.30kW
1.0nF
1/2
OPA2652
1.0nF
-5V 24.9W
143W
180pF
+5V
200W
2.7nF
VOUT
100pF
1/2
OPA2652
18pF
20.0W
225W
158W
12pF
150pF
100W
-5V
24.9W
107W
Figure 31. Bandpass Filter
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Video Line Driver
C2
Figure 32 shows the OPA2652 used as a video line
driver. Its outstanding differential gain and phase
allow it to be used in studio equipment, while its low
cost and SOT23-8 package option also support
consumer applications.
C1
402W
402W
VIN
VOUT
+5V
+5V
1/2
OPA2652
Video
Input
75.0W
Video
Output
1/2
OPA2652
75.0W
402W
-5V
-5V
402W
402W
Figure 34. Inverting Bandpass Filter
DESIGN-IN TOOLS
Figure 32. Video Line Driver
Demonstration Fixtures
Pulse Delay Circuit
Figure 33 shows the OPA2652 used in a pulse delay
circuit. This circuit cascades the two op amps in the
OPA2652, each forming a single pole, active allpass
filter. The overall gain is +1, and the overall delay
through the filter is:
Table 1. Demonstration Fixtures for the OPA2652
tGD = n(2RC), overall group delay
n = 2, the number of cascaded stages
+5V
C
Two printed circuit boards (PCBs) are available to
assist in the initial evaluation of circuit performance
using the OPA2652 in its two package options. Both
of these are offered free of charge as unpopulated
PCBs, delivered with a user's guide. The summary
information for these fixtures is shown in Table 1.
PRODUCT
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
OPA2652U
SO-8
DEM-OPA-SO-2A
SBOU003
OPA2652E
SOT23-8
DEM-OPA-SOT-2A
SBOU001
+5V
C
VIN
R
RG
402W
1/2
OPA2652
R
-5V
RF
402W
1/2
OPA2652
VO
-5V
402W
402W
Figure 33. Pulse Delay Circuit
RF and RG need to be equal to maintain a constant
gain magnitude. The rise and fall times of the input
pulses, tr(IN), should be slow enough to prevent
pre-shoot artifacts in the response.
tr(IN) ≥ 5RC, minimal pre-shoot
Simple Bandpass Filter
Figure 34 shows the OPA2652 used as simple
bandpass filter. The OPA2652 is well-suited for this
type of circuit because it is very stable at a noise
gain of +1.
12
The demonstration fixtures can be requested at the
Texas Instruments web site (www.ti.com) through the
OPA2652 product folder.
Macromodels and Applications Support
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
method is particularly true for video and RF amplifier
circuits where parasitic capacitance and inductance
can have a major effect on circuit performance.
Check the Texas Instruments web site (www.ti.com)
for available SPICE products (note that not all parts
have models). These models do a good job of
predicting small-signal AC and transient performance
under a wide variety of operating conditions. They do
not do as well in predicting the harmonic distortion or
dG/dφ characteristics. These models do not attempt
to distinguish between the package types in
small-signal AC performance.
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OPERATING SUGGESTIONS
Optimizing Resistor Values
Because the OPA2652 is a unity gain stable voltage
feedback op amp, a wide range of resistor values
may be used for the feedback and gain setting
resistors. The primary limits on these values are set
by dynamic range (noise and distortion) and parasitic
capacitance considerations. For a noninverting unity
gain follower application, the feedback connection
should be made with a 25Ω resistor, not a direct
short. This configuration isolates the inverting input
capacitance from the output pin and improves the
frequency response flatness. Usually, the feedback
resistor value should be between 200Ω and 1.5kΩ.
Below 200Ω, the feedback network presents
additional output loading that can degrade the
harmonic distortion performance of the OPA2652.
Above 1.5kΩ, the typical parasitic capacitance
(approximately 0.2pF) across the feedback resistor
may cause unintentional bandlimiting in the amplifier
response.
A good rule of thumb is to target the parallel
combination of RF and RG (see Figure 28) to be less
than approximately 300Ω. The combined impedance
RF || RG interacts with the inverting input
capacitance, placing an additional pole in the
feedback network, and thus a zero in the forward
response. Assuming a 2pF total parasitic on the
inverting node, holding RF || RG < 300Ω keeps this
pole above 250MHz. By itself, this constraint implies
that the feedback resistor RF can increase to several
kΩ at high gains. This increase is acceptable as long
as the pole formed by RF and any parasitic
capacitance appearing in parallel is kept out of the
frequency range of interest.
Bandwidth vs Gain: Noninverting Operation
Voltage feedback op amps exhibit decreasing
closed-loop bandwidth as the signal gain is
increased. In theory, this relationship is described by
the Gain Bandwidth Product (GBP) shown in the
specifications. Ideally, dividing GBP by the
noninverting signal gain (also called the Noise Gain,
or NG) predicts the closed-loop bandwidth. In
practice, this prediction only holds true when the
phase margin approaches 90°, as it does in high
gain configurations. At low gains (increased
feedback factor), most amplifiers exhibit a wider
bandwidth and lower phase margin. The OPA2652 is
compensated to give a flat response in a
noninverting gain of 1 (see Figure 28). This
configuration results in a typical gain of +1 bandwidth
of 700MHz, far exceeding that predicted by dividing
the 200MHz GBP by NG = 1. Increasing the gain
causes the phase margin to approach 90° and the
bandwidth to more closely approach the predicted
value of (GBP/NG). At a gain of +5, the 45MHz
bandwidth shown in the Electrical Characteristics is
close to that predicted using this simple formula.
Inverting Amplifier Operation
Because the OPA2652 is a general-purpose,
wideband voltage feedback op amp, all of the
familiar op amp application circuits are available to
the designer. Inverting operation is one of the more
common
requirements
and
offers
several
performance benefits. Figure 29 shows a typical
inverting configuration.
In the inverting configuration, three key design
consideration must be noted. First, the gain resistor
(RG) becomes part of the signal channel input
impedance. If input impedance matching is desired
(which is beneficial whenever the signal is coupled
through a cable, twisted pair, long PCB trace or other
transmission line conductor), RG may be set equal to
the required termination value and RF adjusted to
give the desired gain. This approach is the simplest,
and results in optimum bandwidth and noise
performance. However, at low inverting gains, the
resulting feedback resistor value can present a
significant load to the amplifier output. For an
inverting gain of –1, setting RG to 50Ω for input
matching eliminates the need for RM but requires a
50W feedback resistor. This configuration has the
interesting advantage that the noise gain becomes
equal to 2 for a 50Ω source impedance—the same
as the noninverting circuits considered above.
However, the amplifier output now sees the 50Ω
feedback resistor in parallel with the external load. In
general, the feedback resistor should be limited to
the 200Ω to 1.5kΩ range. In this case, it is preferable
to increase both the RF and RG values as shown in
Figure 29, and then achieve the input matching
impedance with a third resistor (RM) to ground. The
total input impedance becomes the parallel
combination of RG and RM.
The second major consideration, touched on in the
previous paragraph, is that the signal source
impedance becomes part of the noise gain equation
and influences the bandwidth. For the example in
Figure 29, the RM value combines in parallel with the
external 50Ω source impedance, yielding an effective
driving impedance of 50Ω || 57.6Ω = 26.8Ω. This
impedance is added in series with RG for calculating
the noise gain (NG). The resulting NG is 1.94 for
Figure 29 (an ideal source would cause NG = 2.00).
The third important consideration in inverting
amplifier design is setting the bias current
cancellation resistor on the noninverting input (RB). If
this resistor is set equal to the total DC resistance
looking out of the inverting node, the output DC
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error, as a result of the input bias currents, is
reduced to (Input Offset Current) • RF. If the 50Ω
source impedance is DC-coupled in Figure 29, the
total resistance to ground on the inverting input will
be 429Ω. Combining this in parallel with the
feedback resistor gives 208Ω, which is close to the
RB = 205Ω used in Figure 29. To reduce the
additional high-frequency noise introduced by this
resistor, it is sometimes bypassed with a capacitor.
As long as RB <300Ω, the capacitor is not required
since its total noise contribution is much less than
that of the op amp input noise voltage.
Output Current and Voltage
The OPA2652 specifications in the spec table,
though familiar in the industry, consider voltage and
current limits separately. In many applications, it is
the voltage • current, or VI product, that is more
relevant to circuit operation. Refer to the Output
Voltage and Current Limitations plot in the Typical
Characteristics. The X and Y axes of this graph show
the zero-voltage output current limit and the zero
current output voltage limit, respectively. The four
quadrants give a more detailed view of the device
output drive capabilities, noting that the graph is
bounded by a Safe Operating Area of 1W maximum
internal power dissipation (500mW for each
channel). Superimposing resistor load lines onto the
plot shows that the OPA2652 can drive ±2.2V into
50Ω or ±2.5V into 100Ω without exceeding the output
capabilities, or the 1W dissipation boundary line.
To maintain maximum output stage linearity, no
output short-circuit protection is provided. This
configuration will not normally be a problem since
most applications include a series matching resistor
at the output that limits the internal power dissipation
if the output side of this resistor is shorted to ground.
However, shorting the output pin directly to the
adjacent positive power supply pin will, in most
cases, destroy the amplifier. Including a small series
resistor (5Ω) in the power-supply line will protect
against this. Always place the 0.1µF decoupling
capacitor directly on the supply pins.
Driving Capacitive Loads
The Typical Characteristics show the recommended
RS versus capacitive load and the resulting
frequency response at the load. Parasitic capacitive
loads greater than 2pF can begin to degrade the
performance of the OPA2652. Long PCB traces,
unmatched cables, and connections to multiple
devices can easily exceed this value. Always
consider this effect carefully, and add the
recommended series resistor as close as possible to
the OPA2652 output pin (see Board Layout
Guidelines).
Distortion Performance
The OPA2652 provides good distortion performance
into a 100Ω load on ±5V supplies. Increasing the
load impedance improves distortion directly.
Remember that the total load includes the feedback
network;
in
the
noninverting
configuration
(Figure 28), this is sum of RF + RG, while in the
inverting configuration, it is only RF. Also, providing
an additional supply decoupling capacitor (0.1µF)
between the supply pins (for bipolar operation)
improves the 2nd-order distortion slightly (3dB to
6dB).
It is also true that increasing the output voltage swing
increases harmonic distortion.
Noise Performance
One of the most demanding and yet very common
load conditions for an op amp is capacitive loading.
Often, the capacitive load is the input of an
analog-to-digital
(A/D)
converter—including
additional external capacitance that may be
recommended to improve A/D linearity. A high-speed
amplifier such as the OPA2652 can be very
susceptible to decreased stability and closed-loop
response peaking when a capacitive load is placed
14
directly on the output pin. When the amplifier
open-loop output resistance is considered, this
capacitive load introduces an additional pole in the
signal path that can decrease the phase margin.
Several external solutions to this problem have been
suggested. When the primary considerations are
frequency response flatness, pulse response fidelity,
and/or distortion, the simplest and most effective
solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor
between the amplifier output and the capacitive load.
This resistor does not eliminate the pole from the
loop response, but rather shifts it and adds a zero at
a higher frequency. The additional zero acts to
cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving
stability.
The
OPA2652
input-referred
voltage
noise
(8nV/√Hz), and the two input-referred current noise
terms (1.4pA/√Hz), combine to give low output noise
under a wide variety of operating conditions.
Figure 35 shows the op amp noise analysis model
with all the noise terms included. In this model, all
noise terms are taken to be noise voltage or current
density terms in either nV/√Hz or pA/√Hz.
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DC Accuracy and Offset Control
ENI
RS
1/2
OPA2652
IBN
EO
ERS
RF
4kTRS
IBI
RG
4kT
RG
4kTRF
-20
4kT = 1.6 x 10
at 290°K
J
Figure 35. Op Amp Noise Analysis Model
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 1 shows the
general form for the output noise voltage using the
terms shown in Figure 35.
EN +
Ǹ
ENI
2
F
(1)
Dividing this expression by the noise gain (NG = 1 +
RF/RG) gives the equivalent input-referred spot noise
voltage at the noninverting input, as shown in
Equation 2.
EO +
ǸǒE
2
NI
2
Ǔ
+ " (1.94 @ 7.0mV) " (402W @ 1.0mA)
+ " 14.0mV
ǒNG + noninverting signal gainǓ
2
ǒ Ǔ ) 4kTR
NG
2
I R
)ǒI BNRSǓ )4kTRS) BI F
NG
The balanced input stage of a wideband voltage
feedback op amp allows good output DC accuracy in
a wide variety of applications. Although the
high-speed input stage does require relatively high
input bias current (typically 4µA out of each input
terminal), the close matching between them may be
used to significantly reduce the output DC error
caused by this current. This reduction is done by
matching the DC source resistances appearing at the
two inputs. This matching reduces the output DC
error resulting from the input bias currents to the
offset current times the feedback resistor. Evaluating
the configuration of Figure 28, using worst-case
+25°C input offset voltage and current specifications,
gives a worst-case output offset voltage equal to:
" ǒNG @ V OS(MAX)Ǔ " ǒR F @ I OS(MAX)Ǔ
2
)ǒI BNRSǓ )4kTRS NG 2)(I BIRF) )4kTRFNG
(2)
Evaluating these two equations for the OPA2652
circuit and component values shown in Figure 28
gives a total output spot noise voltage of 17nV/√Hz
and a total equivalent input spot noise voltage of
8.4nV/√Hz. This noise includes the noise added by
the bias current cancellation resistor (205Ω) on the
noninverting input. This total input-referred spot
noise voltage is only slightly higher than the 8nV/√Hz
specification for the op amp voltage noise alone.
This result will be the case as long as the
impedances appearing at each op amp input are
limited to the previously recommend maximum value
of 300Ω. Keeping both (RF || RG) and the
noninverting input source impedance less than 300Ω
satisfies both noise and frequency response flatness
considerations. Since the resistor-induced noise is
relatively negligible, additional capacitive decoupling
across the bias current cancellation resistor (RB) for
the inverting op amp configuration of Figure 29 is not
required.
A fine scale output offset null, or DC operating point
adjustment, is often required. Numerous techniques
are available for introducing DC offset control into an
op amp circuit. Most of these techniques add a DC
current through the feedback resistor. In selecting an
offset trim method, one key consideration is the
impact on the desired signal path frequency
response. If the signal path is intended to be
noninverting, the offset control is best applied as an
inverting summing signal to avoid interaction with the
signal source. If the signal path is intended to be
inverting, applying the offset control to the
noninverting input may be considered. However, the
DC offset voltage on the summing junction sets up a
DC current back into the source which must be
considered. Applying an offset adjustment to the
inverting op amp input can change the noise gain
and frequency response flatness. For a DC-coupled
inverting amplifier, Figure 36 shows one example of
an offset adjustment technique that has minimal
impact on the signal frequency response. In this
case, the DC offset current is brought into the
inverting input node through resistor values that are
much larger than the signal path resistors. This
configuration ensures that the adjustment circuit has
minimal effect on the loop gain, and therefore on the
frequency response as well.
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+5V
Supply Decoupling
Not Shown
0.1mF
328W
1/2
OPA2652
This absolute worst-case condition meets the
specified maximum junction temperature. Actual PDL
will almost always be less than that considered here.
Carefully consider maximum TJ in your application.
VO
BOARD LAYOUT GUIDELINES
Achieving
optimum
performance
with
a
high-frequency amplifier such as the OPA2652
requires careful attention to board layout parasitics
and external component types. Recommendations
that will optimize performance include:
-5V
+5V
RG
500W
5kW
RF
1kW
VI
20kW
±200mV Output Adjustment
10kW
0.1mF
5kW
R
VO
= - F = -2
RG
VI
-5V
Figure 36. DC-Coupled, Inverting Gain of –2, with
Offset Adjustment
Thermal Analysis
Heatsinking or forced airflow may be required under
extreme operating conditions. Maximum desired
junction temperature will set the maximum allowed
internal power dissipation as described below. In no
case should the maximum junction temperature be
allowed to exceed 175°C.
Operating junction temperature (TJ) is given by TA +
PD • θJA. The total internal power dissipation (PD) is
the sum of quiescent power (PDQ) and additional
power dissipated in the output stage (PDL) to deliver
load power. Quiescent power is simply the specified
no-load supply current times the total supply voltage
across the part. PDL depends on the required output
signal and load; for a grounded resistive load, PDL is
at a maximum when the output is fixed at a voltage
equal to 1/2 of either supply voltage (for equal
bipolar supplies). Under this condition, PDL =
VS2/(4 • RL) where RL includes feedback network
loading.
Note that it is the power in the output stage, and not
into the load, that determines internal power
dissipation.
As an example, compute the maximum TJ using an
OPA2652E (SOT23-8 package) in the circuit of
Figure 28 operating at the maximum specified
ambient temperature of +85°C and with both outputs
driving 2.5VDC into a grounded 100Ω load.
PD = 10V • 15.5mA + 2 [52/(4 • [100Ω 804Ω])] =
296mW
Maximum TJ = +85°C + (0.30W • 150°C/W) = 130°C
16
a) Minimize parasitic capacitance to any AC
ground for all of the signal I/O pins. Parasitic
capacitance on the output and inverting input pins
can cause instability: on the noninverting input, it can
react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted
capacitance, a window around the signal I/O pins
should be opened in all of the ground and power
planes around those pins. Otherwise, ground and
power planes should be unbroken elsewhere on the
board.
b) Minimize the distance (< 0.25") from the
power-supply pins to high-frequency 0.1µF
decoupling capacitors. At the device pins, the ground
and power plane layout should not be in close
proximity to the signal I/O pins. Avoid narrow power
and ground traces to minimize inductance between
the pins and the decoupling capacitors. The
power-supply connections should always be
decoupled with these capacitors. An optional supply
decoupling capacitor (0.1µF) across the two power
supplies (for bipolar operation) will improve 2nd
harmonic distortion performance. Larger (2.2µF to
6.8µF) decoupling capacitors, effective at lower
frequency, should also be used on the main supply
pins. These capacitors may be placed somewhat
farther from the device and may be shared among
several devices in the same area of the PCB.
c) Careful selection and placement of external
components will preserve the high frequency
performance of the OPA2652. Resistors should be
a very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal
film or carbon composition axiallyleaded resistors
can also provide good high frequency performance.
Again, keep resistor leads and PCB traces as short
as possible. Never use wirewound type resistors in a
high-frequency application. Since the output pin and
inverting input pin are the most sensitive to parasitic
capacitance, always position the feedback and series
output resistor, if any, as close as possible to the
output pin. Other network components, such as
noninverting input termination resistors, should also
be placed close to the package. Where double-side
component mounting is allowed, place the feedback
resistor directly under the package on the other side
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of the board between the output and inverting input
pins. Even with a low parasitic capacitance shunting
the external resistors, excessively high resistor
values can create significant time constants that can
degrade performance. Good axial metal film or
surface-mount resistors have approximately 0.2pF in
shunt with the resistor. For resistor values >1.5kΩ,
this parasitic capacitance can add a pole and/or zero
below 500MHz that can effect circuit operation. Keep
resistor values as low as possible consistent with
load driving considerations. The 402Ω feedback
used in the typical performance specifications is a
good starting point for design. Note that a 25Ω
feedback resistor, rather than a direct short, is
suggested for the unity gain follower application. This
effectively isolates the inverting input capacitance
from the output pin that would otherwise cause
additional peaking in the gain of +1 frequency
response.
d) Connections to other wideband devices on the
board may be made with short direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to the
next device as a lumped capacitive load. Relatively
wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up
around them. Estimate the total capacitive load and
set RS from the plot of Recommended RS vs
Capacitive Load (Figure 17). Low parasitic capacitive
loads (< 5pF) may not need an RS since the
OPA2652 is nominally compensated to operate with
a 2pF parasitic load. Higher parasitic capacitive
loads without an RS are allowed as the signal gain
increases (increasing the unloaded phase margin) If
a long trace is required, and the 6dB signal loss
intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched impedance
transmission line using microstrip or stripline
techniques (consult an ECL design handbook for
microstrip and stripline layout techniques). A 50Ω
environment is normally not necessary on board, and
in fact, a higher impedance environment will improve
distortion as shown in the distortion versus load
plots. With a characteristic board trace impedance
defined (based on board material and trace
dimensions), a matching series resistor into the trace
from the output of the OPA2652 is used as well as a
terminating shunt resistor at the input of the
destination device. Remember also that the
terminating impedance will be the parallel
combination of the shunt resistor and the input
impedance of the destination device; this total
effective impedance should be set to match the trace
impedance. The high output voltage and current
capability of the OPA2652 allows multiple destination
devices to be handled as separate transmission
lines, each with respective series and shunt
terminations. If the 6dB attenuation of a
doubly-terminated transmission line is unacceptable,
a long trace can be series-terminated at the source
end only. Treat the trace as a capacitive load in this
case and set the series resistor value as shown in
the plot of Recommended RS vs Capacitive Load
(Figure 17). This configuration will not preserve
signal integrity as well as a doubly-terminated line. If
the input impedance of the destination device is low,
there will be some signal attenuation due to the
voltage divider formed by the series output into the
terminating impedance.
e) Socketing a high-speed part like the OPA2652
is not recommended. The additional lead length
and pin-to-pin capacitance introduced by the socket
can create an extremely troublesome parasitic
network that can make it almost impossible to
achieve a smooth, stable frequency response. Best
results are obtained by soldering the OPA2652
directly onto the board.
Input and ESD Protection
The OPA2652 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are
reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD
protection diodes to the power supplies as shown in
Figure 37.
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA
continuous current. Where higher currents are
possible (for example, in systems with ±15V supply
parts driving into the OPA2652), current-limiting
series resistors should be added into the two inputs.
Keep these resistor values as low as possible since
high values degrade both noise performance and
frequency response.
+VCC
External
Pin
Internal
Circuitry
-VCC
Figure 37. Internal ESD Protection
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Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (June 2000) to A Revision ........................................................................................................ Page
•
•
•
•
18
Changed format of data sheet. Updated to XML from PageMaker. ..................................................................................... 1
Changed input voltage axis values to correct units. ............................................................................................................ 8
Changed reference to alternate part numbers.................................................................................................................... 10
Changed information regarding available demonstration fixtures....................................................................................... 12
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PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
OPA2652E/250
ACTIVE
SOT-23
DCN
8
250
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
OPA2652E/250G4
ACTIVE
SOT-23
DCN
8
250
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
OPA2652E/3K
ACTIVE
SOT-23
DCN
8
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
OPA2652E/3KG4
ACTIVE
SOT-23
DCN
8
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
OPA2652U
ACTIVE
SOIC
D
8
100
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
OPA2652U-1/2K5
ACTIVE
SOIC
D
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
OPA2652U-1/2K5G4
ACTIVE
SOIC
D
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
OPA2652U/2K5
ACTIVE
SOIC
D
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
OPA2652U/2K5G4
ACTIVE
SOIC
D
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
OPA2652UG4
ACTIVE
SOIC
D
8
100
CU NIPDAU
Level-2-260C-1 YEAR
Green (RoHS &
no Sb/Br)
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
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