LINER LTC3407AEMSE-2-PBF Dual synchronous 800ma,2.25mhz step-down dc/dc regulator Datasheet

LTC3407A-2
Dual Synchronous 800mA,
2.25MHz Step-Down
DC/DC Regulator
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FEATURES
DESCRIPTIO
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The LTC®3407A-2 is a dual, constant frequency, synchronous step-down DC/DC converter. Intended for low power
applications, it operates from a 2.5V to 5.5V input voltage
range and has a constant 2.25MHz switching frequency,
enabling the use of tiny, low cost capacitors and inductors
1mm or less in height. Each output voltage is adjustable
from 0.6V to 5V. Internal synchronous 0.35Ω, 1A power
switches provide high efficiency without the need for
external Schottky diodes.
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■
■
■
■
■
■
■
■
■
■
High Efficiency: Up to 95%
Low Ripple (<35mVPK-PK) Burst Mode Operation;
IQ = 40μA
2.25MHz Constant Frequency Operation
No Schottky Diodes Required
Low RDS(ON) Internal Switches: 0.35Ω
Current Mode Operation for Excellent Line
and Load Transient Response
Short-Circuit Protected
Low Dropout Operation: 100% Duty Cycle
Ultralow Shutdown Current: IQ <1μA
Output Voltages from 5V down to 0.6V
Power-On Reset Output
Externally Synchronizable Oscillator
Optional External Soft-Start
Small Thermally Enhanced MSOP and 3mm × 3mm
DFN Packages
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APPLICATIO S
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To further maximize battery life, the P-channel MOSFETs
are turned on continuously in dropout (100% duty cycle),
and both channels draw a total quiescent current of only
40μA. In shutdown, the device draws <1μA.
, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by
U.S. Patents including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131.
PDAs/Palmtop PCs
Digital Cameras
Cellular Phones
Wireless and DSL Modems
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A user selectable mode input is provided to allow the user
to trade-off ripple noise for low power efficiency. Burst
Mode® operation provides the highest efficiency at light
loads, while Pulse Skip Mode provides the lowest ripple
noise at light loads.
TYPICAL APPLICATIO
Efficiency/Power Loss Curve
1mm High 2.5V/1.8V at 800mA Step-Down Regulators
1
100
VIN = 2.5V
TO 5.5V
RUN/SS2
VIN
RUN/SS1
RESET
LTC3407A-2
2.2μH
2.2μH
SW2
0.1
70
POR
SW1
22pF
22pF
VOUT1 = 1.8V
AT 800mA
60
0.01
50
40
30
0.001
20
10μF
VFB1
VFB2
887k
280k
GND
VIN = 3.3V
Burst Mode OPERATION
NO LOAD ON OTHER CHANNEL
10
887k
442k
0
10μF
POWER LOSS (W)
MODE/SYNC
1.8V
80
100k
EFFICIENCY (%)
10μF
VOUT2 = 2.5V
AT 800mA
2.5V
90
1
10
100
LOAD CURRENT (mA)
0.0001
1000
3407A2 TA02
3407A2 TA01
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LTC3407A-2
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ABSOLUTE
AXI U RATI GS
(Note 1)
VIN Voltage ..................................................– 0.3V to 6V
VFB1, VFB2 Voltages...................................– 0.3V to 1.5V
RUN/SS1, RUN/SS2 Voltages ..................... – 0.3V to VIN
MODE/SYNC Voltage .................................. – 0.3V to VIN
SW1, SW2 Voltages ....................... – 0.3V to VIN + 0.3V
POR Voltage ................................................– 0.3V to 6V
Operating Temperature Range (Note 2)
LTC3407AE-2 ..................................... – 40°C to 85°C
LTC3407AI-2 .................................... – 40°C to 125°C
Junction Temperature (Note 5) ............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)
LTC3407AEMSE-2 ............................................ 300°C
LTC3407AIMSE-2 ............................................. 300°C
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PI CO FIGURATIO
TOP VIEW
VFB1
1
RUN/SS1
2
TOP VIEW
10 VFB2
9 RUN/SS2
11
VIN
3
SW1
4
7 SW2
GND
5
6 MODE/SYNC
8 POR
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS PGND, MUST BE CONNECTED TO GND
VFB1
RUN/SS1
VIN
SW1
GND
1
2
3
4
5
11
10
9
8
7
6
VFB2
RUN/SS2
POR
SW2
MODE/SYNC
MSE PACKAGE
10-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS PGND, MUST BE CONNECTED TO GND
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ORDER I FOR ATIO
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3407AEDD-2#PBF
LTC3407AEDD-2#TRPBF
LDDH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3407AEMSE-2#PBF
LTC3407AEMSE-2#TRPBF
LTDDJ
10-Lead Plastic MSOP
–40°C to 85°C
LTC3407AIDD-2#PBF
LTC3407AIDD-2#TRPBF
LDDH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3407AIMSE-2#PBF
LTC3407AIMSE-2#TRPBF
LTDDJ
10-Lead Plastic MSOP
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3407AEDD-2
LTC3407AEDD-2#TR
LDDH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3407AEMSE-2
LTC3407AEMSE-2#TR
LTDDJ
10-Lead Plastic MSOP
–40°C to 85°C
LTC3407AIDD-2
LTC3407AIDD-2#TR
LDDH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3407AIMSE-2
LTC3407AIMSE-2#TR
LTDDJ
10-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3407A-2
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V, unless otherwise specified. (Note 2)
SYMBOL
PARAMETER
VIN
Operating Voltage Range
IFB
Feedback Pin Input Current
VFB
Feedback Voltage (Note 3)
CONDITIONS
MIN
●
TYP
2.5
●
0°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C (Note 2)
●
0.588
0.585
MAX
UNITS
5.5
V
30
nA
0.6
0.6
0.612
0.612
V
V
0.5
ΔVLINE REG
Reference Voltage Line Regulation
VIN = 2.5V to 5.5V (Note 3)
0.3
ΔVLOAD REG
Output Voltage Load Regulation
MODE/SYNC = 0V (Note 3)
0.5
IS
Input DC Supply Current
Active Mode
Sleep Mode
Shutdown
(Note 4)
VFB1 = VFB2 = 0.5V
VFB1 = VFB2 = 0.63V, MODE/SYNC = 3.6V
RUN = 0V, VIN = 5.5V, MODE/SYNC = 0V
700
40
0.1
950
60
1
fOSC
Oscillator Frequency
VFBX = 0.6V
2.25
2.7
fSYNC
Synchronization Frequency
ILIM
Peak Switch Current Limit
VIN = 3V, VFBX = 0.5V, Duty Cycle <35%
1.2
1.6
A
RDS(ON)
Top Switch On-Resistance
Bottom Switch On-Resistance
(Note 6)
(Note 6)
0.35
0.30
0.45
0.45
Ω
Ω
ISW(LKG)
Switch Leakage Current
VIN = 5V, VRUN = 0V, VFBX = 0V
0.01
1
μA
POR
Power-On Reset Threshold
VFBX Ramping Up, MODE/SYNC = 0V
VFBX Ramping Down, MODE/SYNC = 0V
8.5
–8.5
●
1.8
%
2.25
1
Power-On Reset On-Resistance
100
Power-On Reset Delay
%/V
μA
μA
μA
MHz
MHz
%
%
200
65,536
Ω
Cycles
VRUN
RUN/SS Threshold Low
RUN/SS Threshold High
●
●
IRUN
RUN/SS Leakage Current
●
VMODE
MODE Threshold Low
0
MODE Threshold High
VIN – 0.5
VIN
V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3407AE-2 is guaranteed to meet specified performance
from 0°C to 85°C. Specifications over the – 40°C and 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3407AI-2 is guaranteed over the
full –40°C to 125°C operating temperature range.
0.3
1
0.01
1.5
2
V
V
1
μA
0.5
V
Note 3: The LTC3407A-2 is tested in a proprietary test mode that connects
VFB to the output of the error amplifier.
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 5: TJ is calculated from the ambient TA and power dissipation PD
according to the following formula: TJ = TA + (PD • θJA).
Note 6: The DFN switch on-resistance is guaranteed by correlation to
wafer level measurements.
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LTC3407A-2
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise specified.
Burst Mode Operation
Pulse Skipping Mode
SW
5V/DIV
SW
5V/DIV
VOUT
50mV/DIV
VOUT
10mV/DIV
IL
200mA/DIV
IL
100mA/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 50mA
CIRCUIT OF FIGURE 3
VIN = 3.6V
VOUT = 1.8V
ILOAD = 50mA
CIRCUIT OF FIGURE 3
3407A2 G01
2μs/DIV
Load Step
3407A2 G02
1μs/DIV
Soft Start
VOUT1
200mV/DIV
VIN
2V/DIV
VOUT2
100mV/DIV
VOUT1
1V/DIV
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 50mA TO 600mA
CIRCUIT OF FIGURE 3
20μs/DIV
Oscillator Frequency vs
Temperature
Efficiency vs Input Voltage
2.5
100mA
90
1mA
10mA
8
2.4
70
FREQUENCY (MHz)
EFFICIENCY (%)
10
VIN = 3.6V
800mA
80
60
50
40
30
2.3
2.2
20
2.1
10 VOUT = 1.8V
CIRCUIT OF FIGURE 3
0
4
3
2
5
INPUT VOLTAGE (V)
2.0
–50 –25
3407A2 G16
1ms/DIV
Oscillator Frequency vs Supply
Voltage
FREQUENCY DEVIATION (%)
100
VIN = 3.6V
VOUT = 1.8V
ILOAD = 500mA
CIRCUIT OF FIGURE 4
3407A2 G03
6
4
2
0
–2
–4
–6
–8
6
3407A2 G04
–10
50
25
75
0
TEMPERATURE (°C)
100
125
2
3
4
5
6
SUPPLY VOLTAGE (V)
3407A2 G05
3407A2 G06
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LTC3407A-2
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage vs
Temperature
TA = 25°C unless otherwise specified.
RDS(ON) vs Input Voltage
RDS(ON) vs Temperature
500
0.615
550
VIN = 3.6V
450
0.610
VIN = 4.2V
450
0.600
0.595
400
350
300
0.590
250
0.585
–50 –25
200
50
25
75
0
TEMPERATURE (°C)
100
2
3
4
VIN (V)
80
3.3V
5
50
40
Load Regulation
Burst Mode OPERATION
3
2
60
PULSE SKIP MODE
50
40
30
30
20
20
10 VOUT = 2.5V Burst Mode OPERATION
CIRCUIT OF FIGURE 3
0
1
10
100
LOAD CURRENT (mA)
10 VIN = 3.6V, VOUT = 1.8V
NO LOAD ON OTHER CHANNEL
0
1
10
100
LOAD CURRENT (mA)
1000
Efficiency vs Load Current
3.3V
–3
–4
1
1000
3.3V
VOUT ERROR (%)
EFFICIENCY (%)
40
20
20
10
10
10
100
LOAD CURRENT (mA)
0.3
4.2V
50
30
3407A2 G13
0.2
0.1
0
–0.1
–0.2
–0.3
–0.4
VOUT = 1.5V Burst Mode OPERATION
0
1000
VOUT = 1.8V
IOUT = 200mA
0.4
60
30
VOUT = 1.2V Burst Mode OPERATION
1000
Line Regulation
2.7V
70
40
10
100
LOAD CURRENT (mA)
3407A2 G12
80
4.2V
50
VIN = 3.6V, VOUT = 1.8V
NO LOAD ON OTHER CHANNEL
0.5
90
60
PULSE SKIP MODE
–1
Efficiency vs Load Current
2.7V
1
0
–2
100
0
1 Burst Mode OPERATION
3407A2 G11
100
70
25 50 75 100 125 150
TEMPERATURE (°C)
3407A2 G09
0
3407A2 G08
VOUT ERROR(%)
EFFICIENCY (%)
60
MAIN SWITCH
SYNCHRONOUS SWITCH
4
3407A2 G10
EFFICIENCY (%)
7
6
70
80
250
100
–50 –25
80
70
90
300
Efficiency vs Load Current
90
4.2V
350
200
100
2.7V
VIN = 3.6V
400
150
1
125
Efficiency vs Load Current
EFFICIENCY (%)
SYNCHRONOUS
SWITCH
3407A2 G07
100
90
MAIN
SWITCH
RDS(ON) (mΩ)
0.605
RDS(ON) (mΩ)
REFERENCE VOLTAGE (V)
VIN = 2.7V
500
1
10
100
LOAD CURRENT (mA)
1000
–0.5
2
3
4
VIN (V)
5
6
3407A2 G15
3407A2 G14
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LTC3407A-2
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PI FU CTIO S
VFB1 (Pin 1): Output Feedback. Receives the feedback
voltage from the external resistive divider across the
output. Nominal voltage for this pin is 0.6V.
be synchronized to an external oscillator applied to this pin
and pulse skipping mode is automatically selected.
SW2 (Pin 7): Regulator 2 Switch Node Connection to the
Inductor. This pin swings from VIN to GND.
RUN/SS1 (Pin 2): Regulator 1 Enable and Soft-Start Input.
Forcing this pin to VIN enables regulator 1, while forcing it
to GND causes regulator 1 to shut down. Connect external
RC-network with desired time-constant to enable softstart feature. This pin must be driven; do not float.
POR (Pin 8): Power-On Reset. This common-drain logic
output is pulled to GND when the output voltage is not
within ±8.5% of regulation and goes high after 216 clock
cycles when both channels are within regulation.
VIN (Pin 3): Main Power Supply. Must be closely decoupled
to GND.
RUN/SS2 (Pin 9): Regulator 2 Enable and Soft-Start Input.
Forcing this pin to VIN enables regulator 2, while forcing it
to GND causes regulator 2 to shut down. Connect external
RC-Network with desired time-constant to enable softstart feature. This pin must be driven; do not float.
SW1 (Pin 4): Regulator 1 Switch Node Connection to the
Inductor. This pin swings from VIN to GND.
GND (Pin 5): Main Ground. Connect to the (–) terminal of
COUT, and (–) terminal of CIN.
VFB2 (Pin 10): Output Feedback. Receives the feedback
voltage from the external resistive divider across the
output. Nominal voltage for this pin is 0.6V.
MODE/SYNC (Pin 6): Combination Mode Selection and
Oscillator Synchronization. This pin controls the operation
of the device. When tied to VIN or GND, Burst Mode
operation or pulse skipping mode is selected, respectively. Do not float this pin. The oscillation frequency can
Exposed Pad (GND) (Pin 11): Power Ground. Connect to
the (–) terminal of COUT, and (–) terminal of CIN. Must be
soldered to electrical ground on PCB.
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BLOCK DIAGRA
REGULATOR 1
MODE/SYNC
6
BURST
CLAMP
VIN
SLOPE
COMP
0.6V
EA
VFB1
SLEEP
ITH
–
+
5Ω
ICOMP
+
0.65V
–
1
EN
–
+
BURST
S
0.55V
–
UVDET
Q
RS
LATCH
R
Q
UV
+
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
ANTI
SHOOTTHRU
4 SW1
+
–
IRCMP
–
0.65V
OV
+
OVDET
SHUTDOWN
PGOOD1
11 GND
VIN
3 VIN
8 POR
RUN/SS1 2
0.6V REF
POR
COUNTER
OSC
RUN/SS2 9
OSC
5 GND
PGOOD2
REGULATOR 2 (IDENTICAL TO REGULATOR 1)
VFB2 10
7 SW2
3407A2 BD
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LTC3407A-2
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OPERATIO
The LTC3407A-2 uses a constant frequency, current mode
architecture. The operating frequency is set at 2.25MHz
and can be synchronized to an external oscillator. Both
channels share the same clock and run in-phase. To suit
a variety of applications, the selectable MODE/SYNC pin
allows the user to trade-off noise for efficiency.
The output voltage is set by an external divider returned to
the VFB pins. An error amplifier compares the divided
output voltage with a reference voltage of 0.6V and adjusts
the peak inductor current accordingly. Overvoltage and
undervoltage comparators will pull the POR output low if
the output voltage is not within ±8.5%. The POR output
will go high after 65,536 clock cycles (about 29ms in pulse
skipping mode) of achieving regulation.
Main Control Loop
During normal operation, the top power switch (P-channel
MOSFET) is turned on at the beginning of a clock cycle
when the VFB voltage is below the reference voltage. The
current into the inductor and the load increases until the
current limit is reached. The switch turns off and energy
stored in the inductor flows through the bottom switch (Nchannel MOSFET) into the load until the next clock cycle.
The peak inductor current is controlled by the internally
compensated ITH voltage, which is the output of the error
amplifier.This amplifier compares the VFB pin to the 0.6V
reference. When the load current increases, the VFB voltage decreases slightly below the reference. This
decrease causes the error amplifier to increase the ITH
voltage until the average inductor current matches the new
load current.
automatically switches into Burst Mode operation in which
the PMOS switch operates intermittently based on load
demand with a fixed peak inductor current. By running
cycles periodically, the switching losses which are dominated by the gate charge losses of the power MOSFETs are
minimized. The main control loop is interrupted when the
output voltage reaches the desired regulated value. A
voltage comparator trips when ITH is below 0.65V, shutting off the switch and reducing the power. The output
capacitor and the inductor supply the power to the load
until ITH exceeds 0.65V, turning on the switch and the main
control loop which starts another cycle.
For lower ripple noise at low currents, the pulse skipping
mode can be used. In this mode, the LTC3407A-2 continues to switch at a constant frequency down to very low
currents, where it will begin skipping pulses.
Dropout Operation
When the input supply voltage decreases toward the
output voltage, the duty cycle increases to 100% which is
the dropout condition. In dropout, the PMOS switch is
turned on continuously with the output voltage being
equal to the input voltage minus the voltage drops across
the internal P-channel MOSFET and the inductor.
An important design consideration is that the RDS(ON) of
the P-channel switch increases with decreasing input
supply voltage (See Typical Performance Characteristics).
Therefore, the user should calculate the power dissipation
when the LTC3407A-2 is used at 100% duty cycle with low
input voltage (See Thermal Considerations in the Applications Information Section).
The main control loop is shut down by pulling the RUN/SS
pin to ground.
Low Supply Operation
Low Current Operation
The LTC3407A-2 incorporates an undervoltage lockout
circuit which shuts down the part when the input voltage
drops below about 1.65V to prevent unstable operation.
Two modes are available to control the operation of the
LTC3407A-2 at low currents. Both modes automatically
switch from continuous operation to the selected mode
when the load current is low.
To optimize efficiency, the Burst Mode operation can be
selected. When the load is relatively light, the LTC3407A-2
A general LTC3407A-2 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement, and begins with the selection of the
inductor L. Once the inductor is chosen, CIN and COUT can
be selected.
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LTC3407A-2
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APPLICATIO S I FOR ATIO
Inductor Selection
Although the inductor does not influence the operating
frequency, the inductor value has a direct effect on ripple
current. The inductor ripple current ΔIL decreases with
higher inductance and increases with higher VIN or VOUT:
ΔIL =
VOUT ⎛ VOUT ⎞
• 1–
fO • L ⎜⎝
VIN ⎟⎠
Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple,
greater core losses, and lower output current capability.
A reasonable starting point for setting ripple current is
ΔIL = 0.3 • ILIM, where ILIM is the peak switch current limit.
The largest ripple current ΔIL occurs at the maximum
input voltage. To guarantee that the ripple current stays
below a specified maximum, the inductor value should be
chosen according to the following equation:
L=
⎞
VOUT ⎛
V
• ⎜ 1– OUT ⎟
fO • ΔIL ⎝ VIN(MAX ) ⎠
The inductor value will also have an effect on Burst Mode
operation. The transition from low current operation begins when the peak inductor current falls below a level set
by the burst clamp. Lower inductor values result in higher
ripple current which causes this transition to occur at
lower load currents. This causes a dip in efficiency in the
upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to increase.
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with
similar electrical characterisitics. The choice of which
style inductor to use often depends more on the price vs
size requirements and any radiated field/EMI requirements than on what the LTC3407A-2 requires to operate.
Table 1 shows some typical surface mount inductors that
work well in LTC3407A-2 applications.
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is
a square wave with a duty cycle of approximately VOUT/
VIN. To prevent large voltage transients, a low equivalent
series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS
capacitor current is given by:
IRMS ≈ IMAX
VOUT ( VIN – VOUT )
VIN
where the maximum average output current IMAX equals
the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL/2.
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case is commonly used
to design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature
than required. Several capacitors may also be paralleled to
meet the size or height requirements of the design. An
additional 0.1μF to 1μF ceramic capacitor is also recommended on VIN for high frequency decoupling, when not
using an all ceramic capacitor solution.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER
Taiyo Yuden CB2016T2R2M
CB2012T2R2M
CB2016T3R3M
MAX DC
VALUE CURRENT DCR
2.2μH
510mA
0.13Ω
2.2μH
530mA
0.33Ω
3.3μH
410mA
0.27Ω
HEIGHT
1.6mm
1.25mm
1.6mm
Panasonic
ELT5KT4R7M
4.7μH
950mA
0.2Ω
1.2mm
Sumida
CDRH2D18/LD
4.7μH
630mA
0.086Ω
2mm
Murata
LQH32CN4R7M23 4.7μH
450mA
0.2Ω
2mm
Taiyo Yuden NR30102R2M
NR30104R7M
2.2μH
4.7μH
1100mA
750mA
0.1Ω
0.19Ω
1mm
1mm
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7μH
3.3μH
2.2μH
1100mA
1200mA
1300mA
0.11Ω
0.1Ω
0.08Ω
1mm
1mm
1mm
TDK
VLF3010AT4R7MR70
VLF3010AT3R3MR87
VLF3010AT2R2M1R0
4.7μH
700mA
0.28Ω
1mm
3.3μH
870mA
0.17Ω
1mm
2.2μH
1000mA
0.12Ω
1mm
3407a2f
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APPLICATIO S I FOR ATIO
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required ESR to
minimize voltage ripple and load step transients. Typically,
once the ESR requirement is satisfied, the capacitance is
adequate for filtering. The output ripple (ΔVOUT) is determined by:
⎛
⎞
1
ΔVOUT ≈ ΔIL ⎜ ESR +
8 fO COUT ⎟⎠
⎝
where fO = operating frequency, COUT = output capacitance
and ΔIL = ripple current in the inductor. The output ripple
is highest at maximum input voltage since ΔIL increases
with input voltage. With ΔIL = 0.3 • ILIM the output ripple
will be less than 100mV at maximum VIN and fO = 2.25MHz
with:
ESRCOUT < 150mΩ
Once the ESR requirements for COUT have been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement, except for an all ceramic solution.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Aluminum electrolytic, special polymer, ceramic and dry tantulum
capacitors are all available in surface mount packages. The
OS-CON semiconductor dielectric capacitor available from
Sanyo has the lowest ESR(size) product of any aluminum
electrolytic at a somewhat higher price. Special polymer
capacitors, such as Sanyo POSCAP, offer very low ESR,
but have a lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density.
However, they also have a larger ESR and it is critical that
they are surge tested for use in switching power supplies.
An excellent choice is the AVX TPS series of surface mount
tantalums, available in case heights ranging from 2mm to
4mm. Aluminum electrolytic capacitors have a significantly larger ESR, and are often used in extremely costsensitive applications provided that consideration is given
to ripple current ratings and long term reliability. Ceramic
capacitors have the lowest ESR and cost, but also have the
lowest capacitance density, a high voltage and temperature coefficient, and exhibit audible piezoelectric effects.
In addition, the high Q of ceramic capacitors along with
trace inductance can lead to significant ringing. Other
capacitor types include the Panasonic Special Polymer
(SP) capacitors.
In most cases, 0.1μF to 1μF of ceramic capacitors should
also be placed close to the LTC3407A-2 in parallel with the
main capacitors for high frequency decoupling.
VIN = 2.5V TO 5.5V
CIN
R7
VIN
BURST*
R6
PULSESKIP*
MODE/SYNC
R5
LTC3407A-2
RUN/SS2
RUN/SS1
L1
L2
VOUT2
POWER-ON
RESET
POR
SW2
VOUT1
SW1
C2
C1
C4
C3
R4
COUT2
VFB1
VFB2
GND
R3
R2
R1
*MODE/SYNC = 0V: PULSE SKIP
MODE/SYNC = VIN: Burst Mode
COUT1
3407A2 F01
Figure 1. LTC3407A-2 General Schematic
Ceramic Input and Output Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
Also, ceramic caps are prone to temperature effects which
requires the designer to check loop stability over the
operating temperature range. To minimize their large
temperature and voltage coefficients, only X5R or X7R
ceramic capacitors should be used. A good selection of
ceramic capacitors is available from Taiyo Yuden, TDK,
and Murata.
Great care must be taken when using only ceramic input
and output capacitors. When a ceramic capacitor is used
at the input and the power is being supplied through long
wires, such as from a wall adapter, a load step at the output
can induce ringing at the VIN pin. At best, this ringing can
couple to the output and be mistaken as loop instability.
3407a2f
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At worst, the ringing at the input can be large enough to
damage the part.
Since the ESR of a ceramic capacitor is so low, the input
and output capacitor must instead fulfill a charge storage
requirement. During a load step, the output capacitor must
instantaneously supply the current to support the load
until the feedback loop raises the switch current enough to
support the load. The time required for the feedback loop
to respond is dependent on the compensation and the
output capacitor size. Typically, 3-4 cycles are required to
respond to a load step, but only in the first cycle does the
output drop linearly. The output droop, VDROOP, is usually
about 3 times the linear drop of the first cycle. Thus, a good
place to start is with the output capacitor size of approximately:
COUT ≈ 3
ΔIOUT
fO • VDROOP
More capacitance may be required depending on the duty
cycle and load step requirements.
In most applications, the input capacitor is merely required to supply high frequency bypassing, since the
impedance to the supply is very low. A 10μF ceramic
capacitor is usually enough for these conditions.
Setting the Output Voltage
The LTC3407A-2 develops a 0.6V reference voltage between the feedback pin, VFB, and ground as shown in
Figure 1. The output voltage is set by a resistive divider
according to the following formula:
⎛ R2 ⎞
VOUT = 0.6 V ⎜ 1+ ⎟
⎝ R1⎠
Keeping the current small (<5μA) in these resistors maximizes efficiency, but making them too small may allow
stray capacitance to cause noise problems and reduce the
phase margin of the error amp loop.
To improve the frequency response, a feed-forward capacitor CF may also be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
Power-On Reset
The POR pin is an open-drain output which pulls low when
either regulator is out of regulation. When both output
voltages are within ±8.5% of regulation, a timer is started
which releases POR after 216 clock cycles (about 29ms in
pulse skipping mode). This delay can be significantly
longer in Burst Mode operation with low load currents,
since the clock cycles only occur during a burst and there
could be milliseconds of time between bursts. This can be
bypassed by tying the POR output to the MODE/SYNC
input, to force pulse skipping mode during a reset. In
addition, if the output voltage faults during Burst Mode
sleep, POR could have a slight delay for an undervoltage
output condition and may not respond to an overvoltage
output. This can be avoided by using pulse skipping mode
instead. When either channel is shut down, the POR
output is pulled low, since one or both of the channels are
not in regulation.
Mode Selection & Frequency Synchronization
The MODE/SYNC pin is a multipurpose pin which provides
mode selection and frequency synchronization. Connecting this pin to VIN enables Burst Mode operation, which
provides the best low current efficiency at the cost of a
higher output voltage ripple. When this pin is connected to
ground, pulse skipping operation is selected which provides the lowest output ripple, at the cost of low current
efficiency.
The LTC3407A-2 can also be synchronized to another
LTC3407A-2 by the MODE/SYNC pin. During synchronization, the mode is set to pulse skipping and the top switch
turn-on is synchronized to the rising edge of the external
clock.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ΔILOAD • ESR, where ESR is the effective series
resistance of COUT. ΔILOAD also begins to charge or
discharge COUT generating a feedback error signal used by
the regulator to return VOUT to its steady-state value. During
3407a2f
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this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem.
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard secondorder overshoot/DC ratio cannot be used to determine
phase margin. In addition, a feed-forward capacitor can be
added to improve the high frequency response, as shown
in Figure 1. Capacitors C1 and C2 provide phase lead by
creating high frequency zeros with R2 and R4 respectively, which improve the phase margin.
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to Application Note 76.
In some applications, a more severe transient can be
caused by switching in loads with large (>1μF) input
capacitors. The discharged input capacitors are effectively
put in parallel with COUT, causing a rapid drop in VOUT. No
regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance
and is driven quickly. The solution is to limit the turn-on
speed of the load switch driver. A Hot SwapTM controller is
designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and softstarting.
Soft-Start
The RUN/SS pins provide a means to separately run or
shut down the two regulators. In addition, they can optionally be used to externally control the rate at which each
regulator starts up and shuts down. Pulling the RUN/SS1
pin below 1V shuts down regulator 1 on the LTC3407A-2.
Forcing this pin to VIN enables regulator 1. In order to
control the rate at which each regulator turns on and off,
connect a resistor and capacitor to the RUN/SS pins as
shown in Figure 1. The soft-start duration can be calculated by using the following formula:
⎛ V −1 ⎞
(s)
t SS = RSSCSSIn ⎜ IN
⎝ VIN − 1.6 ⎟⎠
For approximately a 1ms ramp time, use RSS = 4.7MΩ and
CSS = 680pF at VIN = 3.3V.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% - (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC3407A-2 circuits: 1)VIN quiescent current, 2)
switching losses, 3) I2R losses, 4) other losses.
1) The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET driver
and control currents. VIN current results in a small (<0.1%)
loss that increases with VIN, even at no load.
2) The switching current is the sum of the MOSFET driver
and control currents. The MOSFET driver current results
from switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from VIN to
ground. The resulting dQ/dt is a current out of VIN that is
typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the
gate charges of the internal top and bottom MOSFET
switches. The gate charge losses are proportional to VIN
and thus their effects will be more pronounced at higher
supply voltages.
3) I2R losses are calculated from the DC resistances of the
internal switches, RSW, and external inductor, RL. In
continuous mode, the average output current flows through
inductor L, but is “chopped” between the internal top and
bottom switches. Thus, the series resistance looking into
Hot Swap is a registered trademark of Linear Technology Corporation.
3407a2f
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the SW pin is a function of both top and bottom MOSFET
RDS(ON) and the duty cycle (D) as follows:
RSW = (RDS(ON)TOP)(D) + (RDS(ON)BOT)(1 – D)
The RDS(ON) for both the top and bottom MOSFETs can be
obtained from the Typical Performance Characteristics
curves. Thus, to obtain I2R losses:
I2R losses = IOUT2(RSW + RL)
4) Other ‘hidden’ losses such as copper trace and internal
battery resistances can account for additional efficiency
degradations in portable systems. It is very important to
include these “system” level losses in the design of a
system. The internal battery and fuse resistance losses
can be minimized by making sure that CIN has adequate
charge storage and very low ESR at the switching frequency. Other losses including diode conduction losses
during dead-time and inductor core losses generally account for less than 2% total additional loss.
Thermal Considerations
In a majority of applications, the LTC3407A-2 does not
dissipate much heat due to its high efficiency. However, in
applications where the LTC3407A-2 is running at high
ambient temperature with low supply voltage and high
duty cycles, such as in dropout, the heat dissipated may
exceed the maximum junction temperature of the part. If
the junction temperature reaches approximately 150°C,
both power switches will be turned off and the SW node
will become high impedance.
To prevent the LTC3407A-2 from exceeding the maximum
junction temperature, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated exceeds the
maximum junction temperature of the part. The temperature rise is given by:
TRISE = PD • θJA
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature, TJ, is given by:
TJ = TRISE + TAMBIENT
As an example, consider the case when the LTC3407A-2
is in dropout on both channels at an input voltage of 2.7V
with a load current of 800mA and an ambient temperature
of 70°C. From the Typical Performance Characteristics
graph of Switch Resistance, the RDS(ON) resistance of the
main switch is 0.425Ω. Therefore, power dissipated by
each channel is:
PD = I2 • RDS(ON) = 272mW
The MS package junction-to-ambient thermal resistance,
θJA, is 45°C/W. Therefore, the junction temperature of the
regulator operating in a 70°C ambient temperature is
approximately:
TJ = 2 • 0.272 • 45 + 70 = 94.5°C
which is below the absolute maximum junction temperature of 125°C.
Design Example
As a design example, consider using the LTC3407A-2 in a
portable application with a Li-Ion battery. The battery
provides a VIN = 2.8V to 4.2V. The load requires a maximum of 800mA in active mode and 2mA in standby mode.
The output voltage is VOUT = 2.5V. Since the load still
needs power in standby, Burst Mode operation is selected
for good low load efficiency.
First, calculate the inductor value for about 30% ripple
current at maximum VIN:
L=
2.5V
⎛ 2.5V ⎞
• ⎜ 1–
= 1.25μH
2.25MHz • 360mA ⎝ 4.2V ⎟⎠
3407a2f
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APPLICATIO S I FOR ATIO
Choosing the next highest standardized inductor value of
2.2μH, results in a maximum ripple current of:
ΔIL =
2.5V
⎛ 2.5V ⎞
• ⎜ 1−
= 204mA
2.25MHz • 2.2μH ⎝ 4.2V ⎟⎠
For cost reasons, a ceramic capacitor will be used. COUT
selection is then based on load step droop instead of ESR
requirements. For a 5% output droop:
COUT ≈ 2.5
800mA
= 7.1μF
2.25MHz • (5% • 2.5V)
The closest standard value is 10μF. Since the output
impedance of a Li-Ion battery is very low, CIN is typically
10μF.
The output voltage can now be programmed by choosing
the values of R1 and R2. To maintain high efficiency, the
current in these resistors should be kept small. Choosing
2μA with the 0.6V feedback voltage makes R1~300k. A
close standard 1% resistor is 280k, and R2 is then 887k.
The POR pin is a common drain output and requires a pullup resistor. A 100k resistor is used for adequate speed.
Figure 3 shows the complete schematic for this design
example. The specific passive components chosen allow
for a 1mm height power supply that maintains a high
efficiency across load.
Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3407A-2. These items are also illustrated graphically
in the layout diagram of Figure 2. Check the following in
your layout:
1. Does the capacitor CIN connect to the power VIN (Pin 3)
and GND (exposed pad) as closely as possible? This
capacitor provides the AC current to the internal power
MOSFETs and their drivers.
2. Are COUT and L1 closely connected? The (–) plate of
COUT returns current to GND and the (–) plate of CIN.
3. The resistor divider formed by R1 and R2 must be
connected between the (+) plate of COUT and a ground
sense line terminated near GND (exposed pad). The feedback signals VFB1 and VFB2 should be routed away from
noisy components and traces, such as the SW lines (Pins
4 and 7), and their traces should be minimized.
4. Keep sensitive components away from the SW pins. The
input capacitor CIN and the resistors R1 to R4 should be
routed away from the SW traces and the inductors.
5. A ground plane is preferred, but if not available keep the
signal and power grounds segregated with small signal
components returning to the GND pin at one point.
Addtionally the two grounds should not share the high
current paths of CIN or COUT.
6. Flood all unused areas on all layers with copper.
Flooding with copper will reduce the temperature rise of
power components. These copper areas should be connected to VIN or GND.
VIN
CIN
RUN/SS2 VIN RUN/SS1
MODE/SYNC
POR
LTC3407A-2
L1
L2
VOUT2
SW1
SW2
C5
VFB1
VFB2
R4
COUT2
VOUT1
C4
GND
R3
R2
R1
COUT1
3407A2 F02
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 2. LTC3407A-2 Layout Diagram (See Board Layout Checklist)
3407a2f
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LTC3407A-2
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VIN = 2.5V
TO 5.5V
RUN/SS2
VIN
RUN/SS1
LTC3407A-2
L2
2.2μH
VOUT2 = 2.5V
AT 800mA
L1
2.2μH
C5, 22pF
R4
887k
C3
10μF
C4, 22pF
VOUT1 = 1.8V
AT 800mA
VFB1
VFB2
R2
R1 604k
301k
GND
R3
280k
2.5V
90
1.8V
80
SW1
SW2
100
POWER-ON
RESET
POR
MODE/SYNC
Efficiency vs Load Current
R5
100k
C2
10μF
70
EFFICIENCY (%)
C1
10μF
60
50
40
30
20
VIN = 3.3V
Burst Mode OPERATION
NO LOAD ON OTHER CHANNEL
10
C1, C2, C3: TAIYO YUDEN JMK316BJ106MD
L1, L2: TDK VLF3010AT-2R2M1R0
3407A2 TA03
0
10
100
LOAD CURRENT (mA)
1
Figure 3. 1mm Height Core Supply
1000
3407A2 TA03b
Efficiency vs Load Current
VIN = 2.5V TO 5.5V
100
R7
100k
VIN
LTC3407A-2 POR
RUN/SS2
VOUT2 = 2.5V
AT 800mA
R5
4.7MΩ
POWER-ON
RESET
L1
4.7μH
SW2
SW1
C2, 22pF
C1, 22pF
VOUT1 = 1.2V
AT 800mA
C4
680pF
C3
680pF
COUT2
10μF
R4
887k
VFB1
VFB2
R3
280k
MODE/SYNC
GND
R1
604k
R2
604k
2.5V
80
1.2V
70
RUN/SS1
L2
4.7μH
90
EFFICIENCY (%)
CIN
10μF
R6
4.7MΩ
60
50
40
30
20
VIN = 3.3V
Burst Mode OPERATION
NO LOAD ON OTHER CHANNEL
10
COUT1
10μF
0
1
CIN, COUT1, COUT2: TAIYO YUDEN JMK316BJ106ML
L1, L2: TDK VLF3012AT-4R7M74
3407A2 TA04
10
100
LOAD CURRENT (mA)
1000
3407A2 TA04b
Figure 4. Low Ripple Buck Regulators with Soft-Start
3407a2f
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LTC3407A-2
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PACKAGE DESCRIPTIO
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
R = 0.115
TYP
0.38 ± 0.10
6
10
5
1
0.675 ±0.05
3.50 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
PACKAGE
OUTLINE
0.25 ± 0.05
(DD) DFN 1103
0.75 ±0.05
0.200 REF
0.50
BSC
2.38 ±0.05
(2 SIDES)
0.25 ± 0.05
0.50 BSC
2.38 ±0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
MSE Package
10-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1664)
2.794 ± 0.102
(.110 ± .004)
5.23
(.206)
MIN
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.889 ± 0.127
(.035 ± .005)
0.497 ± 0.076
(.0196 ± .003)
REF
10 9 8 7 6
2.083 ± 0.102 3.20 – 3.45
(.082 ± .004) (.126 – .136)
0.254
(.010)
DETAIL “A”
0° – 6° TYP
BOTTOM VIEW OF
EXPOSED PAD OPTION
1
2.06 ± 0.102
(.081 ± .004)
1.83 ± 0.102
(.072 ± .004)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
GAUGE PLANE
0.50
0.305 ± 0.038
(.0197)
(.0120 ± .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
10
0.86
(.034)
REF
1.10
(.043)
MAX
DETAIL “A”
0.18
(.007)
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
1 2 3 4 5
0.50
(.0197)
BSC
0.127 ± 0.076
(.005 ± .003)
MSOP (MSE) 0603
3407a2f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3407A-2
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TYPICAL APPLICATIO
2mm Height Lithium-Ion Single Inductor Buck-Boost Regulator and a Buck Regulator
VIN = 2.8V
TO 4.2V
C1
10μF
RUN/SS2
VIN
MODE/SYNC
POWER-ON
RESET
POR
LTC3407A-2
L2
10μH
D1
VOUT2 = 3.3V
AT 200mA
R5
100k
RUN/SS1
SW2
L1
2.2μH
C4, 22pF
M1
+
C6
47μF
C3
10μF
R4
887k R3
196k
VFB1
VFB2
C1, C2, C3: TAIYO YUDEN JMK316BJ106ML
C6: SANYO 6TPB47M
D1: PHILIPS PMEG2010
R2
R1 887k
442k
GND
C2
10μF
L1: MURATA LQH32CN2R2M33
L2: TOKO A914BYW-100M (D52LC SERIES)
M1: SILICONIX Si2302
Efficiency vs Load Current
3407A TA05
Efficiency vs Load Current
90
100
80
90
3.6V
2.8V
4.2V
3.6V
70
EFFICIENCY (%)
60
2.8V
80
4.2V
70
EFFICIENCY (%)
VOUT1 = 1.8V
AT 800mA
SW1
50
40
30
60
50
40
30
20
20
VOUT = 3.3V
Burst Mode OPERATION
NO LOAD ON OTHER CHANNEL
10
0
1
10
100
LOAD CURRENT (mA)
VOUT = 1.8V
Burst Mode OPERATION
NO LOAD ON OTHER CHANNEL
10
0
1000
1
3407A2 TA05a
10
100
LOAD CURRENT (mA)
1000
3407A2 TA05b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3405/LTC3405A
300mA (IOUT), 1.5MHz,
Synchronous Step-Down DC/DC Converter
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20μA,
ISD <1μA, ThinSOT Package
LTC3406/LTC3406B
600mA (IOUT), 1.5MHz,
Synchronous Step-Down DC/DC Converter
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20μA,
ISD <1μA, ThinSOT Package
LTC3407/LTC3407-2
600mA/800mA (IOUT), 1.5MHz/2.25MHz,
LTC3407-3/LTC3407-4/ Dual Synchronous Step-Down DC/DC Converter
LTC3407A
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA,
ISD <1μA, MS10E Package, DFN Package
LTC3410/LTC3410B
300mA (IOUT), 2.25MHz,
Synchronous Step-Down DC/DC Converter in SC70
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26μA,
ISD <1μA, SC70 Package
LTC3411
1.25A (IOUT), 4MHz,
Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60μA,
ISD <1μA, MSOP-10 Package
LTC3412/LTC3412A
2.5A (IOUT), 4MHz,
Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60μA,
ISD <1μA, TSSOP-16E Package
LTC3414
4A (IOUT), 4MHz,
Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64μA,
ISD <1μA, TSSOP-28E Package
LTC3440/LTC3441
600mA/1.2A (IOUT), 2MHz/1MHz,
Synchronous Buck-Boost DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V, IQ = 25μA,
ISD <1μA, MSOP-10 Package/DFN Package
LTC3548/
LTC3548-1/LTC3548-2
400mA/800mA (IOUT), 2.25MHz,
Dual Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA,
ISD <1μA, MS10E Package/DFN Package
3407a2f
16
Linear Technology Corporation
LT 0907 • PRINTED IN USA
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