DAC902 902 DAC DAC 902 SBAS094B – MAY 2002 12-Bit, 165MSPS DIGITAL-TO-ANALOG CONVERTER TM FEATURES APPLICATIONS ● SINGLE +5V OR +3V OPERATION ● COMMUNICATION TRANSMIT CHANNELS: WLL, Cellular Base Station Digital Microwave Links Cable Modems ● HIGH SFDR: 5MHz Output at 100MSPS: 67dBc ● LOW GLITCH: 3pV-s ● LOW POWER: 170mW at +5V ● WAVEFORM GENERATION: Direct Digital Synthesis (DDS) Arbitrary Waveform Generation (ARB) ● INTERNAL REFERENCE: Optional Ext. Reference Adjustable Full-Scale Range Multiplying Option ● MEDICAL/ULTRASOUND ● HIGH-SPEED INSTRUMENTATION AND CONTROL ● VIDEO, DIGITAL TV DESCRIPTION For noncontinuous operation of the DAC902, a power-down mode results in only 45mW of standby power. The DAC902 is a high-speed, Digital-to-Analog Converter (DAC) offering a 12-bit resolution option within the SpeedPlus Family of high-performance converters. Featuring pin compatibility among family members, the DAC908, DAC900, and DAC904 provide a component selection option to an 8-, 10-, and 14-bit resolution, respectively. All models within this family of DACs support update rates in excess of 165MSPS with excellent dynamic performance, and are especially suited to fulfill the demands of a variety of applications. The DAC902 comes with an integrated 1.24V bandgap reference and edge-triggered input latches, offering a complete converter solution. Both +3V and +5V CMOS logic families can be interfaced to the DAC902. The advanced segmentation architecture of the DAC902 is optimized to provide a high Spurious-Free Dynamic Range (SFDR) for single-tone, as well as for multi-tone signals—essential when used for the transmit signal path of communication systems. The DAC902 has a high impedance (200kΩ) current output with a nominal range of 20mA and an output compliance of up to 1.25V. The differential outputs allow for both a differential or singleended analog signal interface. The close matching of the current outputs ensures superior dynamic performance in the differential configuration, which can be implemented with a transformer. Utilizing a small geometry CMOS process, the monolithic DAC902 can be operated on a wide, single-supply range of +2.7V to +5.5V. Its low power consumption allows for use in portable and batteryoperated systems. Further optimization can be realized by lowering the output current with the adjustable full-scale option. The reference structure of the DAC902 allows for additional flexibility by utilizing the on-chip reference, or applying an external reference. The full-scale output current can be adjusted over a span of 2mA to 20mA, with one external resistor, while maintaining the specified dynamic performance. The DAC902 is available in the SO-28 and TSSOP-28 packages. +VA +VD BW DAC902 FSA Current Sources REFIN IOUT LSB Switches IOUT BYP Segmented Switches INT/EXT Latches PD +1.24V Ref. 12-Bit Data Input AGND CLK D11...D0 DGND Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. Copyright © 2002, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com ABSOLUTE MAXIMUM RATINGS ELECTROSTATIC DISCHARGE SENSITIVITY +VA to AGND ........................................................................ –0.3V to +6V +VD to DGND ........................................................................ –0.3V to +6V AGND to DGND ................................................................. –0.3V to +0.3V +VA to +VD ............................................................................... –6V to +6V CLK, PD to DGND ..................................................... –0.3V to VD + 0.3V D0-D11 to DGND ....................................................... –0.3V to VD + 0.3V IOUT, IOUT to AGND ........................................................ –1V to VA + 0.3V BW, BYP to AGND ..................................................... –0.3V to VA + 0.3V REFIN, FSA to AGND ................................................. –0.3V to VA + 0.3V INT/EXT to AGND ...................................................... –0.3V to VA + 0.3V Junction Temperature .................................................................... +150°C Case Temperature ......................................................................... +100°C Storage Temperature .................................................................... +125°C This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER DAC902U SO-28 217 –40°C to +85°C DAC902U DAC902U Rails " " " " DAC902U/1K Tape and Reel TSSOP-28 360 –40°C to +85°C DAC902E DAC902E Rails " " " " DAC902E/2K5 Tape and Reel " DAC902E " SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER(1) TRANSPORT MEDIA NOTE: (1) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of “DAC902E/2K5” will get a single 2500-piece Tape and Reel. DEMO BOARD ORDERING INFORMATION PRODUCT DEMO BOARD ORDERING NUMBER DAC902U DAC902E DEM-DAC90xU DEM-DAC902E COMMENT Populated evaluation board without the DAC. Order sample of desired DAC90x model separately. Populated evaluation board including the DAC902E. ELECTRICAL CHARACTERISTICS At TA = full specified temperature range, +VA = +5V, +VD = +5V, differential transformer coupled output, 50Ω doubly terminated, unless otherwise specified. DAC902U/E PARAMETER CONDITIONS MIN MAX UNITS 12 Bits 125 165 –40 165 200 +85 MSPS MSPS °C –1.75 –2.5 ±0.5 ±1.0 +1.75 +2.5 LSB LSB 71 77 75 68 67 61 61 57 60 53 dBc dBc dBc dBc dBc dBc dBc dBc dBc 80 80 dBc dBc –74 –75 dBc dBc 64 dBc RESOLUTION OUTPUT UPDATE RATE Output Update Rate (fCLOCK) Full Specified Temperature Range, Operating STATIC ACCURACY(1) Differential Nonlinearity (DNL) Integral Nonlinearity (INL) DYNAMIC PERFORMANCE Spurious-Free Dynamic Range (SFDR) fOUT = 1MHz, fCLOCK = 25MSPS fOUT = 2.1MHz, fCLOCK = 50MSPS fOUT = 5.04MHz, fCLOCK = 50MSPS fOUT = 5.04MHz, fCLOCK = 100MSPS fOUT = 20.2MHz, fCLOCK = 100MSPS fOUT = 25.3MHz, fCLOCK = 125MSPS fOUT = 41.5MHz, fCLOCK = 125MSPS fOUT = 27.4MHz, fCLOCK = 165MSPS fOUT = 54.8MHz, fCLOCK = 165MSPS Spurious-Free Dynamic Range within a Window fOUT = 5.04MHz, fCLOCK = 50MSPS fOUT = 5.04MHz, fCLOCK = 100MSPS Total Harmonic Distortion (THD) fOUT = 2.1MHz, fCLOCK = 50MSPS fOUT = 2.1MHz, fCLOCK = 125MSPS Two Tone fOUT1 = 13.5MHz, fOUT2 = 14.5MHz, fCLOCK = 100MSPS 2 2.7V to 3.3V 4.5V to 5.5V Ambient, TA fCLOCK TA = +25°C = 25MSPS, fOUT = 1.0MHz TYP TA = +25°C To Nyquist 2MHz Span 4MHz Span DAC902 SBAS094B ELECTRICAL CHARACTERISTICS (Cont.) At TA = full specified temperature range, +VA = +5V, +VD = +5V, differential transformer coupled output, 50Ω doubly terminated, unless otherwise specified. DAC902U/E PARAMETER DYNAMIC PERFORMANCE (Cont.) Output Settling Time(2) Output Rise Time(2) Output Fall Time(2) Glitch Impulse DC-ACCURACY Full-Scale Output Range(3)(FSR) Output Compliance Range Gain Error Gain Error Gain Drift Offset Error Offset Drift Power-Supply Rejection, +VA Power-Supply Rejection, +VD Output Noise Output Resistance Output Capacitance CONDITIONS to 0.1% 10% to 90% 10% to 90% All Bits High, IOUT With Internal Reference With External Reference With Internal Reference With Internal Reference With Internal Reference POWER SUPPLY Supply Voltages +VA +VD Supply Current(6) IVA IVA, Power-Down Mode IVD Power Dissipation Power Dissipation, Power-Down Mode Thermal Resistance, θJA SO-28 TSSOP-28 TYP MAX 30 2 2 3 2.0 –1.0 –10 –10 ±1 ±2 ±120 –0.025 IOUT = 20mA, RLOAD = 50Ω 20.0 +1.25 +10 +10 +0.025 +0.2 +0.025 50 200 12 IOUT, IOUT to Ground +1.24 ±5 ±50 10 1 0.1 1.25 1.3 +VD +VD +VD +VD +VD +VD = = = = = = +5V +5V +3V +3V +5V +5V 3.5 2 +2.7 +2.7 +5V, IOUT = 20mA +3V, IOUT = 2mA Straight Binary Rising Edge of Clock 5 0 3 0 ±20 ±20 5 UNITS ns ns ns pV-s ±0.1 –0.2 –0.025 REFERENCE Reference Voltage Reference Tolerance Reference Voltage Drift Reference Output Current Reference Input Resistance Reference Input Compliance Range Reference Small-Signal Bandwidth(4) DIGITAL INPUTS Logic Coding Latch Command Logic High Voltage, VIH Logic Low Voltage, VIL Logic High Voltage, VIH Logic Low Voltage, VIL Logic High Current, IIH(5) Logic Low Current, IIL Input Capacitance MIN mA V %FSR %FSR ppmFSR/°C %FSR ppmFSR/°C %FSR/V %FSR/V pA/√Hz kΩ pF V % ppmFSR/°C µA MΩ V MHz 0.8 V V V V µA µA pF +5 +5 +5.5 +5.5 V V 24 1.1 8 170 50 45 30 2 15 230 mA mA mA mW mW mW 75 50 1.2 °C/W °C/W NOTES: (1) At output IOUT, while driving a virtual ground. (2) Measured single-ended into 50Ω Load. (3) Nominal full-scale output current is 32 • IREF; see Application Section for details. (4) Reference bandwidth depends on size of external capacitor at the BW pin and signal level. (5) Typically 45µA for the PD pin, which has an internal pull-down resistor. (6) Measured at fCLOCK = 50MSPS and fOUT = 1.0MHz. DAC902 SBAS094B 3 PIN CONFIGURATION PIN DESCRIPTIONS Top View SO, TSSOP Bit 1 1 28 CLK Bit 2 2 27 +VD Bit 3 3 26 DGND Bit 4 4 25 NC Bit 5 5 24 +VA Bit 6 6 23 BYP Bit 7 7 22 IOUT Bit 8 8 21 IOUT Bit 9 9 20 AGND Bit 10 10 19 BW Bit 11 11 18 FSA Bit 12 12 17 REFIN NC 13 16 INT/EXT NC 14 15 PD DAC902 PIN DESIGNATOR 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 Bit 1 Bit 2 Bit 3 Bit 4 Bit 5 Bit 6 Bit 7 Bit 8 Bit 9 Bit 10 Bit 11 Bit 12 NC NC PD 16 INT/EXT 17 REFIN 18 19 FSA BW 20 21 22 23 24 25 26 27 28 AGND IOUT IOUT BYP +VA NC DGND +VD CLK DESCRIPTION Data Bit 1 (D11), MSB Data Bit 2 (D10) Data Bit 3 (D9) Data Bit 4 (D8) Data Bit 5 (D7) Data Bit 6 (D6) Data Bit 7 (D5) Data Bit 8 (D4) Data Bit 9 (D3) Data Bit 10 (D2) Data Bit 11 (D1) Data Bit 12 (D0), LSB No Connection No Connection Power Down, Control Input; Active HIGH. Contains internal pull-down circuit; may be left unconnected if not used. Reference Select Pin; Internal ( = 0) or External ( = 1) Reference Operation. Reference Input/Ouput. See Applications section for further details. Full-Scale Output Adjust Bandwidth/Noise Reduction Pin: Bypass with 0.1µF to +VA for Optimum Performance. Analog Ground Complementary DAC Current Output DAC Current Output Bypass Node: Use 0.1µF to AGND Analog Supply Voltage, 2.7V to 5.5V No Connection Digital Ground Digital Supply Voltage, 2.7V to 5.5V Clock Input TYPICAL CONNECTION CIRCUIT +5V +5V 0.1µF +VA +VD BW DAC902 IOUT LSB Switches FSA Current Sources REFIN RSET 0.1µF 1:1 IOUT BYP Segmented MSB Switches 0.1µF 50Ω 20pF 50Ω 20pF INT/EXT PD Latches +1.24V Ref. 12-Bit Data Input AGND 4 CLK D11.......D0 DGND DAC902 SBAS094B TIMING DIAGRAM t2 t1 CLOCK tS D13 D0 Data Changes tH Stable Valid Data Data Changes tPD tSET Iout or Iout SYMBOL t1 t2 tS tH tPD tSET DAC902 SBAS094B DESCRIPTION Clock Pulse HIGH Time Clock Pulse LOW Time Data Setup Time Data Hold Time Propagation Delay Time Output Settling Time to 0.1% MIN TYP 3.0 3.0 1.0 1.5 1 30.0 MAX UNITS ns ns ns ns ns ns 5 TYPICAL CHARACTERISTICS: VD = VA = +5V At TA = +25°C, differential transformer coupled output, 50Ω doubly terminated, and SFDR up to Nyquist, unless otherwise noted. TYPICAL DNL 2.0 2.0 1.5 1.5 1.0 1.0 0 DAC Code 85 80 80 –6dBFS 75 0dBFS 65 75 –6dBFS 70 65 0dBFS 60 60 55 0 2.0 4.0 6.0 8.0 Frequency (MHz) 10.0 12.0 0 5.0 SFDR vs fOUT AT 100MSPS 10.0 15.0 Frequency (MHz) 20.0 25.0 SFDR vs fOUT AT 125MSPS 85 85 80 80 75 75 SFDR (dBc) SFDR (dBc) 4000 4096 SFDR vs fOUT AT 50MSPS 85 SFDR (dBc) SFDR (dBc) SFDR vs fOUT AT 25MSPS 70 –6dBFS 65 60 55 70 –6dBFS 65 60 55 0dBFS 0dBFS 50 50 45 45 0 6 3500 DAC Code 90 70 3000 0 4000 4096 3500 1000 3000 –2.5 2500 –2.0 –2.5 2000 –1.5 –2.0 1500 –1.0 –1.5 500 –1.0 2500 –0.5 2000 –0.5 0.5 1500 0 1000 0.5 500 Error (LSBs) 2.5 0 Error (LSBs) TYPICAL INL 2.5 10.0 20.0 30.0 Frequency (MHz) 40.0 50.0 0 10.0 20.0 30.0 40.0 Frequency (MHz) 50.0 60.0 DAC902 SBAS094B TYPICAL CHARACTERISTICS: VD = VA = +5V (Cont.) At TA = +25°C, differential transformer coupled output, 50Ω doubly terminated, and SFDR up to Nyquist, unless otherwise noted. SFDR vs fOUT AT 165MSPS SFDR vs fOUT AT 200MSPS 80 80 75 75 70 70 SFDR (dBc) SFDR (dBc) –6dBFS 65 60 55 65 –6dBFS 60 55 0dBFS 0dBFS 50 50 45 45 40 40 0 10.0 20.0 30.0 40.0 50.0 Frequency (MHz) 60.0 70.0 80.0 0 DIFFERENTIAL vs SINGLE-ENDED SFDR vs fOUT AT 100MSPS 10.0 20.0 30.0 40.0 50.0 60.0 70.0 80.0 Frequency (MHz) 90.0 SFDR vs IOUTFS and fOUT AT 100MSPS, 0dBFS 85 80 2.1MHz 80 X 70 IOUT (–6dBFS) SFDR (dBc) 75 SFDR (dBc) 75 X 70 Diff (–6dBFS) 65 X X X 60 Diff (0dBFS) 20.2MHz 60 10.1MHz 55 40.4MHz X X X 55 50 IOUT (0dBFS) 50 65 X X 2 5 X 45 45 40 0 10.0 20.0 30.0 Frequency (MHz) 40.0 50.0 10 SFDR vs TEMPERATURE AT 100MSPS, 0dBFS THD vs fCLOCK AT fOUT = 2.1MHz 85 –70 2HD 80 –75 2.1MHz –80 SFDR (dBc) 75 THD (dBc) 20 IOUTFS (mA) 3HD –85 X X X 4HD X –90 70 10.1MHz 65 60 55 –95 50 –100 0 DAC902 SBAS094B 25 50 100 fCLOCK (MSPS) 125 150 40.4MHz X 45 –40 X X –20 0 X X 25 50 Temperature (°C) X X 70 85 7 TYPICAL CHARACTERISTICS: VD = VA = +5V (Cont.) At TA = +25°C, differential transformer coupled output, 50Ω doubly terminated, and SFDR up to Nyquist, unless otherwise noted. SINGLE-TONE OUTPUT SPECTRUM DUAL-TONE OUTPUT SPECTRUM 0 0 –10 –10 fCLOCK = 100MSPS fOUT = 2.1MHz SFDR = 74dBc Amplitude = 0dBFS –30 –40 fCLOCK = 100MSPS fOUT1 = 13.5MHz fOUT2 = 14.5MHz SFDR = 64dBc Amplitude = 0dBFS –20 Magnitude (dBm) Magnitude (dBm) –20 –50 –60 –70 –80 –30 –40 –50 –60 –70 –80 –90 –90 –100 –100 0 5 10 15 20 25 30 35 40 45 50 0 5 10 Frequency (MHz) 15 20 25 30 35 40 45 50 Frequency (MHz) FOUR-TONE OUTPUT SPECTRUM 0 –10 fCLOCK = 50MSPS fOUT1 = 6.25MHz fOUT2 = 6.75MHz fOUT3 = 7.25MHz fOUT4 = 7.75MHz SFDR = 66dBc Amplitude = 0dBFS Magnitude (dBm) –20 –30 –40 –50 –60 –70 –80 –90 –100 0 5 10 15 20 25 Frequency (MHz) 8 DAC902 SBAS094B TYPICAL CHARACTERISTICS: VD = VA = +3V At TA = +25°C, differential transformer coupled output, 50Ω doubly terminated, and SFDR up to Nyquist, unless otherwise noted. SFDR vs fOUT AT 25MSPS SFDR vs fOUT AT 50MSPS 85 85 80 80 SFDR (dBc) SFDR (dBc) –6dBFS 75 70 0dBFS 65 75 –6dBFS 70 65 60 60 55 55 0dBFS 0 2.0 4.0 6.0 8.0 Frequency (MHz) 10.0 12.0 0 5.0 10.0 15.0 Frequency (MHz) SFDR vs fOUT AT 100MSPS 20.0 25.0 SFDR vs fOUT AT 125MSPS 85 85 80 80 75 75 SFDR (dBc) SFDR (dBc) –6dBFS 70 –6dBFS 65 60 70 65 60 0dBFS 55 55 0dBFS 50 50 45 45 0 10.0 20.0 30.0 Frequency (MHz) 40.0 50.0 0 10.0 20.0 30.0 40.0 Frequency (MHz) 50.0 60.0 DIFFERENTIAL vs SINGLE-ENDED SFDR vs fOUT AT 100MSPS SFDR vs fOUT AT 165MSPS 80 85 75 80 70 75 Diff (–6dBFS) –6dBFS SFDR (dBc) SFDR (dBc) IOUT (–6dBFS) 65 60 55 70 Diff (0dBFS) 65 60 55 50 0dBFS 45 50 40 45 0 DAC902 SBAS094B 10.0 20.0 30.0 40.0 50.0 Frequency (MHz) 60.0 70.0 80.0 IOUT (0dBFS) 0 10.0 20.0 30.0 Frequency (MHz) 40.0 50.0 9 TYPICAL CHARACTERISTICS: VD = VA = +3V (Cont.) At TA = +25°C, differential transformer coupled output, 50Ω doubly terminated, and SFDR up to Nyquist, unless otherwise noted. SFDR vs IOUTFS and fOUT AT 100MSPS, 0dBFS THD vs fCLOCK AT fOUT = 2.1MHz –70 80 2.1MHz 2HD 75 –75 –80 10.1MHz 65 60 THD (dBc) SFDR (dBc) 70 20.2MHz 55 –85 –90 4HD X 50 X 45 3HD 40.4MHz X X –95 –100 40 2 5 10 0 20 25 50 100 fCLOCK (MSPS) IOUTFS (mA) SFDR vs TEMPERATURE AT 100MSPS, 0dBFS 0 –10 80 2.1MHz Magnitude (dBm) SFDR (dBc) 70 10.1MHz 65 60 55 40.4MHz X 45 –40 –30 –40 –50 –60 –70 –80 X X X X X X fCLOCK = 100MSPS fOUT = 2.1MHz SFDR = 76dBc Amplitude = 0dBFS –20 75 –90 –100 –20 0 25 50 Temperature (°C) 70 85 0 DUAL-TONE OUTPUT SPECTRUM 10 15 20 25 30 35 40 45 50 FOUR-TONE OUTPUT SPECTRUM 0 –10 –10 fCLOCK = 100MSPS fOUT1 = 13.5MHz fOUT2 = 14.5MHz SFDR = 68dBc Amplitude = 0dBFS –30 –40 –50 fCLOCK = 50MSPS fOUT1 = 6.25MHz fOUT2 = 6.75MHz fOUT3 = 7.25MHz fOUT4 = 7.75MHz SFDR = 66dBc Amplitude = 0dBFS –20 Magnitude (dBm) –20 Magnitude (dBm) 5 Frequency (MHz) 0 –60 –70 –80 –30 –40 –50 –60 –70 –80 –90 –90 –100 –100 0 5 10 15 20 25 30 Frequency (MHz) 10 150 SINGLE-TONE OUTPUT SPECTRUM 85 50 125 35 40 45 50 0 5 10 15 20 25 Frequency (MHz) DAC902 SBAS094B APPLICATION INFORMATION DAC TRANSFER FUNCTION THEORY OF OPERATION The total output current, IOUTFS, of the DAC902 is the summation of the two complementary output currents: The architecture of the DAC902 uses the current steering technique to enable fast switching and a high update rate. The core element within the monolithic DAC is an array of segmented current sources that are designed to deliver a fullscale output current of up to 20mA, as shown in Figure 1. An internal decoder addresses the differential current switches each time the DAC is updated and a corresponding output current is formed by steering all currents to either output summing node, IOUT or IOUT. The complementary outputs deliver a differential output signal that improves the dynamic performance through reduction of even-order harmonics, common-mode signals (noise), and double the peak-to-peak output signal swing by a factor of two, compared to singleended operation. IOUTFS = IOUT + IOUT (1) The individual output currents depend on the DAC code and can be expressed as: IOUT = IOUTFS • (Code/4096) (2) IOUT = IOUTFS • (4095 – Code/4096) (3) where ‘Code’ is the decimal representation of the DAC data input word. Additionally, IOUTFS is a function of the reference current IREF, which is determined by the reference voltage and the external setting resistor, RSET. The segmented architecture results in a significant reduction of the glitch energy, improves the dynamic performance (SFDR), and DNL. The current outputs maintain a very high output impedance of greater than 200kΩ. IOUTFS = 32 • IREF = 32 • VREF /RSET The full-scale output current is determined by the ratio of the internal reference voltage (1.24V) and an external resistor, RSET. The resulting IREF is internally multiplied by a factor of 32 to produce an effective DAC output current that can range from 2mA to 20mA, depending on the value of RSET. (4) In most cases the complementary outputs will drive resistive loads or a terminated transformer. A signal voltage will develop at each output according to: The DAC902 is split into a digital and an analog portion, each of which is powered through its own supply pin. The digital section includes edge-triggered input latches and the decoder logic, while the analog section comprises the current source array with its associated switches, and the reference circuitry. VOUT = IOUT • RLOAD (5) VOUT = IOUT • RLOAD (6) +3V to +5V Digital +3V to +5V Analog 0.1µF Bandwidth Control +VA DAC902 RSET 2kΩ BW +VD IOUT Full-Scale Adjust Resistor FSA Ref Control Amp Ref Input REFIN 400pF PMOS Current Source Array 0.1µF LSB Switches 1:1 VOUT IOUT Segmented MSB Switches 50Ω 0.1µF 20pF 50Ω 20pF BYP INT/EXT Ref Buffer Latches and Switch Decoder Logic PD Power Down (internal pull-down) +1.24V Ref AGND Analog Ground 12-Bit Data Input CLK Clock Input DGND D11...D0 Digital Ground NOTE: Supply bypassing not shown. FIGURE 1. Functional Block Diagram of the DAC902. DAC902 SBAS094B 11 The value of the load resistance is limited by the output compliance specification of the DAC902. To maintain specified linearity performance, the voltage for IOUT and IOUT should not exceed the maximum allowable compliance range. The two single-ended output voltages can be combined to find the total differential output swing: VOUTDIFF = VOUT – VOUT = (2 • Code – 4095) • I OUTFS • R LOAD (7) 4096 ANALOG OUTPUTS The DAC902 provides two complementary current outputs, IOUT and IOUT. The simplified circuit of the analog output stage representing the differential topology is shown in Figure 2. The output impedance of 200kΩ || 12pF for IOUT and IOUT results from the parallel combination of the differential switches, along with the current sources and associated parasitic capacitances. IOUT and IOUT. Furthermore, using the differential output configuration in combination with a transformer will be instrumental for achieving excellent distortion performance. Common-mode errors, such as even-order harmonics or noise, can be substantially reduced. This is particularly the case with high output frequencies and/or output amplitudes below full-scale. For those applications requiring the optimum distortion and noise performance, it is recommended to select a full-scale output of 20mA. A lower full-scale range down to 2mA may be considered for applications that require a low power consumption, but can tolerate a reduced performance level. INPUT CODE (D11 - D0) IOUT IOUT 1111 1111 1111 20mA 0mA 1000 0000 0000 10mA 10mA 0000 0000 0000 0mA 20mA TABLE I. Input Coding vs Analog Output Current. OUTPUT CONFIGURATIONS +VA DAC902 IOUT IOUT RL RL The current output of the DAC902 allows for a variety of configurations, some of which are illustrated below. As mentioned previously, utilizing the converter’s differential outputs will yield the best dynamic performance. Such a differential output circuit may consist of an RF transformer or a differential amplifier configuration. The transformer configuration is ideal for most applications with ac coupling, while op amps will be suitable for a DC-coupled configuration. The single-ended configuration may be considered for applications requiring a unipolar output voltage. Connecting a resistor from either one of the outputs to ground will convert the output current into a ground-referenced voltage signal. To improve on the DC linearity, an I-to-V converter can be used instead. This will result in a negative signal excursion and, therefore, requires a dual supply amplifier. FIGURE 2. Equivalent Analog Output. DIFFERENTIAL WITH TRANSFORMER The signal voltage swing that may develop at the two outputs, IOUT and IOUT, is limited by a negative and positive compliance. The negative limit of –1V is given by the breakdown voltage of the CMOS process, and exceeding it will compromise the reliability of the DAC902, or even cause permanent damage. With the full-scale output set to 20mA, the positive compliance equals 1.25V, operating with +VD = 5V. Note that the compliance range decreases to about 1V for a selected output current of IOUTFS = 2mA. Care should be taken that the configuration of DAC902 does not exceed the compliance range to avoid degradation of the distortion performance and integral linearity. Best distortion performance is typically achieved with the maximum full-scale output signal limited to approximately 0.5V. This is the case for a 50Ω doubly-terminated load and a 20mA full-scale output current. A variety of loads can be adapted to the output of the DAC902 by selecting a suitable transformer while maintaining optimum voltage levels at 12 Using an RF transformer provides a convenient way of converting the differential output signal into a single-ended signal while achieving excellent dynamic performance (see Figure 3). The appropriate transformer should be carefully selected based on the output frequency spectrum and impedance requirements. The differential transformer configuration has the benefit of significantly reducing common-mode signals, thus improving the dynamic performance over a wide range of frequencies. Furthermore, by selecting a suitable impedance ratio (winding ratio), the transformer can be used to provide optimum impedance matching while controlling the compliance voltage for the converter outputs. The model shown in Figure 3 has a 1:1 ratio and may be used to interface the DAC902 to a 50Ω load. This results in a 25Ω load for each of the outputs, IOUT and IOUT. The output signals are ac coupled and inherently isolated because of its magnetic coupling. DAC902 SBAS094B As shown in Figure 3, the transformer’s center tap is connected to ground. This forces the voltage swing on IOUT and IOUT to be centered at 0V. In this case the two resistors, RS, may be replaced with one, RDIFF, or omitted altogether. This approach should only be used if all components are close to each other, and if the VSWR is not important. A complete power transfer from the DAC output to the load can be realized, but the output compliance range should be observed. Alternatively, if the center tap is not connected, the signal swing will be centered at RS • IOUTFS/2. However, in this case, the two resistors (RS) must be used to enable the necessary DC-current flow for both outputs. ADT1-1WT (Mini-Circuits) 1:1 IOUT DAC902 RS 50Ω Optional RDIFF RL IOUT RS 50Ω The OPA680 is configured for a gain of two. Therefore, operating the DAC902 with a 20mA full-scale output will produce a voltage output of ±1V. This requires the amplifier to operate off of a dual power supply (±5V). The tolerance of the resistors typically sets the limit for the achievable common-mode rejection. An improvement can be obtained by fine tuning resistor R4. This configuration typically delivers a lower level of ac performance than the previously discussed transformer solution because the amplifier introduces another source of distortion. Suitable amplifiers should be selected based on their slew-rate, harmonic distortion, and output swing capabilities. High-speed amplifiers like the OPA680 or OPA687 may be considered. The ac performance of this circuit may be improved by adding a small capacitor, CDIFF, between the outputs IOUT and IOUT (as shown in Figure 4). This will introduce a real pole to create a low-pass filter in order to slew-limiting the DACs fast output signal steps that otherwise could drive the amplifier into slew-limitations or into an overload condition; both would cause excessive distortion. The difference amplifier can easily be modified to add a level shift for applications requiring the single-ended output voltage to be unipolar, i.e., swing between 0V and +2V. DUAL TRANSIMPEDANCE OUTPUT CONFIGURATION FIGURE 3. Differential Output Configuration Using an RF Transformer. DIFFERENTIAL CONFIGURATION USING AN OP AMP If the application requires a DC-coupled output, a difference amplifier may be considered, as shown in Figure 4. Four external resistors are needed to configure the voltage-feedback op amp OPA680 as a difference amplifier performing the differential to single-ended conversion. Under the shown configuration, the DAC902 generates a differential output signal of 0.5Vp-p at the load resistors, RL. The resistor values shown were selected to result in a symmetric 25Ω loading for each of the current outputs since the input impedance of the difference amplifier is in parallel to resistors RL, and should be considered. The circuit example of Figure 5 shows the signal output currents connected into the summing junction of the OPA2680, which is set up as a transimpedance stage, or I-to-V converter. With this circuit, the DAC’s output will be kept at a virtual ground, minimizing the effects of output impedance variations, which results in the best DC linearity (INL). However, as mentioned previously, the amplifier may be driven into slew-rate limitations, and produce unwanted distortion. This may occur especially at high DAC update rates. +5V 50Ω 1/2 OPA2680 RF1 DAC902 R2 402Ω IOUT R1 200Ω –VOUT = IOUT • RF CD1 IOUT CF1 RF2 DAC902 IOUT OPA680 COPT RL 26.1Ω R3 200Ω RL 28.7Ω VOUT IOUT CD2 CF2 –5V +5V R4 402Ω 1/2 OPA2680 –VOUT = IOUT • RF 50Ω FIGURE 4. Difference Amplifier Provides Differential to Single-Ended Conversion and DC-Coupling. –5V FIGURE 5. Dual, Voltage-Feedback Amplifier OPA2680 Forms Differential Transimpedance Amplifier. DAC902 SBAS094B 13 The DC gain for this circuit is equal to feedback resistor RF. At high frequencies, the DAC output impedance (CD1, CD2) will produce a zero in the noise gain for the OPA2680 that may cause peaking in the closed-loop frequency response. CF is added across RF to compensate for this noise-gain peaking. To achieve a flat transimpedance frequency response, the pole in each feedback network should be set to: 1 GBP = 2 πR F C F 4 πR F C D The DAC902 has an on-chip reference circuit that comprises a 1.24V bandgap reference and a control amplifier. Grounding pin 16, INT/EXT, enables the internal reference operation. The full-scale output current, IOUTFS, of the DAC902 is determined by the reference voltage, VREF, and the value of resistor RSET. IOUTFS can be calculated by: IOUTFS = 32 • IREF = 32 • VREF / RSET (8) with GBP = Gain Bandwidth Product of OPA which will give a corner frequency f-3dB of approximately: f−3dB = INTERNAL REFERENCE OPERATION GBP 2πR F C D (10) As shown in Figure 7, the external resistor RSET connects to the FSA pin (Full-Scale Adjust). The reference control amplifier operates as a V-to-I converter producing a reference current, IREF, which is determined by the ratio of VREF and RSET, as shown in Equation 10. The full-scale output current, IOUTFS, results from multiplying IREF by a fixed factor of 32. (9) CCOMPEXT +5V 0.1µF The full-scale output voltage is simply defined by the product of IOUTFS • RF, and has a negative unipolar excursion. To improve on the ac performance of this circuit, adjustment of RF and/or IOUTFS should be considered. Further extensions of this application example may include adding a differential filter at the OPA2680’s output followed by a transformer, in order to convert to a single-ended signal. IREF = FSA RSET 2kΩ Using a single load resistor connected to the one of the DAC outputs, a simple current-to-voltage conversion can be accomplished. The circuit in Figure 6 shows a 50Ω resistor connected to IOUT, providing the termination of the further connected 50Ω cable. Therefore, with a nominal output current of 20mA, the DAC produces a total signal swing of 0V to 0.5V into the 25Ω load. +VA VREF RSET REFIN SINGLE-ENDED CONFIGURATION BW DAC902 Ref Control Amp Current Sources CCOMP 400pF 0.1µF INT/EXT +1.24V Ref. FIGURE 7. Internal Reference Configuration. IOUTFS = 20mA VOUT = 0V to +0.5V IOUT DAC902 50Ω IOUT 50Ω 25Ω FIGURE 6. Driving a Doubly-Terminated 50Ω Cable Directly. Different load resistor values may be selected as long as the output compliance range is not exceeded. Additionally, the output current, IOUTFS, and the load resistor may be mutually adjusted to provide the desired output signal swing and performance. 14 Using the internal reference, a 2kΩ resistor value results in a 20mA full-scale output. Resistors with a tolerance of 1% or better should be considered. Selecting higher values, the converter output can be adjusted from 20mA down to 2mA. Operating the DAC902 at lower than 20mA output currents may be desirable for reasons of reducing the total power consumption, improving the distortion performance, or observing the output compliance voltage limitations for a given load condition. It is recommended to bypass the REFIN pin with a ceramic chip capacitor of 0.1µF or more. The control amplifier is internally compensated, and its small signal bandwidth is approximately 3MHz. To improve the ac performance, an additional capacitor (CCOMPEXT) should be applied between the BW pin and the analog supply, +VA, as shown in Figure 7. Using a 0.1µF capacitor, the small-signal bandwidth and output impedance of the control amplifier is further diminished, reducing the noise that is fed into the current source array. This also helps shunting feedthrough signals more effectively, and improving the noise performance of the DAC902. DAC902 SBAS094B EXTERNAL REFERENCE OPERATION The internal reference can be disabled by applying a logic HIGH (+VA) to pin INT/EXT. An external reference voltage can then be driven into the REFIN pin, which in this case functions as an input, as shown in Figure 8. The use of an external reference may be considered for applications that require higher accuracy and drift performance, or to add the ability of dynamic gain control. While a 0.1µF capacitor is recommended to be used with the internal reference, it is optional for the external reference operation. The reference input, REFIN, has a high input impedance (1MΩ) and can easily be driven by various sources. Note that the voltage range of the external reference should stay within the compliance range of the reference input (0.1V to 1.25V). DIGITAL INPUTS The digital inputs, D0 (LSB) through D11 (MSB) of the DAC902 accepts standard-positive binary coding. The digital input word is latched into a master-slave latch with the rising edge of the clock. The DAC output becomes updated with the following falling clock edge (refer to the specification table and timing diagram for details). The best performance will be achieved with a 50% clock duty cycle, however, the duty cycle may vary as long as the timing specifications are met. Additionally, the setup and hold times may be chosen within their specified limits. All digital inputs are CMOS compatible. The logic thresholds depend on the applied digital supply voltage such that they are set to approximately half the supply voltage; Vth = +VD/2 (±20% tolerance). The DAC902 is designed to operate over a supply range of 2.7V to 5.5V. POWER-DOWN MODE The DAC902 features a power-down function that can be used to reduce the supply current to less than 9mA over the specified supply range of 2.7V to 5.5V. Applying a logic HIGH to the PD pin will initiate the power-down mode, while a logic LOW enables normal operation. When left unconnected, an internal active pull-down circuit will enable the normal operation of the converter. GROUNDING, DECOUPLING AND LAYOUT INFORMATION Proper grounding and bypassing, short lead length, and the use of ground planes are particularly important for high frequency designs. Multilayer pc-boards are recommended for best performance since they offer distinct advantages such as minimization of ground impedance, separation of signal layers by ground layers, etc. The DAC902 uses separate pins for its analog and digital supply and ground connections. The placement of the decoupling capacitor should be such that the analog supply (+VA) is bypassed to the analog ground (AGND), and the digital supply bypassed to the digital ground (DGND). In most cases 0.1uF ceramic chip capacitors at each supply pin are adequate to provide a low impedance decoupling path. Keep in mind that their effectiveness largely depends on the proximity to the individual supply and ground pins. Therefore, they should be located as close as physically possible to those device leads. Whenever possible, the capacitors should be located immediately under each pair of supply/ ground pins on the reverse side of the pc-board. This layout approach will minimize the parasitic inductance of component leads and pcb runs. CCOMPEXT +5V 0.1µF BW DAC902 IREF = +VA VREF RSET FSA REFIN External Reference Ref Control Amp Current Sources CCOMP 400pF RSET +5V INT/EXT +1.24V Ref. FIGURE 8. External Reference Configuration. DAC902 SBAS094B 15 Further supply decoupling with surface mount tantalum capacitors (1uF to 4.7uF) may be added as needed in proximity of the converter. Low noise is required for all supply and ground connections to the DAC902. It is recommended to use a multilayer pcboard utilizing separate power and ground planes. Mixed signal designs require particular attention to the routing of the different supply currents and signal traces. Generally, analog supply and ground planes should only extend into analog signal areas, such as the DAC output signal and the reference signal. Digital supply and ground planes must be confined to areas covering digital circuitry, including the digital input lines connecting to the converter, as well as the clock signal. The analog and digital ground planes should be joined together at one point underneath the DAC. This can be realized with a short track of approximately 1/8" (3mm). 16 The power to the DAC902 should be provided through the use of wide pcb runs or planes. Wide runs will present a lower trace impedance, further optimizing the supply decoupling. The analog and digital supplies for the converter should only be connected together at the supply connector of the pc-board. In the case of only one supply voltage being available to power the DAC, ferrite beads along with bypass capacitors may be used to create an LC filter. This will generate a low-noise analog supply voltage that can then be connected to the +VA supply pin of the DAC902. While designing the layout, it is important to keep the analog signal traces separate from any digital line, in order to prevent noise coupling onto the analog signal path. DAC902 SBAS094B PACKAGE OPTION ADDENDUM www.ti.com 21-May-2010 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp (3) DAC902E ACTIVE TSSOP PW 28 50 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902E/2K5 ACTIVE TSSOP PW 28 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902E/2K5G4 ACTIVE TSSOP PW 28 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902EG4 ACTIVE TSSOP PW 28 50 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902U ACTIVE SOIC DW 28 20 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902U/1K ACTIVE SOIC DW 28 1000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902U/1KG4 ACTIVE SOIC DW 28 1000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR DAC902UG4 ACTIVE SOIC DW 28 20 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Addendum-Page 1 PACKAGE OPTION ADDENDUM www.ti.com 21-May-2010 Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. 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Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant DAC902E/2K5 TSSOP PW 28 2500 330.0 16.4 6.9 10.2 1.8 12.0 16.0 Q1 DAC902U/1K SOIC DW 28 1000 330.0 32.4 11.35 18.67 3.1 16.0 32.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) DAC902E/2K5 DAC902U/1K TSSOP PW 28 2500 367.0 367.0 38.0 SOIC DW 28 1000 367.0 367.0 55.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46C and to discontinue any product or service per JESD48B. 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