AAT1121 1.5MHz, 250mA Step-Down Converter General Description Features The AAT1121 SwitchReg is a 1.5MHz step-down converter with an input voltage range of 2.7V to 5.5V and output as low as 0.6V. Its low supply current, small size, and high switching frequency make the AAT1121 the ideal choice for portable applications. • • • • • • • • The AAT1121 delivers 250mA of load current, while maintaining a low 30µA no load quiescent current. The 1.5MHz switching frequency minimizes the size of external components, while keeping switching losses low. The AAT1121 feedback and control delivers excellent load regulation and transient response with a small output inductor and capacitor. • • • • The AAT1121 is available in a Pb-free, 8-pin, 2x2mm TDFN or STDFN package and is rated over the -40°C to +85°C temperature range. • SwitchReg™ VIN Range: 2.7V to 5.5V VOUT Range: 0.6V to VIN 250mA Max Output Current Up to 96% Efficiency 30µA Typical Quiescent Current 1.5MHz Switching Frequency Soft-Start Control Over-Temperature and Current Limit Protection 100% Duty Cycle Low-Dropout Operation <1µA Shutdown Current Small External Components Ultra-Small TDFN22-8 or STDFN22-8 Package Temperature Range: -40°C to +85°C Applications • • • • • • Bluetooth™ Headsets Cellular Phones Digital Cameras Handheld Instruments Portable Music Players USB Devices Typical Application VIN VO = 1.8V AAT1121 VP LX VIN C1 4.7µF EN GND 1121.2007.03.1.2 FB PGND 250mA L1 3.0μH R1 118kΩ R2 59kΩ C2 4.7µF 1 AAT1121 1.5MHz, 250mA Step-Down Converter Pin Descriptions Pin # Symbol 1 VP Input power pin; connected to the source of the P-channel MOSFET. Connect to the input capacitor. 2 VIN Input bias voltage for the converter. 3 GND Non-power signal ground pin. 4 FB Feedback input pin. Connect this pin to an external resistive divider for adjustable output. 5 N/C No connect. 6 EN Enable pin. A logic high enables normal operation. A logic low shuts down the converter. 7 LX Switching node. Connect the inductor to this pin. It is connected internally to the drain of both high- and low-side MOSFETs. 8 PGND Input power return pin; connected to the source of the N-channel MOSFET. Connect to the output and input capacitor return. EP Function Exposed paddle (bottom): connect to ground directly beneath the package. Pin Configuration TDFN22-8/STDFN22-8 (Top View) VP VIN GND FB 2 1 8 2 7 3 6 4 5 PGND LX EN N/C 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Absolute Maximum Ratings1 Symbol VIN VLX VOUT VEN TJ TLEAD Description Input Voltage and Bias Power to GND LX to GND FB to GND EN to GND Operating Junction Temperature Range Maximum Soldering Temperature (at leads, 10 sec) Value Units 6.0 -0.3 to VIN + 0.3 -0.3 to VIN + 0.3 -0.3 to 6.0 -40 to 150 300 V V V V °C °C Value Units 2 50 W °C/W Thermal Information Symbol PD θJA Description Maximum Power Dissipation Thermal Resistance2 1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at conditions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time. 2. Mounted on an FR4 board. 1121.2007.03.1.2 3 AAT1121 1.5MHz, 250mA Step-Down Converter Electrical Characteristics1 VIN = 3.6V, TA = -40°C to +85°C, unless otherwise noted; typical values are TA = 25°C. Symbol Description VIN Input Voltage VUVLO UVLO Threshold VOUT Output Voltage Tolerance2 VOUT IQ ISHDN ILIM Output Voltage Range Quiescent Current Shutdown Current P-Channel Current Limit High-Side Switch On Resistance Low-Side Switch On Resistance LX Leakage Current Line Regulation Feedback Threshold Voltage Accuracy FB Leakage Current Oscillator Frequency RDS(ON)H RDS(ON)L ILXLEAK ΔVLinereg/ΔVIN VFB IFB FOSC TS TSD THYS VEN(L) VEN(H) IEN Startup Time Over-Temperature Shutdown Threshold Over-Temperature Shutdown Hysteresis Enable Threshold Low Enable Threshold High Input Low Current Conditions Min Typ 2.7 VIN Rising Hysteresis VIN Falling IOUT = 0 to 250mA, VIN = 2.7V to 5.5V Max Units 5.5 2.6 V V mV V % 250 2.0 -3.0 3.0 0.6 No Load EN = GND VIN 1.5 V µA µA mA Ω Ω µA %/V V µA MHz 100 µs 140 15 °C °C V V µA 30 1.0 600 0.59 0.42 VIN = 5.5V, VLX = 0 to VIN VIN = 2.7V to 5.5V VIN = 3.6V VOUT = 1.0V 1.0 0.591 From Enable to Output Regulation 0.2 0.600 0.609 0.2 0.6 VIN = VEN = 5.5V 1.4 -1.0 1.0 1. The AAT1121 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured by design, characterization, and correlation with statistical process controls. 2. Output voltage tolerance is independent of feedback resistor network accuracy. 4 1121.2007.03.1.2 1121.2006.10.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Typical Characteristics Efficiency vs. Load DC Load Regulation (VOUT = 1.2V; L = 1.5µH) (VOUT = 1.2V; L = 1.5µH) 100 1.0 Efficiency (%) Output Error (%) VIN = 2.7V 90 VIN = 3.6V 80 70 VIN = 4.2V 60 VIN = 5.0V 50 0.5 VIN = 3.6V 0.0 VIN = 2.7V -0.5 VIN = 4.2V 40 30 0.1 1 10 100 -1.0 1000 0.1 1 Output Current (mA) (VOUT = 1.8V; L = 3.3µH) 100 1.0 VIN = 2.7V VIN = 3.6V Output Error (%) Efficiency (%) 90 80 70 VIN = 4.2V 60 50 40 0.1 1 10 100 0.5 VIN = 3.6V 0.0 VIN = 2.7V -0.5 -1.0 0.1 1000 VIN = 4.2V 1 Output Current (mA) 10 100 1000 Output Current (mA) Efficiency vs. Load DC Load Regulation (VOUT = 3.0V; L = 4.7µH) (VOUT = 3.0V; L = 4.7µH) 100 1.0 VIN = 3.6V Output Error (%) Efficiency (%) 1000 DC Load Regulation (VOUT = 1.8V; L = 3.3µH) VIN = 4.2V 80 70 VIN = 5.0V 60 50 40 100 Output Current (mA) Efficiency vs. Load 90 10 0.5 VIN = 4.2V VIN = 3.6V 0.0 VIN = 5.0V -0.5 -1.0 0.1 1 10 Output Current (mA) 1121.2007.03.1.2 100 1000 0.1 1 10 100 1000 Output Current (mA) 5 AAT1121 1.5MHz, 250mA Step-Down Converter Typical Characteristics Soft Start Line Regulation (VOUT = 1.8V) 5.0 0.6 0.5 VEN IOUT = 0mA 0.4 3.0 2.0 1.0 0.8 0.0 0.6 VO 0.4 0.2 0.0 IL Accuracy (%) 4.0 Inductor Current (bottom) (A) Enable and Output Voltage (top) (V) (VIN = 3.6V; VOUT = 1.8V; IOUT = 250mA; CFF = 100pF) 0.3 IOUT = 50mA 0.2 IOUT = 150mA 0.1 0.0 -0.1 IOUT = 10mA IOUT = 250mA -0.2 -0.3 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 Time (100µs/div) Input Voltage (V) Output Voltage Error vs. Temperature Switching Frequency Variation vs. Temperature (VIN = 3.6V; VOUT = 1.8V; IOUT = 250mA) (VIN = 3.6V; VOUT = 1.8V) 3.0 2.0 8.0 Variation (%) Output Error (%) 10.0 1.0 0.0 -1.0 6.0 4.0 2.0 0.0 -2.0 -4.0 -6.0 -2.0 -8.0 -3.0 -40 -10.0 -20 0 20 40 60 80 100 -40 -20 0 Temperature (°°C) 80 100 50 VOUT = 1.8V 1.0 Supply Current (µA) Frequency Variation (%) 60 No Load Quiescent Current vs. Input Voltage 2.0 0.0 -1.0 -2.0 VOUT = 3.0V -3.0 2.7 3.1 3.5 3.9 4.3 Input Voltage (V) 6 40 Temperature (°°C) Frequency Variation vs. Input Voltage -4.0 20 4.7 5.1 5.5 45 40 35 85°C 30 25°C 25 -40°C 20 15 10 2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5 Input Voltage (V) 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Typical Characteristics P-Channel RDS(ON) vs. Input Voltage N-Channel RDS(ON) vs. Input Voltage 750 1000 120°C 700 100°C 700 600 25°C 500 85°C 550 500 450 25°C 350 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 300 2.5 3.0 Input Voltage (V) 3.5 4.0 4.5 5.0 5.5 6.0 Input Voltage (V) Load Transient Response Load Transient Response (10mA to 250mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF; CFF = 100pF) (10mA to 250mA; VIN = 3.6V; VOUT = 1.8V; COUT = 4.7µF) 1.7 IO 250mA 1.6 ILX 10mA 0.2 0.0 -0.2 Time (25µs/div) 1.9 Output Voltage (top) (V) VO 1.8 2.0 1.8 VO 1.7 1.6 250mA IO 10mA ILX 0.2 0.0 -0.2 Load and Inductor Current (bottom) (200mA/div) 1.9 Load and Inductor Current (bottom) (200mA/div) 2.0 Output Voltage (top) (V) 100°C 600 400 400 300 120°C 650 85°C 800 RDS(ON)L (mΩ Ω) RDS(ON)H (mΩ Ω) 900 Time (25µs/div) Line Response (VOUT = 1.8V @ 250mA; CFF = 100pF) 1.90 1.80 VO 1.75 1.70 5.0 4.5 VIN 4.0 Input Voltage (bottom) (V) Output Voltage (top) (V) 1.85 3.5 3.0 Time (25µs/div) 1121.2007.03.1.2 7 AAT1121 1.5MHz, 250mA Step-Down Converter Typical Characteristics Output Ripple Output Ripple (VIN = 3.6V; VOUT = 1.8V; IOUT = 1mA) (VIN = 3.6V; VOUT = 1.8V; IOUT = 250mA) VO -20 0.04 0.03 0.02 0.01 IL 0.00 20 0 VO -20 0.3 0.2 IL 0.1 0.0 -0.01 Time (2µs/div) 8 Inductor Current (bottom) (A) 0 40 Output Voltage (AC Coupled) (top) (mV) 20 Inductor Current (bottom) (A) Output Voltage (AC Coupled) (top) (mV) 40 Time (200ns/div) 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Functional Block Diagram FB VP VIN Err Amp DH Voltage Reference LX Logic EN INPUT DL PGND GND Functional Description The AAT1121 is a high performance 250mA, 1.5MHz monolithic step-down converter designed to operate with an input voltage range of 2.7V to 5.5V. The converter operates at 1.5MHz, which minimizes the size of external components. Typical values are 3.3µH for the output inductor and 4.7µF for the ceramic output capacitor. The device is designed to operate with an output voltage as low as 0.6V. Power devices are sized for 250mA current capability while maintaining over 1121.2007.03.1.2 90% efficiency at full load. Light load efficiency is maintained at greater than 80% down to 1mA of load current. At dropout, the converter duty cycle increases to 100% and the output voltage tracks the input voltage minus the RDS(ON) drop of the P-channel highside MOSFET. A high-DC gain error amplifier with internal compensation controls the output. It provides excellent transient response and load/line regulation. Soft start eliminates any output voltage overshoot when the enable or the input voltage is applied. 9 AAT1121 1.5MHz, 250mA Step-Down Converter Control Loop The AAT1121 is a 250mA current mode step-down converter. The current through the P-channel MOSFET (high side) is sensed for current loop control, as well as short-circuit and overload protection. A fixed slope compensation signal is added to the sensed current to maintain stability for duty cycles greater than 50%. The peak current mode loop appears as a voltage-programmed current source in parallel with the output capacitor. The output of the voltage error amplifier programs the current mode loop for the necessary peak switch current to force a constant output voltage for all load and line conditions. Internal loop compensation terminates the transconductance voltage error amplifier output. The error amplifier reference is fixed at 0.6V. Soft Start / Enable Soft start increases the inductor current limit point in discrete steps when the input voltage or enable input is applied. It limits the current surge seen at the input and eliminates output voltage overshoot. When pulled low, the enable input forces the AAT1121 into a low-power, non-switching state. The total input current during shutdown is less than 1µA. Current Limit and Over-Temperature Protection For overload conditions, the peak input current is limited. As load impedance decreases and the output voltage falls closer to zero, more power is dissipated internally, raising the device temperature. Thermal protection completely disables switching when internal dissipation becomes excessive, protecting the device from damage. The junction over-temperature threshold is 140°C with 15°C of hysteresis. Under-Voltage Lockout Internal bias of all circuits is controlled via the VIN power. Under-voltage lockout (UVLO) guarantees sufficient VIN bias and proper operation of all internal circuits prior to activation. 10 Applications Information Inductor Selection The step-down converter uses peak current mode control with slope compensation to maintain stability for duty cycles greater than 50%. The output inductor value must be selected so the inductor current down slope meets the internal slope compensation requirements. The internal slope compensation for the adjustable and low-voltage fixed versions of the AAT1121 is 0.45A/µsec. This equates to a slope compensation that is 75% of the inductor current down slope for a 1.8V output and 3.0µH inductor. m= 0.75 ⋅ VO 0.75 ⋅ 1.8V A = = 0.45 L 3.0µH µsec This is the internal slope compensation for the AAT1121. When externally programming to 3.0V, the calculated inductance is 5.0µH. L= 0.75 ⋅ VO = m = 1.67 0.75 ⋅ VO µsec ≈ 1.67 A ⋅ VO A 0.45A µsec µsec ⋅ 3.0V = 5.0µH A In this case, a standard 4.7µH value is selected. For most designs, the AAT1121 operates with an inductor value of 1µH to 4.7µH. Table 1 displays inductor values for the AAT1121 with different output voltage options. Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive losses due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Output Voltage (V) L1 (µH) 1.0 1.2 1.5 1.8 2.5 3.0 3.3 1.5 2.2 2.7 3.0 3.9 4.7 5.6 The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current. VO ⎛ V ⎞ · 1- O = VIN ⎝ VIN ⎠ D · (1 - D) = Input Capacitor Select a 4.7µF to 10µF X7R or X5R ceramic capacitor for the input. To estimate the required input capacitor size, determine the acceptable input ripple level (VPP) and solve for CIN. The calculated value varies with input voltage and is a maximum when VIN is double the output voltage. CIN = VO ⎛ V ⎞ · 1- O VIN ⎝ VIN ⎠ ⎛ VPP ⎞ - ESR · FS ⎝ IO ⎠ VO ⎛ V ⎞ 1 · 1 - O = for VIN = 2 × VO VIN ⎝ VIN ⎠ 4 CIN(MIN) = 1 ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO ⎠ Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the proper value. For example, the capacitance of a 10µF, 6.3V, X5R ceramic capacitor with 5.0V DC applied is actually about 6µF. The maximum input capacitor RMS current is: IRMS = IO · 1121.2007.03.1.2 VO ⎛ V ⎞ · 1- O VIN ⎝ VIN ⎠ 1 2 for VIN = 2 x VO Table 1: Inductor Values. The 3.0µH CDRH2D09 series inductor selected from Sumida has a 150mΩ DCR and a 470mA DC current rating. At full load, the inductor DC loss is 9.375mW which gives a 2.08% loss in efficiency for a 250mA, 1.8V output. 0.52 = IRMS(MAX) = VO IO 2 ⎛ V ⎞ · 1- O The term VIN ⎝ VIN ⎠ appears in both the input voltage ripple and input capacitor RMS current equations and is a maximum when VO is twice VIN. This is why the input voltage ripple and the input capacitor RMS current ripple are a maximum at 50% duty cycle. The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the AAT1121. Low ESR/ESL X7R and X5R ceramic capacitors are ideal for this function. To minimize stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C1) can be seen in the evaluation board layout in Figure 2. A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low-ESR ceramic input capacitor, can create a high Q network that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. 11 AAT1121 1.5MHz, 250mA Step-Down Converter In applications where the input power source lead inductance cannot be reduced to a level that does not affect the converter performance, a high ESR tantalum or aluminum electrolytic should be placed in parallel with the low ESR, ESL bypass ceramic. This dampens the high Q network and stabilizes the system. Output Capacitor The output capacitor limits the output ripple and provides holdup during large load transitions. A 4.7µF to 10µF X5R or X7R ceramic capacitor typically provides sufficient bulk capacitance to stabilize the output during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. For enhanced transient response and low temperature operation application, a 10µF (X5R, X7R) ceramic capacitor is recommended to stabilize extreme pulsed load conditions. The output voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current, the ceramic output capacitor alone supplies the load current until the loop responds. Within two or three switching cycles, the loop responds and the inductor current increases to match the load current demand. The relationship of the output voltage droop during the three switching cycles to the output capacitance can be estimated by: COUT = 3 · ΔILOAD VDROOP · FS Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the minimum output capacitor value to 4.7µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, and gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: IRMS(MAX) = 1 VOUT · (VIN(MAX) - VOUT) L · F · VIN(MAX) 2· 3 · Dissipation due to the RMS current in the ceramic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot-spot temperature. Adjustable Output Resistor Selection Resistors R1 and R2 of Figure 1 program the output to regulate at a voltage higher than 0.6V. To limit the bias current required for the external feedback resistor string while maintaining good noise immunity, the suggested value for R2 is 59kΩ. Decreased resistor values are necessary to maintain noise immunity on the FB pin, resulting in increased quiescent current. Table 2 summarizes the resistor values for various output voltages. ⎛ VOUT ⎞ ⎛ 3.3V ⎞ R1 = V -1 · R2 = 0.6V - 1 · 59kΩ = 267kΩ ⎝ REF ⎠ ⎝ ⎠ With enhanced transient response for extreme pulsed load application, an external feed-forward capacitor, (C3 in Figure 1), can be added. R2 = 59kΩ R2 = 221kΩ VOUT (V) R1 (kΩ) R1 (kΩ) 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 3.3 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 267 75 113 150 187 221 261 301 332 442 464 523 715 1000 Table 2: Adjustable Resistor Values For Step-Down Converter. 12 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter For the condition where the step-down converter is in dropout at 100% duty cycle, the total device dissipation reduces to: Thermal Calculations There are three types of losses associated with the AAT1121 step-down converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the RDS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified form of the losses is given by: PTOTAL = 2 O I PTOTAL = IO2 · RDSON(H) + IQ · VIN Since RDS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. Given the total losses, the maximum junction temperature can be derived from the θJA for the TDFN22-8 package which is 50°C/W. · (RDSON(H) · VO + RDSON(L) · [VIN - VO]) VIN TJ(MAX) = PTOTAL · ΘJA + TAMB + (tsw · F · IO + IQ) · VIN IQ is the step-down converter quiescent current. The term tsw is used to estimate the full load stepdown converter switching losses. U1 1 VIN 2 3 4 C1 4.7μF VP PGND VIN LX GND EN FB N/C 8 7 LX L1 +VOUT 6 5 AAT1121 C2 4.7μF R1 Adj. C3 (optional) 100pF R2 59kΩ GND GND Figure 1: AAT1121 Schematic. 1121.2007.03.1.2 13 AAT1121 1.5MHz, 250mA Step-Down Converter Layout The suggested PCB layout for the AAT1121 is shown in Figures 2, 3, and 4. The following guidelines should be used to help ensure a proper layout. 1. The input capacitor (C1) should connect as closely as possible to VP (Pin 1), PGND (Pin 8), and GND (Pin 3) 2. C2 and L1 should be connected as closely as possible. The connection of L1 to the LX pin should be as short as possible. Do not make the node small by using narrow trace. The trace should be kept wide, direct and short. 3. The feedback pin (Pin 4) should be separate from any power trace and connect as closely as possible to the load point. Sensing along a Figure 2: AAT1121 Evaluation Board Top Side Layout. high-current load trace will degrade DC load regulation. Feedback resistors should be placed as closely as possible to the FB pin (Pin 4) to minimize the length of the high impedance feedback trace. If possible, they should also be placed away from the LX (switching node) and inductor to improve noise immunity. 4. The resistance of the trace from the load return to PGND (Pin 8) and GND (Pin 3) should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal signal ground and the power ground. 5. A high density, small footprint layout can be achieved using an inexpensive, miniature, nonshielded, high DCR inductor. Figure 3: Exploded View of AAT1121 Evaluation Board Top Side Layout. Figure 4: AAT1121 Evaluation Board Bottom Side Layout. 14 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Step-Down Converter Design Example Specifications VO = 1.8V @ 250mA, Pulsed Load ΔILOAD = 200mA VIN = 2.7V to 4.2V (3.6V nominal) FS = 1.5MHz TAMB = 85°C 1.8V Output Inductor L1 = 1.67 µsec µsec ⋅ VO2 = 1.67 ⋅ 1.8V = 3µH A A (use 3.0µH; see Table 1) For Sumida inductor CDRH2D09-3R0, 3.0µH, DCR = 150mΩ. ΔIL1 = ⎛ VO V ⎞ 1.8V 1.8V ⎞ ⎛ ⋅ 1- O = ⋅ 1= 228mA L1 ⋅ F ⎝ VIN ⎠ 3.0µH ⋅ 1.5MHz ⎝ 4.2V ⎠ IPKL1 = IO + ΔIL1 = 250mA + 114mA = 364mA 2 PL1 = IO2 ⋅ DCR = 250mA2 ⋅ 150mΩ = 9.375mW 1.8V Output Capacitor VDROOP = 0.1V COUT = 3 · ΔILOAD 3 · 0.2A = = 4µF (use 4.7µF) 0.1V · 1.5MHz VDROOP · FS IRMS = (VO) · (VIN(MAX) - VO) 1 1.8V · (4.2V - 1.8V) · = 66mArms = 3.0µH · 1.5MHz · 4.2V · V L1 · F 2· 3 2· 3 S IN(MAX) 1 · Pesr = esr · IRMS2 = 5mΩ · (66mA)2 = 21.8µW 1121.2007.03.1.2 15 AAT1121 1.5MHz, 250mA Step-Down Converter Input Capacitor Input Ripple VPP = 25mV CIN = IRMS = 1 ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO ⎠ = 1 = 1.38µF (use 4.7µF) ⎛ 25mV ⎞ - 5mΩ · 4 · 1.5MHz ⎝ 0.2A ⎠ IO = 0.1Arms 2 P = esr · IRMS2 = 5mΩ · (0.1A)2 = 0.05mW AAT1121 Losses PTOTAL = IO2 · (RDSON(HS) · VO + RDSON(LS) · [VIN -VO]) VIN + (tsw · F · IO + IQ) · VIN = 0.22 · (0.59Ω · 1.8V + 0.42Ω · [4.2V - 1.8V]) 4.2V + (5ns · 1.5MHz · 0.2A + 30µA) · 4.2V = 26.14mW TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 26.14mW = 86.3°C 16 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Output Voltage VOUT (V) R2 = 59kΩ R1 (kΩ) R2 = 221kΩ1 R1 (kΩ) L1 (µH) — 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 267 — 75 113 150 187 221 261 301 332 442 464 523 715 1000 1.5 1.5 1.5 1.5 1.5 1.5 1.5 2.2 2.7 3.0/3.3 3.0/3.3 3.0/3.3 3.9/4.2 5.6 2 0.6 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 3.3 Table 3: Evaluation Board Component Values. Manufacturer Sumida Sumida Sumida Sumida Sumida Sumida Sumida Sumida Sumida Sumida Sumida Taiyo Yuden Taiyo Yuden Taiyo Yuden Taiyo Yuden FDK FDK FDK FDK Part Number Inductance (µH) Max DC Current (mA) DCR (mΩ) Size (mm) LxWxH Type CDRH2D09-1R5 CDRH2D09-2R2 CDRH2D09-2R5 CDRH2D09-3R0 CDRH2D09-3R9 CDRH2D09-4R7 CDRH2D09-5R6 CDRH2D11-1R5 CDRH2D11-2R2 CDRH2D11-3R3 CDRH2D11-4R7 NR3010 NR3010 NR3010 NR3010 MIPWT3226D-1R5 MIPWT3226D-2R2 MIPWT3226D-3R0 MIPWT3226D-4R2 1.5 2.2 2.5 3 3.9 4.7 5.6 1.5 2.2 3.3 4.7 1.5 2.2 3.3 4.7 1.5 2.2 3 4.2 730 600 530 470 450 410 370 900 780 600 500 1200 1100 870 750 1200 1100 1000 900 88 115 135 150 180 230 260 54 78 98 135 80 95 140 190 90 100 120 140 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.2x3.2x1.2 3.2x3.2x1.2 3.2x3.2x1.2 3.2x3.2x1.2 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.0x3.0x1.0 3.2x2.6x0.8 3.2x2.6x0.8 3.2x2.6x0.8 3.2x2.6x0.8 Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Shielded Chip shielded Chip shielded Chip shielded Chip shielded Table 4: Suggested Inductors and Suppliers. 1. For reduced quiescent current, R2 = 221kΩ. 2. R2 is opened, R1 is shorted. 1121.2007.03.1.2 17 AAT1121 1.5MHz, 250mA Step-Down Converter Manufacturer Murata Murata Part Number Value (µF) Voltage Rating Temp. Co. Case Size GRM118R60J475KE19B GRM188R60J106ME47D 4.7 10 6.3 6.3 X5R X5R 0603 0603 Table 5: Surface Mount Capacitors. 18 1121.2007.03.1.2 AAT1121 1.5MHz, 250mA Step-Down Converter Ordering Information Output Voltage Package Marking1 Part Number (Tape and Reel)2 0.6V 0.6V TDFN22-8 STDFN22-8 RWXYY RWXYY AAT1121IPS-0.6-T1 AAT1121IES-0.6-T1 All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. Package Information3 TDFN22-8 0.600 ± 0.050 0.1 REF (optional) 0.450 ± 0.050 C0.3 Bottom View 2.000 ± 0.050 0.350 ± 0.100 Detail "A" 1.270 ± 0.050 2.000 ± 0.050 Index Area Pin 1 Indicator (optional) 0.230 ± 0.050 Top View 0.050 ± 0.050 0.229 ± 0.051 0.850 MAX 4x Detail "A" Side View All dimensions in millimeters. 1. XYY = assembly and date code. 2. Sample stock is generally held on all part numbers listed in BOLD. 3. The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection. 1121.2007.03.1.2 19 AAT1121 1.5MHz, 250mA Step-Down Converter STDFN22-8 Index Area (D/2 x E/2) 0.80 ± 0.05 Detail "A" 1.45 ± 0.05 2.00 ± 0.05 2.00 ± 0.05 Top View Bottom View Side View Pin 1 Indicator (optional) 0.45 ± 0.05 0.23 ± 0.05 0.05 ± 0.05 0.15 ± 0.025 0.55 ± 0.05 0.35 ± 0.05 Detail "A" All dimensions in millimeters. © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service without notice. Except as provided in AnalogicTech’s terms and conditions of sale, AnalogicTech assumes no liability whatsoever, and AnalogicTech disclaims any express or implied warranty relating to the sale and/or use of AnalogicTech products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate design and operating safeguards must be provided by the customer to minimize inherent or procedural hazards. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders. Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737- 4600 Fax (408) 737- 4611 20 1121.2007.03.1.2