80W Operational Amplifier
General Description
The LM12 is a power op amp capable of driving ± 25V at
± 10A while operating from ± 30V supplies. The monolithic IC
can deliver 80W of sine wave power into a 4Ω load with
0.01% distortion. Power bandwidth is 60 kHz. Further, a
peak dissipation capability of 800W allows it to handle reactive loads such as transducers, actuators or small motors
without derating. Important features include:
• input protection
• controlled turn on
• thermal limiting
• overvoltage shutdown
• output-current limiting
• dynamic safe-area protection
The IC delivers ± 10A output current at any output voltage
yet is completely protected against overloads, including
shorts to the supplies. The dynamic safe-area protection is
provided by instantaneous peak-temperature limiting within
the power transistor array.
The turn-on characteristics are controlled by keeping the
output open-circuited until the total supply voltage reaches
14V. The output is also opened as the case temperature ex-
ceeds 150˚C or as the supply voltage approaches the
BVCEO of the output transistors. The IC withstands overvoltages to 80V.
This monolithic op amp is compensated for unity-gain feedback, with a small-signal bandwidth of 700 kHz. Slew rate is
9V/µs, even as a follower. Distortion and capacitive-load stability rival that of the best designs using complementary output transistors. Further, the IC withstands large differential
input voltages and is well behaved should the
common-mode range be exceeded.
The LM12 establishes that monolithic ICs can deliver considerable output power without resorting to complex switching
schemes. Devices can be paralleled or bridged for even
greater output capability. Applications include operational
power supplies, high-voltage regulators, high-quality audio
amplifiers, tape-head positioners, x-y plotters or other
servo-control systems.
The LM12 is supplied in a four-lead, TO-3 package with V−
on the case. A gold-eutectic die-attach to a molybdenum interface is used to avoid thermal fatigue problems. The LM12
is specified for either military or commercial temperature
Typical Application*
Connection Diagram
*Low distortion (0.01%) audio amplifier
4-pin glass epoxy TO-3
socket is available from
Part number 8112-AG7
Bottom View
Order Number LM12CLK
See NS Package Number K04A
© 1999 National Semiconductor Corporation
LM12CL 80W Operational Amplifier
May 1999
Absolute Maximum Ratings (Note 1)
Storage Temperature Range
Lead Temperature
(Soldering, 10 seconds)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Total Supply Voltage (Note 1)
Input Voltage
Output Current
Junction Temperature
−65˚C to 150˚C
Operating Ratings
(Note 2)
Internally Limited
(Note 3)
Total Supply Voltage
Case Temperature (Note 4)
15V to 60V
0˚C to 70˚C
Electrical Characteristics (Note 4)
Input Offset Voltage
± 10V ≤ VS ≤ ± 0.5 VMAX, VCM = 0
mV (max)
Input Bias Current
V− + 4V ≤ VCM ≤ V+ −2V
µA (max)
Input Offset Current
V− +4V ≤ VCM ≤ V+ −2V
µA (max)
Common Mode
V− +4V ≤ VCM ≤ V+ −2V
dB (min)
Power Supply
V+ = 0.5 VMAX,
dB (min)
−6V ≥ V− ≥ −0.5 VMAX
V− = −0.5 VMAX,
dB (min)
Output Saturation
Large Signal Voltage
Thermal Gradient
6V ≤ V+ ≤ 0.5 VMAX
tON = 1 ms,
∆VIN = 5 (10 ) mV,
V (max)
V (max)
tON = 2 ms,
VSAT = 2V, IOUT = 0
VSAT = 8V, RL = 4Ω
V (max)
V/mV (min)
V/mV (min)
PDISS = 50W, tON = 65 ms
µV/W (max)
tON = 10 ms, VDISS = 10V
tON = 100 ms, VDISS = 58V
A (max)
A (min)
A (max)
W (min)
W (min)
˚C/W (max)
Output-Current Limit
Power Dissipation
DC Thermal Resistance
AC Thermal Resistance
Supply Current
tON = 100 ms, VDISS = 20V
(Note 5) VDISS = 20V
˚C/W (max)
(Note 5)
VOUT = 0, IOUT = 0
˚C/W (max)
mA (max)
Note 1: Absolute maximum ratings indicate limits beyond which damage to the device may occur. The maximum voltage for which the LM12 is guaranteed to operate
is given in the operating ratings and in Note 4. With inductive loads or output shorts, other restrictions described in applications section apply.
Note 2: Neither input should exceed the supply voltage by more than 50 volts nor should the voltage between one input and any other terminal exceed 60 volts.
Note 3: Operating junction temperature is internally limited near 225˚C within the power transistor and 160˚C for the control circuitry.
Note 4: The supply voltage is ± 30V (VMAX = 60V), unless otherwise specified. The voltage across the conducting output transistor (supply to output) is VDISS and
internal power dissipation is PDISS. Temperature range is 0˚C ≤ TC ≤ 70˚C where TC is the case temperature. Standard typeface indicates limits at 25˚C while boldface type refers to limits or special conditions over full temperature range. With no heat sink, the package will heat at a rate of 35˚C/sec per 100W of internal
Note 5: This thermal resistance is based upon a peak temperature of 200˚C in the center of the power transistor and a case temperature of 25˚C measured at the
center of the package bottom. The maximum junction temperature of the control circuitry can be estimated based upon a dc thermal resistance of 0.9˚C/W or an ac
thermal resistance of 0.6˚C/W for any operating voltage.
Although the output and supply leads are resistant to electrostatic discharges from handling, the input leads are not.
The part should be treated accordingly.
Output-Transistor Ratings (guaranteed)
Safe Area
DC Thermal Resistance
Pulse Thermal Resistance
Typical Performance Characteristics
Pulse Power Limit
Pulse Power Limit
Output Saturation Voltage
Peak Output Current
Large Signal Response
Large Signal Gain
Follower Pulse Response
Thermal Response
Total Harmonic Distortion
Typical Performance Characteristics
Frequency Response
Output Impedance
Input Bias Current
Input Noise Voltage
Supply Current
Common Mode Rejection
Supply Current
Cross-Supply Current
are realized, will prompt their use in applications that might
now seem trivial. Replacing single power transistors with an
op amp will become economical because of improved performance, simplification of attendant circuitry, vastly improved fault protection, greater reliability and the reduction of
design time.
Power op amps introduce new factors into the design equation. With current transients above 10A, both the inductance
and resistance of wire interconnects become important in a
number of ways. Further, power ratings are a crucial factor in
determining performance. But the power capability of the IC
cannot be realized unless it is properly mounted to an adequate heat sink. Thus, thermal design is of major importance with power op amps.
Application Information
Twenty five years ago the operational amplifier was a specialized design tool used primarily for analog computation.
However, the availability of low cost IC op amps in the late
1960’s prompted their use in rather mundane applications,
replacing a few discrete components. Once a few basic principles are mastered, op amps can be used to give exceptionally good results in a wide range of applications while minimizing both cost and design effort.
The availability of a monolithic power op amp now promises
to extend these advantages to high-power designs. Some
conventional applications are given here to illustrate op amp
design principles as they relate to power circuitry. The inevitable fall in prices, as the economies of volume production
Power Supply Rejection
This application summary starts off by identifying the origin
of strange problems observed while using the LM12 in a
Application Information
Many problems unrelated to system performance can be
traced to the grounding of line-operated test equipment used
for system checkout. Hidden paths are particularly difficult to
sort out when several pieces of test equipment are used but
can be minimized by using current probes or the new isolated oscilloscope pre-amplifiers. Eliminating any direct
ground connection between the signal generator and the oscilloscope synchronization input solves one common problem.
wide variety of designs with all sorts of fault conditions. A few
simple precautions will eliminate these problems. One
would do well to read the section on supply bypassing,
lead inductance, output clamp diodes, ground loops and
reactive loading before doing any experimentation.
Should there be problems with erratic operation,
blow-outs, excessive distortion or oscillation, another
look at these sections is in order.
The management and protection circuitry can also affect operation. Should the total supply voltage exceed ratings or
drop below 15–20V, the op amp shuts off completely. Case
temperatures above 150˚C also cause shut down until the
temperature drops to 145˚C. This may take several seconds,
depending on the thermal system. Activation of the dynamic
safe-area protection causes both the main feedback loop to
lose control and a reduction in output power, with possible
oscillations. In ac applications, the dynamic protection will
cause waveform distortion. Since the LM12 is well protected
against thermal overloads, the suggestions for determining
power dissipation and heat sink requirements are presented
When a push-pull amplifier goes into power limit while driving an inductive load, the stored energy in the load inductance can drive the output outside the supplies. Although the
LM12 has internal clamp diodes that can handle several amperes for a few milliseconds, extreme conditions can cause
destruction of the IC. The internal clamp diodes are imperfect in that about half the clamp current flows into the supply
to which the output is clamped while the other half flows
across the supplies. Therefore, the use of external diodes to
clamp the output to the power supplies is strongly recommended. This is particularly important with higher supply
Experience has demonstrated that hard-wire shorting the
output to the supplies can induce random failures if these external clamp diodes are not used and the supply voltages are
above ± 20V. Therefore it is prudent to use outputclamp diodes even when the load is not particularly inductive. This
also applies to experimental setups in that blowouts have
been observed when diodes were not used. In packaged
equipment, it may be possible to eliminate these diodes, providing that fault conditions can be controlled.
All op amps should have their supply leads bypassed with
low-inductance capacitors having short leads and located
close to the package terminals to avoid spurious oscillation
problems. Power op amps require larger bypass capacitors.
The LM12 is stable with good-quality electrolytic bypass capacitors greater than 20 µF. Other considerations may require larger capacitors.
The current in the supply leads is a rectified component of
the load current. If adequate bypassing is not provided, this
distorted signal can be fed back into internal circuitry. Low
distortion at high frequencies requires that the supplies be
bypassed with 470 µF or more, at the package terminals.
With ordinary op amps, lead-inductance problems are usually restricted to supply bypassing. Power op amps are also
sensitive to inductance in the output lead, particularly with
heavy capacitive loading. Feedback to the input should be
taken directly from the output terminal, minimizing common
inductance with the load. Sensing to a remote load must be
accompanied by a high-frequency feedback path directly
from the output terminal. Lead inductance can also cause
voltage surges on the supplies. With long leads to the power
source, energy stored in the lead inductance when the output is shorted can be dumped back into the supply bypass
capacitors when the short is removed. The magnitude of this
transient is reduced by increasing the size of the bypass capacitor near the IC. With 20 µF local bypass, these voltage
surges are important only if the lead length exceeds a couple
feet ( > 1 µH lead inductance). Twisting together the supply
and ground leads minimizes the effect.
Heat sinking of the clamp diodes is usually unimportant in
that they only clamp current transients. Forward drop with
15A fault transients is of greater concern. Usually, these
transients die out rapidly. The clamp to the negative supply
can have somewhat reduced effectiveness under worst case
conditions should the forward drop exceed 1.0V. Mounting
this diode to the power op amp heat sink improves the situation. Although the need has only been demonstrated with
some motor loads, including a third diode (D3 above) will
eliminate any concern about the clamp diodes. This diode,
however, must be capable of dissipating continuous power
as determined by the negative supply current of the op amp.
With fast, high-current circuitry, all sorts of problems can
arise from improper grounding. In general, difficulties can be
avoided by returning all grounds separately to a common
point. Sometimes this is impractical. When compromising,
special attention should be paid to the ground returns for the
supply bypasses, load and input signal. Ground planes also
help to provide proper grounding.
The LM12 is normally stable with resistive, inductive or
smaller capacitive loads. Larger capacitive loads interact
with the open-loop output resistance (about 1Ω) to reduce
the phase margin of the feedback loop, ultimately causing
oscillation. The critical capacitance depends upon the feedback applied around the amplifier; a unity-gain follower can
handle about 0.01 µF, while more than 1 µF does not cause
problems if the loop gain is ten. With loop gains greater than
unity, a speedup capacitor across the feedback resistor will
Application Information
The LM12 is prone to low-amplitude oscillation bursts coming out of saturation if the high-frequency loop gain is near
unity. The voltage follower connection is most susceptible.
This glitching can be eliminated at the expense of
small-signal bandwidth using input compensation. Input
compensation can also be used in combination with LR load
isolation to improve capacitive load stability.
aid stability. In all cases, the op amp will behave predictably
only if the supplies are properly bypassed, ground loops are
controlled and high-frequency feedback is derived directly
from the output terminal, as recommended earlier.
So-called capacitive loads are not always capacitive. A
high-Q capacitor in combination with long leads can present
a series-resonant load to the op amp. In practice, this is not
usually a problem; but the situation should be kept in mind.
An example of a voltage follower with input compensation is
shown here. The R2C2 combination across the input works
with R1 to reduce feedback at high frequencies without
greatly affecting response below 100 kHz. A lead capacitor,
C1, improves phase margin at the unity-gain crossover frequency. Proper operation requires that the output impedance
of the circuitry driving the follower be well under 1 kΩ at frequencies up to a few hundred kilohertz.
Large capacitive loads (including series-resonant) can be
accommodated by isolating the feedback amplifier from the
load as shown above. The inductor gives low output impedance at lower frequencies while providing an isolating impedance at high frequencies. The resistor kills the Q of series resonant circuits formed by capacitive loads. A low
inductance, carbon-composition resistor is recommended.
Optimum values of L and R depend upon the feedback gain
and expected nature of the load, but are not critical. A 4 µH
inductor is obtained with 14 turns of number 18 wire, close
spaced, around a one-inch-diameter form.
Extending input compensation to the integrator connection is
shown here. Both the follower and this integrator will handle
1 µF capacitive loading without LR output isolation.
The LM12 can be made stable for all loads with a large capacitor on the output, as shown above. This compensation
gives the lowest possible closed-loop output impedance at
high frequencies and the best load-transient response. It is
appropriate for such applications as voltage regulators.
A feedback capacitor, C1, is connected directly to the output
pin of the IC. The output capacitor, C2, is connected at the
output terminal with short leads. Single-point grounding to
avoid dc and ac ground loops is advised.
The impedance, Z1, is the wire connecting the op amp output
to the load capacitor. About 3-inches of number-18 wire
(70 nH) gives good stability and 18-inches (400 nH) begins
to degrade load-transient response. The minimum load capacitance is 47 µF, if a solid-tantalum capacitor with an
equivalent series resistance (ESR) of 0.1Ω is used. Electrolytic capacitors work as well, although capacitance may have
to be increased to 200 µF to bring ESR below 0.1Ω.
Loop stability is not the only concern when op amps are operated with reactive loads. With time-varying signals, power
dissipation can also increase markedly. This is particularly
true with the combination of capacitive loads and
high-frequency excitation.
This circuit provides an output current proportional to the input voltage. Current drive is sometimes preferred for servo
motors because it aids in stabilizing the servo loop by reducing phase lag caused by motor inductance. In applications
requiring high output resistance, such as operational power
supplies running in the current mode, matching of the feedback resistors to 0.01% is required. Alternately, an adjustable resistor can be used for trimming.
Application Information
proached. This will not damage the LM12. It can be avoided
in both cases by connecting A1 as an inverting amplifier and
restricting bandwidth with C1.
Output drive beyond the capability of one power amplifier
can be provided as shown here. The power op amps are
wired as followers and connected in parallel with the outputs
coupled through equalization resistors. A standard,
high-voltage op amp is used to provide voltage gain. Overall
feedback compensates for the voltage dropped across the
equalization resistors.
With parallel operation, there may be an increase in unloaded supply current related to the offset voltage across the
equalization resistors. More output buffers, with individual
equalization resistors, may be added to meet even higher
drive requirements.
Although op amps are usually operated from dual supplies,
single-supply operation is practical. This bridge amplifier
supplies bi-directional current drive to a servo motor while
operating from a single positive supply. The output is easily
converted to voltage drive by shorting R6 and connecting R7
to the output of A2, rather than A1.
Either input may be grounded, with bi-directional drive provided to the other. It is also possible to connect one input to
a positive reference, with the input signal varying about this
voltage. If the reference voltage is above 5V, R2 and R3 are
not required.
This connection allows increased output capability without
requiring a separate control amplifier. The output buffer, A2,
provides load current through R5 equal to that supplied by
the main amplifier, A1, through R4. Again, more output buffers can be added.
Current sharing among paralleled amplifiers can be affected
by gain error as the power-bandwidth limit is approached. In
the first circuit, the operating current increase will depend
upon the matching of high-frequency characteristics. In the
second circuit, however, the entire input error of A2 appears
across R4 and R5. The supply current increase can cause
power limiting to be activated as the slew limit is ap-
The voltage swing delivered to the load can be doubled by
using the bridge connection shown here. Output clamping to
the supplies can be provided by using a bridge-rectifier assembly.
Application Information
One limitation of the standard bridge connection is that the load cannot be returned to ground. This can be circumvented by operating the bridge with floating supplies, as shown above. For single-ended drive, either input can be grounded.
This circuit shows how two amplifiers can be cascaded to double output swing. The advantage over the bridge is that the output
can be increased with any number of stages, although separate supplies are required for each.
Discrete transistors can be used to increase output drive to ± 70V at ± 10A as shown above. With proper thermal design, the IC
will provide safe-area protection for the external transistors. Voltage gain is about thirty.
Application Information
Note: Supply voltages for the LM318s are ± 15V
External current limit can be provided for a power op amp as shown above. The positive and negative current limits can be set
precisely and independently. Fast response is assured by D1 and D2. Adjustment range can be set down to zero with potentiometers R3 and R7. Alternately, the limit can be programmed from a voltage supplied to R2 and R6. This is the set up required for
an operational power supply or voltage-programmable power source.
drive to the motor. Current drive eliminates loop phase shift
due to motor inductance and makes high-performance servos easier to stabilize.
When making servo systems with a power op amp, there is
a temptation to use it for frequency shaping to stabilize the
servo loop. Sometimes this works; other times there are better ways; and occasionally it just doesn’t fly. Usually it’s a
matter of how quickly and to what accuracy the servo must
This position servo uses an op amp to develop the rate signal electrically instead of using a tachometer. In
high-performance servos, rate signals must be developed
with large error signals well beyond saturation of the motor
drive. Using a separate op amp with a feedback clamp allows the rate signal to be developed properly with position
errors more than an order of magnitude beyond the
loop-saturation level as long as the photodiode sensors are
positioned with this in mind.
This motor/tachometer servo gives an output speed proportional to input voltage. A low-level op amp is used for frequency shaping while the power op amp provides current
Application Information
An op amp can be used as a positive or negative regulator.
Unlike most regulators, it can sink current to absorb energy
dumped back into the output. This positive regulator has a
0–50V output range.
A power amplifier suitable for use in high-quality audio equipment is shown above. Harmonic distortion is about
0.01-percent. Intermodulation distortion (60 Hz/7 kHz, 4:1)
measured 0.015-percent. Transient response and saturation
recovery are clean, and the 9 V/µs slew rate of the LM12 virtually eliminates transient intermodulation distortion. Using
separate amplifiers to drive low- and high-frequency speakers gets rid of high-level crossover networks and attenuators. Further, it prevents clipping on the low-frequency channel from distorting the high frequencies.
It is a simple matter to establish power requirements for an
op amp driving a resistive load at frequencies well below
10 Hz. Maximum dissipation occurs when the output is at
one-half the supply voltage with high-line conditions. The individual output transistors must be rated to handle this power
continuously at the maximum expected case temperature.
The power rating is limited by the maximum junction temperature as determined by
where TC is the case temperature as measured at the center
of the package bottom, PDISS is the maximum power dissipation and θJC is the thermal resistance at the operating voltage of the output transistor. Recommended maximum junction temperatures are 200˚C within the power transistor and
150˚C for the control circuitry.
If there is ripple on the supply bus, it is valid to use the average value in worst-case calculations as long as the peak rating of the power transistor is not exceeded at the ripple peak.
With 120 Hz ripple, this is 1.5 times the continuous power
Dissipation requirements are not so easily established with
time varying output signals, especially with reactive loads.
Both peak and continuous dissipation ratings must be taken
into account, and these depend on the signal waveform as
well as load characteristics.
Dual supplies are not required to use an op amp as a voltage
regulator if zero output is not required. This 4V to 50V regulator operates from a single supply. Should the op amp not
be able to absorb enough energy to control an overvoltage
condition, a SCR will crowbar the output.
With a sine wave output, analysis is fairly straightforward.
With supply voltages of ± VS, the maximum average power
dissipation of both output transistors is
Remote sensing as shown above allows the op amp to correct for dc drops in cables connecting the load. Even so,
cable drop will affect transient response. Degradation can be
minimized by using twisted, heavy-gauge wires on the output line. Normally, common and one input are connected together at the sending end.
Application Information
sulting in even higher peak dissipation than a
permanent-magnet motor having the same locked-rotor resistance.
where ZL is the magnitude of the load impedance and θ its
phase angle. Maximum average dissipation occurs below
maximum output swing for θ < 40˚.
The pass transistor dissipation of a voltage regulator is easily determined in the operating mode. Maximum continuous
dissipation occurs with high line voltage and maximum load
current. As discussed earlier, ripple voltage can be averaged
if peak ratings are not exceeded; however, a higher average
voltage will be required to insure that the pass transistor
does not saturate at the ripple minimum.
Conditions during start-up can be more complex. If the input
voltage increases slowly such that the regulator does not go
into current limit charging output capacitance, there are no
problems. If not, load capacitance and load characteristics
must be taken into account. This is also the case if automatic
restart is required in recovering from overloads.
Automatic restart or start-up with fast-rising input voltages
cannot be guaranteed unless the continuous dissipation rating of the pass transistor is adequate to supply the load current continuously at all voltages below the regulated output
voltage. In this regard, the LM12 performs much better than
IC regulators using foldback current limit, especially with
high-line input voltage above 20V.
The instantaneous power dissipation over the conducting
half cycle of one output transistor is shown here. Power dissipation is near zero on the other half cycle. The output level
is that resulting in maximum peak and average dissipation.
Plots are given for a resistive and a series RL load. The latter
is representative of a 4Ω loudspeaker operating below resonance and would be the worst case condition in most audio
applications. The peak dissipation of each transistor is about
four times average. In ac applications, power capability is often limited by the peak ratings of the power transistor.
The pulse thermal resistance of the LM12 is specified for
constant power pulse duration. Establishing an exact
equivalency between constant-power pulses and those encountered in practice is not easy. However, for sine waves,
reasonable estimates can be made at any frequency by assuming a constant power pulse amplitude given by:
where φ = 60˚ and θ is the absolute value of the phase angle
of ZL. Equivalent pulse width is tON ≅ 0.4τ for θ = 0 and tON
≅ 0.2τ for θ ≥ 20˚, where τ is the period of the output waveform.
Should the power ratings of the LM12 be exceeded, dynamic
safe-area protection is activated. Waveforms with this power
limiting are shown for the LM12 driving ± 26V at 30 Hz into
3Ω in series with 24 mH (θ = 45˚). With an inductive load, the
output clamps to the supplies in power limit, as above. With
resistive loads, the output voltage drops in limit. Behavior
with more complex RCL loads is between these extremes.
Secondary thermal limit is activated should the case temperature exceed 150˚C. This thermal limit shuts down the IC
completely (open output) until the case temperature drops to
about 145˚C. Recovery may take several seconds.
A motor with a locked rotor looks like an inductance in series
with a resistance, for purposes of determining driver dissipation. With slow-response servos, the maximum signal amplitude at frequencies where motor inductance is significant
can be so small that motor inductance does not have to be
taken into account. If this is the case, the motor can be
treated as a simple, resistive load as long as the rotor speed
is low enough that the back emf is small by comparison to
the supply voltage of the driver transistor.
A permanent-magnet motor can build up a back emf that is
equal to the output swing of the op amp driving it. Reversing
this motor from full speed requires the output drive transistor
to operate, initially, along a loadline based upon the motor
resistance and total supply voltage. Worst case, this loadline
will have to be within the continuous dissipation rating of the
drive transistor; but system dynamics may permit taking advantage of the higher pulse ratings. Motor inductance can
cause added stress if system response is fast.
Shunt- and series-wound motors can generate back emf’s
that are considerably more than the total supply voltage, re-
Power op amps do not require regulated supplies. However,
the worst-case output power is determined by the low-line
supply voltage in the ripple trough. The worst-case power
dissipation is established by the average supply voltage with
high-line conditions. The loss in power output that can be
guaranteed is the square of the ratio of these two voltages.
Relatively simple off-line switching power supplies can provide voltage conversion, line isolation and 5-percent regulation while reducing size and weight.
The regulation against ripple and line variations can provide
a substantial increase in the power output that can be guar-
Application Information
pound. Experience has shown that these rubber washers
deteriorate and must be replaced should the IC be dismounted.
“Isostrate” insulating pads for four-lead TO-3 packages are
available from Power Devices, Inc. Thermal grease is not required, and the insulators should not be reused.
anteed under worst-case conditions. In addition, switching
power supplies can convert low-voltage power sources such
as automotive batteries up to regulated, dual, high-voltage
supplies optimized for powering power op amps.
A semiconductor manufacturer has no control over heat sink
design. Temperature rating can only be based upon case
temperature as measured at the center of the package bottom. With power pulses of longer duration than 100 ms, case
temperature is almost entirely dependent on heat sink design and the mounting of the IC to the heat sink.
Definition of Terms
Input offset voltage: The absolute value of the voltage between the input terminals with the output voltage and current
at zero.
Input bias current: The absolute value of the average of the
two input currents with the output voltage and current at
Input offset current: The absolute value of the difference in
the two input currents with the output voltage and current at
Common-mode rejection: The ratio of the input voltage
range to the change in offset voltage between the extremes.
Supply-voltage rejection: The ratio of the specified
supply-voltage change to the change in offset voltage between the extremes.
Output saturation threshold: The output swing limit for a
specified input drive beyond that required for zero output. It
is measured with respect to the supply to which the output is
Large signal voltage gain: The ratio of the output voltage
swing to the differential input voltage required to drive the
output from zero to either swing limit. The output swing limit
is the supply voltage less a specified quasi-saturation voltage. A pulse of short enough duration to minimize thermal effects is used as a measurement signal.
Thermal gradient feedback: The input offset voltage
change caused by thermal gradients generated by heating of
the output transistors, but not the package. This effect is delayed by several milliseconds and results in increased gain
error below 100 Hz.
Output-current limit: The output current with a fixed output
voltage and a large input overdrive. The limiting current
drops with time once the protection circuitry is activated.
Power dissipation rating: The power that can be dissipated for a specified time interval without activating the protection circuitry. For time intervals in excess of 100 ms, dissipation capability is determined by heat sinking of the IC
package rather than by the IC itself.
Thermal resistance: The peak, junction-temperature rise,
per unit of internal power dissipation, above the case temperature as measured at the center of the package bottom.
The dc thermal resistance applies when one output transistor is operating continuously. The ac thermal resistance applies with the output transistors conducting alternately at a
high enough frequency that the peak capability of neither
transistor is exceeded.
Supply current: The current required from the power
source to operate the amplifier with the output voltage and
current at zero.
The design of heat sink is beyond the scope of this work.
Convection-cooled heat sinks are available commercially,
and their manufacturers should be consulted for ratings. The
preceding figure is a rough guide for temperature rise as a
function of fin area (both sides) available for convection cooling.
Proper mounting of the IC is required to minimize the thermal
drop between the package and the heat sink. The heat sink
must also have enough metal under the package to conduct
heat from the center of the package bottom to the fins without excessive temperature drop.
A thermal grease such as Wakefield type 120 or Thermalloy
Thermacote should be used when mounting the package to
the heat sink. Without this compound, thermal resistance will
be no better than 0.5˚C/W, and probably much worse. With
the compound, thermal resistance will be 0.2˚C/W or less,
assuming under 0.005 inch combined flatness runout for the
package and heat sink. Proper torquing of the mounting
bolts is important. Four to six inch-pounds is recommended.
Should it be necessary to isolate V− from the heat sink, an
insulating washer is required. Hard washers like berylium oxide, anodized aluminum and mica require the use of thermal
compound on both faces. Two-mil mica washers are most
common, giving about 0.4˚C/W interface resistance with the
compound. Silicone-rubber washers are also available. A
0.5˚C/W thermal resistance is claimed without thermal com-
Equivalent Schematic
(excluding active protection circuitry)
LM12CL 80W Operational Amplifier
Physical Dimensions
inches (millimeters) unless otherwise noted
4-Lead TO-3 Steel Package (K)
Order Number LM12CLK
NS Package Number K04A
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significant injury to the user.
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