LT1074/LT1076 Step-Down Switching Regulator tions allow this device to be used as a positive-to-negative converter, a negative boost converter, and as a flyback converter. The switch output is specified to swing 40V below ground, allowing the LT1074 to drive a tappedinductor in the buck mode with output currents up to 10A. FEATURES ■ ■ ■ ■ ■ ■ ■ ■ 5A On-Board Switch (LT1074) 100kHz Switching Frequency Greatly Improved Dynamic Behavior Available in Low Cost 5 and 7-Lead Packages Only 8.5mA Quiescent Current Programmable Current Limit Operates Up to 60V Input Micropower Shutdown Mode The LT1074 uses a true analog multiplier in the feedback loop. This makes the device respond nearly instantaneously to input voltage fluctuations and makes loop gain independent of input voltage. As a result, dynamic behavior of the regulator is significantly improved over previous designs. UO APPLICATI ■ ■ ■ ■ ■ S Buck Converter with Output Voltage Range of 2.5V to 50V Tapped-Inductor Buck Converter with 10A Output at 5V Positive-to-Negative Converter Negative Boost Converter Multiple Output Buck Converter U DESCRIPTIO The LT1074 is a 5A (LT1076 is rated at 2A) monolithic bipolar switching regulator which requires only a few external parts for normal operation. The power switch, all oscillator and control circuitry, and all current limit components, are included on the chip. The topology is a classic positive “buck” configuration but several design innova- On-chip pulse by pulse current limiting makes the LT1074 nearly bust-proof for output overloads or shorts. The input voltage range as a buck converter is 8V to 60V, but a selfboot feature allows input voltages as low as 5V in the inverting and boost configurations. The LT1074 is available in low cost TO-220 or TO-3 packages with frequency pre-set at 100kHz and current limit at 6.5A (LT1076 = 2.6A). A 7-pin TO-220 package is also available which allows current limit to be adjusted down to zero. In addition, full micropower shutdown can be programmed. See Application Note 44 for design details. A fixed 5V output, 2A version is also available. See LT1076-5. UO TYPICAL APPLICATI Buck Converter Efficiency Basic Positive Buck Converter LT1074 L1** 50 µH (LT1074) 100 µH (LT1076) VIN MBR745* LT1074 GND + VC FB R3 2.7k C3 † 200 µF 5V 5A VSW C2 0.01µ F R1 2.8k 1% R2 2.21k 1% + C1 500 µF 25V *USE MBR340 FOR LT1076 **COILTRONICS #50-2-52 (LT1074) #100-1-52 (LT1076) PULSE ENGINEERING, INC. #PE-92114 (LT1074) #PE-92102 (LT1076) HURRICANE #HL-AK147QQ (LT1074) #HL-AG210LL (LT1076) † RIPPLE CURRENT RATING ≥ I OUT / 2 EFFICIENCY (%) 10V TO 40V 100 VOUT = 12V, VIN = 20V 90 80 VOUT = 5V, VIN = 15V 70 L = 50 µ H TYPE 52 CORE DIODE = MBR735 60 50 0 LT1074 • TA01 1 2 3 5 4 6 OUTPUT LOAD CURRENT (A) LT1074 • TPC27 1 LT1074/LT1076 W W W AXI U U ABSOLUTE RATI GS Input Voltage LT1074/ LT1076 .................................................. 45V LT1074HV/76HV.................................................. 64V Switch Voltage with Respect to Input Voltage LT1074/ 76 .......................................................... 64V LT1074HV/76HV.................................................. 75V Switch Voltage with Respect to Ground Pin (VSW Negative) LT1074/76 (Note 6) ............................................. 35V LT1074HV/76HV (Note 6).................................... 45V Feedback Pin Voltage ..................................... –2V, +10V Shutdown Pin Voltage (Not to Exceed VIN) .............. 40V ILIM Pin Voltage (Forced) ............................................ 5.5V Maximum Operating Ambient Temperature Range LT1074C/76C, LT1074HVC/76HVC ............ 0°C to 70°C LT1074I/76I, LT1074HVI/76HVI ............. –40°C to 85°C LT1074M/76M, LT1074HVM/76HVM ... –55°C to 125°C Maximum Operating Junction Temperature Range LT1074C/76C, LT1074HVC/76HVC .......... 0°C to 125°C LT1074I/76I, LT1074HVI/76HVI ........... –40°C to 125°C LT1074M/76M, LT1074HVM/76HVM ... – 55°C to 150°C Maximum Storage Temperature ................ –65°C to 150°C Lead Temperature (Soldering, 10 sec) ..................... 300°C U W U PACKAGE/ORDER I FOR ATIO FRONT VIEW VIN VSW GND VC FB/SENSE 5 4 3 2 1 ORDER PART NUMBER ORDER PART NUMBER BOTTOM VIEW LT1076CQ VC 1 Q PACKAGE 5-LEAD PLASTIC DD LT1076: θJC = 4°C/W, θJA = 30°C/W* 7 6 5 4 3 2 1 SHDN VC FB/SENSE GND ILIM VSW VIN LT1076CR LT1076HVCR R PACKAGE 7-LEAD PLASTIC DD FRONT VIEW 4 2 7 5 3 1 VC SHUTDOWN FB GND ILIM VSW VIN Y PACKAGE, 7-LEAD TO-220 LT1074: θJC = 2.5°C/W, θJA = 50°C/W LT1076: θJC = 4°C/W, θJA = 50°C/W CASE IS GND VSW K PACKAGE, 4-LEAD TO-3 METAL CAN LT1074: θJC = 2.5°C/W, θJA = 35°C/W LT1076: θJC = 4°C/W, θJA = 35°C/W FRONT VIEW LT1076: θJC = 4°C/W, θJA = 30°C/W* 6 2 3 4 FB FRONT VIEW LT1074CK LT1074HVCK LT1074MK LT1074HVMK LT1076CK LT1076HVCK LT1076MK LT1076HVMK VIN LT1074CY LT1074HVCY LT1074IY LT1074HVIY LT1076CY LT1076HVCY 5 4 3 2 1 LT1074CT LT1074HVCT LT1074IT LT1074HVIT LT1076CT LT1076HVCT LT1076IT VIN VSW GND VC FB T PACKAGE, 5-LEAD T0-220 LEADS ARE FORMED STANDARD FOR STRAIGHT LEADS, ORDER FLOW 06 LT1074: θJC = 2.5°C/W, θJA = 50°C/W LT1076: θJC = 4°C/W, θJA = 50°C/W * Assumes package is soldered to 0.5 IN2 of 1 oz. copper over internal ground plane or over back side plane. ELECTRICAL CHARACTERISTICS Tj = 25°C, VIN = 25V, unless otherwise noted. PARAMETER CONDITIONS Switch “On” Voltage (Note 1) LT1074 ISW = 1A, Tj ≥ 0°C ISW = 1A, Tj < 0°C ISW = 5A, Tj ≥ 0°C ISW = 5A, Tj < 0°C LT1076 ISW = 0.5A ISW = 2A 2 MIN ● ● TYP MAX UNITS 1.85 2.1 2.3 2.5 V V V V 1.2 1.7 V V LT1074/LT1076 ELECTRICAL CHARACTERISTICS Tj = 25°C, VIN = 25V, unless otherwise noted. PARAMETER CONDITIONS Switch “Off” Leakage LT1074 VIN ≤ 25V, VSW = 0 VIN = VMAX, VSW = 0 (Note 7) MIN LT1076 VIN = 25V, VSW = 0 VIN = VMAX, VSW = 0 (Note 7) TYP MAX 5 10 300 500 UNITS µA µA 150 250 µA µA Supply Current (Note 2) VFB = 2.5V, VIN ≤ 40V 40V < VIN < 60V VSHUT = 0.1V (Device Shutdown) (Note 8) ● ● ● 8.5 9 140 11 12 300 mA mA µA Minimum Supply Voltage Normal Mode Startup Mode (Note 3) ● ● 7.3 3.5 8 4.8 V V Switch Current Limit (Note 4) LT1074 ILIM Open RLIM = 10k (Note 5) RLIM = 7k (Note 5) ● 5.5 6.5 4.5 3 8.5 A A A LT1076 ILIM Open RLIM = 10k (Note 5) RLIM = 7k (Note 5) ● 2 2.6 1.8 1.2 3.2 A A A ● 85 90 Tj ≤ 125°C Tj > 125°C VFB = 0V through 2kΩ (Note 4) 90 85 85 100 ● ● Switching Frequency Line Regulation 8V ≤ VIN ≤ VMAX (Note 7) ● Error Amplifier Voltage Gain (Note 6) 1V ≤ VC ≤ 4V Maximum Duty Cycle Switching Frequency % 110 120 125 kHz kHz kHz kHz 0.1 %/V 20 0.03 2000 Error Amplifier Transconductance V/V µmho 3700 5000 8000 100 0.7 140 1 225 1.6 µA mA 0.5 2 µA 2.155 2.21 2.265 Error Amplifier Source and Sink Current Source (VFB = 2V) Sink (VFB = 2.5V) Feedback Pin Bias Current VFB = VREF ● Reference Voltage VC = 2V ● Reference Voltage Tolerance VREF (Nominal) = 2.21V All Conditions of Input Voltage, Output Voltage, Temperature and Load Current ● 8V ≤ VIN ≤ VMAX (Note 7) ● Over Temperature ● Shutdown Pin Current VSH = 5V VSH ≤ VTHRESHOLD (≅2.5V) ● ● 5 10 20 50 µA µA Shutdown Thresholds Switch Duty Cycle = 0 Fully Shut Down ● ● 2.2 0.1 2.45 0.3 2.7 0.5 V V Thermal Resistance Junction to Case LT1074 LT1076 2.5 4.0 °C/W °C/W Reference Voltage Line Regulation VC Voltage at 0% Duty Cycle ± 0.5 ±1 0.005 ± 1.5 ± 2.5 0.02 1.5 –4 Multiplier Reference Voltage V % % %/V V mV/°C 24 The ● denotes the specifications which apply over the full operating temperature range. Note 1: To calculate maximum switch “on” voltage at currents between low and high conditions, a linear interpolation may be used. Note 2: A feedback pin voltage (VFB) of 2.5V forces the VC pin to its low clamp level and the switch duty cycle to zero. This approximates the zero load condition where duty cycle approaches zero. Note 3: Total voltage from VIN pin to ground pin must be ≥ 8V after startup for proper regulation. V Note 4: Switch frequency is internally scaled down when the feedback pin voltage is less than 1.3V to avoid extremely short switch on times. During testing, VFB is adjusted to give a minimum switch on time of 1µs. Note 5: ILIM ≈ RLIM – 1k R – 1k (LT1074), ILIM ≈ LIM (LT1076). 2k 5.5k Note 6: Switch to input voltage limitation must also be observed. Note 7: VMAX = 40V for the LT1074/76 and 60V for the LT1074HV/76HV. Note 8: Does not include switch leakage. 3 LT1074/LT1076 W BLOCK DIAGRA INPUT SUPPLY LT1074 320 µ A 10 µ A 0.3V + µ-POWER SHUTDOWN – 6V REGULATOR AND BIAS 500 Ω 6V TO ALL CIRCUITRY CURRENT LIMIT COMP CURRENT LIMIT SHUTDOWN 2.35V + 0.04 + C2 250 Ω – – I LIM* SHUTDOWN* 4.5V FREQ SHIFT 10k R R/S Q S LATCH R 100kHz OSCILLATOR SYNC G1 3V(p-p) VIN + + 2.21V – FB Z ANALOG X MULTIPLIER XY Z Y A1 ERROR AMP VC 400 Ω 15 Ω C1 – PULSE WIDTH COMPARATOR SWITCH OUTPUT (VSW ) 24V (EQUIVALENT) LT1076 0.1 Ω *AVAILABLE ON PACKAGES WITH PIN COUNTS GREATER THAN 5. 100 Ω SWITCH OUTPUT (VSW ) LT1074 • BD01 4 LT1074/LT1076 U W BLOCK DIAGRA DESCRIPTIO A switch cycle in the LT1074 is initiated by the oscillator setting the R/S latch. The pulse that sets the latch also locks out the switch via gate G1. The effective width of this pulse is approximately 700ns, which sets the maximum switch duty cycle to approximately 93% at 100kHz switching frequency. The switch is turned off by comparator C1, which resets the latch. C1 has a sawtooth waveform as one input and the output of an analog multiplier as the other input. The multiplier output is the product of an internal reference voltage, and the output of the error amplifier, A1, divided by the regulator input voltage. In standard buck regulators, this means that the output voltage of A1 required to keep a constant regulated output is independent of regulator input voltage. This greatly improves line transient response, and makes loop gain independent of input voltage. The error amplifier is a transconductance type with a GM at null of approximately 5000µmho. Slew current going positive is 140µA, while negative slew current is about 1.1mA. This asymmetry helps prevent overshoot on start-up. Overall loop frequency compensation is accomplished with a series RC network from VC to ground. Switch current is continuously monitored by C2, which resets the R/S latch to turn the switch off if an overcurrent condition occurs. The time required for detection and switch turn off is approximately 600ns. So minimum switch “on” time in current limit is 600ns. Under dead shorted output conditions, switch duty cycle may have to be as low as 2% to maintain control of output current. This would require switch on time of 200ns at 100kHz switching frequency, so frequency is reduced at very low output voltages by feeding the FB signal into the oscillator and creating a linear frequency downshift when the FB signal drops below 1.3V. Current trip level is set by the voltage on the ILIM pin which is driven by an internal 320µA current source. When this pin is left open, it self-clamps at about 4.5V and sets current limit at 6.5A for the LT1074 and 2.6A for the LT1076. In the 7-pin package an external resistor can be connected from the ILIM pin to ground to set a lower current limit. A capacitor in parallel with this resistor will soft start the current limit. A slight offset in C2 guarantees that when the ILIM pin is pulled to within 200mV of ground, C2 output will stay high and force switch duty cycle to zero. The “Shutdown” pin is used to force switch duty cycle to zero by pulling the ILIM pin low, or to completely shut down the regulator. Threshold for the former is approximately 2.35V, and for complete shutdown, approximately 0.3V. Total supply current in shutdown is about 150µA. A 10µA pull-up current forces the shutdown pin high when left open. A capacitor can be used to generate delayed startup. A resistor divider will program “undervoltage lockout” if the divider voltage is set at 2.35V when the input is at the desired trip point. The switch used in the LT1074 is a Darlington NPN (single NPN for LT1076) driven by a saturated PNP. Special patented circuitry is used to drive the PNP on and off very quickly even from the saturation state. This particular switch arrangement has no “isolation tubs” connected to the switch output, which can therefore swing to 40V below ground. 5 LT1074/LT1076 U W TYPICAL PERFOR A CE CHARACTERISTICS VC Pin Characteristics 200 2.0 150 1.5 VC Pin Characteristics Feedback Pin Characteristics 500 400 300 VFB ADJUSTED FOR IC = 0 AT VC = 2V 1.0 50 0 SLOPE ≈ 400k Ω –50 VFB ≥ 2.5V 200 CURRENT (µA) CURRENT (mA) CURRENT (mA) 100 0.5 0 –0.5 START OF FREQUENCY SHIFTING 100 0 –100 –200 –100 –1.0 VFB ≤ 2V –300 –1.5 –150 –400 –500 –2.0 –200 0 1 2 3 5 4 6 7 8 0 9 1 2 3 5 4 7 6 30 –5 20 –10 –10 –20 Tj = 25°C CURRENT FLOWS OUT OF SHUTDOWN PIN –15 0 SHUTDOWN THRESHOLD –20 –25 7 8 9 10 Tj = 25°C –100 –150 –200 –250 –300 –350 50 60 70 0 80 0.5 1.0 1.5 VOLTAGE (V) 2.0 2.5 3.0 3.5 4.0 VOLTAGE (V) LT1074 • TPC04 Supply Current 18 INPUT CURRENT (mA) 16 14 DEVICE NOT SWITCHING 12 VC = 1V 10 8 6 4 2 0 0 10 20 30 40 –400 –2 –1 0 1 2 3 4 5 6 7 8 VOLTAGE (V) LT1074 • TPC05 20 50 60 INPUT VOLTAGE (V) LT1074 • TPC11 6 6 50 –40 40 5 ILIM Pin Characteristics –35 –40 30 4 100 –30 DETAILS OF THIS AREA SHOWN IN OTHER GRAPH 20 3 LT1074 • TPC03 CURRENT ( µ A) THIS POINT MOVES WITH VIN 10 2 –50 VIN = 50V 0 1 VOLTAGE (V) Shutdown Pin Characteristics 0 CURRENT ( µ A) CURRENT ( µ A) Shutdown Pin Characteristics 40 –30 0 LT1074 • TPC02 LT1074 • TPC01 0 9 VOLTAGE (V) VOLTAGE (V) 10 8 LT1074 • TPC06 LT1074/LT1076 U W TYPICAL PERFOR A CE CHARACTERISTICS Reference Voltage vs Temperature Supply Current (Shutdown) 300 3.0 Tj = 25° C 2.24 250 2.5 200 150 "ON" VOLTAGE (V) 2.23 2.22 VOLTAGE (V) 2.21 2.20 100 LT1074 1.5 LT1076 1.0 2.18 2.17 –50 –25 0 0 10 20 30 40 50 60 INPUT VOLTAGE (V) 0 25 50 75 8k 10 TRI WAVE –20 SQUARE WAVE –40 –50 –60 θ 5k 120 150 115 100 110 0 GM 2k –100 90 1k –150 85 0 10k 100k –200 10M 1M 95 80 –50 –25 0 25 50 75 100 125 150 JUNCTION TEMPERATURE ( ° C) FREQUENCY (Hz) PEAK-TO-PEAK RIPPLE AT FB PIN (mV) 6 100 –50 1k 5 105 3k 60 80 100 120 140 160 180 200 LT1074 • TPC16 LT1074 • TPC18 LT1074 • TPC17 Feedback Pin Frequency Shift Current Limit vs Temperature* 160 8 140 7 OUTPUT CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) 200 50 4k –80 120 100 80 150°C –55°C 40 4 Switching Frequency vs Temperature Error Amplifier Phase and GM 6k –70 60 3 LT1074 • TPC28 PHASE ( °) TRANSCONDUCTANCE ( µ mho) 0 20 40 2 SWITCH CURRENT (A) 7k 0 1 LT1074 • TPC14 20 –30 0 JUNCTION TEMPERATURE (°C) Reference Shift with Ripple Voltage –10 0.5 100 125 150 LT1074 • TPC13 CHANGE IN REFERENCE VOLTAGE (mV) 2.0 2.19 50 FREQUENCY (kHz) INPUT CURRENT ( µ A) Switch “On” Voltage 2.25 25°C I LIM PIN OPEN 6 5 R LIM = 10kΩ 4 3 2 RLIM = 5k Ω 1 20 0 0 0.5 1.0 1.5 2.0 2.5 3.0 FEEDBACK PIN VOLTAGE (V) LT1074 • TPC19 *MULTIPLY CURRENTS BY 0.4 FOR LT1076 0 –50 –25 0 25 50 75 100 125 150 JUNCTION TEMPERATURE (° C) LT1074 • TPC22 7 LT1074/LT1076 U U PI DESCRIPTIO S VIN PIN The VIN pin is both the supply voltage for internal control circuitry and one end of the high current switch. It is important, especially at low input voltages, that this pin be bypassed with a low ESR, and low inductance capacitor to prevent transient steps or spikes from causing erratic operation. At full switch current of 5A, the switching transients at the regulator input can get very large as shown in Figure 1. Place the input capacitor very close to the regulator and connect it with wide traces to avoid extra inductance. Use radial lead capacitors. ∆VOUT = (∆VGND ) (VOUT ) 2 .21 To ensure good load regulation, the ground pin must be connected directly to the proper output node, so that no high currents flow in this path. The output divider resistor should also be connected to this low current connection line as shown in Figure 2. LT1074 GND ( dIdt)(L P) FB R2 STEP = ( I SW )(ESR) RAMP = ( I SW )(TON ) C HIGH CURRENT RETURN PATH NEGATIVE OUTPUT NODE WHERE LOAD REGULATION WILL BE MEASURED LT1074 • PD01 Figure 1. Input Capacitor Ripple LP = Total inductance in input bypass connections and capacitor. “Spike” height (dI/dt • LP) is approximately 2V per inch of lead length for LT1074 and 0.8V per inch for LT1076. “Step” for ESR = 0.05Ω and ISW = 5A is 0.25V. “Ramp” for C = 200µF, TON = 5µs, and ISW = 5A, is 0.12V. Input current on the VIN Pin in shutdown mode is the sum of actual supply current (≈140µA, with a maximum of 300µA), and switch leakage current. Consult factory for special testing if shutdown mode input current is critical. GROUND PIN It might seem unusual to describe a ground pin, but in the case of regulators, the ground pin must be connected properly to ensure good load regulation. The internal reference voltage is referenced to the ground pin; so any error in ground pin voltage will be multiplied at the output; 8 LT1074 • PD02 Figure 2. Proper Ground Pin Connection FEEDBACK PIN The feedback pin is the inverting input of an error amplifier which controls the regulator output by adjusting duty cycle. The non-inverting input is internally connected to a trimmed 2.21V reference. Input bias current is typically 0.5µA when the error amplifier is balanced (IOUT = 0). The error amplifier has asymmetrical GM for large input signals to reduce startup overshoot. This makes the amplifier more sensitive to large ripple voltages at the feedback pin. 100mVp-p ripple at the feedback pin will create a 14mV offset in the amplifier, equivalent to a 0.7% output voltage shift. To avoid output errors, output ripple (P-P) should be less than 4% of DC output voltage at the point where the output divider is connected. See the “Error Amplifier” section for more details. Frequency Shifting at the Feedback Pin The error amplifier feedback pin (FB) is used to downshift the oscillator frequency when the regulator output voltage LT1074/LT1076 U U PI DESCRIPTIO S is low. This is done to guarantee that output short circuit current is well controlled even when switch duty cycle must be extremely low. Theoretical switch “on” time for a buck converter in continuous mode is; V + VD t ON = OUT VIN • f VD = Catch diode forward voltage ( ≈ 0.5V) f = Switching frequency At f = 100kHz, tON must drop to 0.2µs when VIN = 25V and the output is shorted (VOUT = 0V). In current limit, the LT1074 can reduce tON to a minimum value of ≈ 0.6µs, much too long to control current correctly for V OUT = 0. To correct this problem, switching frequency is lowered from 100kHz to 20kHz as the FB pin drops from 1.3V to 0.5V. This is accomplished by the circuitry shown in Figure 3. TO OSCILLATOR VOUT +2V VC + ERROR AMPLIFIER – 2.21V Q1 R1 R3 3k EXTERNAL DIVIDER FB R2 2.21k SHUTDOWN PIN The shutdown pin is used for undervoltage lockout, micropower shutdown, soft start, delayed start, or as a general purpose on/off control of the regulator output. It controls switching action by pulling the ILIM pin low, which forces the switch to a continuous “off” state. Full micropower shutdown is initiated when the shutdown pin drops below 0.3V. The V/I characteristics of the shutdown pin are shown in Figure 4. For voltages between 2.5V and ≈VIN, a current of 10µA flows out of the shutdown pin. This current increases to ≈25µA as the shutdown pin moves through the 2.35V threshold. The current increases further to ≈ 30µA at the 0.3V threshold, then drops to ≈15µA as the shutdown voltage falls below 0.3V. The 10µA current source is included to pull the shutdown pin to its high or default state when left open. It also provides a convenient pullup for delayed start applications with a capacitor on the shutdown pin. When activated, the typical collector current of Q1 in Figure 5, is ≈ 2mA. A soft start capacitor on the ILIM pin will delay regulator shutdown in response to C1, by ≈(5V)(CLIM)/2mA. Soft start after full micropower shutdown is ensured by coupling C2 to Q1. 0 Tj = 25°C CURRENT FLOWS OUT OF SHUTDOWN PIN LT1074 • PD03 –5 Figure 3. Frequency Shifting CURRENT ( µ A) Q1 is off when the output is regulating (VFB = 2.21V). As the output is pulled down by an overload, VFB will eventually reach 1.3V, turning on Q1. As the output continues to drop, Q1 current increases proportionately and lowers the frequency of the oscillator. Frequency shifting starts when the output is ≈ 60% of normal value, and is down to its minimum value of ≅ 20kHz when the output is ≅ 20% of normal value. The rate at which frequency is shifted is determined by both the internal 3k resistor R3 and the external divider resistors. For this reason, R2 should not be increased to more than 4kΩ, if the LT1074 will be subjected to the simultaneous conditions of high input voltage and output short circuit. –10 –15 SHUTDOWN THRESHOLD –20 –25 –30 –35 –40 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 VOLTAGE (V) LT1074 • TPC05 Figure 4. Shutdown Pin Characteristics 9 LT1074/LT1076 U U PI DESCRIPTIO S Hysteresis in undervoltage lockout may be accomplished by connecting a resistor (R3) from the ILIM pin to the shutdown pin as shown in Figure 7. D1 prevents the shutdown divider from altering current limit. V IN 300 µ A 10 µ A SHUTDOWN PIN – ILIM PIN C1 2.3V VIN R1 + SHUT Q1 6V – EXTERNAL CLIM R3 I LIM R2 C2 0.3V LT1074 D1* OPTIONAL CURRENT LIMIT RESISTOR + LT1074 • PD09 *1N4148 TO TOTAL REGULATOR SHUTDOWN Figure 7. Adding Hysteresis LT1074 • PD07 Figure 5. Shutdown Circuitry Undervoltage Lockout Undervoltage lockout point is set by R1 and R2 in Figure 6. To avoid errors due to the 10µA shutdown pin current, R2 is usually set at 5k, and R1 is found from: R1 = R 2 (VTP – VSH ) VSH R1 Trip Point = VTP = 2 . 35 V 1 + R 2 If R3 is added, the lower trip point (VIN descending) will be the same. The upper trip point (VUTP) will be; R1 R1 R1 VUTP = VSH 1 + + – 0. 8 V R 3 R 2 R 3 If R1 and R2 are chosen, R3 is given by VTP = Desired undervoltage lockout voltage. VSH = Threshold for lockout on the shutdown pin = 2.45V. If quiescent supply current is critical, R2 may be increased up to 15kΩ, but the denominator in the formula for R2 should replace VSH with VSH – (10µA)(R2). R1 VIN R3 = (VSH – 0.8 V) (R1) R1 VUTP – VSH 1 + R 2 Example: An undervoltage lockout is required such that the output will not start until VIN = 20V, but will continue to operate until VIN drops to 15V. Let R2 = 2.32k. SHUT LT1074 R2 5k GND LT1074 • PD08 Figure 6. Undervoltage Lockout 10 ( R1 = 2 . 32k R3 = ( ) (15V – 2.35V) = 12.5k 2 . 35 V 2 . 35 – 0. 8 12 . 5 )( ) 12 . 5 20 – 2 . 35 1 + 2 . 32 = 3. 9 k LT1074/LT1076 U U PI DESCRIPTIO S ILIM PIN The ILIM pin is used to reduce current limit below the preset value of 6.5A. The equivalent circuit for this pin is shown in Figure 8. TO LIMIT CIRCUIT from forcing current back into the ILIM pin. To calculate a value for RFB, first calculate RLIM, then RFB; RFB = VIN *Change 0.44 to 0.16, and 0.5 to 0.18 for LT1076. 320 µ A D2 Example: ILIM = 4A, ISC = 1.5A, RLIM = (4)(2k) + 1k = 9k Q1 D1 R1 8K (ISC − 0.44*) (RL ) (R inkΩ) L 0.5 * (RL – 1kΩ) – ISC 4.3V D3 6V R FB = (15. − 0.44) (9kΩ) . 0.5 (9k − 1k ) − 15 I LIM LT1047 • PD12 VOUT Figure 8. ILIM Pin Circuit LT1074 When ILIM is left open, the voltage at Q1 base clamps at 5V through D2. Internal current limit is determined by the current through Q1. If an external resistor is connected between ILIM and ground, the voltage at Q1 base can be reduced for lower current limit. The resistor will have a voltage across it equal to (320µA) (R), limited to ≈ 5V when clamped by D2. Resistance required for a given current limit is RLIM = ILIM (2kΩ) + 1kΩ (LT1074) RLIM = ILIM (5.5kΩ) + 1kΩ (LT1076) As an example, a 3A current limit would require 3A (2k) + 1k = 7kΩ for the LT1074. The accuracy of these formulas is ±25% for 2A ≤ ILIM ≤ 5A (LT1074) and 0.7A ≤ ILIM ≤ 1.8A (LT1076), so ILIM should be set at least 25% above the peak switch current required. Foldback current limiting can be easily implemented by adding a resistor from the output to the ILIM pin as shown in Figure 9. This allows full desired current limit (with or without RLIM) when the output is regulating, but reduces current limit under short circuit conditions. A typical value for RFB is 5kΩ, but this may be adjusted up or down to set the amount of foldback. D2 prevents the output voltage FB I LIM R FB R LIM D2 1N4148 LT1074 • PD13 Figure 9. Foldback Current Limit ERROR AMPLIFIER The error amplifier in Figure 10 is a single stage design with added inverters to allow the output to swing above and below the common mode input voltage. One side of the amplifier is tied to a trimmed internal reference voltage of 2.21V. The other input is brought out as the FB (feedback) pin. This amplifier has a GM (voltage “in” to current “out”) transfer function of ≈5000µmho. Voltage gain is determined by multiplying GM times the total equivalent output loading, consisting of the output resistance of Q4 and Q6 in parallel with the series RC external frequency compensation network. At DC, the external RC is ignored, and with a parallel output impedance for Q4 and Q6 of 400kΩ, voltage gain is ≈ 2000. At frequencies above a few hertz, voltage gain is determined by the external compensation, RC and CC. 11 LT1074/LT1076 U U PI DESCRIPTIO S 5.8V Q4 90 µ A 90 µ A Q3 50 µ A Q1 Q2 X1.8 VC D1 FB 50 µ A 90 µ A D2 Q6 2.21V EXTERNAL FREQUENCY COMPENSATION RC 140 µ A CC 300 Ω ALL CURRENTS SHOWN ARE AT NULL CONDITION LT1074 • PD11 Figure 10. Error Amplifier AV = Gm at midfrequencies 2π • f • CC A V = Gm• RC at high frequencies Phase shift from the FB pin to the VC pin is 90° at midfrequencies where the external CC is controlling gain, then drops back to 0° (actually 180° since FB is an inverting input) when the reactance of CC is small compared to RC. The low frequency “pole” where the reactance of CC is equal to the output impedance of Q4 and Q6 (rO), is 1 fPOLE = r0 ≈ 400kΩ 2 π • r0 • C Although fPOLE varies as much as 3:1 due to rO variations, mid-frequency gain is dependent only on GM, which is specified much tighter on the data sheet. The higher frequency “zero” is determined solely by RC and CC. fZERO = 12 1 2π • RC • CC The error amplifier has asymmetrical peak output current. Q3 and Q4 current mirrors are unity gain, but the Q6 mirror has a gain of 1.8 at output null and a gain of 8 when the FB pin is high (Q1 current = 0). This results in a maximum positive output current of 140µA and a maximum negative (sink) output current of ≅ 1.1mA. The asymmetry is deliberate — it results in much less regulator output overshoot during rapid start-up or following the release of an output overload. Amplifier offset is kept low by area scaling Q1 and Q2 at 1.8:1. Amplifier swing is limited by the internal 5.8V supply for positive outputs and by D1 and D2 when the output goes low. Low clamp voltage is approximately one diode drop (≈ 0.7V – 2mV/°C). Note that both the FB pin and the VC pin have other internal connections. Refer to the frequency shifting and sychronizing discussions. LT1074/LT1076 UO TYPICAL APPLICATI S Tapped-Inductor Buck Converter L2 5µH L1* † VIN VIN VSW 20V - 35V + VC C3 200 µ F 50V R1 2.8k D1** + FB R3 1k C2 0.2 µ F VOUT 5V, 10A† 1 D2 35V 5W LT1074HV GND 3 D3 1N5819 R2 2.21k C1 4400 µF (2 EA 2200 µF, 16V) + C4 390 µF 16V 0.01µF *PULSE ENGINEERING #PE–65282 **MOTOROLA MBR2030CTL †IF INPUT VOLTAGE IS BELOW 20V, MAXIMUM OUTPUT CURRENT WILL BE REDUCED. SEE AN44 LT1074 • TA02 Positive-to-Negative Converter with 5V Output + + VIN 4.5V to 40V C1 220µF 50V L1 25µH 5A ✝ ✝ VIN + LT1074 GND VC R3* 2.74k R1** 5.1k VSW R2** 10k C2 1000 µ F 10V OPTIONAL FILTER VFB ✝ 5µH D1 MBR745 C3 0.1µF C4** 0.01µF R4 1.82k* – 200µF + 10V –5V,1A*** * = 1% FILM RESISTORS D1 = MOTOROLA-MBR745 C1 = NICHICON-UPL1C221MRH6 C2 = NICHICON-UPL1A102MRH6 L1 = COILTRONICS-CTX25-5-52 ✝ ✝✝ LOWER REVERSE VOLTAGE RATING MAY BE USED FOR LOWER INPUT VOLTAGES. LOWER CURRENT RATING IS ALLOWED FOR LOWER OUTPUT CURRENT. SEE AN44. LOWER CURRENT RATING MAY BE USED FOR LOWER OUTPUT CURRENT. SEE AN44. ** R1, R2, AND C4 ARE USED FOR LOOP FREQUENCY COMPENSATION WITH LOW INPUT VOLTAGE, BUT R1 AND R2 MUST BE INCLUDED IN THE CALCULATION FOR OUTPUT VOLTAGE DIVIDER VALUES. FOR HIGHER OUTPUT VOLTAGES, INCREASE R1, R2, AND R3 PROPORTIONATELY. FOR INPUT VOLTAGE > 10V, R1, R2, AND C4 CAN BE ELIMINATED, AND COMPENSATION IS DONE TOTALLY ON THE VC PIN. R3 = VOUT –2.37 (K Ω) R1 = (R3) (1.86) R2 = (R3) (3.65) * ** MAXIMUM OUTPUT CURRENT OF 1A IS DETERMINED BY MINIMUM INPUT VOLTAGE OF 4.5V. HIGHER MINIMUM INPUT VOLTAGE WILL ALLOW MUCH HIGHER OUTPUT CURRENTS. SEE AN44. LT1074 • TA03 13 LT1074/LT1076 UO TYPICAL APPLICATI S Negative Boost Converter R1 12.7k 100pF VIN FB LT1074 GND 200µF 15V R2 2.21k V VC SW + + C2 C3 L1 25µH 1nF R3 750Ω 0.01µF + C1 1000 µ F 25V D1* VOUT** –15V –VIN –5 TO –15V *MBR735 ** IOUT (MAX) = 1A-3A DEPENDING ON INPUT VOLTAGE. SEE AN44 + 5µH 100µF OPTIONAL OUTPUT FILTER LT1074 • TA04 U PACKAGE DESCRIPTIO Dimensions in inches (milimeters) unless otherwise noted. R Package, 7-Lead PLASTIC DD Q Package, 5-Lead PLASTIC DD 0.401 ± 0.015 (10.185 ± 0.381) 0.175 ± 0.008 (4.445 ± 0.203) 0.060 (1.524) 15° TYP ( +0.012 0.331 –0.020 +0.305 8.407 –0.508 0.059 (1.499) TYP ) 0.060 (1.524) 0.401 ± 0.015 (10.185 ± 0.381) 0.050 ± 0.008 (1.270 ± 0.203) ( +0.008 0.004 –0.004 +0.203 0.102 –0.102 ) 0.175 ± 0.008 (4.445 ± 0.203) 15° TYP ( +0.012 0.331 –0.020 +0.305 8.407 –0.508 0.059 (1.499) TYP ) ( +0.305 3.632 –0.508 14 ) 0.067 ± 0.010 (1.702 ± 0.254) 0.032 ± 0.008 (0.813 ± 0.203) 0.022 ± 0.005 (0.559 ± 0.127) 0.050 ± 0.012 (1.270 ± 0.305) ( +0.012 0.143 –0.020 +0.305 3.632 –0.508 DD5 0693 ( +0.008 0.004 –0.004 +0.203 0.102 –0.102 ) 0.105 ± 0.008 (2.667 ± 0.203) 0.105 ± 0.008 (2.667 ± 0.203) +0.012 0.143 –0.020 0.050 ± 0.008 (1.270 ± 0.203) ) 0.050 ± 0.010 (1.270 ± 0.254) 0.030 ± 0.008 (0.762 ± 0.203) 0.022 ± 0.005 (0.559 ± 0.127) 0.050 ± 0.012 (1.270 ± 0.305) DD7 0693 LT1074/LT1076 U PACKAGE DESCRIPTIO Dimensions in inches (milimeters) unless otherwise noted. K Package, 4-Lead TO-3 Metal Can 0.760 – 0.775 (19.30 – 19.69) 0.320 – 0.350 (8.13 – 8.89) 1.177 – 1.197 (29.90 – 30.40) 0.060 – 0.135 (1.524 – 3.429) 0.655 – 0.675 (16.64 – 19.05) 0.470 TP P.C.D. 0.151 – 0.161 (3.84 – 4.09) DIA 2 PLC 0.420 – 0.480 (10.67 – 12.19) 0.167 – 0.177 (4.24 – 4.49) R TYP 0.038 – 0.043 (0.965 – 1.09) 0.495 – 0.525 (12.57 – 13.34) R 72° 18° K4 0592 T Package, 5-Lead TO-220 0.380 – 0.420 (9.652 – 10.668) 0.139 – 0.153 (3.531 – 3.886) DIA 0.079 – 0.135 (2.007 – 3.429) 0.560 – 0.650 (14.224 – 16.510) 0.169 – 0.185 (4.293 – 4.699) 0.460 – 0.500 (11.68 – 12.70) 0.035 – 0.055 (0.889 – 1.397) 0.620 ± 0.020 (15.75 ± 0.508) 0.866 – 0.913 (21.996 – 23.190) 0.700 – 0.728 (17.780 – 18.491) 0.970 – 1.050 (24.64 – 26.67) 0.057 – 0.077 (1.448 – 1.956) 0.015 – 0.025 (0.381 – 0.635) 0.210 – 0.240 (5.334 – 6.096) 0.028 – 0.035 (0.711 – 0.889) 0.304 – 0.380 (7.722 – 9.652) 0.055 – 0.090 (1.397 – 2.286) 0.079 – 0.115 (2.007 – 2.921) 0.149 – 0.230 (3.785 – 5.842) T5 (FORMED) 0993 Y Package, 7-Lead Molded TO-220 0.390 – 0.410 (9.91 – 10.41) 0.147 – 0.155 (3.73 – 3.94) DIA 0.169 – 0.185 (4.29 – 4.70) 0.045 – 0.055 (1.14 – 1.40) 0.235 – 0.258 (5.97 – 6.55) 0.103 – 0.113 (2.62 – 2.87) 0.560 – 0.590 (14.22 – 14.99) 0.620 (15.75) TYP 0.700 – 0.728 (17.78 – 18.49) 0.152 – 0.202 (3.86 – 5.13) 0.260 – 0.320 (6.60 – 8.13) 0.026 – 0.036 (0.66 – 0.91) 0.045 – 0.055 (1.14 – 1.40) 0.016 – 0.022 (0.41 – 0.56) 0.135 – 0.165 (3.43 – 4.19) 0.095 – 0.115 (2.41 – 2.92) 0.155 – 0.195 (3.94 – 4.95) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. Y7 0893 15 LT1074/LT1076 U.S. Area Sales Offices NORTHEAST REGION Linear Technology Corporation One Oxford Valley 2300 E. Lincoln Hwy.,Suite 306 Langhorne, PA 19047 Phone: (215) 757-8578 FAX: (215) 757-5631 SOUTHEAST REGION Linear Technology Corporation 17060 Dallas Parkway Suite 208 Dallas, TX 75248 Phone: (214) 733-3071 FAX: (214) 380-5138 SOUTHWEST REGION Linear Technology Corporation 22141 Ventura Blvd. Suite 206 Woodland Hills, CA 91364 Phone: (818) 703-0835 FAX: (818) 703-0517 Linear Technology Corporation 266 Lowell St., Suite B-8 Wilmington, MA 01887 Phone: (508) 658-3881 FAX: (508) 658-2701 CENTRAL REGION Linear Technology Corporation Chesapeake Square 229 Mitchell Court, Suite A-25 Addison, IL 60101 Phone: (708) 620-6910 FAX: (708) 620-6977 NORTHWEST REGION Linear Technology Corporation 782 Sycamore Dr. Milpitas, CA 95035 Phone: (408) 428-2050 FAX: (408) 432-6331 International Sales Offices FRANCE Linear Technology S.A.R.L. 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The Coliseum, Riverside Way Camberley, Surrey GU15 3YL United Kingdom Phone: 44-276-677676 FAX: 44-276-64851 JAPAN Linear Technology KK 5F YZ Bldg. 4-4-12 Iidabashi, Chiyoda-Ku Tokyo, 102 Japan Phone: 81-3-3237-7891 FAX: 81-3-3237-8010 World Headquarters Linear Technology Corporation 1630 McCarthy Blvd. Milpitas, CA 95035-7487 Phone: (408) 432-1900 FAX: (408) 434-0507 0294 16 Linear Technology Corporation BA/GP 0494 2K REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7487 (408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977 LINEAR TECHNOLOGY CORPORATION 1994