AD AD9765AST

a
FEATURES
12-Bit Dual Transmit DAC
125 MSPS Update Rate
Excellent SFDR to Nyquist @ 5 MHz Output: 75 dBc
Excellent Gain and Offset Matching: 0.1%
Fully Independent or Single Resistor Gain Control
Dual Port or Interleaved Data
On-Chip 1.2 V Reference
Single 5 V or 3 V Supply Operation
Power Dissipation: 380 mW @ 5 V
Power-Down Mode: 50 mW @ 5 V
48-Lead LQFP
APPLICATIONS
Communications
Base Stations
Digital Synthesis
Quadrature Modulation
PRODUCT DESCRIPTION
The AD9765 is a dual port, high speed, two channel, 12-bit
CMOS DAC. It integrates two high quality 12-bit TxDAC+
cores, a voltage reference and digital interface circuitry into a small
48-lead LQFP package. The AD9765 offers exceptional ac and
dc performance while supporting update rates up to 125 MSPS.
12-Bit, 125 MSPS
Dual TxDAC+® D/A Converter
AD97651
FUNCTIONAL BLOCK DIAGRAM
DVDD
DCOM
WRT2
ACOM
“1”
LATCH
PORT1
WRT1
AVDD
DIGITAL
INTERFACE
“1”
DAC
MODE
IOUTA1
IOUTB1
REFERENCE
REFIO
FSADJ1
FSADJ2
GAINCTRL
BIAS
GENERATOR
SLEEP
“2”
DAC
IOUTA2
AD9765
“2”
LATCH
PORT2
CLK1
IOUTB2
CLK2
and provide a nominal full-scale current of 20 mA. The fullscale currents between each DAC are matched to within 0.1%.
The AD9765 is manufactured on an advanced low cost CMOS
process. It operates from a single supply of 3.0 V to 5.0 V and
consumes 380 mW of power.
The AD9765 has been optimized for processing I and Q data in
communications applications. The digital interface consists of
two double-buffered latches as well as control logic. Separate
write inputs allow data to be written to the two DAC ports
independent of one another. Separate clocks control the update
rate of the DACs.
PRODUCT HIGHLIGHTS
A mode control pin allows the AD9765 to interface to two separate
data ports, or to a single interleaved high speed data port. In interleaving mode the input data stream is demuxed into its original
I and Q data and then latched. The I and Q data is then converted by the two DACs and updated at half the input data rate.
3. Matching: Gain matching is typically 0.1% of full scale, and
offset error is better than 0.02%.
The GAINCTRL pin allows two modes for setting the full-scale
current (IOUTFS) of the two DACs. IOUTFS for each DAC can be
set independently using two external resistors, or IOUTFS for both
DACs can be set by using a single external resistor.2
The DACs utilize a segmented current source architecture combined with a proprietary switching technique to reduce glitch
energy and to maximize dynamic accuracy. Each DAC provides
differential current output thus supporting single-ended or differential applications. Both DACs can be simultaneously updated
1. The AD9765 is a member of a pin-compatible family of dual
TxDACs providing 8-, 10-, 12-, and 14-bit resolution.
2. Dual 12-Bit, 125 MSPS DACs: A pair of high performance
DACs optimized for low distortion performance provide for
flexible transmission of I and Q information.
4. Low Power: Complete CMOS Dual DAC function operates
on 380 mW from a 3.0 V to 5.0 V single supply. The DAC
full-scale current can be reduced for lower power operation,
and a sleep mode is provided for low power idle periods.
5. On-Chip Voltage Reference: The AD9765 includes a 1.20 V
temperature-compensated bandgap voltage reference.
6. Dual 12-Bit Inputs: The AD9765 features a flexible dualport interface allowing dual or interleaved input data.
TxDAC+ is a registered trademark of Analog Devices, Inc.
1
Patent pending.
2
Please see GAINCTRL Mode section, for important date code information on
this feature.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2000
AD9765–SPECIFICATIONS
DC SPECIFICATIONS (T
MIN
to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, unless otherwise noted.)
Parameter
Min
RESOLUTION
Typ
Max
12
Units
Bits
1
DC ACCURACY
Integral Linearity Error (INL)
TA = +25°C
TMIN to TMAX
Differential Nonlinearity (DNL)
TA = +25°C
TMIN to TMAX
ANALOG OUTPUT
Offset Error
Gain Error (Without Internal Reference)
Gain Error (With Internal Reference)
Gain Match
Full-Scale Output Current2
Output Compliance Range
Output Resistance
Output Capacitance
REFERENCE OUTPUT
Reference Voltage
Reference Output Current3
REFERENCE INPUT
Input Compliance Range
Reference Input Resistance
Small Signal Bandwidth
–1.5
–2.0
± 0.4
+1.5
+2.0
LSB
LSB
–0.75
–1.0
± 0.3
+0.75
+1.0
LSB
LSB
+0.02
+2
+5
+1.6
+0.14
20.0
+1.25
% of FSR
% of FSR
% of FSR
% of FSR
dB
mA
V
kΩ
pF
1.26
V
nA
1.25
1
0.5
V
MΩ
MHz
0
± 50
± 100
± 50
ppm of FSR/°C
ppm of FSR/°C
ppm of FSR/°C
ppm/°C
–0.02
–2
–5
–1.6
–0.14
2.0
–1.0
100
5
1.14
OPERATING RANGE
1.20
100
0.1
TEMPERATURE COEFFICIENTS
Offset Drift
Gain Drift (Without Internal Reference)
Gain Drift (With Internal Reference)
Reference Voltage Drift
POWER SUPPLY
Supply Voltages
AVDD
DVDD
Analog Supply Current (IAVDD)
Digital Supply Current (IDVDD)4
Digital Supply Current (IDVDD)5
Supply Current Sleep Mode (IAVDD)
Power Dissipation4 (5 V, IOUTFS = 20 mA)
Power Dissipation5 (5 V, IOUTFS = 20 mA)
Power Dissipation6 (5 V, IOUTFS = 20 mA)
Power Supply Rejection Ratio7—AVDD
Power Supply Rejection Ratio7—DVDD
± 0.25
±1
0.1
3
2.7
5
5
71
5
–0.4
–0.025
+0.4
+0.025
V
V
mA
mA
mA
mA
mW
mW
mW
% of FSR/V
% of FSR/V
–40
+85
°C
8
380
420
450
5.5
5.5
75
7
15
12.0
410
450
NOTES
1
Measured at I OUTA, driving a virtual ground.
2
Nominal full-scale current, I OUTFS, is 32 times the I REF current.
3
An external buffer amplifier with input bias current <100 nA should be used to drive any external load.
4
Measured at f CLOCK = 25 MSPS and fOUT = 1.0 MHz.
5
Measured at f CLOCK = 100 MSPS and f OUT = 1 MHz.
6
Measured as unbuffered voltage output with I OUTFS = 20 mA and 50 Ω RLOAD at IOUTA and IOUTB, fCLOCK = 100 MSPS and f OUT = 40 MHz.
7
± 10% Power supply variation.
Specifications subject to change without notice.
–2–
REV. B
(TMIN to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, Differential
DYNAMIC SPECIFICATIONS Transformer Coupled Output, 50 ⍀ Doubly Terminated, unless otherwise noted.)
Parameter
Min
DYNAMIC PERFORMANCE
Maximum Output Update Rate (fCLOCK)
Output Settling Time (tST) (to 0.1%)1
Output Propagation Delay (tPD)
Glitch Impulse
Output Rise Time (10% to 90%)1
Output Fall Time (90% to 10%)1
Output Noise (IOUTFS = 20 mA)
Output Noise (IOUTFS = 2 mA)
Max
125
AC LINEARITY
Spurious-Free Dynamic Range to Nyquist
fCLOCK = 100 MSPS; fOUT = 1.00 MHz
0 dBFS Output
–6 dBFS Output
–12 dBFS Output
–18 dBFS Output
fCLOCK = 65 MSPS; fOUT = 1.00 MHz
fCLOCK = 65 MSPS; fOUT = 2.51 MHz
fCLOCK = 65 MSPS; fOUT = 5.02 MHz
fCLOCK = 65 MSPS; fOUT = 14.02 MHz
fCLOCK = 65 MSPS; fOUT = 25 MHz
fCLOCK = 125 MSPS; fOUT = 25 MHz
fCLOCK = 125 MSPS; fOUT = 40 MHz
Spurious-Free Dynamic Range Within a Window
fCLOCK = 100 MSPS; fOUT = 1.00 MHz; 2 MHz Span
fCLOCK = 50 MSPS; fOUT = 5.02 MHz; 10 MHz Span
fCLOCK = 65 MSPS; fOUT = 5.03 MHz; 10 MHz Span
fCLOCK = 125 MSPS; fOUT = 5.04 MHz; 10 MHz Span
Total Harmonic Distortion
fCLOCK = 100 MSPS; fOUT = 1.00 MHz
fCLOCK = 50 MSPS; fOUT = 2.00 MHz
fCLOCK = 125 MSPS; fOUT = 4.00 MHz
fCLOCK = 125 MSPS; fOUT = 10.00 MHz
Multitone Power Ratio (Eight Tones at 110 kHz Spacing)
fCLOCK = 65 MSPS; fOUT = 2.00 MHz to 2.99 MHz
0 dBFS Output
–6 dBFS Output
–12 dBFS Output
–18 dBFS Output
Channel Isolation
fCLOCK = 125 MSPS; fOUT = 10 MHz
fCLOCK = 125 MSPS; fOUT = 40 MHz
Specifications subject to change without notice.
–3–
Units
35
1
5
2.5
2.5
50
30
MSPS
ns
ns
pV-s
ns
ns
pA/√Hz
pA/√Hz
70
81
77
72
70
81
79
78
68
55
67
60
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
80
90
88
88
88
dBc
dBc
dBc
dBc
–80
–78
–75
–75
NOTES
1
Measured single-ended into 50 Ω load.
REV. B
Typ
AD9765
–70
dBc
dBc
dBc
dBc
80
79
77
75
dBc
dBc
dBc
dBc
85
77
dBc
dBc
AD9765–SPECIFICATIONS
DIGITAL SPECIFICATIONS (T
MIN
to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, unless otherwise noted.)
Parameter
DIGITAL INPUTS
Logic “1” Voltage @ DVDD = +5 V
Logic “1” @ DVDD = 3
Logic “0” Voltage @ DVDD = +5 V
Logic “0” @ DVDD = 3
Logic “1” Current
Logic “0” Current
Input Capacitance
Input Setup Time (tS)
Input Hold Time (tH)
Latch Pulsewidth (tLPW, tCPW)
Min
Typ
3.5
2.1
5
3
0
Max
Units
V
V
V
V
µA
µA
pF
ns
ns
ns
1.3
0.9
+10
+10
0
–10
–10
5
2.0
1.5
3.5
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS*
Parameter
AVDD
DVDD
ACOM
AVDD
MODE, CLK1, CLK2, WRT1, WRT2
Digital Inputs
IOUTA1/IOUTA2, IOUTB1/IOUTB2
REFIO, FSADJ1, FSADJ2
GAINCTRL, SLEEP
Junction Temperature
Storage Temperature
Lead Temperature (10 sec)
With
Respect to
Min
Max
Units
ACOM
DCOM
DCOM
DVDD
DCOM
DCOM
ACOM
ACOM
ACOM
–0.3
–0.3
–0.3
–6.5
–0.3
–0.3
–1.0
–0.3
–0.3
+6.5
+6.5
+0.3
+6.5
DVDD + 0.3
DVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
+150
+150
+300
V
V
V
V
V
V
V
V
V
°C
°C
°C
–65
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings
for extended periods may affect device reliability.
ORDERING GUIDE
tS
Temperature
Range
Package
Description
–40°C to +85°C
48-Lead LQFP
ST-48
Evaluation Board
Model
tH
DATA IN
Package
Option*
(WRT2) (WRT1 / IQWRT)
AD9765AST
AD9765-EB
t LPW
(CLK2) (CLK1/ IQCLK)
t CPW
*ST = Thin Plastic Quad Flatpack.
IOUTA
OR
IOUTB
THERMAL CHARACTERISTICS
Thermal Resistance
t PD
Figure 1. Timing Diagram for Dual and Interleaved
Modes
48-Lead LQFP
θJA = 91°C/W
See Dynamic and Digital sections for timing specifications.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD9765 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–4–
WARNING!
ESD SENSITIVE DEVICE
REV. B
AD9765
PIN FUNCTION DESCRIPTIONS
Pin No.
Name
Description
1–12
13, 14, 35, 36
15, 21
16, 22
17
18
19
20
23–34
37
38
39, 40
41
42
43
44
45, 46
47
48
PORT1
NC
DCOM1, DCOM2
DVDD1, DVDD2
WRT1/IQWRT
CLK1/IQCLK
CLK2/IQRESET
WRT2/IQSEL
PORT2
SLEEP
ACOM
IOUTA2, IOUTB2
FSADJ2
GAINCTRL
REFIO
FSADJ1
IOUTB1, IOUTA1
AVDD
MODE
Data Bits DB11–P1 to DB0–P1.
No Connect.
Digital Common.
Digital Supply Voltage.
Input write signal for PORT 1 (IQWRT in interleaving mode).
Clock input for DAC1 (IQCLK in interleaving mode).
Clock input for DAC2 (IQRESET in interleaving mode).
Input write signal for PORT 2 (IQSEL in interleaving mode).
Data Bits DB11–P2 to DB0–P2.
Power-Down Control Input.
Analog Common.
“PORT 2” differential DAC current outputs.
Full-scale current output adjust for DAC2.
GAINCTRL Mode (0 = 2 resistor, 1 = 1 resistor.)
Reference Input/Output.
Full-scale current output adjust for DAC1.
“PORT 1” differential DAC current outputs.
Analog Supply Voltage.
Mode Select (1 = Dual Port, 0 = Interleaved).
SLEEP
ACOM
IOUTA2
IOUTB2
FSADJ2
GAINCTRL
REFIO
FSADJ1
IOUTB1
IOUTA1
AVDD
MODE
PIN CONFIGURATION
48 47 46 45 44 43 42 41 40 39 38 37
DB11-P1 (MSB) 1
36 NC
PIN 1
IDENTIFIER
DB10-P1 2
35 NC
DB9-P1 3
34 DB0-P2
DB8-P1 4
33 DB1-P2
DB7-P1 5
DB6-P1 6
AD9765
DB5-P1 7
TOP VIEW
(Not to Scale)
DB4-P1 8
32 DB2-P2
31 DB3-P2
30 DB4-P2
29 DB5-P2
DB3-P1 9
28 DB6-P2
DB2-P1 10
27 DB7-P2
DB1-P1 11
26 DB8-P2
DB0-P1 12
25 DB9-P2
REV. B
–5–
DB10-P2
DB11-P2 (MSB)
DVDD2
DCOM2
WRT2/IQSEL
CLK2/IQRESET
CLK1/IQCLK
DVDD1
DCOM1
WRT1/IQWRT
NC = NO CONNECT
NC
NC
13 14 15 16 17 18 19 20 21 22 23 24
AD9765
DEFINITIONS OF SPECIFICATIONS
Linearity Error (Also Called Integral Nonlinearity or INL)
Temperature Drift
Temperature drift is specified as the maximum change from the
ambient (+25°C) value to the value at either TMIN or TMAX. For
offset and gain drift, the drift is reported in ppm of full-scale
range (FSR) per degree C. For reference drift, the drift is
reported in ppm per degree C.
Linearity error is defined as the maximum deviation of the
actual analog output from the ideal output, determined by a
straight line drawn from zero to full scale.
Differential Nonlinearity (or DNL)
DNL is the measure of the variation in analog value, normalized
to full scale, associated with a 1 LSB change in digital input
code.
Power Supply Rejection
The maximum change in the full-scale output as the supplies
are varied from nominal to minimum and maximum specified
voltages.
Monotonicity
A D/A converter is monotonic if the output either increases or
remains constant as the digital input increases.
Settling Time
The time required for the output to reach and remain within a
specified error band about its final value, measured from the
start of the output transition.
Offset Error
The deviation of the output current from the ideal of zero is
called offset error. For IOUTA, 0 mA output is expected when the
inputs are all 0s. For IOUTB, 0 mA output is expected when all
inputs are set to 1s.
Glitch Impulse
Asymmetrical switching times in a DAC give rise to undesired
output transients that are quantified by a glitch impulse. It is
specified as the net area of the glitch in pV-s.
Gain Error
The difference between the actual and ideal output span. The
actual span is determined by the output when all inputs are set
to 1s minus the output when all inputs are set to 0s.
Spurious-Free Dynamic Range
Output Compliance Range
Total Harmonic Distortion
The range of allowable voltage at the output of a current-output
DAC. Operation beyond the maximum compliance limits may
cause either output stage saturation or breakdown resulting in
nonlinear performance.
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured input signal. It is
expressed as a percentage or in decibels (dB).
The difference, in dB, between the rms amplitude of the output
signal and the peak spurious signal over the specified bandwidth.
5V
CLK1/IQCLK CLK2/IQRESET
AVDD
FSADJ1
RSET1
2k⍀
PMOS
CURRENT
SOURCE
ARRAY
REFIO
IOUTA1
PMOS
CURRENT
SOURCE
ARRAY
RSET2
2k⍀
1.2V REF
AD9765
SEGMENTED
LSB
IOUTB1
SWITCHES FOR SWITCH
DAC1
DAC 1
LATCH
0.1␮F
FSADJ2
MINI
CIRCUITS
T1-1T
SLEEP
CLK
DIVIDER
IOUTA2
SEGMENTED
LSB
SWITCHES FOR SWITCH IOUTB2
DAC2
DAC 2
LATCH
WRT1/
IQWRT
CHANNEL 1 LATCH
LECROY 9210
PULSE
GENERATOR
5V
CHANNEL 2 LATCH
ACOM DCOM
DB0 – DB11
DVDD
50⍀
DCOM
*RETIMED CLOCK OUTPUT
50⍀
MODE
MULTIPLEXING LOGIC
DVDD
GAINCTRL
50⍀
TO HP3589A
SPECTRUM/
NETWORK
ANALYZER
DB0 – DB11
WRT2/
IQSEL
DIGITAL
DATA
TEKTRONIX
AWG-2021
w/OPTION 4
*AWG2021 CLOCK RETIMED SUCH THAT
DIGITAL DATA TRANSITIONS ON FALLING
EDGE OF 50% DUTY CYCLE CLOCK
Figure 2. Basic AC Characterization Test Setup for AD9765, Testing Port 1 in Dual Port Mode, Using Independent
GAINCTRL Resistors on FSADJ1 and FSADJ2
–6–
REV. B
AD9765
Typical Characterization Curves
(AVDD = +5 V, DVDD = +3.3 V, IOUTFS = 20 mA, 50 ⍀ Doubly Terminated Load, Differential Output, TA = +25ⴗC, SFDR up to Nyquist, unless
otherwise noted.)
90
95
90
5MSPS
85
25MSPS
70
125MSPS
85
–6dBFS
–12dBFS
80
–6dBFS
75
–12dBFS
70
80
60
65
65MSPS
1
10
fOUT – MHz
100
Figure 3. SFDR vs. fOUT @ 0 dBFS
60
75
1.00
50
1.25
1.50
1.75
fOUT – MHz
2.00
2.25
Figure 4. SFDR vs. fOUT @ 5 MSPS
85
80
8
6
fOUT – MHz
10
12
80
IOUTFS = 10mA
0dBFS
75
–12dBFS
65
IOUTFS = 20mA
75
SFDR – dBc
SFDR – dBc
75
–6dBFS
70
70
–6dBFS
65
70
IOUTFS = 5mA
65
–12dBFS
60
60
60
55
55
55
5
10
15
20
fOUT – MHz
25
30
35
Figure 6. SFDR vs. fOUT @ 65 MSPS
50
0
50
10
20
30
40
fOUT – MHz
50
60
70
Figure 7. SFDR vs. fOUT @ 125 MSPS
0
85
2.27MHz/25MSPS
10
15
20
fOUT – MHz
25
30
80
1MHz/5MSPS
0.91MHz/10MSPS
5
Figure 8. SFDR vs. fOUT and IOUTFS
@ 65 MSPS and 0 dBFS
90
90
85
4
2
85
0dBFS
50
0
0
Figure 5. SFDR vs. fOUT @ 25 MSPS
85
80
SFDR – dBc
0dBFS
0dBFS
SFDR – dBc
90
SFDR – dBc
SFDR – dBc
80
3.38/3.36MHz@25MSPS
0.965/1.035MHz@7MSPS
2MHz/10MSPS
75
5MHz/25MSPS
75
SFDR – dBc
SFDR – dBc
SFDR – dBc
80
80
75
13MHz/65MSPS
70
70
65
6.75/7.25MHz@65MSPS
70
65
25MHz/125MSPS
11.37MHz/125MSPS
65
60
60
16.9/18.1MHz@125MSPS
5.91MHz/65MSPS
60
–20
–15
–10
AOUT – dBFS
–5
0
Figure 9. Single-Tone SFDR vs. AOUT
@ fOUT = fCLOCK/11
REV. B
55
–20
–15
–10
AOUT – dBFS
–5
0
Figure 10. Single-Tone SFDR vs.
AOUT @ fOUT = fCLOCK/5
–7–
55
–20
–15
–10
AOUT – dBFS
–5
0
Figure 11. Dual-Tone SFDR vs. AOUT
@ fOUT = fCLOCK/7
AD9765
75
IOUTFS = 20mA
0.05
0.5
0
0.4
IOUTFS = 10mA
–0.05
65
DNL – LSBs
0.3
INL – LSBs
0.2
0.1
0
–0.1
60
–0.4
60
80
100
fCLOCK – MSPS
–0.30
120
140
–0.35
0
1000
2000
CODE
10
1.0
0
fOUT = 10MHz
fOUT = 25MHz
65
fOUT = 40MHz
60
55
–10
0.03
0.5
OFFSET ERROR
0.00
0.00
–0.03
–0.5
GAIN ERROR – % FS
OFFSET ERROR – % FS
GAIN ERROR
70
500 1000 1500 2000 2500 3000 3500 4000
CODE
Figure 14. Typical DNL
0.05
75
SFDR – dBc
0
4000
fOUT = 1MHz
80
–20
–30
–40
–50
–60
–70
–80
50
fOUT = 60MHz
45
0
20 40
60
–60 –40 –20
TEMPERATURE – ⴗC
80
–0.05
–40
100
Figure 15. SFDR vs. Temperature @
125 MSPS, 0 dBFS
0
0
–10
–10
–20
–20
–30
–30
–40
–50
–60
–80
–90
Figure 18. Dual-Tone SFDR
@ fCLK = 125 MSPS
40
–1.0
–90
0
10
30
20
FREQUENCY – MHz
40
Figure 17. Single-Tone SFDR
@ fCLK = 125 MSPS
–60
–70
20
10
30
FREQUENCY – MHz
80
–50
–80
0
0
20
40
60
TEMPERATURE – ⴗC
–40
–70
–90
–20
Figure 16. Reference Voltage Drift
vs. Temperature
SFDR – dBm
SFDR – dBm
3000
Figure 13. Typical INL
Figure 12. SINAD vs. fCLOCK and
IOUTFS @ fOUT = 5 MHz and 0 dBFS
85
–0.20
–0.25
–0.3
40
–0.15
–0.2
IOUTFS = 5mA
55
20
–0.10
SFDR – dBm
SINAD – dBc
70
0.6
0
20
10
30
FREQUENCY – MHz
40
Figure 19. Four-Tone SFDR
@ fCLK = 125 MSPS
–8–
REV. B
AD9765
5V
CLK1/IQCLK
CLK2/IQRESET
AVDD
RSET1
2k⍀
IREF1
0.1␮F
RSET2
2k⍀
FSADJ1
REFIO
FSADJ2
IREF 2
1.2V REF
SLEEP
PMOS
CURRENT
SOURCE
ARRAY
PMOS
CURRENT
SOURCE
ARRAY
AD9765
ACOM
VDIFF = VOUT A – VOUT B
IOUTA1
SEGMENTED
LSB
I
SWITCHES FOR
SWITCH OUTB1
DAC1
VOUT 1B
CLK
DIVIDER
DAC 1
LATCH
DAC 2
LATCH
VOUT 2B
DVDD
CHANNEL 2 LATCH
RL2A
50⍀
RL2B
50⍀
MODE
RL1A
50⍀
RL1B
50⍀
VOUT 2A
IOUTA2
SEGMENTED
LSB
SWITCHES FOR
SWITCH IOUTB2
DAC2
MULTIPLEXING LOGIC
CHANNEL 1 LATCH
VOUT 1A
5V
DCOM
GAINCTRL
WRT1/
IQWRT
DB0 – DB11
DB0 – DB11
DIGITAL DATA INPUTS
WRT2/
IQSEL
Figure 20. Simplified Block Diagram
FUNCTIONAL DESCRIPTION
REFERENCE OPERATION
Figure 20 shows a simplified block diagram of the AD9765.
The AD9765 consists of two DACs, each one with its own
independent digital control logic and full-scale output current
control. Each DAC contains a PMOS current source array
capable of providing up to 20 mA of full-scale current (IOUTFS).
The array is divided into 31 equal currents that make up the five
most significant bits (MSBs). The next four bits, or middle bits,
consist of 15 equal current sources whose value is 1/16th of an
MSB current source. The remaining LSB is a binary weighted
fraction of the middle bit current sources. Implementing the
middle and lower bits with current sources, instead of an R-2R
ladder, enhances the dynamic performance for multitone or low
amplitude signals and helps maintain the DAC’s high output
impedance (i.e., >100 kΩ).
The AD9765 contains an internal 1.20 V bandgap reference.
This can easily be overridden by an external reference with no
effect on performance. REFIO serves as either an input or output, depending on whether the internal or an external reference
is used. To use the internal reference, simply decouple the
REFIO pin to ACOM with a 0.1 µF capacitor. The internal
reference voltage will be present at REFIO. If the voltage at
REFIO is to be used elsewhere in the circuit, an external buffer
amplifier with an input bias current of less than 100 nA should
be used. An example of the use of the internal reference is
shown in Figure 21.
All of these current sources are switched to one or the other of
the two output nodes (i.e., IOUTA or IOUTB) via PMOS differential current switches. The switches are based on a new architecture that drastically improves distortion performance. This new
switch architecture reduces various timing errors and provides
matching complementary drive signals to the inputs of the differential current switches.
OPTIONAL
EXTERNAL
REFERENCE
BUFFER
AVDD
GAINCTRL
+1.2V
REF
AD9765
REFERENCE
SECTION
REFIO
ADDITIONAL
EXTERNAL
LOAD
0.1␮F
IREF
CURRENT
SOURCE
ARRAY
FSADJ
2k⍀
ACOM
Figure 21. Internal Reference Configuration
The analog and digital sections of the AD9765 have separate
power supply inputs (i.e., AVDD and DVDD) that can operate
independently over a 3 V to 5.5 V range. The digital section,
which is capable of operating up to a 125 MSPS clock rate,
consists of edge-triggered latches and segment decoding logic
circuitry. The analog section includes the PMOS current sources,
the associated differential switches, a 1.20 V bandgap voltage
reference and two reference control amplifiers.
An external reference can be applied to REFIO as shown in
Figure 22. The external reference may provide either a fixed
reference voltage to enhance accuracy and drift performance or
a varying reference voltage for gain control. Note that the 0.1 µF
compensation capacitor is not required since the internal reference is overridden, and the relatively high input impedance of
REFIO minimizes any loading of the external reference.
The full-scale output current of each DAC is regulated by separate reference control amplifiers and can be set from 2 mA to
20 mA via an external resistor, RSET, connected to the Full
Scale Adjust (FSADJ) pin. The external resistor, in combination
with both the reference control amplifier and voltage reference
VREFIO, sets the reference current IREF, which is replicated to the
segmented current sources with the proper scaling factor. The
full-scale current, IOUTFS, is 32 × IREF.
AVDD
GAINCTRL
AVDD
+1.2V
REF
REFIO
EXTERNAL
REFERENCE
FSADJ
IREF
2k⍀
AD9765
REFERENCE
SECTION
CURRENT
SOURCE
ARRAY
ACOM
Figure 22. External Reference Configuration
REV. B
–9–
AD9765
MASTER/SLAVE RESISTOR MODE, GAINCTRL
The AD9765 allows the gain of each channel to be independently set by connecting one RSET resistor to FSADJ1 and
another RSET resistor to FSADJ2. To add flexibility and reduce
system cost, a single RSET resistor can be used to set the gain of
both channels simultaneously.
When GAINCTRL is low (i.e., connected to AGND), the independent channel gain control mode using two resistors is enabled.
In this mode, individual RSET resistors should be connected to
FSADJ1 and FSADJ2. When GAINCTRL is high (i.e., connected to AVDD), the master/slave channel gain control mode
using one resistor is enabled. In this mode, a single RSET resistor
is connected to FSADJ1 and the resistor on FSADJ2 must be
removed.
The two current outputs will typically drive a resistive load
directly or via a transformer. If dc coupling is required, IOUTA
and IOUTB should be directly connected to matching resistive
loads, RLOAD, that are tied to analog common, ACOM. Note,
RLOAD may represent the equivalent load resistance seen by
IOUTA or IOUTB as would be the case in a doubly terminated 50
Ω or 75 Ω cable. The single-ended voltage output appearing at
the IOUTA and IOUTB nodes is simply:
VOUTA = IOUTA × RLOAD
(5)
VOUTB = IOUTB × RLOAD
(6)
Note the full-scale value of VOUTA and VOUTB should not exceed
the specified output compliance range to maintain specified
distortion and linearity performance.
VDIFF = (IOUTA – IOUTB) × RLOAD
NOTE: Only parts with date code of 9930 or later have the
Master/Slave GAINCTRL function. For parts with a date code
before 9930, Pin 42 must be connected to AGND, and the part
will operate in the two resistor, independent gain control mode.
(7)
Substituting the values of IOUTA, IOUTB and IREF; VDIFF can be
expressed as:
VDIFF = {(2 × DAC CODE – 4095)/4096} ×
(32 × RLOAD/RSET) × VREFIO
REFERENCE CONTROL AMPLIFIER
(8)
Both of the DACs in the AD9765 contain a control amplifier
that is used to regulate the full-scale output current, IOUTFS.
The control amplifier is configured as a V-I converter as shown
in Figure 21, so that its current output, IREF, is determined
by the ratio of the VREFIO and an external resistor, RSET, as
stated in Equation 4. IREF is copied to the segmented current
sources with the proper scale factor to set IOUTFS as stated in
Equation 3.
These last two equations highlight some of the advantages of
operating the AD9765 differentially. First, the differential
operation will help cancel common-mode error sources associated with IOUTA and IOUTB such as noise, distortion and dc
offsets. Second, the differential code-dependent current and
subsequent voltage, VDIFF, is twice the value of the single-ended
voltage output (i.e., VOUTA or VOUTB), thus providing twice the
signal power to the load.
The control amplifier allows a wide (10:1) adjustment span of
IOUTFS from 2 mA to 20 mA by setting IREF between 62.5 µA
and 625 µA. The wide adjustment range of IOUTFS provides
several benefits. The first relates directly to the power dissipation of the AD9765, which is proportional to IOUTFS (refer to
the Power Dissipation section). The second relates to the 20 dB
adjustment, which is useful for system gain control purposes.
Note, the gain drift temperature performance for a single-ended
(VOUTA and VOUTB) or differential output (VDIFF) of the AD9765
can be enhanced by selecting temperature tracking resistors for
RLOAD and RSET due to their ratiometric relationship as shown in
Equation 8.
The small signal bandwidth of the reference control amplifier is
approximately 500 kHz and can be used for low frequency,
small signal multiplying applications.
DAC TRANSFER FUNCTION
Both DACs in the AD9765 provide complementary current
outputs, IOUTA and IOUTB. IOUTA will provide a near full-scale
current output, IOUTFS, when all bits are high (i.e., DAC CODE
= 4095) while IOUTB, the complementary output, provides no
current. The current output appearing at IOUTA and I OUTB
is a function of both the input code and IOUTFS and can be
expressed as:
IOUTA = (DAC CODE/4096) × IOUTFS
(1)
IOUTB = (4095 – DAC CODE/4096) × IOUTFS
(2)
where DAC CODE = 0 to 4095 (i.e., Decimal Representation).
As previously mentioned, IOUTFS is a function of the reference
current IREF, which is nominally set by a reference voltage,
VREFIO and external resistor RSET. It can be expressed as:
IOUTFS = 32 × IREF
(3)
where
IREF = VREFIO /RSET
(4)
ANALOG OUTPUTS
The complementary current outputs in each DAC, IOUTA and
IOUTB, may be configured for single-ended or differential operation.
IOUTA and IOUTB can be converted into complementary singleended voltage outputs, VOUTA and VOUTB, via a load resistor,
RLOAD, as described in the DAC Transfer Function section by
Equations 5 through 8. The differential voltage, VDIFF, existing
between VOUTA and VOUTB can also be converted to a single-ended
voltage via a transformer or differential amplifier configuration.
The ac performance of the AD9765 is optimum and specified
using a differential transformer coupled output in which the
voltage swing at IOUTA and IOUTB is limited to ± 0.5 V. If a singleended unipolar output is desirable, IOUTA should be selected.
The distortion and noise performance of the AD9765 can be
enhanced when it is configured for differential operation. The
common-mode error sources of both IOUTA and IOUTB can be
significantly reduced by the common-mode rejection of a transformer or differential amplifier. These common-mode error
sources include even-order distortion products and noise. The
enhancement in distortion performance becomes more significant as the frequency content of the reconstructed waveform
increases. This is due to the first order cancellation of various
dynamic common-mode distortion mechanisms, digital feedthrough and noise.
–10–
REV. B
AD9765
Performing a differential-to-single-ended conversion via a transformer also provides the ability to deliver twice the reconstructed
signal power to the load (i.e., assuming no source termination).
Since the output currents of IOUTA and IOUTB are complementary, they become additive when processed differentially. A
properly selected transformer will allow the AD9765 to provide
the required power and voltage levels to different loads.
The output impedance of IOUTA and IOUTB is determined by the
equivalent parallel combination of the PMOS switches associated with the current sources and is typically 100 kΩ in parallel
with 5 pF. It is also slightly dependent on the output voltage
(i.e., VOUTA and VOUTB) due to the nature of a PMOS device.
As a result, maintaining IOUTA and/or IOUTB at a virtual ground
via an I-V op amp configuration will result in the optimum dc
linearity. Note the INL/DNL specifications for the AD9765 are
measured with IOUTA maintained at a virtual ground via an
op amp.
DAC TIMING
The AD9765 can operate in two timing modes, dual and interleaved, which are described below. The block diagram in Figure
25 represents the latch architecture in the interleaved timing mode.
DUAL PORT MODE TIMING
When the mode pin is at Logic 1, the AD9765 operates in dual
port mode. The AD9765 functions as two distinct DACs. Each
DAC has its own completely independent digital input and control lines.
The AD9765 features a double buffered data path. Data enters
the device through the channel input latches. This data is then
transferred to the DAC latch in each signal path. Once the data
is loaded into the DAC latch, the analog output will settle to its
new value.
For general consideration, the WRT lines control the channel
input latches and the CLK lines control the DAC latches. Both
sets of latches are updated on the rising edge of their respective
control signals.
IOUTA and IOUTB also have a negative and positive voltage compliance range that must be adhered to in order to achieve optimum performance. The negative output compliance range of
–1.0 V is set by the breakdown limits of the CMOS process.
Operation beyond this maximum limit may result in a breakdown of the output stage and affect the reliability of the AD9765.
The positive output compliance range is slightly dependent on
the full-scale output current, IOUTFS. It degrades slightly from
its nominal 1.25 V for an IOUTFS = 20 mA to 1.00 V for an
IOUTFS = 2 mA. The optimum distortion performance for a
single-ended or differential output is achieved when the maximum full-scale signal at IOUTA and IOUTB does not exceed 0.5 V.
Applications requiring the AD9765’s output (i.e., VOUTA and/or
VOUTB) to extend its output compliance range should size RLOAD
accordingly. Operation beyond this compliance range will adversely affect the AD9765’s linearity performance and subsequently degrade its distortion performance.
The rising edge of CLK should occur before or simultaneously
with the rising edge of WRT. Should the rising edge of CLK
occur after the rising edge of WRT, a 2 ns minimum delay should
be maintained from the rising edge of WRT to the rising edge
of CLK.
Timing specifications for dual port mode are given in Figures 23
and 24.
DIGITAL INPUTS
The AD9765’s digital inputs consist of two independent channels. For the dual port mode, each DAC has its own dedicated
12-bit data port, WRT line and CLK line. In the interleaved
timing mode, the function of the digital control pins changes as
described in the Interleaved Mode Timing section. The 12-bit
parallel data inputs follow straight binary coding where DB11 is
the Most Significant Bit (MSB) and DB0 is the Least Significant Bit (LSB). IOUTA produces a full-scale output current
when all data bits are at Logic 1. IOUTB produces a complementary output with the full-scale current split between the two
outputs as a function of the input code.
The digital interface is implemented using an edge-triggered
master slave latch. The DAC outputs are updated following
either the rising edge, or every other rising edge of the clock,
depending on whether dual or interleaved mode is being used.
The DAC outputs are designed to support a clock rate as high
as 125 MSPS. The clock can be operated at any duty cycle that
meets the specified latch pulsewidth. The setup and hold times
can also be varied within the clock cycle as long as the specified
minimum times are met, although the location of these transition edges may affect digital feedthrough and distortion performance. Best performance is typically achieved when the input
data transitions on the falling edge of a 50% duty cycle clock.
REV. B
tS
tH
DATA IN
WRT1/WRT2
t LPW
CLK1/CLK2
t CPW
IOUTA
OR
IOUTB
t PD
Figure 23. Dual Mode Timing
DATA IN
D1
D2
D3
D4
WRT1/WRT2
CLK1/CLK2
IOUTA
OR
IOUTB
xx
D1
D2
Figure 24. Dual Mode Timing
–11–
D5
D3
D4
AD9765
INTERLEAVED MODE TIMING
INTERLEAVED
DATA
For the following section, refer to Figure 25.
When the mode pin is at Logic 0, the AD9765 operates in interleaved mode. WRT1 now functions as IQWRT and CLK1
functions as IQCLK. WRT2 functions as IQSEL and CLK2
functions as IQRESET.
PORT 2
INPUT
LATCH
DAC1
LATCH
DAC1
IQCLK
IQRESET
DEINTERLEAVED
DATA OUT
DAC2
LATCH
DAC2
ⴜ2
Figure 25. Latch Structure Interleaved Mode
Timing specifications for interleaved mode are given in Figures
26 and 27.
The digital inputs are CMOS-compatible with logic thresholds,
VTHRESHOLD, set to approximately half the digital positive supply
(DVDD) or
tS
D4
D5
IQWRT
DAC OUTPUT
PORT 1
XX
DAC OUTPUT
PORT 2
XX
D1
D3
D4
D2
Figure 27. Interleaved Mode Timing
The internal digital circuitry of the AD9765 is capable of
operating over a digital supply range of 3 V to 5.5 V. As a
result, the digital inputs can also accommodate TTL levels
when DVDD is set to accommodate the maximum high level
voltage of the TTL drivers VOH(MAX). A DVDD of 3 V to
3.3 V will typically ensure proper compatibility with most
TTL logic families. Figure 28 shows the equivalent digital
input circuit for the data and clock inputs. The sleep mode
input is similar with the exception that it contains an active
pull-down circuit, thus ensuring that the AD9765 remains
enabled if this input is left disconnected.
Since the AD9765 is capable of being updated up to 125 MSPS,
the quality of the clock and data input signals are important in
achieving the optimum performance. Operating the AD9765
with reduced logic swings and a corresponding digital supply
(DVDD) will result in the lowest data feedthrough and on-chip
digital noise. The drivers of the digital data interface circuitry
should be specified to meet the minimum setup and hold times
of the AD9765 as well as its required min/max input logic level
thresholds.
Digital signal paths should be kept short and run lengths matched
to avoid propagation delay mismatch. The insertion of a low
value resistor network (i.e., 20 Ω to 100 Ω) between the AD9765
digital inputs and driver outputs may be helpful in reducing any
overshooting and ringing at the digital inputs that contribute to
digital feedthrough. For longer board traces and high data update rates, stripline techniques with proper impedance and
termination resistors should be considered to maintain “clean”
digital inputs.
tH
DATA IN
IQSEL
IQWRT
D3
IQRESET
As with the dual port mode, IQCLK should occur before or
simultaneously with IQWRT.
IQWRT
IQSEL
D2
IQCLK
When IQRESET is high, IQCLK is disabled. When IQRESET
goes low, the following rising edge on IQCLK will update both
DAC latches with the data present at their inputs. In the interleaved mode IQCLK is divided by 2 internally. Following this
first rising edge, the DAC latches will only be updated on every
other rising edge of IQCLK. In this way, IQRESET can be used
to synchronize the routing of the data to the DACs.
PORT 1
INPUT
LATCH
D1
IQSEL
Data enters the device on the rising edge of IQWRT. The logic
level of IQSEL will steer the data to either Channel Latch 1
(IQSEL = 1) or to Channel Latch 2 (IQSEL = 0). For proper
operation, IQSEL should only change state when IQWRT and
IQCLK are low.
INTERLEAVED
DATA IN, PORT 1
XX
t H*
t LPW
IQCLK
The external clock driver circuitry should provide the AD9765
with a low jitter clock input meeting the min/max logic levels
while providing fast edges. Fast clock edges will help minimize
any jitter that will manifest itself as phase noise on a reconstructed waveform. Thus, the clock input should be driven by
the fastest logic family suitable for the application.
t PD
IOUTA
OR
IOUTB
* APPLIES TO FALLING EDGE OF IQCLK/IQWRT AND IQSEL ONLY
Figure 26. Interleaved Mode Timing
–12–
REV. B
AD9765
Note that the clock input could also be driven via a sine wave,
which is centered around the digital threshold (i.e., DVDD/2)
and meets the min/max logic threshold. This will typically result
in a slight degradation in the phase noise, which becomes more
noticeable at higher sampling rates and output frequencies.
Also, at higher sampling rates, the 20% tolerance of the digital
logic threshold should be considered since it will affect the effective clock duty cycle and, subsequently, cut into the required
data setup and hold times.
DVDD
DIGITAL
INPUT
equal to 0.5 × AVDD. This digital input also contains an active
pull-down circuit that ensures the AD9765 remains enabled if
this input is left disconnected. The AD9765 takes less than
50 ns to power down and approximately 5 µs to power back up.
POWER DISSIPATION
The power dissipation, PD, of the AD9765 is dependent on
several factors that include: (1) The power supply voltages
(AVDD and DVDD), (2) the full-scale current output IOUTFS,
(3) the update rate fCLOCK, (4) and the reconstructed digital
input waveform. The power dissipation is directly proportional
to the analog supply current, IAVDD, and the digital supply current, IDVDD. IAVDD is directly proportional to IOUTFS as shown
in Figure 30 and is insensitive to fCLOCK.
80
70
Figure 28. Equivalent Digital Input
60
INPUT CLOCK AND DATA TIMING RELATIONSHIP
50
40
30
20
10
0
5
15
10
20
25
IOUTFS
Figure 30. IAVDD vs. IOUTFS
Conversely, IDVDD is dependent on both the digital input waveform, fCLOCK, and digital supply DVDD. Figures 31 and 32
show IDVDD as a function of full-scale sine wave output ratios
(fOUT/fCLOCK) for various update rates with DVDD = 5 V and
DVDD = 3 V, respectively. Note how IDVDD is reduced by more
than a factor of 2 when DVDD is reduced from 5 V to 3 V.
70
60
50
SNR – dBc
IAVDD
SNR in a DAC is dependent on the relationship between the
position of the clock edges and the point in time at which the
input data changes. The AD9765 is rising edge triggered, and so
exhibits SNR sensitivity when the data transition is close to this
edge. In general, the goal when applying the AD9765 is to make
the data transition close to the falling clock edge. This becomes
more important as the sample rate increases. Figure 29 shows
the relationship of SNR to clock placement with different sample
rates. Note that at the lower sample rates, much more tolerance
is allowed in clock placement, while much more care must be
taken at higher rates.
40
35
30
IDVDD – mA
20
10
0
–4
–3
–2
–1
0
1
2
3
4
TIME OF DATA CHANGE RELATIVE TO
RISING CLOCK EDGE – ns
Figure 29. SNR vs. Clock Placement @ fOUT = 20 MHz and
fCLK = 125 MSPS
30
125MSPS
25
100MSPS
20
65MSPS
15
10
25MSPS
5
5MSPS
0
0
SLEEP MODE OPERATION
The AD9765 has a power-down function that turns off the output
current and reduces the supply current to less than 8.5 mA
over the specified supply range of 3.0 V to 5.5 V and temperature range. This mode can be activated by applying a Logic
Level “1” to the SLEEP pin. The SLEEP pin logic threshold is
REV. B
0.10
0.20
0.30
RATIO – fOUT/fCLK
0.40
0.50
Figure 31. IDVDD vs. Ratio @ DVDD = 5 V
–13–
AD9765
electrical isolation and the ability to deliver twice the power to
the load. Transformers with different impedance ratios may also
be used for impedance matching purposes. Note that the transformer provides ac coupling only.
18
125MSPS
16
14
100MSPS
IDVDD – mA
12
AD9765
10
65MSPS
IOUTA
MINI-CIRCUITS
T1-1T
8
RLOAD
6
25MSPS
IOUTB
4
0
OPTIONAL
RDIFF
5MSPS
2
Figure 33. Differential Output Using a Transformer
0
0.10
0.20
0.30
RATIO – fOUT/fCLK
0.40
0.50
Figure 32. IDVDD vs. Ratio @ DVDD = 3 V
APPLYING THE AD9765
Output Configurations
The following sections illustrate some typical output configurations for the AD9765. Unless otherwise noted, it is assumed
that IOUTFS is set to a nominal 20 mA. For applications requiring the optimum dynamic performance, a differential output
configuration is suggested. A differential output configuration
may consist of either an RF transformer or a differential op
amp configuration. The transformer configuration provides the
optimum high frequency performance and is recommended for
any application allowing for ac coupling. The differential op
amp configuration is suitable for applications requiring dc
coupling, a bipolar output, signal gain and/or level-shifting, within
the bandwidth of the chosen op amp.
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage will
result if IOUTA and/or IOUTB is connected to an appropriatelysized load resistor, RLOAD, referred to ACOM. This configuration may be more suitable for a single-supply system requiring a dc coupled, ground referred output voltage. Alternatively,
an amplifier could be configured as an I-V converter, thus
converting IOUTA or IOUTB into a negative unipolar voltage. This
configuration provides the best dc linearity since IOUTA or IOUTB
is maintained at a virtual ground. Note that IOUTA provides
slightly better performance than IOUTB.
DIFFERENTIAL COUPLING USING A TRANSFORMER
An RF transformer can be used to perform a differential-tosingle-ended signal conversion as shown in Figure 33. A
differentially coupled transformer output provides the optimum distortion performance for output signals whose spectral
content lies within the transformer’s passband. An RF transformer such as the Mini-Circuits T1-1T provides excellent
rejection of common-mode distortion (i.e., even-order harmonics) and noise over a wide frequency range. It also provides
The center tap on the primary side of the transformer must be
connected to ACOM to provide the necessary dc current path
for both IOUTA and IOUTB. The complementary voltages appearing at IOUTA and IOUTB (i.e., VOUTA and VOUTB) swing symmetrically around ACOM and should be maintained with the specified
output compliance range of the AD9765. A differential resistor,
RDIFF, may be inserted in applications where the output of the
transformer is connected to the load, RLOAD, via a passive reconstruction filter or cable. RDIFF is determined by the transformer’s
impedance ratio and provides the proper source termination
that results in a low VSWR. Note that approximately half the
signal power will be dissipated across RDIFF.
DIFFERENTIAL COUPLING USING AN OP AMP
An op amp can also be used to perform a differential-to-singleended conversion as shown in Figure 34. The AD9765 is configured with two equal load resistors, RLOAD, of 25 Ω. The
differential voltage developed across IOUTA and IOUTB is converted to a single-ended signal via the differential op amp configuration. An optional capacitor can be installed across IOUTA
and IOUTB, forming a real pole in a low-pass filter. The addition
of this capacitor also enhances the op amps distortion performance by preventing the DACs high slewing output from overloading the op amp’s input.
The common-mode rejection of this configuration is typically
determined by the resistor matching. In this circuit, the differential op amp circuit using the AD8047 is configured to provide
some additional signal gain. The op amp must operate from a
dual supply since its output is approximately ± 1.0 V. A high
speed amplifier capable of preserving the differential performance of the AD9765 while meeting other system level objectives (i.e., cost, power) should be selected. The op amp’s
differential gain, its gain setting resistor values, and full-scale
output swing capabilities should all be considered when optimizing this circuit.
–14–
REV. B
AD9765
SINGLE-ENDED, BUFFERED VOLTAGE OUTPUT
CONFIGURATION
500⍀
AD9765
225⍀
IOUTA
AD8047
225⍀
IOUTB
COPT
500⍀
25⍀
25⍀
Figure 34. DC Differential Coupling Using an Op Amp
The differential circuit shown in Figure 35 provides the necessary level-shifting required in a single supply system. In this
case AVDD, which is the positive analog supply for both the
AD9765 and the op amp, is also used to level-shift the differential output of the AD9765 to midsupply (i.e., AVDD/2). The
AD8055 is a suitable op amp for this application.
500⍀
Figure 37 shows a buffered single-ended output configuration
in which the op amp U1 performs an I-V conversion on the
AD9765 output current. U1 maintains IOUTA (or IOUTB) at a
virtual ground, thus minimizing the nonlinear output impedance effect on the DAC’s INL performance as discussed in
the Analog Output section. Although this single-ended configuration typically provides the best dc linearity performance, its ac distortion performance at higher DAC update
rates may be limited by U1’s slewing capabilities. U1 provides a negative unipolar output voltage and its full-scale
output voltage is simply the product of RFB and IOUTFS. The
full-scale output should be set within U1’s voltage output
swing capabilities by scaling IOUTFS and/or RFB. An improvement in ac distortion performance may result with a reduced
IOUTFS since the signal current U1 will be required to sink
will be subsequently reduced.
COPT
AD9765
225⍀
IOUTA
225⍀
IOUTB
RFB
200⍀
AD8055
COPT
AD9765
1k⍀
AVDD
25⍀
25⍀
IOUTFS = 10mA
IOUTA
U1
500⍀
IOUTB
VOUT = IOUTFS ⴛ RFB
200⍀
Figure 35. Single Supply DC Differential Coupled Circuit
Figure 37. Unipolar Buffered Voltage Output
SINGLE-ENDED UNBUFFERED VOLTAGE OUTPUT
Figure 36 shows the AD9765 configured to provide a unipolar
output range of approximately 0 V to +0.5 V for a doubly terminated 50 Ω cable since the nominal full-scale current, IOUTFS,
of 20 mA flows through the equivalent RLOAD of 25 Ω. In this
case, RLOAD represents the equivalent load resistance seen by
IOUTA or IOUTB. The unused output (IOUTA or IOUTB) can be
connected to ACOM directly or via a matching RLOAD. Different values of IOUTFS and RLOAD can be selected as long as the
positive compliance range is adhered to. One additional consideration in this mode is the integral nonlinearity (INL) as
discussed in the Analog Output section of this data sheet. For
optimum INL performance, the single-ended, buffered voltage
output configuration is suggested.
AD9765
IOUTFS = 20mA
VOUTA = 0 TO +0.5V
IOUTA
50⍀
50⍀
IOUTB
25⍀
Figure 36. 0 V to 0.5 V Unbuffered Voltage Output
REV. B
POWER AND GROUNDING CONSIDERATIONS, POWER
SUPPLY REJECTION
Many applications seek high speed and high performance under
less than ideal operating conditions. In these application circuits, the implementation and construction of the printed circuit
board is as important as the circuit design. Proper RF techniques must be used for device selection, placement and routing, as well as power supply bypassing and grounding to ensure
optimum performance. Figures 45 to 52 illustrate the recommended printed circuit board ground, power and signal plane
layouts which are implemented on the AD9765 evaluation board.
One factor that can measurably affect system performance is the
ability of the DAC output to reject dc variations or ac noise
superimposed on the analog or digital dc power distribution.
This is referred to as the Power Supply Rejection Ratio. For dc
variations of the power supply, the resulting performance of the
DAC directly corresponds to a gain error associated with the
DAC’s full-scale current, IOUTFS. AC noise on the dc supplies is
common in applications where the power distribution is generated by a switching power supply. Typically, switching power
supply noise will occur over the spectrum from tens of kHz to
several MHz. The PSRR vs. frequency of the AD9765 AVDD
supply over this frequency range is shown in Figure 38.
–15–
AD9765
FERRITE
BEADS
90
TTL/CMOS
LOGIC
CIRCUITS
ELECTROLYTIC
CERAMIC
AVDD
10␮F–22␮F
100␮F
0.1␮F
ACOM
TANTALUM
+5V
POWER SUPPLY
80
Figure 39. Differential LC Filter for Single +5 V and +3 V
Applications
75
70
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FREQUENCY – MHz
0.9
1.0
APPLICATIONS
VDSL Applications Using the AD9765
1.1
Figure 38. Power Supply Rejection Ratio of AD9765
Note that the units in Figure 38 are given in units of (amps out/
volts in). Noise on the analog power supply has the effect of
modulating the internal current sources, and therefore the output current. The voltage noise on AVDD, therefore, will be
added in a nonlinear manner to the desired IOUT. PSRR is very
code dependent thus producing mixing effects which can modulate low frequency power supply noise to higher frequencies.
Worst case PSRR for either one of the differential DAC outputs
will occur when the full-scale current is directed towards that
output. As a result, the PSRR measurement in Figure 38 represents a worst case condition in which the digital inputs remain
static and the full-scale output current of 20 mA is directed to
the DAC output being measured.
An example serves to illustrate the effect of supply noise on the
analog supply. Suppose a switching regulator with a switching
frequency of 250 kHz produces 10 mV of noise and, for simplicity sake (i.e., ignore harmonics), all of this noise is concentrated
at 250 kHz. To calculate how much of this undesired noise will
appear as current noise superimposed on the DAC’s full-scale
current, IOUTFS, one must determine the PSRR in dB using
Figure 38 at 250 kHz. To calculate the PSRR for a given RLOAD,
such that the units of PSRR are converted from A/V to V/V,
adjust the curve in Figure 38 by the scaling factor 20 × Log
(RLOAD ). For instance, if RLOAD is 50 Ω, the PSRR is reduced
by 34 dB (i.e., PSRR of the DAC at 250 kHz, which is 85 dB in
Figure 38, becomes 51 dB VOUT/VIN).
Very High Frequency Digital Subscriber Line (VDSL) technology is growing rapidly in applications requiring data transfer
over relatively short distances. By using QAM modulation and
transmitting the data in Discrete Multiple Tones (DMT), high
data rates can be achieved.
As with other multitone applications, each VDSL tone is capable of transmitting a given number of bits, depending on the
signal-to-noise ratio (SNR) in a narrow band around that tone.
For a typical VDSL application, the tones are evenly spaced
over the range of several kHz to 10 MHz. At the high frequency
end of this range, performance is generally limited by cable
characteristics and environmental factors, such as external interferers. Performance at the lower frequencies is much more dependent on the performance of the components in the signal
chain. In addition to in-band noise, intermodulation from other
tones can also potentially interfere with the data recovery for a
given tone. The two graphs in Figure 40 represent a 500-tone
missing bin test vector, with frequencies evenly spaced from
400 Hz to 10 MHz. This test is very commonly done to determine if distortion will limit the number of bits that can be transmitted in a tone. The test vector has a series of missing tones
around 750 kHz, which is represented in Figure 40a, and a
series of missing tones around 5 MHz, which is represented in
Figure 40b. In both cases, the spurious free dynamic range
(SFDR) between the transmitted tones and the empty bins is
greater than 60 dB.
Proper grounding and decoupling should be a primary objective
in any high speed, high resolution system. The AD9765 features
separate analog and digital supply and ground pins to optimize
the management of analog and digital ground currents in a
system. In general, AVDD, the analog supply, should be decoupled to ACOM, the analog common, as close to the chip as
physically possible. Similarly, DVDD, the digital supply, should
be decoupled to DCOM as close to the chip as physically possible.
For those applications that require a single +5 V or +3 V supply
for both the analog and digital supplies, a clean analog supply
may be generated using the circuit shown in Figure 39. The
circuit consists of a differential LC filter with separate power
supply and return lines. Lower noise can be attained by using
low ESR type electrolytic and tantalum capacitors.
–16–
–20
–30
–40
–50
–60
dBm
PSRR – dB
85
–70
–80
–90
–100
–110
–120
0.665 0.685 0.705 0.725 0.745 0.765 0.785 0.805 0.825
FREQUENCY – MHz
Figure 40a. Notch in Missing Bin at 750 kHz Is Down
>60 dB (Peak Amplitude = 0 dBm)
REV. B
AD9765
quadrature mixer. The matching Nyquist filters shape and limit
each component’s spectral envelope while minimizing intersymbol interference. The DAC is typically updated at the QAM
symbol rate or possibly a multiple of it if an interpolating filter
precedes the DAC. The use of an interpolating filter typically
eases the implementation and complexity of the analog filter,
which can be a significant contributor to mismatches in gain and
phase between the two baseband channels. A quadrature mixer
modulates the I and Q components with the in-phase and
quadrature carrier frequency and then sums the two outputs to
provide the QAM signal.
–30
–40
–50
dBm
–60
–70
–80
–90
–100
–110
12
–120
4.85
4.90
4.95
5.00
5.05
FREQUENCY – MHz
5.10
DAC
5.15
DSP
OR
ASIC
Figure 40b. Notch in Missing Bin at 5 MHz Is Down
>60 dB (Peak Amplitude = 0 dBm)
A common and traditional implementation of a QAM modulator is shown in Figure 41. The modulation is performed in the
analog domain in which two DACs are used to generate the
baseband I and Q components. Each component is then typically applied to a Nyquist filter before being applied to a
NYQUIST
FILTERS
Figure 41. Typical Analog QAM Architecture
AVDD
DVDD
RHODE &
SCHWARZ
RFSEA30B
0.1␮F
RL
PORT I
DIGITAL INTERFACE
PORT Q
“I”
DAC
“I” DAC
LATCH
CA
IOUTB
RL
BBIP
RB
VOUT
BBIN
+
RL
LA
QOUTA
“Q”
DAC
FS ADJ I
RSET
3.9k⍀
FS ADJ Q
RSET
3.9k⍀
REFIO
0.1␮F
DAC'S FULL-SCALE OUTPUT CURRENT = IOUTFS
NOTE: RA, RB, AND RL ARE THIN FILM RESISTOR NETWORKS
WITH 0.1% MATCHING, 1% ACCURACY AVAILABLE FROM
OHMTEK ORNXXXXD SERIES
LOIP
RA
RB
BBQP
PHASE
SPLITTER
LOIN
CA
QOUTB RL
IQSEL
RA
RL
RL
“Q” DAC
MODE
LA
VPBF
AD9765
LATCH
SLEEP
RB
CB
CB
LA
RB
CFILTER
DIFFERENTIAL
RLC FILTER
RL = 200⍀
RA = 2500⍀
RB = 500⍀
RP = 200⍀
CA = 280pf
CB = 45pf
LA = 10␮H
OUIFS = 11mA
AVDD = 5.0V
VCM = 1.2V
AD8346
BBQN
RL
VDIFF = 1.82V p–p
RHODE & SCHWARZ
AVDD
AD976X
RL
RB
RA
–17–
SIGNAL GENERATOR
AD8346
VMOD
0 TO IOUTFS
VDAC
Figure 42. Baseband QAM Implementation Using an AD9765 and AD8346
REV. B
SPECTRUM
ANALYZER
RA
RA
RL
LA
IOUTA
IQCLK
QUADRATURE
MODULATOR
In this implementation, it is much more difficult to maintain
proper gain and phase matching between the I and Q channels.
The circuit implementation shown in Figure 42 helps improve
upon the matching between the I and Q channels, as well as
showing a path for up-conversion using the AD8346 quadrature
modulator. The AD9765 provides both I and Q DACs as well
as a common reference that will improve the gain matching and
stability. RCAL can be used to compensate for any mismatch in
gain between the two channels. The mismatch may be attributed to the mismatch between RSET1 and RSET2, effective load
resistance of each channel, and/or the voltage offset of the control amplifier in each DAC. The differential voltage outputs of
both DACs in the AD9765 are fed into the respective differential inputs of the AD8346 via matching networks.
ACOM AVDD
IQWRT
TO
MIXER
DAC
QAM is one of the most widely used digital modulation schemes
in digital communications systems. This modulation technique
can be found in FDM as well as spread spectrum (i.e., CDMA)
based systems. A QAM signal is a carrier frequency that is
modulated in both amplitude (i.e., AM modulation) and phase
(i.e., PM modulation). It can be generated by independently
modulating two carriers of identical frequency but with a 90°
phase difference. This results in an in-phase (I) carrier component and a quadrature (Q) carrier component at a 90° phase
shift with respect to the I component. The I and Q components
are then summed to provide a QAM signal at the specified carrier frequency.
TEKTRONICS
AWG2021
W/OPTION 4
Σ
90
12
Using the AD9765 for Quadrature Amplitude Modulation
DCOM
0
CARRIER
FREQUENCY
AD9765
I and Q digital data can be fed into the AD9765 in two different
ways. In dual port mode, The digital I information drives one
input port, while the digital Q information drives the other input
port. If no interpolation filter precedes the DAC, the symbol
rate will be the rate at which the system clock drives the CLK
and WRT pins on the AD9765. In interleaved mode, the digital
input stream at Port 1 contains the I and the Q information in
alternating digital words. Using IQSEL and IQRESET, the
AD9765 can be synchronized to the I and Q data stream. The
internal timing of the AD9765 routes the selected I and Q data
to the correct DAC output. In interleaved mode, if no interpolation filter precedes the AD9765, the symbol rate will be half that
of the system clock driving the digital data stream and the
IQWRT and IQCLK pins on the AD9765.
and the ACP must fall under this mask. If distortion in the
transmit path causes the ACP to be above the spectral mask,
then filtering, or different component selection, is needed to
meet the mask requirements.
Figure 43 shows the AD9765, when used with the AD8346,
reconstructing a wideband CDMA signal at 2.4 GHz. The
baseband signal is being sampled at 65 MSPS and has a chip
rate of 8M chips.
–30
–40
–50
–60
==
–70
CDMA
dB
Carrier Division Multiple Access, or CDMA, is an air transmit/
receive scheme where the signal in the transmit path is modulated with a pseudorandom digital code (sometimes referred to
as the spreading code). The effect of this is to spread the transmitted signal across a wide spectrum. Similar to a DMT waveform, a CDMA waveform containing multiple subscribers can
be characterized as having a high peak to average ratio (i.e.,
crest factor), thus demanding highly linear components in the
transmit signal path. The bandwidth of the spectrum is defined
by the CDMA standard being used, and in operation is implemented by using a spreading code with particular characteristics.
–80
–90
–100
–110
c11
c11
cu1
–120
cu1
C0
C0
–130
CENTER 2.4GHz
3MHz
FREQUENCY
SPAN 30MHz
Figure 43. CDMA Signal, 8 M Chips Sampled at 65 MSPS,
Recreated at 2.4 GHz, Adjacent Channel Power > 60 dBm
Distortion in the transmit path can lead to power being transmitted out of the defined band. The ratio of power transmitted
in-band to out-of-band is often referred to as Adjacent Channel
Power (ACP). This is a regulatory issue due to the possibility of
interference with other signals being transmitted by air. Regulatory bodies define a spectral mask outside of the transmit band,
Figure 44 shows an example of the AD9765 used in a W-CDMA
transmitter application using the AD6122 CDMA 3 V IF subsystem. The AD6122 has functions, such as external gain control and low distortion characteristics, needed for the superior
Adjacent Channel Power (ACP) requirements of W-CDMA.
+3V
DVDD
AVDD
CLK1
FSADJ1
U1
(“I DAC”)
RSET1
2k⍀
I DATA
INPUT
634⍀
INPUT
LATCHES
WRT1
500⍀
IOUTA
DAC DAC
LATCH
500⍀
IOUTB
50⍀
IIPN
LOIPP
LOIPN
500⍀
INPUT
LATCHES
FSADJ2
RSET2
1.9k⍀
(“Q DAC”)
RCAL
220⍀
REFIO SLEEP
U2
DAC DAC
LATCH
AD6122
IIPP
50⍀
AD9765
WRT2
Q DATA
INPUT
500⍀
500⍀
ⴜ2
PHASE
SPLITTER
500⍀
QOUTA
500⍀
IIQP
QOUTB
500⍀
IIQN
MODOPP
MODOPN
ACOM
DCOM
50⍀
50⍀
TEMPERATURE
COMPENSATION
CLK2
REFIN
0.1␮F
GAIN
CONTROL
VGAIN
VCC
VCC
GAIN
CONTROL
SCALE
FACTOR
TXOPP
TXOPN
Figure 44. CDMA Transmit Application Using AD9765 and AD6122
–18–
REV. B
AD9765
EVALUATION BOARD
General Description
best performance is obtained by running the Digital Supply
(DVDD) at +3 V and the Analog Supply (AVDD) at +5 V.
The AD9765-EB is an evaluation board for the AD9765 12-bit
dual D/A converter. Careful attention to layout and circuit
design, combined with a prototyping area, allow the user to
easily and effectively evaluate the AD9765 in any application
where high resolution, high speed conversion is required.
This board allows the user the flexibility to operate the AD9765
in various configurations. Possible output configurations include
transformer coupled, resistor terminated, and single and differential outputs. The digital inputs can be used in dual port or
POWER DECOUPLING AND INPUT CLOCKS
interleaved mode, and are designed to be driven from various
RED
RED
word generators, with
TP10the on-board option to add a resistor
TP11
B1
L1
L2
network for proper
load termination.
When operating the AD9765,B3
DVDDIN
DVDD
BEAD
BAN-JACK
AVDDIN
1
C9
BLK
10␮F
TP37
2 25V
B2
BAN-JACK
BLK
TP38
BAN-JACK
BLK
TP39
TP43
BLK
1
B4
C10
BLK
10␮F
TP40
25V
2
BAN-JACK
DGND
DVDD
AVDD
BEAD
BLK
TP41
BLK
TP42
TP44
BLK
1
2
1
C7
0.1␮F
2
C8
0.01␮F
AGND
JP9
3
A B
DCLKIN2
JP6
JP16
2
1
DVDD
WHT
TP29
WRT1IN S1
IQWRT
2
1
DCLKIN1
JP2
4
DGND;3,4,5
WHT
TP30
CLK1IN S2
IQCLK
JP5
A 2B
1
I
DGND;3,4,5
WHT
TP31
CLK2IN S3
RESET
I
WHT
TP32
1
2
1
R1
50⍀
2
1
R2
50⍀
2
R3
50⍀
1
2
I
J
1
3
U1
Q
K
Q
CLR
15
3
5
11
13
CLK
2
DVDD
C
JP3
A2 B
1
DGND;3,4,5
JP1
10
PRE
3
C
JP4
A 2B
1
DGND;3,4,5
WRT2IN S4
IQSEL
3
A B
6
12
PRE
J
9
Q
U2
CLK
K
TSSOP112
14
DGND;8
DVDD;16
7
Q
CLR
TSSOP112
DGND;8
DVDD;16
A B
DVDD
1
3
2
JP7
/2 CLOCK DIVIDER
3
C
WRT1
R4
50⍀
CLK1
CLK2
WHT
TP33
WRT2
SLEEP
SLEEP
1
2
R13
50⍀
RP16
R1
22⍀
RCOM
1
2
INP1
R2
22⍀
3
INP2
R3
22⍀
4
INP3
R4
22⍀
5
INP4
R5
22⍀
6
INP5
R6
22⍀
7
INP6
R7
22⍀
8
INP7
R8
22⍀
9
RP9
R9
22⍀
R1
22⍀
RCOM
10
1
INP8
2
R2
22⍀
3
R3
22⍀
4
R4
22⍀
5
R5
22⍀
6
R6
22⍀
7
R7
22⍀
8
INP9 INP10 INP11 INP12 INP13 INP14
R8
22⍀
9
R1
22⍀
1
2
R2
22⍀
3
R3
22⍀
4
R4
22⍀
5
R5
22⍀
6
R6
22⍀
7
R7
22⍀
8
R8
22⍀
9
RP15
R9
22⍀
R1
22⍀
RCOM
10
1
INP23 INP24 INP25 INP26 INP27 INP28 INP29 INP30
2
R2
22⍀
3
R3
22⍀
4
R4
22⍀
5
R5
22⍀
6
R6
22⍀
7
INP31 INP32 INP33 INP34 INP35 INP36
Figure 45. Power Decoupling and Clocks on AD9765 Evaluation Board
REV. B
10
INCK1
RP10
RCOM
R9
22⍀
–19–
R7
22⍀
8
R8
22⍀
9
INCK2
R9
22⍀
10
AD9765
DIGITAL INPUT SIGNAL CONDITIONING
RP3
RP1
RCOM R1
R9
22⍀
2
P1
P1
1
4
P1
P1
3
6
P1
P1
5
8
P1
P1
7
10
P1
P1
9
12
P1
P1 11
14
P1
P1 13
16
P1
P1 15
18
P1
P1 17
20
P1
P1 19
22
P1
P1 21
24
P1
P1 23
26
P1
P1 25
28
P1
P1
30
P1
P1 29
32
P1
P1
34
P1
P1 33
36
P1
P1 35
38
P1
P1 37
40
P1
P1
27
INP2
INP3
INP4
INP5
INP6
INP7
INP8
INP9
INP10
INP11
INP12
INP13
INP14
DVDD
RP5, 10⍀
1
16
14
2
12
4
10
6
16
8
14
2
12
DUTP2
DUTP3
DUTP4
13
DUTP5
DUTP6
11
DUTP7
DUTP8
9
DUTP9
DUTP10
15
DUTP11
RP6, 10⍀
4
RP6, 10⍀
5
DUTP1
15
RP6, 10⍀
RP6, 10⍀
3
2 3 4 5 6 7 8 9 10
1
DVDD
RP5, 10⍀
RP6, 10⍀
1
2 3 4 5 6 7 8 9 10
1
R9
33⍀
RP5, 10⍀
RP5, 10⍀
7
RCOM R1
RP5, 10⍀
RP5, 10⍀
5
R9
33⍀
2 3 4 5 6 7 8 9 10
1
RP11
RCOM R1
RP5, 10⍀
RP5, 10⍀
3
R9
22⍀
2 3 4 5 6 7 8 9 10
1
INP1
RP13
RCOM R1
DUTP12
13
DUTP13
RP6, 10⍀
DUTP14
11
6
31
RP6, 10⍀
INCK1
8
DCLKIN1
9
39
RP4
RP2
RCOM R1
R9
22⍀
1
2
P2
P2
1
4
P2
P2
3
6
P2
P2
5
8
P2
P2
7
10
P2
P2
9
12
P2
P2 11
14
P2
P2 13
16
P2
P2 15
18
P2
P2 17
20
P2
P2 19
22
P2
P2 21
24
P2
P2 23
26
P2
P2 25
28
P2
P2
30
P2
P2 29
32
P2
P2
34
P2
P2 33
36
P2
P2 35
38
P2
P2 37
40
P2
P2
27
INP23
INP24
INP25
INP26
INP27
INP28
INP29
INP30
INP31
INP32
INP33
INP34
INP35
INP36
1
16
RP7, 10⍀
3
14
RP7, 10⍀
5
12
RP7, 10⍀
7
10
RP8, 10⍀
1
16
RP8, 10⍀
3
14
RP8, 10⍀
5
12
R9
22⍀
2 3 4 5 6 7 8 9 10
DVDD
RP7, 10⍀
RP14
RCOM R1
1
RP12
RCOM R1
33⍀
2 3 4 5 6 7 8 9 10
1
2 3 4 5 6 7 8 9 10
DUTP25
DUTP26
13
DUTP27
DUTP28
11
DUTP29
DUTP30
9
DUTP31
RP8, 10⍀
2
DUTP32
15
DUTP33
RP8, 10⍀
4
DUTP34
13
DUTP35
RP8, 10⍀
6
2 3 4 5 6 7 8 9 10
DUTP24
15
RP7, 10⍀
8
1
DUTP23
RP7, 10⍀
6
R9
DVDD
RP7, 10⍀
4
RCOM R1
33⍀
RP7, 10⍀
2
R9
DUTP36
11
31
39
INCK2
RP8, 10⍀
8
DCLKIN2
9
SPARES
RP5, 10⍀
7
10
RP8, 10⍀
7
10
Figure 46. Digital Input Signal Conditioning
–20–
REV. B
AD9765
DUT AND ANALOG OUTPUT SIGNAL CONDITIONING
BL1
TP34
WHT
ACOM
DVDD
1
C1
2 VAL
1
C2
2 0.01␮F
NC = 5
3
C3
2 0.1␮F
1
AVDD
2
2
R11
VAL
3
A B
1:1
6
T1
JP8
1
1
DB13P1MSB
MODE 48
DUTP2
2
DB12P1
AVDD
DUTP3
3
DB11P1
4
DB9P1
FSADJ1 44
DUTP6
6
DB8P1
REFIO 43
DUTP7
7
DB7P1
GAINCTRL 42
DUTP9
9
DB5P1
DUTP10
10
DB4P1
2
C4 2
10pF 1
1
R5
50⍀
2
R6
50⍀
C5 2
10pF 1
TP45
WHT
R9
1.92k⍀
1
C16
22nF
IB1 45
5
DB6P1
1
IA1 46
DB10P1
8
BL2
3
47
DUTP5
DUTP8
2
A B
DUTP1
DUTP4
2
1
C17
22nF
2
1
FSADJ2 41
R10
1.92k⍀
IB2 40
U2
IA2 39
C15 2
DUTP11
11
DB3P1
ACOM 38
DUTP12
12
DB2P1
SLEEP 37
SLEEP
DUTP13
13
DB1P1
DB0P2 36
DUTP36
DUTP14
14
DB0P1
DB1P2 35
DUTP35
15
DCOM1
DB2P2 34
DUTP34
16
DVDD1
DB3P2 33
DUTP33
WRT1
17
WRT1
DB4P2 32
DUTP32
CLK1
18
CLK2
WRT2
CLK1
DB5P2
31
DUTP31
19
CLK2
DB6P2 30
DUTP30
20
WRT2
DB7P2 29
DUTP29
21
DCOM2
DB8P2 28
DUTP28
22
DVDD2
DB9P2 27
DUTP27
DUTP23
23
DB13P2MSB
DB10P2 26
DUTP26
DUTP24
24
DB12P2
DB11P2 25
DUTP25
1
C6 2
10pF 1
10pF 1
1
2
1
R7
50⍀
2
R8
50⍀
C11
2 1␮F
2
R15
256⍀
1
WHT
TP46
R14
256⍀
1
1
2
C14
0.1␮F
JP10
2
NC = 5
BL4
C13
2 0.1␮F
Figure 47. AD9765 and Output Signal Conditioning
S11
OUT2
AGND;3,4,5
6
T2
1
4
1:1
1
–21–
2
TP35
WHT
R12
VAL
C12
2 0.01␮F
TP36
WHT
BL3
3
1
REFIO
2
2
AVDD
1
REV. B
S6
OUT1
AGND;3,4,5
1
MODE
DVDD
4
JP15
1
AD9765
Figure 48. Assembly, Top Side
–22–
REV. B
AD9765
Figure 49. Assembly, Bottom Side
REV. B
–23–
AD9765
Figure 50. Layer 1, Top Side
–24–
REV. B
AD9765
Figure 51. Layer 2, Ground Plane
REV. B
–25–
AD9765
Figure 52. Layer 3, Power Plane
–26–
REV. B
AD9765
Figure 53. Layer 4, Bottom Side
REV. B
–27–
AD9765
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
0.063 (1.60)
MAX
0.354 (9.00) BSC SQ
0.030 (0.75)
0.018 (0.45)
37
48
36
1
0.276
(7.00)
BSC
SQ
TOP VIEW
(PINS DOWN)
0ⴗ
MIN
12
25
13
0.019 (0.5)
BSC
0.008 (0.2)
0.004 (0.09)
24
0.011 (0.27)
0.006 (0.17)
0.057 (1.45)
0.053 (1.35)
7ⴗ
0ⴗ
0.006 (0.15) SEATING
0.002 (0.05) PLANE
PRINTED IN U.S.A.
COPLANARITY
0.003 (0.08)
C3584a–0–5/00 (rev. B) 00619
48-Lead Thin Plastic Quad Flatpack
(ST-48)
–28–
REV. B