LTC3410 2.25MHz, 300mA Synchronous Step-Down Regulator in SC70 U FEATURES DESCRIPTIO ■ The LTC ®3410 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. The device is available in adjustable and fixed output voltage versions. Supply current during operation is only 26µA, dropping to <1µA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3410 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ High Efficiency: Up to 96% Low Ripple (20mVP-P) Burst Mode Operation: IQ 26µA Low Output Voltage Ripple 300mA Output Current at VIN = 3V 380mA Minimum Peak Switch Current 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle Stable with Ceramic Capacitors 0.8V Reference Allows Low Output Voltages Shutdown Mode Draws < 1µA Supply Current ±2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Available in Low Profile SC70 Package U APPLICATIO S ■ ■ ■ ■ Cellular Telephones Wireless and DSL Modems Digital Cameras MP3 Players Portable Instruments The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.8V feedback reference voltage. The LTC3410 is available in a tiny, low profile SC70 package. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131, 5994885. U ■ Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors. The LTC3410 is specifically designed to work well with ceramic output capacitors, achieving very low output voltage ripple and a small PCB footprint. TYPICAL APPLICATIO Efficiency and Power Loss vs Output Current 100 4.7µH SW LTC3410 10pF RUN VFB GND 887k 412k VOUT 2.5V COUT 4.7µF CER 80 EFFICIENCY (%) CIN 4.7µF CER VIN 70 0.1 EFFICIENCY 60 0.01 50 40 POWER LOSS 30 3410 TA01a 20 10 0 0.1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 10 100 OUTPUT CURRENT (mA) POWER LOSS (W) VIN 2.7V TO 5.5V 1 90 0.001 0.0001 1000 3410 TA01b 3410fb 1 LTC3410 W W W AXI U U ABSOLUTE RATI GS (Note 1) Input Supply Voltage .................................. – 0.3V to 6V RUN, VFB Voltages ..................................... – 0.3V to VIN SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 500mA N-Channel Switch Sink Current (DC) ................. 500mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Notes 3, 5) ...................... 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C U U W PACKAGE/ORDER I FOR ATIO TOP VIEW TOP VIEW RUN 1 6 VFB RUN 1 6 VOUT GND 2 5 GND GND 2 5 GND SW 3 4 VIN SW 3 4 VIN SC6 PACKAGE 6-LEAD PLASTIC SC70 SC6 PACKAGE 6-LEAD PLASTIC SC70 TJMAX = 125°C, θJA = 250°C/ W TJMAX = 125°C, θJA = 250°C/ W ORDER PART NUMBER LTC3410ESC6 SC6 PART MARKING LBSD ORDER PART NUMBER LTC3410ESC6-1.2 LTC3410ESC6-1.5 LTC3410ESC6-1.65 LTC3410ESC6-1.8 LTC3410ESC6-1.875* SC6 PART MARKING LCHV LCNB LCJF LCNC LCFQ Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges. *A separate data sheet is available for the LT3410-1.875. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN IVFB Feedback Current Adjustable Output Voltage ● IVOUT Output Voltage Feedback Current Fixed Output Voltage ● IPK Peak Inductor Current VIN = 3V, VFB = 0.7V or VOUT = 90%, Duty Cycle < 35% VFB Regulated Feedback Voltage Adjustable Output Voltage (LTC3410E) ● ∆VFB Reference Voltage Line Regulation VIN = 2.5V to 5.5V ● VOUT Regulated Output Voltage LTC3410-1.2, IOUT = 100mA LTC3410-1.5, IOUT = 100mA LTC3410-1.65, IOUT = 100mA LTC3410-1.8, IOUT = 100mA LTC3410-1.875, IOUT = 100mA ● ● ● ● ● ∆VOUT Output Voltage Line Regulation VIN = 2.5V to 5.5V ● VLOADREG Output Voltage Load Regulation ILOAD = 50mA to 250mA VIN Input Voltage Range TYP 3.3 380 0.784 1.176 1.47 1.617 1.764 1.837 MAX ±30 nA 6 µA 490 mA 0.8 0.816 0.04 0.4 1.2 1.5 1.65 1.8 1.875 1.224 1.53 1.683 1.836 1.913 0.04 0.4 0.5 ● 2.5 UNITS V %/V V V V V V %/V % 5.5 V 3410fb 2 LTC3410 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER CONDITIONS VUVLO Undervoltage Lockout Threshold IS MIN TYP MAX UNITS VIN Rising VIN Falling 2 1.94 2.3 V V Input DC Bias Current Burst Mode® Operation Shutdown (Note 4) VFB = 0.83V or VOUT = 104%, ILOAD = 0A VRUN = 0V 26 0.1 35 1 µA µA fOSC Oscillator Frequency VFB = 0.8V or VOUT = 100% VFB = 0V or VOUT = 0V 2.25 310 2.7 MHz kHz RPFET RDS(ON) of P-Channel FET ISW = 100mA 0.75 0.9 Ω RNFET RDS(ON) of N-Channel FET ISW = –100mA 0.55 0.7 Ω ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V ±0.01 ±1 µA VRUN RUN Threshold ● 1 1.5 V IRUN RUN Leakage Current ● ±0.01 ±1 µA ● 1.8 0.3 Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3410E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3410: TJ = TA + (PD)(250°C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure1 Except for the Resistive Divider Resistor Values) Efficiency vs Input Voltage IOUT = 100mA IOUT = 250mA 70 EFFICIENCY (%) EFFFICIENCY (%) IOUT = 10mA 80 IOUT = 1mA 60 100 90 90 80 80 70 70 60 50 40 30 IOUT = 0.1mA 50 100 20 40 10 VOUT = 1.8V 30 2.5 3 4.5 4 3.5 INPUT VOLTAGE (V) EFFICIENCY (%) 100 90 Efficiency vs Output Current Efficiency vs Output Current 5 5.5 3410 G02 60 50 40 30 VIN = 2.7V VIN = 3.6V VIN = 4.2V VOUT = 1.8V 0 1 10 100 0.1 OUTPUT CURRENT (mA) 1000 3410 G03 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V VOUT = 1.2V 0 1 10 100 0.1 OUTPUT CURRENT (mA) 1000 3410 G04 3410fb 3 LTC3410 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1 Except for the Resistive Divider Resistor Values) Oscillator Frequency vs Temperature Reference Voltage vs Temperature 2.7 0.814 VIN = 3.6V 2.7 VIN = 3.6V 2.6 0.804 0.799 0.794 0.789 2.6 OSCILLATOR FREQUENCY (MHz) OSCILLATOR FREQUENCY (MHz) 0.809 REFERENCE VOLTAGE (V) Oscillator Frequency vs Supply Voltage 2.5 2.4 2.3 2.2 2.1 2.0 1.9 0.784 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 –25 0 25 50 75 TEMPERATURE (°C) 100 3410 G05 2.3 2.2 2.1 2.0 1.8 125 2 3410 G07 RDS(ON) vs Temperature RDS(ON) vs Input Voltage 1.2 1.2 VIN = 3.6V VOUT = 1.8V 1.1 0.8 MAIN SWITCH 0.8 RDS (ON) (Ω) RDS (ON) (Ω) VOUT ERROR (%) –0.5 0.7 0.6 0.5 SYNCHRONOUS SWITCH 0.6 VIN = 4.2V 0.4 0.4 VIN = 2.7V 0.3 –1.0 0.2 0.2 0 100 200 400 300 LOAD CURRENT (mA) 0 500 1 2 5 4 3 INPUT VOLTAGE (V) 3410 G08 3410 G10 3410 G09 Dynamic Supply Current vs Temperature Dynamic Supply Current vs VIN Switch Leakage vs Temperature 110 50 50 VOUT = 1.2V ILOAD = 0A 100 40 30 20 10 VIN = 5.5V RUN = 0V 90 40 SWITCH LEAKAGE (nA) DYNAMIC SUPPLY CURRENT (µA) DYNAMIC SUPPLY CURRENT (µA) 0 –50 –30 –10 10 30 50 70 90 110 130 TEMPERATURE (°C) 7 6 VIN = 3.6V MAIN SWITCH SYNCHRONOUS SWITCH 0.1 –1.5 VIN = 3.6V VIN = 2.7V 0.9 0 VIN = 4.2V 1.0 1.0 0.5 6 3 5 4 SUPPLY VOLTAGE (V) 3410 G06 Output Voltage vs Load Current 1.0 2.4 1.9 1.8 –50 125 2.5 30 20 10 80 70 SYNCHRONOUS SWITCH 60 50 40 30 MAIN SWITCH 20 10 0 1 2 4 3 5 6 VIN (V) 3410 G11 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3410 G12 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3410 G13 3410fb 4 LTC3410 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1 Except for the Resistive Divider Resistor Values) Burst Mode Operation Switch Leakage vs Input Voltage Start-Up from Shutdown 600 550 LEAKAGE CURRENT (pA) 500 RUN 2V/DIV SW 5V/DIV 450 400 350 MAIN SWITCH 300 VOUT 50mV/DIV AC COUPLED 250 VOUT 1V/DIV 200 150 IL 100mA/DIV SYNCHRONOUS SWITCH 100 50 IL 200mA/DIV 0 0 1 4 3 2 INPUT VOLTAGE (V) 5 6 VIN = 3.6V VOUT = 1.8V ILOAD = 10mA 3410 G14 2µs/DIV 3410 G15 VIN = 3.6V VOUT = 1.8V ILOAD = 300mA Load Step Start-Up from Shutdown VOUT 100mV/DIV AC COUPLED VOUT 1V/DIV IL 200mA/DIV IL 200mA/DIV IL 200mA/DIV ILOAD 200mA/DIV ILOAD 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A 200µs/DIV 3410 G19 10µs/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0mA TO 300mA 3410 G16 Load Step VOUT 100mV/DIV AC COUPLED RUN 2V/DIV 200µs/DIV 3410 G17 10µs/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 20mA TO 300mA 3410 G18 3410fb 5 LTC3410 U U U PI FU CTIO S RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1µA supply current. Do not leave RUN floating. GND (Pins 2, 5): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2µF or greater ceramic capacitor. VFB (Pin 6 Adjustable Version ): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. VOUT (Pin 6 Fixed Voltage Versions): Output Voltage Feedback Pin. An internal resistive divider divides the output voltage down for comparison to the internal reference voltage. W FU CTIO AL DIAGRA U U SLOPE COMP 0.65V OSC OSC 4 VIN FREQ SHIFT – VFB/VOUT + 0.8V R1* – EA R2 240k 0.4V EN SLEEP – + S Q R Q RUN 0.8V REF ( SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW + SHUTDOWN V *R1 = 240k OUT – 1 0.8 5Ω + ICOMP BURST RS LATCH VIN 1 – + ) IRCMP – 6 5 2 GND 3410 BD 3410fb 6 LTC3410 U OPERATIO (Refer to Functional Diagram) Main Control Loop Short-Circuit Protection The LTC3410 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. When the output is shorted to ground, the frequency of the oscillator is reduced to about 310kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 2.25MHz when VFB rises above 0V. Burst Mode Operation The LTC3410 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 70mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 26µA. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier’s output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. Dropout Operation As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. Another important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3410 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3410 uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 3410fb 7 LTC3410 U W U U APPLICATIO S I FOR ATIO Inductor Core Selection VIN 2.7V TO 5.5V 4.7µH CIN 4.7µF CER VIN SW LTC3410 10pF RUN VOUT 1.2V COUT 4.7µF CER VFB GND 232k 464k 3410 F01 Figure 1. High Efficiency Step-Down Converter The basic LTC3410 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3410 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3410 applications. Table 1. Representative Surface Mount Inductors MANUFACTURER PART NUMBER Inductor Selection Taiyo Yuden For most applications, the value of the inductor will fall in the range of 2.2µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 120mA (40% of 300mA). ∆IL = ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ ( f)(L) ⎝ VIN ⎠ (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. MAX DC VALUE CURRENT DCR HEIGHT CB2016T2R2M CB2012T2R2M LBC2016T3R3M 2.2µH 2.2µH 3.3µH 510mA 530mA 410mA 0.13Ω 1.6mm 0.33Ω 1.25mm 0.27Ω 1.6mm Panasonic ELT5KT4R7M 4.7µH 950mA 0.2Ω 1.2mm Sumida CDRH2D18/LD 4.7µH 630mA 0.086Ω 2mm Murata LQH32CN4R7M23 4.7µH 450mA Taiyo Yuden NR30102R2M NR30104R7M 2.2µH 4.7µH 1100mA 0.1Ω 1mm 750mA 0.19Ω 1mm FDK FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D 4.7µH 3.3µH 2.2µH 1100mA 0.11Ω 1mm 1200mA 0.1Ω 1mm 1300mA 0.08Ω 1mm 0.2Ω 2mm CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX [VOUT (VIN − VOUT )]1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 3410fb 8 LTC3410 U W U U APPLICATIO S I FOR ATIO 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. The recommended capacitance value to use is 4.7µF for both input and output capacitor. For applications with VOUT greater than 2.5V, the recommended value for output capacitance should be increased. See Table 2. Table 2. Capacitance Selection OUTPUT VOLTAGE RANGE OUTPUT CAPACITANCE INPUT CAPACITANCE 0.8V ≤ VOUT ≤ 2.5V 4.7µF 4.7µF VOUT > 2.5V 10µH or 2x 4.7µF 4.7µF If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Output Voltage Programming (LTC3410 Only) Using Ceramic Input and Output Capacitors The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3410’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. The output voltage is set by a resistive divider according to the following formula: ⎛ R2⎞ VOUT = 0.8V ⎜ 1+ ⎟ ( 2) ⎝ R1⎠ The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 2. Efficiency Considerations Efficiency = 100% – (L1 + L2 + L3 + ...) 0.8V ≤ VOUT ≤ 5.5V R2 However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires VFB LTC3410 R1 GND 3410 F02 Figure 2. Setting the LTC3410 Output Voltage 3410fb 9 LTC3410 U W U U APPLICATIO S I FOR ATIO where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3410 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 3. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3410 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3410 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If 1 VIN = 3.6V POWER LOSS (W) 0.1 0.01 0.001 0.0001 0.00001 0.1 VOUT = 3.3V VOUT = 1.8V VOUT = 1.2V 10 100 1 LOAD CURRENT (mA) 1000 3410 F03 Figure 3. Power Loss vs Load Current 3410fb 10 LTC3410 U W U U APPLICATIO S I FOR ATIO the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3410 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJAis the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: T J = TA + TR where TA is the ambient temperature. As an example, consider the LTC3410 in dropout at an input voltage of 2.7V, a load current of 300mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 1.0Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 90mW For the SC70 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.09)(250) = 92.5°C which is well below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3410. These items are also illustrated graphically in Figures 4 and 5. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 3410fb 11 LTC3410 U W U U APPLICATIO S I FOR ATIO 1 1 RUN 2 – VFB GND + 2 6 COUT VOUT L1 VIN SW 5 6 COUT VOUT R1 4 VOUT GND – R2 3 RUN LTC3410-1.875 LTC3410 3 + L1 CFWD SW VIN 5 CIN 4 CIN VIN VIN 3410 F04b 3410 F04a BOLD LINES INDICATE HIGH CURRENT PATHS BOLD LINES INDICATE HIGH CURRENT PATHS Figure 4a. LTC3410 Layout Diagram Figure 4b. LTC3410-1.875 Layout Diagram VIA TO GND R1 VOUT VOUT VIN VIA TO VIN VIA TO VOUT R2 PIN 1 L1 PIN 1 L1 CFWD LTC3410 VIN VIA TO VIN LTC34101.875 SW SW COUT COUT CIN CIN GND 3410 F05b 3410 F05a Figure 5b. LTC3410 Fixed Output Voltage Suggested Layout Figure 5a. LTC3410 Suggested Layout 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the (–) plates of CIN and COUT as close as possible. 5. Keep the switching node, SW, away from the sensitive VFB node. Design Example As a design example, assume the LTC3410 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.3A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 3V. With this information we can calculate L using Equation (1), L= ⎛ V ⎞ 1 VOUT ⎜ 1− OUT ⎟ ( f)(∆IL ) ⎝ VIN ⎠ (3) 3410fb 12 LTC3410 U W U U APPLICATIO S I FOR ATIO Substituting VOUT = 3V, VIN = 4.2V, ∆IL = 100mA and f = 2.25MHz in Equation (3) gives: L= of less than 0.5Ω. In most cases, a ceramic capacitor will satisfy this requirement. From Table 2, Capacitance Selection, COUT = 10µF and CIN = 4.7µF. 3V 3V ⎞ ⎛ ⎜ 1− ⎟ = 3.8µH 2.25MHz(100mA) ⎝ 4.2V ⎠ For the feedback resistors, choose R1 = 301k. R2 can then be calculated from equation (2) to be: A 4.7µH inductor works well for this application. For best efficiency choose a 350mA or greater inductor with less than 0.3Ω series resistance. ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1= 827.8k ; use 825k ⎝ 0.8 ⎠ Figure 6 shows the complete circuit along with its efficiency curve. CIN will require an RMS current rating of at least 0.125A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR VIN 2.7V TO 4.2V 4 CIN†† 4.7µF CER VIN SW 3 VOUT 3V 10pF LTC3410 1 4.7µH* COUT† 10µF CER RUN VFB 6 825k GND 2, 5 301k 3410 F06a † TAIYO YUDEN JMK212BJ106 TAIYO YUDEN JMK212BJ475 *MURATA LQH32CN4R7M23 †† Figure 6a 100 90 VOUT 100mV/DIV AC COUPLED EFFICIENCY (%) 80 70 60 50 IL 200mA/DIV 40 30 20 VIN = 3.6V VIN = 4.2V 10 0 0.1 1 10 ILOAD (mA) 100 1000 3410 F06b Figure 6b ILOAD 200mA/DIV 20µs/DIV VIN = 3.6V VOUT = 3V ILOAD = 100mA TO 300mA 3410 F06c Figure 6c 3410fb 13 LTC3410 U TYPICAL APPLICATIO S Using Low Profile Components, <1mm Height VIN 2.7V TO 4.2V 4 VIN † CIN 4.7µF SW 3 4.7µH* LTC3410-1.875 1 RUN VOUT 6 COUT† 4.7µF CER VOUT 1.875V GND † TAIYO YUDEN JMK212BJ475 *FDK MIPF2520D 2, 5 3410 TA06a Low Profile Efficiency 100 EFFICIENCY (%) 90 Load Step VIN = 2.7V VIN = 3.6V VIN = 4.2V VOUT 100mV/DIV AC COUPLED 80 IL 200mA/DIV 70 ILOAD 200mA/DIV 60 3410 TA06c 20µs/DIV 50 0.1 VIN = 3.6V ILOAD = 100mA TO 300mA 1 10 LOAD (mA) 100 1000 3410 TA06b 3410fb 14 LTC3410 U PACKAGE DESCRIPTIO SC6 Package 6-Lead Plastic SC70 (Reference LTC DWG # 05-08-1638) 0.47 MAX 0.65 REF 1.80 – 2.20 (NOTE 4) 1.00 REF INDEX AREA (NOTE 6) 1.80 – 2.40 1.15 – 1.35 (NOTE 4) 2.8 BSC 1.8 REF PIN 1 RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.10 – 0.40 0.65 BSC 0.15 – 0.30 6 PLCS (NOTE 3) 0.80 – 1.00 0.00 – 0.10 REF 1.00 MAX GAUGE PLANE 0.15 BSC 0.26 – 0.46 0.10 – 0.18 (NOTE 3) SC6 SC70 1205 REV B NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. DETAILS OF THE PIN 1 INDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE INDEX AREA 7. EIAJ PACKAGE REFERENCE IS EIAJ SC-70 8. JEDEC PACKAGE REFERENCE IS MO-203 VARIATION AB 3410fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3410 U TYPICAL APPLICATIO Using Low Profile Components, <1mm Height Efficiency 4 CIN† 4.7µF VIN SW 3 1 VOUT 1.5V 10pF LTC3410 90 VFB 6 3410 TA02 402k GND 464k 2, 5 VOUT 100mV/DIV AC COUPLED 80 COUT† 4.7µF RUN Load Step 100 4.7µH* EFFICIENCY (%) VIN 2.7V TO 4.2V 70 60 50 IL 200mA/DIV 40 30 VIN = 2.7V VIN = 3.6V VIN = 4.2V 20 † TAIYO YUDEN JMK212BJ475 *FDK MIPF2520D 10 0 0.1 1 10 ILOAD (mA) 100 1000 3410 TA03 ILOAD 200mA/DIV 20µs/DIV 3410 TA04 VIN = 3.6V VOUT = 1.5V ILOAD = 100mA TO 300mA RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1616 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1µA, ThinSOT Package LT1676 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LT1776 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 40V, VOUT = 1.24V, IQ = 3.2mA, ISD = 30µA, N8, S8 Packages LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 15µA, ISD = <1µA, TSSOP-16 Package LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable, IQ = 20µA, ISD = <1µA, DFN Package LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3406 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3409 600mA (IOUT), 1.5MHz/2.25MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 1.6V to 5.5V, VOUT = 0.613V, IQ = 65µA, DD8 Package LTC3410B 300mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter with Burst Disabled 96% Efficiency, VIN = 2.5V to 3.5V, VOUT(MIN) = 0.8V, IQ = 200µA, ISD = <1µA, SC70 Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, MS Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, TSSOP-16E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25µA, ISD = <1µA, MS Package 3410fb 16 Linear Technology Corporation LT 0806 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005