LM3151/LM3152/LM3153 SIMPLE SWITCHER® CONTROLLER, High Input Voltage Synchronous Step-Down General Description Features SWITCHER® The LM3151/2/3 SIMPLE Controller is an easy to use and simplified step down power controller capable of providing up to 12A of output current in a typical application. Operating with an input voltage range from 6V-42V, the LM3151/2/3 features a fixed output voltage of 3.3V, and features switching frequencies of 250 kHz, 500 kHz, and 750 kHz. The synchronous architecture provides for highly efficient designs. The LM3151/2/3 controller employs a Constant On-Time (COT) architecture with a proprietary Emulated Ripple Mode (ERM) control that allows for the use of low ESR output capacitors, which reduces overall solution size and output voltage ripple. The Constant On-Time (COT) regulation architecture allows for fast transient response and requires no loop compensation, which reduces external component count and reduces design complexity. Fault protection features such as thermal shutdown, undervoltage lockout, over-voltage protection, short-circuit protection, current limit, and output voltage pre-bias startup allow for a reliable and robust solution. The LM3151/2/3 SIMPLE SWITCHER® concept provides for an easy to use complete design using a minimum number of external components and National’s WEBENCH® online design tool. WEBENCH® provides design support for every step of the design process and includes features such as external component calculation with a new MOSFET selector, electrical simulation, thermal simulation, and Build-It boards for prototyping. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ PowerWise® step-down controller 6V to 42V Wide input voltage range Fixed output voltage of 3.3V Fixed switching frequencies of 250 kHz/500 kHz/750 kHz No loop compensation required Fully WEBENCH® enabled Low external component count Constant On-Time control Ultra-Fast transient response Stable with low ESR capacitors Output voltage pre-bias startup Valley current limit Programmable soft-start Typical Applications ■ ■ ■ ■ ■ Telecom Networking Equipment Routers Security Surveillance Power Modules Typical Application 30053201 SIMPLE SWITCHER® is a registered trademark of National Semiconductor Corporation © 2009 National Semiconductor Corporation 300532 www.national.com LM3151/LM3152/LM3153 SIMPLE SWITCHER® CONTROLLER, High Input Voltage Synchronous Step-Down August 26, 2009 LM3151/LM3152/LM3153 Connection Diagram 30053202 eTSSOP-14 Ordering Information Order Number Package Type NSC Package Drawing Input Voltage Range Output Voltage Switching Frequency LM3151MH-3.3 LM3151MHE-3.3 Supplied As 94 Units per Anti-Static Tube eTSSOP-14 MXA14A 6V - 42V 3.3V 250KHz 250 Units in Tape and Reel LM3151MHX-3.3 2500 Units in Tape and Reel LM3152MH-3.3 94 Units per Anti-Static Tube LM3152MHE-3.3 eTSSOP-14 MXA14A 6V - 33V 3.3V 500KHz 250 Units in Tape and Reel LM3152MHX-3.3 2500 Units in Tape and Reel LM3153MH-3.3 94 Units per Anti-Static Tube LM3153MHE-3.3 eTSSOP-14 MXA14A 8V - 18V LM3153MHX-3.3 www.national.com 3.3V 750KHz 250 Units in Tape and Reel 2500 Units in Tape and Reel 2 Pin Name Description 1 VCC Supply Voltage for FET Drivers Function 2 VIN Input Supply Voltage 3 EN Enable To enable the IC apply a logic high signal to this pin greater than 1.26V typical or leave floating. To disable the part, ground the EN pin. 4 FB Feedback Internally connected to the resistor divider network which sets the fixed output voltage. This pin also senses the output voltage faults such a over-voltage and short circuit conditions. 5,9 SGND Signal Ground Ground for all internal bias and reference circuitry. Should be connected to PGND at a single point. 6 SS Soft-Start An internal 7.7 µA current source charges an external capacitor to provide the soft-start function. 7,8 N/C Not Connected Internally not electrically connected. These pins may be left unconnected or connected to ground. 10 SW Switch Node Switch pin of controller and high-gate driver lower supply rail. A boost capacitor is also connected between this pin and BST pin 11 HG High-Side Gate Drive 12 BST 13 LG Low-Side Gate Drive 14 PGND Power Ground Synchronous rectifier MOSFET source connection. Tie to power ground plane. Should be tied to SGND at a single point. EP EP Exposed Pad Exposed die attach pad should be connected directly to SGND. Also used to help dissipate heat out of the IC. Nominally regulated to 5.95V. Connect a 1 µF to 2.2 µF decoupling capacitor from this pin to ground. Supply pin to the device. Nominal input range is 6V to 42V. See ordering information for Vin limitations. Gate drive signal to the high-side NMOS switch. The high-side gate driver voltage is supplied by the differential voltage between the BST pin and SW pin. High-gate driver upper supply rail. Connect a 0.33 µF-0.47 µF capacitor from SW pin to Connection for this pin. An internal diode charges the capacitor during the high-side switch off-time. Do Bootstrap Capacitor not connect to an external supply rail. Gate drive signal to the low-side NMOS switch. The low-side gate driver voltage is supplied by VCC. 3 www.national.com LM3151/LM3152/LM3153 Pin Descriptions LM3151/LM3152/LM3153 Absolute Maximum Ratings (Note 1) Operating Ratings If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN Junction Temperature Range (TJ) EN VIN to GND SW to GND BST to SW BST to GND All Other Inputs to GND ESD Rating (Note 2) Storage Temperature Range (Note 1) 6V to 42V −40°C to + 125°C 0V to 5V -0.3V to 47V -3V to 47V -0.3V to 7V -0.3V to 52V -0.3V to 7V 2kV -65°C to +150°C Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 18V. Symbol Parameter Conditions Min Typ Max Units CVCC = 1 µF, 0 mA to 40 mA 5.65 5.95 6.25 V Start-Up Regulator, VCC VCC IVCC = 2 mA, Vin = 5.5V 40 IVCC = 30 mA, Vin = 5.5V 330 VIN - VCC VIN - VCC Dropout Voltage IVCCL VCC Current Limit (Note 3) VCC = 0V VCC Under-voltage Lockout threshold (UVLO) VCC Increasing VCC-UVLO-HYS VCC UVLO Hysteresis VCC Decreasing tCC-UVLO-D VCC UVLO Filter Delay IIN Input Operating Current VCCUVLO IIN-SD 65 100 4.75 5.1 No Switching mV mA 5.40 V 475 mV 3 µs 3.6 5.2 32 55 mA Input Operating Current, Device Shutdown VEN = 0V Boost Pin Leakage VBST – VSW = 6V 2 nA HG Drive Pull–Up On-Resistance IHG Source = 200 mA 5 Ω HG Drive Pull–Down On-Resistance IHG Sink = 200 mA 3.4 Ω LG Drive Pull–Up On-Resistance ILG Source = 200 mA 3.4 Ω LG Drive Pull–Down On-Resistance ILG Sink = 200 mA 2 Ω SS Pin Source Current VSS = 0V µA GATE Drive IQ-BST RDS-HG-Pull-Up RDS-HG-Pull-Down RDS-LG-Pull-Up RDS-LG-Pull-Down Soft-Start ISS ISS-DIS 5.9 7.7 9.5 200 SS Pin Discharge Current mA µA Current Limit VCL 175 Current Limit Voltage Threshold 200 225 mV ON/OFF Timer tON-MIN ON Timer Minimum Pulse Width 200 tOFF OFF Timer Minimum Pulse Width 370 525 1.20 1.26 ns ns Enable Input VEN VEN-HYS www.national.com EN Pin Input Threshold Trip Point VEN Rising EN Pin threshold Hysteresis VEN Falling 4 1.14 120 V mV Parameter Conditions Min Typ Max Units Boost Diode Vf Forward Voltage IBST = 2 mA 0.7 V IBST = 30 mA 1 V Thermal Characteristics TSD Thermal Shutdown Rising 165 °C Thermal Shutdown Hysteresis Falling 15 °C 4 Layer JEDEC Printed Circuit Board, 9 Vias, No Air Flow 40 2 Layer JEDEC Printed Circuit Board. No Air Flow 140 θJA Junction to Ambient θJC Junction to Case °C/W 4 No Air Flow °C/W 3.3V Output Option Symbol Parameter VOUT Conditions Min Typ Max Units Output Voltage 3.234 3.3 3.366 V VOUT-OV Output Voltage Over-Voltage Threshold 3.83 4.00 4.17 V VIN-MAX Maximum Input Voltage (Note 4) VIN-MIN fS tON RFB Minimum Input Voltage (Note 4) Switching Frequency On-Time LM3151-3.3 42 LM3152-3.3 33 LM3153-3.3 18 LM3151-3.3 6 LM3152-3.3 6 LM3153-3.3 8 LM3151-3.3, RON = 115 kΩ 250 LM3152-3.3, RON = 51 kΩ 500 LM3153-3.3, RON = 32 kΩ 750 LM3151-3.3, RON = 115 kΩ 730 LM3152-3.3, RON = 51 kΩ 400 LM3153-3.3, RON = 32 kΩ 330 566 FB Resistance to Ground V V kHz ns kΩ Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but does not guarantee specific performance limits. For guaranteed specifications and conditions, see the Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test Method is per JESD-22-A114. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: The input voltage range is dependent on minimum on-time, off-time, and therefore frequency, and is also affected by optimized MOSFET selection. 5 www.national.com LM3151/LM3152/LM3153 Symbol LM3151/LM3152/LM3153 Simplified Block Diagram 30053203 www.national.com 6 LM3151/LM3152/LM3153 Typical Performance Characteristics Boost Diode Forward Voltage vs. Temperature Quiescent Current vs. Temperature 30053240 30053242 Soft-Start Current vs. Temperature VCC Current Limit vs. Temperature 30053243 30053247 VCC Dropout vs. Temperature VCC vs. Temperature 30053248 30053249 7 www.national.com LM3151/LM3152/LM3153 VCL vs. Temperature On-Time vs. Temperature (250 kHz) 30053282 30053283 On-Time vs. Temperature (500 kHz) On-Time vs. Temperature (750 kHz) 30053284 www.national.com 30053286 8 The LM3151/2/3 synchronous step-down SIMPLE SWITCHER® Controller employs a Constant On-Time (COT) architecture which is a derivative of the hysteretic control scheme. COT relies on a fixed switch on-time to regulate the output. The on-time of the high-side switch is set internally by resistor RON. The LM3151/2/3 automatically adjusts the on-time inversely with the input voltage to maintain a constant frequency. Assuming an ideal system and VIN is much greater than 1V, the following approximations can be made: The on-time, tON: Over-Voltage Comparator The over-voltage comparator is provided to protect the output from over-voltage conditions due to sudden input line voltage changes or output loading changes. The over-voltage comparator continuously monitors the attenuated FB voltage versus a 0.72V internal reference. If the voltage at FB rises above 0.72V the on-time pulse is immediately terminated. This condition can occur if the input or the output load changes suddenly. Once the over-voltage protection is activated, the HG and LG signals remain off until the attenuated FB voltage falls below 0.72V. Current Limit Where K = 100 pC, and RON is specified in the electrical characteristics table. Control is based on a comparator and the on-timer, with the output voltage feedback (FB) attenuated and then compared with an internal reference of 0.6V. If the attenuated FB level is below the reference, the high-side switch is turned on for a fixed time, t ON, which is determined by the input voltage and the internal resistor, RON. Following this on-time, the switch remains off for a minimum off-time, tOFF, as specified in the Electrical Characteristics table or until the attenuated FB voltage is less than 0.6V. This switching cycle will continue while maintaining regulation. During continuous conduction mode (CCM), the switching frequency depends only on duty cycle and on-time. The duty cycle can be calculated as: Current limit detection occurs during the off-time by monitoring the current through the low-side switch. If during the offtime the current in the low-side switch exceeds the user defined current limit value, the next on-time cycle is immediately terminated. Current sensing is achieved by comparing the voltage across the low-side switch against an internal reference value, VCL, of 200 mV. If the voltage across the lowside switch exceeds 200 mV, the current limit comparator will trigger logic to terminate the next on-time cycle. The current limit ICL, can be determined as follows: Where IOCL is the user-defined average output current limit value, RDS(ON)max is the resistance value of the low-side FET at the expected maximum FET junction temperature, VCL is the internal current limit reference voltage and Tj is the junction temperature of the LM3151/2/3. Figure 1 illustrates the inductor current waveform. During normal operation, the output current ripple is dictated by the switching of the FETs. The current through the low-side switch, Ivalley, is sampled at the end of each switching cycle and compared to the current limit threshold voltage, VCL. The valley current can be calculated as follows: Where the switching frequency of a COT regulator is: Typical COT hysteretic controllers need a significant amount of output capacitor ESR to maintain a minimum amount of ripple at the FB pin in order to switch properly and maintain efficient regulation. The LM3151/2/3 however utilizes proprietary, Emulated Ripple Mode Control Scheme (ERM) that allows the use of ceramic output capacitors without additional equivalent series resistance (ESR) compensation. Not only does this reduce the need for output capacitor ESR, but also significantly reduces the amount of output voltage ripple seen in a typical hysteretic control scheme. The output ripple voltage can become so low that it is comparable to voltage-mode and current-mode control schemes. Where IOUT is the average output current and ΔIL is the peakto-peak inductor ripple current. If an overload condition occurs, the current through the lowside switch will increase which will cause the current limit comparator to trigger the logic to skip the next on-time cycle. The IC will then try to recover by checking the valley current during each off-time. If the valley current is greater than or equal to ICL, then the IC will keep the low-side FET on and allow the inductor current to further decay. Throughout the whole process, regardless of the load current, the on-time of the controller will stay constant and thereby the positive ripple current slope will remain constant. During each on-time the current ramps up an amount equal to: Regulation Comparator The output voltage is sampled through the FB pin and then divided down by two internal resistors and compared to the internal reference voltage of 0.6V by the error comparator. In normal operation, an on-time period is initiated when the sampled output voltage at the input of the error comparator falls below 0.6V. The high-side switch stays on for the specified on-time, causing the sampled voltage on the error comparator input to rise above 0.6V. After the on-time period, the highside switch stays off for the greater of the following: 9 www.national.com LM3151/LM3152/LM3153 1) Minimum off time as specified in the electrical characteristics table 2) The error comparator sampled voltage falls below 0.6V Theory of Operation LM3151/LM3152/LM3153 the inductor current is forced to decay following any overload conditions. The valley current limit feature prevents current runaway conditions due to propagation delays or inductor saturation since 30053212 FIGURE 1. Inductor Current - Current Limit Operation An internal switch grounds the SS pin if VCC is below the under-voltage lockout threshold, if a thermal shutdown occurs, or if the EN pin is grounded. By using an externally controlled switch, the output voltage can be shut off by grounding the SS pin. During startup the LM3151/2/3 will operate in diode emulation mode, where the low-side gate LG will turn off and remain off when the inductor current falls to zero. Diode emulation mode allows for start up into a pre-biased output voltage. When softstart is greater than 0.7V, the LM3151/2/3 will remain in continuous conduction mode. During diode emulation mode at current limit the low-gate will remain off when the inductor current is off. The soft start time should be greater than the rise time specified by, Short-Circuit Protection The LM3151/2/3 will sense a short-circuit on the output by monitoring the output voltage. When the attenuated feedback voltage has fallen below 60% of the reference voltage, Vref x 0.6 (≈ 0.36V), short-circuit mode of operation will start. During short-circuit operation, the SS pin is discharged and the output voltage will fall to 0V. The SS pin voltage, VSS, is then ramped back up at the rate determined by the SS capacitor and ISS until VSS reaches 0.7V. During this re-ramp phase, if the short-circuit fault is still present the output current will be equal to the set current limit. Once the soft-start voltage reaches 0.7V the output voltage is sensed again and if the attenuated VFB is still below Vref x 0.6 then the SS pin is discharged again and the cycle repeats until the short-circuit fault is removed. tSS ≥ (VOUT x COUT) / (IOCL - IOUT) Soft-Start Enable/Shutdown The soft-start (SS) feature allows the regulator to gradually reach a steady-state operating point, which reduces start-up stresses and current surges. At turn-on, while VCC is below the under-voltage threshold, the SS pin is internally grounded and VOUT is held at 0V. The SS capacitor is used to slowly ramp VFB from 0V to it's final output voltage as programmed by the internal resistor divider. By changing the soft-start capacitor value, the duration of start-up can be changed accordingly. The start-up time can be calculated using the following equation: The EN pin can be activated by either leaving the pin floating due to an internal pull up resistor to VIN or by applying a logic high signal to the EN pin of 1.26V or greater. The LM3151/2/3 can be remotely shut down by taking the EN pin below 1.02V. Low quiescent shutdown is achieved when VEN is less than 0.4V. During low quiescent shutdown the internal bias circuitry is turned off. The LM3151/2/3 has certain fault conditions that can trigger shutdown, such as over-voltage protection, current limit, under-voltage lockout, or thermal shutdown. During shutdown, the soft-start capacitor is discharged. Once the fault condition is removed, the soft-start capacitor begins charging, allowing the part to start up in a controlled fashion. In conditions where there may be an open drain connection to the EN pin, it may be necessary to add a 1000 pF bypass capacitor to this pin. This will help decouple noise from the EN pin and prevent false disabling. Where tSS is measured in seconds, Vref = 0.6V and ISS is the soft-start pin source current, which is typically 7.7 µA (refer to electrical characteristics table). www.national.com 10 The LM3151/2/3 should be operated such that the junction temperature does not exceed the maximum operating junction temperature. An internal thermal shutdown circuit, which activates at 165°C (typical), takes the controller to a low-power reset state by disabling the buck switch and the on-timer, and grounding the SS pin. This feature helps prevent catastrophic failures from accidental device overheating. When the junction temperature falls back below 150°C the SS pin is released and normal operation resumes. 3. Determine Inductor Required Using Figure 2 To use the nomograph below calculate the inductor volt-microsecond constant ET from the following formula: Design Guide The design guide provides the equations required to design with the LM3151/2/3 SIMPLE SWITCHER® Controller. WEBENCH® design tool can be used with or in place of this section for a more complete and simplified design process. 1. Define Power Supply Operating Conditions a. Maximum and Minimum DC Input voltage b. Maximum Expected Load Current during normal operation Where fS is in kHz units. The intersection of the Load Current and the Volt-microseconds lines on the chart below will determine which inductors are capable for use in the design. The chart shows a sample of parts that can be used. The offline calculator tools and WEBENCH® will fully calculate the requirements for the components needed for the design. 30053252 FIGURE 2. Inductor Nomograph 11 www.national.com LM3151/LM3152/LM3153 c. Target Switching Frequency 2. Determine which IC Controller to Use The desired input voltage range will determine which version of the LM3151/2/3 controller will be chosen. The higher switching frequency options allow for physically smaller inductors but efficiency may decrease. Thermal Protection LM3151/LM3152/LM3153 TABLE 1. Inductor Selection Table Inductor Designator Inductance (µH) Current (A) Part Name Vendor L01 47 7-9 L02 33 L03 22 7-9 SER2817H-333KL COILCRAFT 7-9 SER2814H-223KL L04 COILCRAFT 15 7-9 7447709150 WURTH L05 10 7-9 RLF12560T-100M7R5 TDK L06 6.8 7-9 B82477-G4682-M EPCOS L07 4.7 7-9 B82477-G4472-M EPCOS L08 3.3 7-9 DR1050-3R3-R COOPER L09 2.2 7-9 MSS1048-222 COILCRAFT L10 1.5 7-9 SRU1048-1R5Y BOURNS L11 1 7-9 DO3316P-102 COILCRAFT L12 0.68 7-9 DO3316H-681 COILCRAFT L13 33 9-12 L14 22 9-12 SER2918H-223 COILCRAFT L15 15 9-12 SER2814H-153KL COILCRAFT L16 10 9-12 7447709100 WURTH L17 6.8 9-12 SPT50H-652 COILCRAFT L18 4.7 9-12 SER1360-472 COILCRAFT L19 3.3 9-12 MSS1260-332 COILCRAFT L20 2.2 9-12 DR1050-2R2-R COOPER L21 1.5 9-12 DR1050-1R5-R COOPER L22 1 9-12 DO3316H-102 COILCRAFT L23 0.68 9-12 L24 0.47 9-12 L25 22 12-15 SER2817H-223KL COILCRAFT L26 15 12-15 L27 10 12-15 SER2814L-103KL COILCRAFT L28 6.8 12-15 7447709006 WURTH L29 4.7 12-15 7447709004 WURTH L30 3.3 12-15 L31 2.2 12-15 L32 1.5 12-15 MLC1245-152 COILCRAFT L33 1 12-15 L34 0.68 12-15 DO3316H-681 COILCRAFT L35 0.47 12-15 L36 0.33 12-15 DR73-R33-R COOPER L37 22 15- L38 15 15- SER2817H-153KL COILCRAFT L39 10 15- SER2814H-103KL COILCRAFT L40 6.8 15- L41 4.7 15- SER2013-472ML COILCRAFT L42 3.3 15- SER2013-362L COILCRAFT L43 2.2 15- L44 1.5 15- HA3778-AL COILCRAFT L45 1 15- B82477-G4102-M EPCOS L46 0.68 15- L47 0.47 15- L48 0.33 15- www.national.com 12 LM3151/LM3152/LM3153 4. Determine Output Capacitance Typical hysteretic COT converters similar to the LM3151/2/3 require a certain amount of ripple that is generated across the ESR of the output capacitor and fed back to the error comparator. Emulated Ripple Mode control built into the LM3151/2/3 will recreate a similar ripple signal and thus the requirement for output capacitor ESR will decrease compared to a typical Hysteretic COT converter. The emulated ripple is generated by sensing the voltage signal across the low-side FET and is then compared to the FB voltage at the error comparator input to determine when to initiate the next on-time period. COmin = 70 / (fs2 x L) The maximum ESR allowed to prevent over-voltage protection during normal operation is: ESRmax = (80 mV x L) / ETmin 30053281 ETmin is calculated using VIN-MIN FIGURE 3. Typical MOSFET Gate Charge Curve The minimum ESR must meet both of the following criteria: ESRmin ≥ (15 mV x L) / ETmax See following design example for estimated power dissipation calculation. ESRmin ≥ [ETmax / (VIN - VOUT)]/ CO 6. Calculate Input Capacitance The main parameters for the input capacitor are the voltage rating, which must be greater than or equal to the maximum DC input voltage of the power supply, and its rms current rating. The maximum rms current is approximately 50% of the maximum load current. ETmax is calculated using VIN-MAX. Any additional parallel capacitors should be chosen so that their effective impedance will not negatively attenuate the output ripple voltage. 5. MOSFET Selection The high-side and low-side FETs must have a drain to source (VDS) rating of at least 1.2 x VIN. The gate drive current from VCC must not exceed the minimum current limit of VCC. The drive current from VCC can be calculated with: Where, ΔVIN-MAX is the maximum allowable input ripple voltage. A good starting point for the input ripple voltage is 5% of VIN. When using low ESR ceramic capacitors on the input of the LM3151/2/3 a resonant circuit can be formed with the impedance of the input power supply and parasitic impedance of long leads/PCB traces to the LM3151/2/3 input capacitors. It is recommended to use a damping capacitor under these circumstances, such as aluminum electrolytic that will prevent ringing on the input. The damping capacitor should be chosen to be approximately 5 times greater than the parallel ceramic capacitors combination. The total input capacitance should be greater than 10 times the input inductance of the power supply leads/pcb trace. The damping capacitor should also be chosen to handle its share of the rms input current which is shared proportionately with the parallel impedance of the ceramic capacitors and aluminum electrolytic at the LM3151/2/3 switching frequency. The CBYP capacitor should be placed directly at the VIN pin. The recommended value is 0.1 µF. 7. Calculate Soft-Start Capacitor IVCCdrive = Qgtotal x fS Where, Q gtotal is the combined total gate charge of the highside and low-side FETs. Use the following equations to calculate the current limit, ICL, as shown in Figure 1. Tj is the junction temperature of the LM3151/2/3. The plateau voltage of the FET VGS vs Qg curve, as shown in Figure 3 must be less than VCC - 750 mV. Where tSS is the soft-start time in seconds and Vref = 0.6V. 13 www.national.com LM3151/LM3152/LM3153 high-side FET. It is charged during the SW off-time. The recommended value for CBST is 0.47 µF. The EN bypass capacitor, CEN, recommended value is 1000 pF when driving the EN pin from open drain type of signal. 8. CVCC, and CBST and CEN CVCC should be placed directly at the VCC pin with a recommended value of 1 µF to 2.2 µF. For input voltage ranges that include voltages below 8V a 1 µF capacitor must be used for CVCC. CBST creates a voltage used to drive the gate of the Design Example 30053261 FIGURE 4. Design Example Schematic 1. Define Power Supply Operating Conditions a. VOUT = 3.3V b. VIN-MIN = 6V, VIN-TYP = 12V, VIN-MAX = 24V For this design the chosen ripple current ratio, r = 0.3, represents the ratio of inductor peak-to-peak current to load current Iout. A good starting point for ripple ratio is 0.3 but it is acceptable to choose r between 0.25 to 0.5. The nomographs in this datasheet all use 0.3 as the ripple current ratio. c. Typical Load Current = 12A, Max Load Current = 15A d. Soft-Start time tSS = 5 ms 2. Determine which IC Controller to Use The LM3151 and LM3152 allow for the full input voltage range. However, from buck converter basic theory, the higher switching frequency will allow for a smaller inductor. Therefore, the LM3152-3.3 500 kHz part is chosen so that a smaller inductor can be used. 3. Determine Inductor Required Irmsco = 1A a. ET = (24-3.3) x (3.3/24) x (1000/500) = 5.7 V µs tON = (3.3V/12V) / 500 kHz = 550 ns b. From the inductor nomograph a 12A load and 5.7 V µs calculation corresponds to a L44 type of inductor. Minimum output capacitance is: c. Using the inductor designator L44 in Table 1 the Coilcraft HA3778-AL 1.65 µH inductor is chosen. COmin = 70 / (fS2 x L) 4. Determine Output Capacitance The voltage rating on the output capacitor should be greater than or equal to the output voltage. As a rule of thumb most capacitor manufacturers suggests not to exceed 90% of the capacitor rated voltage. In the case of multilayer ceramics the capacitance will tend to decrease dramatically as the applied voltage is increased towards the capacitor rated voltage. The capacitance can decrease by as much as 50% when the applied voltage is only 30% of the rated voltage. The chosen capacitor should also be able to handle the rms current which is equal to: COmin = 70 / (500 kHz2 x 1.65 µH) = 169 µF www.national.com The maximum ESR allowed to prevent over-voltage protection during normal operation is: ESRmax = (80 mV x L) / ET ESRmax = (80 mV x 1.65 µH) / 5.7 V µs ESRmax = 23 mΩ The minimum ESR must meet both of the following criteria: ESRmin ≥ (15 mV x L) / ET 14 tion temperature rise above ambient temperature and θJA = 30°C/W, can be estimated by: ESRmin ≥ (15 mV x 1.65 µH) / 5.7 V µs = 4.3 mΩ Pdmax = 125°C / 30°C/W = 4.1W ESRmin ≥ [5.7 V µs / (12 - 3.3)] / 169 µF = 3.9 mΩ The system calculated Pdh of 0.674W is much less than the FET Pdmax of 4.1W and therefore the RJK0305DPB max allowable power dissipation criteria is met. Based on the above criteria two 150 µF polymer aluminum capacitors with a ESR = 12 mΩ each for a effective ESR in parallel of 6 mΩ was chosen from Panasonic. The part number is EEF-UE0J151P. Low-Side MOSFET Primary loss is conduction loss given by: Pdl = Iout2 x RDS(ON) x (1-D) = 122 x 0.01 x (1-0.275) = 1W 5. MOSFET Selection The LM3151/2/3 are designed to drive N-channel MOSFETs. For a maximum input voltage of 24V we should choose Nchannel MOSFETs with a maximum drain-source voltage, VDS, greater than 1.2 x 24V = 28.8V. FETs with maximum VDS of 30V will be the first option. The combined total gate charge Qgtotal of the high-side and low-side FET should satisfy the following: Pdl is also less than the Pdmax specified on the RJK0305DPB MOSFET datasheet. However, it is not always necessary to use the same MOSFET for both the high-side and low-side. For most applications it is necessary to choose the high-side MOSFET with the lowest gate charge and the low-side MOSFET is chosen for the lowest allowed RDS(ON). The plateau voltage of the FET VGS vs Qg curve must be less than VCC - 750 mV. The current limit, IOCL, is calculated by estimating the RDS (ON) of the low-side FET at the maximum junction temperature of 100°C. Then the following calculation of IOCL is: Qgtotal ≤ IVCCL / fs Qgtotal ≤ 65 mA / 500 kHz Qgtotal ≤ 130 nC IOCL = ICL + ΔIL / 2 Where IVCCL is the minimum current limit of VCC, over the temperature range, specified in the electrical characteristics table. The MOSFET gate charge Qg is gathered from reading the VGS vs Qg curve of the MOSFET datasheet at the VGS = 5V for the high-side, M1, MOSFET and VGS = 6V for the lowside, M2, MOSFET. The Renesas MOSFET RJK0305DPB has a gate charge of 10 nC at VGS = 5V, and 12 nC at VGS = 6V. This combined gate charge for a high-side, M1, and low-side, M2, MOSFET 12 nC + 10 nC = 22 nC is less than 130 nC calculated Qgtotal. ICL = 200 mV / 0.014 = 14.2A IOCL = 14.2A + 3.6 / 2 = 16A 6. Calculate Input Capacitance The input capacitor should be chosen so that the voltage rating is greater than the maximum input voltage which for this example is 24V. Similar to the output capacitor, the voltage rating needed will depend on the type of capacitor chosen. The input capacitor should also be able to handle the input rms current which is approximately 0.5 x IOUT. For this example the rms input current is approximately 0.5 x 12A = 6A. The minimum capacitance with a maximum 5% input ripple ΔVIN-MAX = (0.05 x 12) = 0.6V: The calculated MOSFET power dissipation must be less than the max allowed power dissipation, Pdmax, as specified in the MOSFET datasheet. An approximate calculation of the FET power dissipated Pd, of the high-side and low-side FET is given by: High-Side MOSFET CIN = [12 x 0.275 x (1-0.275)] / [500 kHz x 0.6] = 8 µF To handle the large input rms current 2 ceramic capacitors are chosen at 10 µF each with a voltage rating of 50V and case size of 1210, that can handle 3A of rms current each. A 100 µF aluminum electrolytic is chosen to help dampen input ringing. CBYP = 0.1 µF ceramic with a voltage rating greater than maximum VIN 7. Calculate Soft-Start Capacitor The soft start-time should be greater than the input voltage rise time and also satisfy the following equality to maintain a smooth transition of the output voltage to the programmed regulation voltage during startup. tSS ≥ (VOUT x COUT) / (IOCL - IOUT) 5 ms > (3.3V x 300 µF) / (1.2 x 12A - 12A) The max power dissipation of the RJK0305DPB is rated as 45W for a junction temperature that is 125°C higher than the case temperature and a thermal resistance from the FET junction to case, θJC, of 2.78°C/W. When the FET is mounted onto the PCB, the PCB will have some additional thermal resistance such that the total system thermal resistance of the FET package and the PCB, θJA, is typically in the range of 30° C/W for this type of FET package. The max power dissipation, Pdmax, with the FET mounted onto a PCB with a 125°C junc- 5 ms > 0.412 ms The desired soft-start time, tSS, of 5 ms satisfies the equality as shown above. Therefore, the soft-start capacitor, CSS, is calculated as: CSS = (7.7 µA x 5 ms) / 0.6V = 0.064 µF 15 www.national.com LM3151/LM3152/LM3153 ESRmin ≥ [ET / (VIN - VOUT)] / CO LM3151/LM3152/LM3153 Let CSS = 0.068 µF, which is the next closest standard value. This should be a ceramic cap with a voltage rating greater than 10V. 8. CVCC, CEN, and CBST CVCC = 1µF ceramic with a voltage rating greater than 10V CEN = 1000 pF ceramic with a voltage rating greater than 10V CBST = 0.47 µF ceramic with a voltage rating greater than 10V Bill of Materials Designator Value Parameters Manufacturer Part Number CBST 0.47 µF Ceramic, X7R, 16V, 10% TDK C2012X7R1C474K CBYP 0.1 µF Ceramic, X7R, 50V, 10% TDK C2012X7R1H104K CEN 1000 pF Ceramic, X7R, 50V, 10% TDK C1608X7R1H102K CIN1 100 µF AL, EEV-FK, 63V, 20% Panasonic EEV-FK1J101P CIN2, CIN3 10 µF Ceramic, X5R, 35V, 10% Taiyo Yuden GMK325BJ106KN-T COUT1, COUT2 150 µF AL, UE, 6.3V, 20% Panasonic EEF-UE0J151R C0805C105K4RACTU CSS 0.068 µF Ceramic, 16V, 10% CVCC 1 µF Ceramic, X7R, 16V, 10% Kemet L1 1.65 µH Shielded Drum Core, A, 2.53 mΩ Coilcraft Inc. HA3778-AL M1, M2 30V 8 nC, RDS(ON) @4.5V = 10 mΩ Renesas RJK0305DB National Semiconductor LM3152MH-3.3 U1 www.national.com 16 0603YC683KAT2A It is good practice to layout the power components first, such as the input and output capacitors, FETs, and inductor. The first priority is to make the loop between the input capacitors and the source of the low side FET to be very small and tie the grounds of each directly to each other and then to the ground plane through vias. As shown in the figure below, when the input cap ground is tied directly to the source of the low side FET, parasitic inductance in the power path, along with noise coupled into the ground plane, are reduced. The switch node is the next item of importance. The switch node should be made only as large as required to handle the load current. There are fast voltage transitions occurring in the switch node at a high frequency, and if the switch node is made too large it may act as an antennae and couple switching noise into other parts of the circuit. For high power designs it is recommended to use a multi-layer board. The FET’s are 30053258 FIGURE 5. Schematic of Parasitics 30053280 FIGURE 6. PCB Placement of Power Stage 17 www.national.com LM3151/LM3152/LM3153 going to be the largest heat generating devices in the design, and as such, care should be taken to remove the heat. On multi layer boards using exposed-pad packages for the FET’s such as the power-pak SO-8, vias should be used under the FETs to the same plane on the interior layers to help dissipate the heat and cool the FETs. For the typical single FET PowerPak type FETs the high-side FET DAP is Vin. The Vin plane should be copied to the other interior layers to the bottom layer for maximum heat dissipation. Likewise, the DAP of the lowside FET is connected to the SW node and it’s shape should be duplicated to the interior layers down to the bottom layer for maximum heat dissipation. See the Evaluation Board application note AN-1900 for an example of a typical multilayer board layout, and the Demonstration Board Reference Design App Note for a typical 2 layer board layout. Each design allows for single sided component mounting. PCB Layout Considerations LM3151/LM3152/LM3153 Physical Dimensions inches (millimeters) unless otherwise noted 14-Lead eTSSOP Package NS Package Number MXA14A www.national.com 18 LM3151/LM3152/LM3153 Notes 19 www.national.com LM3151/LM3152/LM3153 SIMPLE SWITCHER® CONTROLLER, High Input Voltage Synchronous Step-Down Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Design Support Amplifiers www.national.com/amplifiers WEBENCH® Tools www.national.com/webench Audio www.national.com/audio App Notes www.national.com/appnotes Clock and Timing www.national.com/timing Reference Designs www.national.com/refdesigns Data Converters www.national.com/adc Samples www.national.com/samples Interface www.national.com/interface Eval Boards www.national.com/evalboards LVDS www.national.com/lvds Packaging www.national.com/packaging Power Management www.national.com/power Green Compliance www.national.com/quality/green Switching Regulators www.national.com/switchers Distributors www.national.com/contacts LDOs www.national.com/ldo Quality and Reliability www.national.com/quality LED Lighting www.national.com/led Feedback/Support www.national.com/feedback Voltage Reference www.national.com/vref Design Made Easy www.national.com/easy www.national.com/powerwise Solutions www.national.com/solutions Mil/Aero www.national.com/milaero PowerWise® Solutions Serial Digital Interface (SDI) www.national.com/sdi Temperature Sensors www.national.com/tempsensors SolarMagic™ www.national.com/solarmagic Wireless (PLL/VCO) www.national.com/wireless www.national.com/training PowerWise® Design University THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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