LTC1474/LTC1475 Low Quiescent Current High Efficiency Step-Down Converters DESCRIPTION U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LT C®1474/LTC1475 series are high efficiency stepdown converters with internal P-channel MOSFET power switches that draw only 10µA typical DC supply current at no load while maintaining output voltage. The LTC1474 uses logic-controlled shutdown while the LTC1475 features pushbutton on/off. High Efficiency: Over 92% Possible Very Low Standby Current: 10µA Typ Available in Space Saving 8-Lead MSOP Package Internal 1.4Ω Power Switch (VIN = 10V) Wide VIN Range: 3V to 18V Operation Very Low Dropout Operation: 100% Duty Cycle Low-Battery Detector Functional During Shutdown Programmable Current Limit with Optional Current Sense Resistor (10mA to 400mA Typ) Short-Circuit Protection Few External Components Required Active Low Micropower Shutdown: IQ = 6µA Typ Pushbutton On/Off (LTC1475 Only) 3.3V, 5V and Adjustable Output Versions The low supply current coupled with Burst ModeTM operation enables the LTC1474/LTC1475 to maintain high efficiency over a wide range of loads. These features, along with their capability of 100% duty cycle for low dropout and wide input supply range, make the LTC1474/LTC1475 ideal for moderate current (up to 300mA) battery-powered applications. The peak switch current is user-programmable with an optional sense resistor (defaults to 325mA minimum if not used) providing a simple means for optimizing the design for lower current applications. The peak current control also provides short-circuit protection and excellent startup behavior. A low-battery detector that remains functional in shutdown is provided . U APPLICATIONS ■ ■ ■ ■ ■ ■ ■ Cellular Telephones and Wireless Modems 4mA to 20mA Current Loop Step-Down Converter Portable Instruments Battery-Operated Digital Devices Battery Chargers Inverting Converters Intrinsic Safety Applications The LTC1474/LTC1475 series availability in 8-lead MSOP and SO packages and need for few additional components provide for a minimum area solution. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U TYPICAL APPLICATION LTC1474 Efficiency 100 VIN 4V TO 18V LOW BATTERY OUT VIN = 5V 90 10µF 25V 0.1µF VIN = 10V 7 VIN 1 SENSE VFB LTC1474-3.3 3 2 LBI LBO 6 LOW BATTERY IN RUN SHDN 100k 8 RUN GND 4 SW VOUT 3.3V AT 250mA L1 100µH + 100µF 6.3V 5 D1 MBR0530 EFFICIENCY (%) + 80 VIN = 15V 70 60 L = 100µH VOUT = 3.3V RSENSE = 0Ω 1474/75 F01 L1 = SUMIDA CDRH74-101 50 Figure 1. High Efficiency Step-Down Converter 0.03 0.3 30 3 LOAD CURRENT (mA) 300 1474/75 TA01 1 LTC1474/LTC1475 W W U W ABSOLUTE MAXIMUM RATINGS Input Supply Voltage (VIN).........................– 0.3V to 20V Switch Current (SW, SENSE) .............................. 750mA Switch Voltage (SW).............. (VIN – 20V) to (VIN + 0.3V) VFB (Adjustable Versions) ..........................– 0.3V to 12V VOUT (Fixed Versions) ................................ –0.3V to 20V LBI, LBO ....................................................– 0.3V to 20V RUN, SENSE .................................. – 0.3V to (VIN + 0.3V) Operating Ambient Temperature Range Commercial ............................................ 0°C to 70°C Industrial ............................................ – 40°C to 85°C Junction Temperature (Note 1) ............................ 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C W U U PACKAGE/ORDER INFORMATION TOP VIEW TOP VIEW VOUT/VFB LBO LBI GND TOP VIEW 8 7 6 5 1 2 3 4 RUN VIN SENSE SW MS8 PACKAGE 8-LEAD PLASTIC MSOP VOUT/VFB LBO LBI/OFF GND 8 RUN VOUT/VFB 1 8 7 6 5 1 2 3 4 ON VIN SENSE SW MS8 PACKAGE 8-LEAD PLASTIC MSOP TOP VIEW LBO 2 7 VIN LBI 3 6 SENSE GND 4 5 SW S8 PACKAGE 8-LEAD PLASTIC SO VOUT/VFB 1 8 ON LBO 2 7 VIN LBI/OFF 3 6 SENSE GND 4 5 SW S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125°C, θJA = 150°C/ W TJMAX = 125°C, θJA = 150°C/ W TJMAX = 125°C, θJA = 110°C/ W TJMAX = 125°C, θJA = 110°C/ W ORDER PART NUMBER ORDER PART NUMBER ORDER PART NUMBER ORDER PART NUMBER LTC1474CMS8 LTC1474CMS8-3.3 LTC1474CMS8-5 LTC1475CMS8 LTC1475CMS8-3.3 LTC1475CMS8-5 LTC1474CS8 LTC1474IS8 LTC1474CS8-3.3 LTC1474CS8-5 LTC1474IS8-3.3 LTC1474IS8-5 LTC1475CS8 LTC1475IS8 LTC1475CS8-3.3 LTC1475CS8-5 MS8 PART MARKING MS8 PART MARKING S8 PART MARKING S8 PART MARKING 1474 1474I 14743 14745 14743I 14745I 1475 1475I 14753 14755 LTBW LTCR LTCS Consult factory for Military grade parts. 2 LTBK LTCP LTCQ LTC1474/LTC1475 ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 10V, VRUN = open, RSENSE = 0, unless otherwise noted. SYMBOL PARAMETER CONDITIONS VFB Feedback Voltage LTC1474/LTC1475 ILOAD = 50mA VOUT Regulated Output Voltage LTC1474-3.3/LTC1475-3.3 LTC1474-5/LTC1475-5 ILOAD = 50mA MIN TYP MAX UNITS ● 1.205 1.230 1.255 V ● ● 3.234 4.900 3.300 5.000 3.366 5.100 V V 0 30 nA IFB Feedback Current LTC1474/LTC1475 Only ISUPPLY No Load Supply Current (Note 3) ILOAD = 0 (Figure 1 Circuit) 10 ∆VOUT Output Voltage Line Regulation VIN = 7V to 12V, ILOAD = 50mA 5 20 mV Output Voltage Load Regulation ILOAD = 0mA to 50mA 2 15 mV Output Ripple ILOAD = 10mA 50 Input DC Supply Current (Note 2) Active Mode (Switch On) Sleep Mode (Note 3) Shutdown (Exclusive of Driver Gate Charge Current) VIN = 3V to 18V VIN = 3V to 18V VIN = 3V to 18V, VRUN = 0V 100 9 6 175 15 12 µA µA µA RON Switch Resistance ISW = 100mA 1.4 1.6 Ω IPEAK Current Comp Max Current Trip Threshold RSENSE = 0Ω RSENSE = 1.1Ω 325 70 400 76 85 mA mA 90 100 110 mV IQ ● µA mVP-P VSENSE Current Comp Sense Voltage Trip Threshold VHYST Voltage Comparator Hysteresis tOFF Switch Off-Time VLBI, TRIP Low Battery Comparator Threshold VRUN Run/ON Pin Threshold 0.4 0.7 1.0 V VLBI, OFF OFF Pin Threshold (LTC1475 Only) 0.4 0.7 1.0 V ILBO, SINK Sink Current into Pin 2 VLBI = 0V, VLBO = 0.4V 0.45 0.70 IRUN, SOURCE Source Current from Pin 8 VRUN = 0V 0.4 ISW, LEAK Switch Leakage Current VIN = 18V, VSW = 0V, VRUN = 0V ILBI, LEAK Leakage Current into Pin 3 ILBO, LEAK Leakage Current into Pin 2 The ● denotes specifications which apply over the full operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1474CS8/LTC1475CS8: TJ = TA + (PD • 110°C/W) LTC1474CMS8/LTC1475CMS8: TJ = TA + (PD • 150°C/W) ● 5 VOUT at Regulated Value VOUT = 0V ● mV 3.5 4.75 65 6.0 µs µs 1.16 1.23 1.27 V mA 0.8 1.2 µA 0.015 1 µA VLBI = 18V, VIN = 18V 0 0.1 µA VLBI = 2V, VLBO = 5V 0 0.5 µA Note 2: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 3: No load supply current consists of sleep mode DC current (9µA typical) plus a small switching component (about 1µA for Figure 1 circuit) necessary to overcome Schottky diode and feedback resistor leakage. 3 LTC1474/LTC1475 U W TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Input Voltage Line Regulation 40 FIGURE 1 CIRCUIT L: CDRH73-101 90 30 ILOAD = 200mA 85 FIGURE 1 CIRCUIT 30 20 RSENSE = 0Ω 20 ILOAD = 25mA ∆VOUT (mV) EFFICIENCY (%) 95 40 FIGURE 1 CIRCUIT ILOAD = 100mA ∆VOUT (mV) 100 Load Regulation 10 RSENSE = 0.33Ω 80 0 70 –10 0 4 –20 16 12 8 INPUT VOLTAGE (V) 10 VIN = 10V 0 VIN = 5V –10 ILOAD = 1mA 75 VIN = 15V –20 –30 0 4 0 16 12 8 INPUT VOLTAGE (V) 1474/75 G01 50 250 150 200 100 LOAD CURRENT (mA) 300 1474/75 G02 1474/75 G03 Switch Resistance vs Input Voltage Current Trip Threshold vs Temperature 500 Supply Current in Shutdown 10.0 5 RSENSE = 0Ω 4 300 200 RSENSE = 1.1Ω 100 0 SUPPLY CURRENT (µA) 400 RDS(ON) (Ω) CURRENT TRIP THRESHOLD (mA) VIN = 10V 3 2 T = 70°C 1 T = 25°C 0 0 20 40 60 80 5 10 2.5 0 20 15 5 10 1474/75 G05 Switch Leakage Current vs Temperature 1474/75 G06 Off-Time vs Output Voltage VIN DC Supply Current 80 120 1.0 20 15 INPUT VOLTAGE (V) INPUT VOLTAGE (V) 1474/75 G04 VIN = 18V VIN = 10V ACTIVE MODE 100 0.6 0.4 0.2 60 80 OFF-TIME (µs) 0.8 SUPPLY CURRENT (µA) LEAKAGE CURRENT (µA) 5.0 0 0 TEMPERATURE (°C) 60 0 20 40 60 TEMPERATURE (°C) 80 100 1474/75 G07 40 40 20 20 0 4 7.5 0 SLEEP MODE 0 0 4 12 8 INPUT VOLTAGE (V) 16 20 1474/75 G08 0 40 100 20 60 80 % OF REGULATED OUTPUT VOLTAGE (%) 1474/75 G09 LTC1474/LTC1475 U U U PIN FUNCTIONS VOUT/VFB (Pin 1): Feedback of Output Voltage. In the fixed versions, an internal resistive divider divides the output voltage down for comparison to the internal 1.23V reference. In the adjustable versions, this divider must be implemented externally. SW (Pin 5): Drain of Internal PMOS Power Switch. Cathode of Schottky diode must be closely connected to this pin. SENSE (Pin 6): Current Sense Input for Monitoring Switch Current and Source of Internal PMOS Power Switch. Maximum switch current is programmed with a resistor between SENSE and VIN pins. LBO (Pin 2): Open Drain Output of the Low Battery Comparator. This pin will sink current when Pin 3 is below 1.23V. VIN (Pin 7): Main Supply Pin. LBI/OFF (Pin 3): Input to Low Battery Comparator. This input is compared to the internal 1.23V reference. For the LTC1475, a momentary ground on this pin puts regulator in shutdown mode. RUN/ON (Pin 8): On LTC1474, voltage level on this pin controls shutdown/run mode (ground = shutdown, open/ high = run). On LTC1475, a momentary ground on this pin puts regulator in run mode. A 100k series resistor must be used between Pin 8 and the switch or control voltage. GND (Pin 4): Ground Pin. W FUNCTIONAL DIAGRA U U LBI/OFF 100mV 1µA VIN – × ON VIN 7 C RSENSE (OPTIONAL) + + VCC – 6 V ON SENSE 5Ω + LTC1474: RUN 8 LTC1475: ON 4.75µs 1× 1-SHOT TRIGGER 20× OUT STRETCH WAKEUP LBO SW 2 5 VOUT + + 3M (5V VERSION) 1.68M (3.3V VERSION) LB 1.23V – READY 3 LTC1474: LBI LTC1475: LBI/OFF × CONNECTION NOT PRESENT IN LTC1474 SERIES CONNECTION PRESENT IN LTC1474 SERIES ONLY VOUT/VFB 1 1.23V REFERENCE 4 GND 1M OUTPUT DIVIDER IS IMPLEMENTED EXTERNALLY IN ADJUSTABLE VERSIONS 1474/75 FD 5 LTC1474/LTC1475 U OPERATIO (Refer to Functional Diagram) The LTC1474/LTC1475 are step-down converters with internal power switches that use Burst Mode operation to keep the output capacitor charged to the proper output voltage while minimizing the quiescent current. Burst Mode operation functions by using short “burst” cycles to ramp the inductor current through the internal power switch and external Schottky diode, followed by a sleep cycle where the power switch is off and the load current is supplied by the output capacitor. During sleep mode, the LTC1474/LTC1475 draw only 9µA typical supply current. At light loads, the burst cycles are a small percentage of the total cycle time; thus the average supply current is very low, greatly enhancing efficiency. Peak Inductor Current Programming Burst Mode Operation Off-Time At the beginning of the burst cycle, the switch is turned on and the inductor current ramps up. At this time, the internal current comparator is also turned on to monitor the switch current by measuring the voltage across the internal and optional external current sense resistors. When this voltage reaches 100mV, the current comparator trips and pulses the 1-shot timer to start a 4.75µs off-time during which the switch is turned off and the inductor current ramps down. At the end of the off-time, if the output voltage is less than the voltage comparator threshold, the switch is turned back on and another cycle commences. To minimize supply current, the current comparator is turned on only during the switch-on period when it is needed to monitor switch current. Likewise, the 1-shot timer will only be on during the 4.75µs off-time period. The off-time duration is 4.75µs when the feedback voltage is close to the reference; however, as the feedback voltage drops, the off-time lengthens and reaches a maximum value of about 65µs when this voltage is zero. This ensures that the inductor current has enough time to decay when the reverse voltage across the inductor is low such as during short circuit. The average inductor current during a burst cycle will normally be greater than the load current, and thus the output voltage will slowly increase until the internal voltage comparator trips. At this time, the LTC1474/LTC1475 go into sleep mode, during which the power switch is off and only the minimum required circuitry is left on: the voltage comparator, reference and low battery comparator. During sleep mode, with the output capacitor supplying the load current, the VFB voltage will slowly decrease until it reaches the lower threshold of the voltage comparator (about 5mV below the upper threshold). The voltage comparator then trips again, signaling the LTC1474/ LTC1475 to turn on the circuitry necessary to begin a new burst cycle. 6 The current comparator provides a means for programming the maximum inductor/switch current for each switch cycle. The 1X sense MOSFET, a portion of the main power MOSFET, is used to divert a sample (about 5%) of the switch current through the internal 5Ω sense resistor. The current comparator monitors the voltage drop across the series combination of the internal and external sense resistors and trips when the voltage exceeds 100mV. If the external sense resistor is not used (Pins 6 and 7 shorted), the current threshold defaults to the 400mA maximum due to the internal sense resistor. Shutdown Mode Both LTC1474 and LTC1475 provide a shutdown mode that turns off the power switch and all circuitry except for the low battery comparator and 1.23V reference, further reducing DC supply current to 6µA typical. The LTC1474’s run/shutdown mode is controlled by a voltage level at the RUN pin (ground = shutdown, open/high = run). The LTC1475’s run/shutdown mode, on the other hand, is controlled by an internal S/R flip-flop to provide pushbutton on/off control. The flip-flop is set (run mode) by a momentary ground at the ON pin and reset (shutdown mode) by a momentary ground at the LBI/OFF pin. Low Battery Comparator The low battery comparator compares the voltage on the LBI pin to the internal reference and has an open drain N-channel MOSFET at its output. If LBI is above the reference, the output FET is off and the LBO output is high impedance. If LBI is below the reference, the output FET is on and sinks current. The comparator is still active in shutdown. LTC1474/LTC1475 U W U U APPLICATIONS INFORMATION The basic LTC1474/LTC1475 application circuit is shown in Figure 1, a high efficiency step-down converter. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, L can be chosen. Finally D1, CIN and COUT are selected. RSENSE Selection The current sense resistor (RSENSE) allows the user to program the maximum inductor/switch current to optimize the inductor size for the maximum load. The LTC1474/ LTC1475 current comparator has a maximum threshold of 100mV/(RSENSE + 0.25). The maximum average output current IMAX is equal to this peak value less half the peakto-peak ripple current ∆IL. Allowing a margin for variations in the LTC1474/LTC1475 and external components, the required RSENSE can be calculated from Figure 2 and the following equation: RSENSE = (0.067/IMAX) – 0.25 (1) for 10mA < IMAX < 200mA. ments. Lower peak currents have the advantage of lower output ripple (∆VOUT = IPEAK • ESR), lower noise, and less stress on alkaline batteries and other circuit components. Also, lower peak currents allow the use of inductors with smaller physical size. Peak currents as low as 10mA can be programmed with the appropriate sense resistor. Increasing RSENSE above 10Ω, however, gives no further reduction of IPEAK. For RSENSE values less than 1Ω, it is recommended that the user parallel standard resistors (available in values ≥ 1Ω) instead of using a special low valued shunt resistor. Although a single resisor could be used with the desired value, these low valued shunt resistor types are much more expensive and are currently not available in case sizes smaller than 1206. Three or four 0603 size standard resistors require about the same area as one 1206 size current shunt resistor at a fraction of the cost. At higher supply voltages and lower inductances, the peak currents may be slightly higher due to current comparator overshoot and can be predicted from the second term in the following equation: 5 0.1 IPEAK = + 0.25 + RSENSE 4 RSENSE (Ω) FOR LOWEST NOISE 3 FOR BEST EFFICIENCY 2 1 0 0 250 100 150 200 50 MAXIMUM OUTPUT CURRENT (mA) 300 1474/75 F02 Figure 2. RSENSE Selection For IMAX above 200mA, RSENSE is set to zero by shorting Pins 6 and 7 to provide the maximum peak current of 400mA (limited by the fixed internal sense resistor). This 400mA default peak current can be used for lower IMAX if desired to eliminate the need for the sense resistor and associated decoupling capacitor. However, for optimal performance, the peak inductor current should be set to no more than what is needed to meet the load current require- (2.5)10−7 (VIN − VOUT) L (2) Note that RSENSE only sets the maximum inductor current peak. At lower dI/dt (lower input voltages and higher inductances), the observed peak current at loads less than IMAX may be less than this calculated peak value due to the voltage comparator tripping before the current ramps up high enough to trip the current comparator. This effect improves efficiency at lower loads by keeping the I2R losses down (see Efficiency Considerations section). Inductor Value Selection Once RSENSE and IPEAK are known, the inductor value can be determined. The minimum inductance recommended as a function of IPEAK and IMAX can be calculated from: ( ) 0.75 VOUT + VD tOFF L MIN ≥ IPEAK − IMAX where tOFF = 4.75µs. (3) 7 LTC1474/LTC1475 U W U U APPLICATIONS INFORMATION If the LMIN calculated is not practical, a larger IPEAK should be used. Although the above equation provides the minimum, better performance (efficiency, line/load regulation, noise) is usually gained with higher values. At higher inductances, peak current and frequency decrease (improving efficiency) and inductor ripple current decreases (improving noise and line/load regulation). For a given inductor type, however, as inductance is increased, DC resistance (DCR) increases, increasing copper losses, and current rating decreases, both effects placing an upper limit on the inductance. The recommended range of inductances for small surface mount inductors as a function of peak current is shown in Figure 3. The values in this range are a good compromise between the trade-offs discussed above. If space is not a premium, inductors with larger cores can be used, which extends the recommended range of Figure 3 to larger values. INDUCTOR VALUE (µH) 1000 500 100 50 10 100 PEAK INDUCTOR CURRENT (mA) 1000 1474/75 F03 Figure 3. Recommended Inductor Values Inductor Core Selection Once the value of L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, as discussed in the previous section, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite and Kool Mµ designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor current above IPEAK and consequent increase in voltage ripple. Do not allow the core to saturate! Coiltronics, Coilcraft, Dale and Sumida make high performance inductors in small surface mount packages with low loss ferrite and Kool Mµ cores and work well in LTC1474/LTC1475 regulators. Catch Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition, the diode must safely handle IPEAK at close to 100% duty cycle. To maximize both low and high current efficiency, a fast switching diode with low forward drop and low reverse leakage should be used. Low reverse leakage current is critical to maximize low current efficiency since the leakage can potentially approach the magnitude of the LTC1474/ LTC1475 supply current. Low forward drop is critical for high current efficiency since loss is proportional to forward drop. These are conflicting parameters (see Table 1), but a good compromise is the MBR0530 0.5A Schottky diode specified in the application circuits. Table 1. Effect of Catch Diode on Performance DIODE (D1) BAS85 LEAKAGE FORWARD NO LOAD DROP SUPPLY CURRENT EFFICIENCY* 200nA 0.6V 9.7µA 77.9% MBR0530 1µA 0.4V 10µA 83.3% MBRS130 20µA 0.3V 16µA 84.6% *Figure 1 circuit with VIN = 15V, IOUT = 0.1A Kool Mµ is a registered trademark of Magnetics, Inc. 8 LTC1474/LTC1475 U U W U APPLICATIONS INFORMATION CIN and COUT Selection At higher load currents, when the inductor current is continuous, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum capacitor current is given by: CIN Required IRMS = [ ( IMAX VOUT VIN − VOUT )] 1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also required on VIN for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR) to meet the output voltage ripple and line regulation requirements. The output voltage ripple during a burst cycle is dominated by the output capacitor ESR and can be estimated from the following relation: 25mV < ∆VOUT, RIPPLE = ∆IL • ESR where ∆IL ≤ IPEAK and the lower limit of 25mV is due to the voltage comparator hysteresis. Line regulation can also vary with COUT ESR in applications with a large input voltage range and high peak currents. ESR is a direct function of the volume of the capacitor. Manufacturers such as Nichicon, AVX and Sprague should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available from SANYO has the lowest ESR for its size at a somewhat higher price. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. For lower current applications with peak currents less than 50mA, 10µF ceramic capacitors provide adequate filtering and are a good choice due to their small size and almost negligible ESR. AVX and Marcon are good sources for these capacitors. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include SANYO OS-CON, Nichicon PL series and Sprague 595D series. Consult the manufacturer for other specific recommendations. To avoid overheating, the output capacitor must be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by: IRMS = IPEAK / 2 Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, three main sources usually account for most of the losses in LTC1474/LTC1475 circuits: VIN current, I2R losses and catch diode losses. 1. The VIN current is due to two components: the DC bias current and the internal P-channel switch gate charge current. The DC bias current is 9µA at no load and increases proportionally with load up to a constant 100µA during continuous mode. This bias current is so 9 LTC1474/LTC1475 U W U U APPLICATIONS INFORMATION small that this loss is negligible at loads above a milliamp but at no load accounts for nearly all of the loss. The second component, the gate charge current, results from switching the gate capacitance of the internal P-channel switch. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN which is typically much larger than the DC bias current. In continuous mode, IGATECHG = fQP where QP is the gate charge of the internal switch. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are predicted from the internal switch, inductor and current sense resistor. At low supply voltages where the switch on-resistance is higher and the switch is on for longer periods due to higher duty cycle, the switch losses will dominate. Keeping the peak currents low with the appropriate RSENSE and with larger inductance helps minimize these switch losses. At higher supply voltages, these losses are proportional to load and result in the flat efficiency curves seen in Figure 1. 3. The catch diode loss is due to the VDID loss as the diode conducts current during the off-time and is more pronounced at high supply voltage where the on-time is short. This loss is proportional to the forward drop. However, as discussed in the Catch Diode section, diodes with lower forward drops often have higher leakage current, so although efficiency is improved, the no load supply current will increase. Adjustable Applications For adjustable versions, the output voltage is programmed with an external divider from VOUT to VFB (Pin 1) as shown in Figure 4. The regulated voltage is determined by: R2 VOUT = 1.23 1+ R1 To minimize no-load supply current, resistor values in the megohm range should be used. The increase in supply current due to the feedback resistors can be calculated from: V V ∆I VIN = OUT OUT R1 + R2 VIN A 10pF feedforward capacitor across R2 is necessary due to the high impedances to prevent stray pickup and improve stability. VOUT 10pF R2 1 LTC1474 V FB LTC1475 R1 GND 4 1474/75 F04 Figure 4. LTC1474/LTC1475 Adjustable Configuration Low Battery Comparator The LTC1474/LTC1475 have an on-chip low battery comparator that can be used to sense a low battery condition when implemented as shown in Figure 5. The resistive divider R3/R4 sets the comparator trip point as follows: R4 VTRIP = 1.23 1 + R3 The divided down voltage at the LBI pin is compared to the internal 1.23V reference. When VLBI < 1.23V, the LBO output sinks current. The low battery comparator is active all the time, even during shutdown mode. VIN R4 LBI (4) R3 LTC1474/LTC1475 LBO – + 1.23V REFERENCE 1474/75 F05 Figure 5. Low Battery Comparator 10 LTC1474/LTC1475 U U W U APPLICATIONS INFORMATION LTC1475 Pushbutton On/Off and Microprocessor Interface The LTC1475 provides pushbutton control of power on/off for use with handheld products. A momentary ground on the ON pin sets an internal S/R latch to run mode while a momentary ground on the LBI/OFF pin resets the latch to shutdown mode. See Figure 6 for a comparsion of on/off operation of the LTC1474 and LTC1475 and Figure 7 for a comparison of the circuit implementations. In the LTC1475, the LBI/OFF pin has a dual function as both the shutdown control pin and the low battery comparator input. Since the “OFF” pushbutton is normally open, it does not affect the normal operation of the low battery comparator. In the unpressed state, the LBI/OFF input is the divided down input voltage from the resistive divider to the internal low battery comparator and will normally be above or just below the trip threshold of 1.23V. When shutdown mode is desired, the LBI/OFF pin is pulled below the 0.7V threshold to invoke shutdown. the depressed switch state is detected by the microcontroller through its input. The microcontroller then pulls the LBI/OFF pin low with the connection to one of its ouputs. With the LBI/OFF pin low, the LTC1475 powers down turning the microcontroller off. Note that since the I/O pins of most microcontrollers have a reversed bias diode between input and supply, a blocking diode with less than 1µA leakage is necessary to prevent the powered down microcontroller from pulling down on the ON pin. Figure 19 in the Typical Applications section shows how to use the low battery comparator to provide a low battery lockout on the “ON” switch. The LBO output disconnects the pushbutton from the ON pin when the comparator has tripped, preventing the LTC1475 from attempting to start up again until VIN is increased. 100k 100k ON RUN VIN RUN LTC1474 ON LTC1475 LBI/OFF RUN OFF LTC1474 MODE RUN SHUTDOWN RUN 1474/75 F07 ON OVERRIDES LBI/OFF WHILE ON IS LOW Figure 7. Simplified Implementation of LTC1474 and LTC1475 On/Off ON Absolute Maximum Ratings and Latchup Prevention LBI/OFF LTC1475 MODE RUN SHUTDOWN RUN 1474/75 F06 Figure 6. Comparison of LTC1474 and LTC1475 Run/Shutdown Operation The ON pin has precedence over the LBI/OFF pin. As seen in Figure 6, if both pins are grounded simultaneously, run mode wins. Figure 18 in the Typical Applications section shows an example for the use of the LTC1475 to control on/off of a microcontroller with a single pushbutton. With both the microcontroller and LTC1475 off, depressing the pushbutton grounds the LTC1475 ON pin and starts up the LTC1475 regulator which then powers up the microcontroller. When the pushbutton is depressed a second time, The absolute maximum ratings specify that SW (Pin 5) can never exceed VIN (Pin 7) by more than 0.3V. Normally this situation should never occur. It could, however, if the output is held up while the supply is pulled down. A condition where this could potentially occur is when a battery is supplying power to an LTC1474 or LTC1475 regulator and also to one or more loads in parallel with the the regulator’s VIN. If the battery is disconnected while the LTC1474 or LTC1475 regulator is supplying a light load and one of the parallel circuits is a heavy load, the input capacitor of the LTC1474 or LTC1475 regulator could be pulled down faster than the output capacitor, causing the absolute maximum ratings to be exceeded. The result is often a latchup which can be destructive if VIN is reapplied. Battery disconnect is possible as a result of mechanical stress, bad battery contacts or use of a lithium-ion battery 11 LTC1474/LTC1475 U W U U APPLICATIONS INFORMATION with a built-in internal disconnect. The user needs to assess his/her application to determine whether this situation could occur. If so, additional protection is necessary. where P is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. Prevention against latchup can be accomplished by simply connecting a Schottky diode across the SW and VIN pins as shown in Figure 8. The diode will normally be reverse biased unless VIN is pulled below VOUT at which time the diode will clamp the (VOUT – VIN) potential to less than the 0.6V required for latchup. Note that a low leakage Schottky should be used to minimize the effect on no-load supply current. Schottky diodes such as MBR0530, BAS85 and BAT84 work well. Another more serious effect of the protection diode leakage is that at no load with nothing to provide a sink for this leakage current, the output voltage can potentially float above the maximum allowable tolerance. To prevent this from occuring, a resistor must be connected between VOUT and ground with a value low enough to sink the maximum possible leakage current. The junction temperature is given by: LATCHUP PROTECTION SCHOTTKY TJ = TA + TR As an example consider the LTC1474/LTC1475 in dropout at an input voltage of 3.5V, a load current of 300mA, and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the on-resistance of the P-channel switch at 70°C is 3.5Ω. Therefore, power dissipated by the part is: P = I2 • RDS(ON) = 0.315W For the MSOP package, the θJA is 150°C/W. Thus the junction temperature of the regulator is: TJ = 70°C + (0.315)(150) = 117°C which is near the maximum junction temperature of 125oC. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance. PC Board Layout Checklist VIN VOUT SW LTC1474 LTC1475 + 1474/75 F08 Figure 8. Preventing Absolute Maximum Ratings from Being Exceeded Thermal Considerations In the majority of the applications, the LTC1474/LTC1475 do not dissipate much heat due to their high efficiency. However, in applications where the switching regulator is running at high ambient temperature with low supply voltage and high duty cycles, such as dropout with the switch on continuously, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated by the regulator exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = P • θJA 12 When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1474/LTC1475. These items are also illustrated graphically in the layout diagram of Figure 9. Check the following in your layout: 1. Is the Schottky diode cathode closely connected to SW (Pin 5)? 2. Is the 0.1µF input decoupling capacitor closely connected between VIN (Pin 7) and ground (Pin 4)? This capacitor carries the high frequency peak currents. 3. When using adjustable version, is the resistive divider closely connected to the (+) and (–) plates of COUT with a 10pF capacitor connected across R2? 4. Is the 1000pF decoupling capacitor for the current sense resistor connected as close as possible to Pins 6 and 7? If no current sense resistor is used, Pins 6 and 7 should be shorted. LTC1474/LTC1475 U U W U APPLICATIONS INFORMATION OUTPUT DIVIDER REQUIRED WITH ADJUSTABLE VERSION ONLY 10pF LTC1474 1 VOUT R2 R1 + 2 3 COUT 4 VFB 8 RUN VIN 7 100k L LBO LBI GND SENSE SW 1000pF 6 RSENSE 5 1474/75 F09 0.1µF CIN D1 + VIN BOLD LINES INDICATE HIGH PATH CURRENTS Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist) 5. Are the signal and power grounds segregated? The signal ground consists of the (–) plate of COUT, Pin 4 of the LTC1474/LTC1475 and the resistive divider. The power ground consists of the Schottky diode anode, the (–) plate of CIN and the 0.1µF decoupling capacitor. 6. Is a 100k resistor connected in series between RUN (Pin 8) and the RUN control voltage? The resistor should be as close as possible to Pin 8. Design Example (Refer to RSENSE and Inductor Selection) As a design example, assume VIN = 10V, VOUT = 3V, and a maximum average output current IMAX = 100mA. With this information, we can easily calculate all the important components: From the equation (1), RSENSE = (0.067/0.1) – 0.25 = 0.42Ω Using the standard resistors (1Ω, 1Ω and 2Ω) in parallel provides 0.4Ω without having to use a more expensive low value current shunt type resistor (see RSENSE Selection section). With RSENSE = 0.4Ω, the peak inductor current IPEAK is calculated from (2), neglecting the second term, to be 150mA. The minimum inductance is, therefore, from the equation (3) and assuming VD = 0.4V, L MIN = ( )( ) = 264µH 0.75 3.3 + 0.4 4.75µs 0.15 − 0.1 From Figure 3, an inductance of 270µH is chosen from the recommended region. The CDRH73-271 or CD54-271 is a good choice for space limited applications. For the feedback resistors, choose R1 = 1M to minimize supply current. R2 can then be calculated from the equation (4) to be: V R2 = OUT − 1 • R1 = 1.43 M 1.23 For the catch diode, the MBR0530 will work well in this application. For the input and output capacitors, AVX 4.7µF and 100µF, respectively, low ESR TPS series work well and meet the RMS current requirement of 100mA/2 = 50mA. They are available in small “C” case sizes with 0.15Ω ESR. The 0.15Ω output capacitor ESR will result in 25mV of output voltage ripple. Figure 10 shows the complete circuit for this example. 13 LTC1474/LTC1475 U TYPICAL APPLICATIONS VIN 3.5V TO 18V 10pF + 4.7µF† 35V 1Ω** 1Ω** 2Ω** 1000pF 6 3 * SUMIDA CDRH73-271 ** 3 PARALLEL STANDARD RESISTORS PROVIDE LEAST EXPENSIVE SOLUTION (SEE R SENSE SELECTION SECTION) † AVX TPSC475M035 †† AVX TPSC107M006 RUN 100k 8 0.1µF 7 VIN SENSE LBI RUN VFB LTC1474 LBO GND SW 1.43M 1% 1 1M 1% 2 L* 270µH + 5 VOUT 3V 100mA 100µF†† 6.3V D1 MBR0530 4 1474/75 F10 Figure 10. High Efficiency 3V/100mA Regulator (Design Example) IN + 4mA TO 20mA D2†† 12V 1µF ×3 2Ω 7.5M SENSE VOUT LTC1474-3.3 3 LBI LBO TO A/D MBR0530 IN 4mA TO 20mA 7 VIN 6 † – 1000pF 1M 100k RUN 8 RUN GND SW VOUT 3.3V 10mA 1 2 L* 330µH 4 D1 MBR0530 * COILCRAFT DO1608-334 ** MARCON THCS50E1E106Z, AVX 1206ZG106Z † OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER † † MOTOROLA MMBZ5242BL Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop 14 10µF** 5 1474/75 F11 LTC1474/LTC1475 U TYPICAL APPLICATIONS MBR0530 VIN 3.5V TO 6V 10pF + + 0.1µF 22µF** 16V 4.7M 1% 7 6 VIN SENSE 3 RUN VOUT –12V 70mA 22µF†† 25V LBI 100k 8 VFB LTC1474 LBO RUN SW GND 1 536k 1% 2 + 22µF†† 25V + 5 VIN (V) 10µF† L* 200µH 25V 4 * COILTRONICS CTX200-4 ** AVX TPSC226M016 † AVX TPSC106M025 †† AVX TPSD226M025 L* 200µH VOUT 12V 70mA D1 MBR0530 I LOAD(MAX) 3.5 30mA 4 50mA 5 70mA 6 90mA 1474/75 F12 Figure 12. 5V to ±12V Regulator + 0.1µF 10µF** 25V 7 6 3 RUN 100k 8 SENSE VIN LTC1474-5 LBI RUN LBO GND 4 * COILTRONICS CTX100-4 ** AVX TPSC106MO25 † AVX TPSC336M010 VOUT SW 1 2 10µF** 25V L* 100µH + 33µF† 10V VOUT 5V 200mA AT VIN = 10V + VIN 3.5V TO 12V 5 VIN (V) L* 100µH D1 MBR0530 3.5 I LOAD(MAX) 70mA 4 95mA 5 125mA 8 180mA 10 200mA 12 225mA 1474/75 F13 Figure 13. 5V Buck-Boost Converter 15 LTC1474/LTC1475 U TYPICAL APPLICATIONS VIN 3.5V TO 12V ON/OFF†† + 0.1µF 10µF** 25V TP0610 7 6 3 8 10M SENSE VIN VOUT LTC1474-5 LBI LBO RUN SW GND 1 2 VIN (V) L* 100µH + 33µF† 10V 5 D1 MBR0530 4 I LOAD(MAX) 3.5 100mA 5 140mA 8 190mA 12 240mA VOUT –5V 140mA AT VIN = 5V * SUMIDA CDRH74-101 ** AVX TPSC106M025 † AVX TPSC336M010 †† RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V IN Figure 14. Positive-to-Negative (– 5V) Converter VIN 8V TO 18V 10pF + 4.7µF** 35V 0.1µF 7 6 3 CHARGER ON/OFF 100k 8 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSD476M016 SENSE LBI RUN VIN VFB LTC1474 LBO GND SW 1 2 1M L* 100µH + 47µF† 16V VOUT 4-NiCd 200mA 5 4 D1 MBR0530 1474/75 F15 Figure 15. 4-NiCd Battery Charger 16 MBR0530 4.69M 1474/75 F14 LTC1474/LTC1475 U TYPICAL APPLICATIONS VIN 4V TO 18V + 4.7µF† 35V 0.1µF 2.2M 7 6 3 1M 100k 8 RUN VIN SENSE VOUT LTC1474-3.3 LBI LBO RUN SW GND 1 2 L* 100µH + 100µF†† 6.3V VOUT 3.3V 250mA 5 D1 MBR0530 4 * SUMIDA CDRH73-101 † AVX TPSC475M035 †† AVX TPSC107M006 1474/75 F16 Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout VIN 5.7V TO 18V + 4.7µF** 35V 0.1µF 7 3.65M 6 3 100k 8 OFF 1M VIN SENSE VOUT LTC1475-5 LBO LBI/OFF ON SW GND ON 1 + 2 L* 100µH 33µF† 10V VOUT 5V 250mA 5 D1 MBR0530 4 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC336M010 1474/75 F17 Figure 17. Pushbutton On/Off 5V/250mA Regulator VCC VIN 4V TO 18V + MMBD914LT1 0.1µF 0.1µF 4.7µF** 35V 100k ON/OFF 2.2M 8 2 3 1M µC ON 7 VIN 6 SENSE VOUT 1 + LTC1475-3.3 L* 100µH LBO LBI/OFF GND SW VOUT 3.3V 250mA 100µF† 6.3V 5 4 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC107M006 D1 MBR0530 1474/75 F18 Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off 17 LTC1474/LTC1475 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 ± 0.004* (3.00 ± 0.102) 8 7 6 5 0.118 ± 0.004** (3.00 ± 0.102) 0.192 ± 0.004 (4.88 ± 0.10) 1 0.040 ± 0.006 (1.02 ± 0.15) 0.007 (0.18) 2 3 4 0.034 ± 0.004 (0.86 ± 0.102) 0° – 6° TYP 0.021 ± 0.006 (0.53 ± 0.015) SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) TYP 0.006 ± 0.004 (0.15 ± 0.102) * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE 18 MSOP (MS8) 1197 LTC1474/LTC1475 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. SO8 0996 19 LTC1474/LTC1475 U TYPICAL APPLICATION 10pF VIN 3.5V to 18V + 0.1µF 4.7µF** 35V 7 1.8M 6 1M 100k MMBT2N2222LT1 8 3 VIN SENSE ON LTC1475 LBI/OFF 1M ON OFF VFB LBO GND SW 1 2 1.02M 1% 1M 1% + L* 100µH VOUT 2.5V 250mA 100µF† 6.3V 5 4 D1 MBR0530 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC107M006 1474/75 F19 Figure 19. Pushbutton On/Off with Low Battery Lockout RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1096/LTC1098 Micropower Sampling 8-Bit Serial I/O A/D Converter IQ = 80µA Max LT1121/LT1121-3.3/LT1121-5 150mA Low Dropout Regulator Linear Regulator, IQ = 30µA LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Selectable IPEAK = 300mA or 600mA LTC1265 1.2A High Efficiency Step-Down DC/DC Converter Burst Mode Operation, Internal MOSFET LT1375/LT1376 1.5A 500kHz Step-Down Switching Regulators 500kHz, Small Inductor, High Efficiency Switchers, 1.5A Switch LTC1440/LTC1441/LTC1442 Ultralow Power Comparator with Reference IQ = 2.8µA Max LT1495/LT1496 1.5µA Precision Rail-to-Rail Op Amps IQ = 1.5µA Max LT1521/LT1521-3/LT1521-3.3/ LT1521-5 300mA Low Dropout Regulator Linear Regulator, IQ = 12µA LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode LT1634-1.25 20 Micropower Precision Shunt Reference Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com IQ(MIN) = 10µA 14745fa LT/TP 0398 4K REV A • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1997