LINER LTR3800 Wide operating range, no rsense step-down controller Datasheet

LTC1778/LTC1778-1
Wide Operating Range,
No RSENSETM Step-Down Controller
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FEATURES
DESCRIPTIO
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The LTC®1778 is a synchronous step-down switching
regulator controller optimized for CPU power. The controller uses a valley current control architecture to deliver
very low duty cycles with excellent transient response
without requiring a sense resistor. Operating frequency is
selected by an external resistor and is compensated for
variations in VIN.
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No Sense Resistor Required
True Current Mode Control
Optimized for High Step-Down Ratios
tON(MIN) ≤ 100ns
Extremely Fast Transient Response
Stable with Ceramic COUT
Dual N-Channel MOSFET Synchronous Drive
Power Good Output Voltage Monitor (LTC1778)
Adjustable On-Time (LTC1778-1)
Wide VIN Range: 4V to 36V
±1% 0.8V Voltage Reference
Adjustable Current Limit
Adjustable Switching Frequency
Programmable Soft-Start
Output Overvoltage Protection
Optional Short-Circuit Shutdown Timer
Micropower Shutdown: IQ < 30µA
Available in a 16-Pin Narrow SSOP Package
Discontinuous mode operation provides high efficiency
operation at light loads. A forced continuous control pin
reduces noise and RF interference, and can assist secondary winding regulation by disabling discontinuous operation when the main output is lightly loaded.
Fault protection is provided by internal foldback current
limiting, an output overvoltage comparator and optional
short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing
capacitor. The regulator current limit level is user programmable. Wide supply range allows operation from 4V to 36V
at the input and from 0.8V up to (0.9)VIN at the output.
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APPLICATIO S
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
No RSENSE is a trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066
Notebook and Palmtop Computers
Distributed Power Systems
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TYPICAL APPLICATIO
RON
1.4MΩ
Efficiency vs Load Current
ION
VIN
RUN/SS
M1
Si4884
TG
CC
500pF
SW
ITH
RC
20k
INTVCC
BG
PGOOD
PGND
+
DB
CMDSH-3
LTC1778
SGND
L1
1.8µH
CB 0.22µF
BOOST
+
M2
Si4874
CVCC
4.7µF
CIN
10µF
50V
×3
D1
B340A
COUT
180µF
4V
×2
VIN
5V TO 28V
VOUT = 2.5V
VIN = 5V
90
VOUT
2.5V
10A
EFFICIENCY (%)
CSS
0.1µF
100
VIN = 25V
80
70
R2
30.1k
VFB
R1
14k
60
0.01
1
0.1
LOAD CURRENT (A)
10
1778 F01b
1778 F01a
Figure 1. High Efficiency Step-Down Converter
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LTC1778/LTC1778-1
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ABSOLUTE
RATI GS
(Note 1)
Input Supply Voltage (VIN, ION)................. 36V to – 0.3V
Boosted Topside Driver Supply Voltage
(BOOST) ................................................... 42V to – 0.3V
SW Voltage .................................................. 36V to – 5V
EXTVCC, (BOOST – SW), RUN/SS,
PGOOD Voltages ....................................... 7V to – 0.3V
FCB, VON, VRNG Voltages .......... INTVCC + 0.3V to – 0.3V
ITH, VFB Voltages...................................... 2.7V to – 0.3V
TG, BG, INTVCC, EXTVCC Peak Currents .................... 2A
TG, BG, INTVCC, EXTVCC RMS Currents .............. 50mA
Operating Ambient Temperature Range (Note 4)
LTC1778E ........................................... – 40°C to 85°C
LTC1778I .......................................... – 40°C to 125°C
Junction Temperature (Note 2) ............................ 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
RUN/SS 1
16 BOOST
PGOOD 2
15 TG
VRNG 3
14 SW
FCB 4
LTC1778EGN
LTC1778IGN
13 PGND
ITH 5
11 INTVCC
ION 7
10 VIN
VFB 8
9
15 TG
VRNG 3
14 SW
FCB 4
SGND 6
GN PART MARKING
EXTVCC
GN PACKAGE
16-LEAD PLASTIC SSOP
1778
1778I
TJMAX = 125°C, θJA = 130°C/ W
16 BOOST
VON 2
ITH 5
12 BG
SGND 6
TOP VIEW
RUN/SS 1
ORDER PART
NUMBER
LTC1778EGN-1
13 PGND
12 BG
11 INTVCC
ION 7
10 VIN
VFB 8
9
GN PART MARKING
EXTVCC
17781
GN PACKAGE
16-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 130°C/ W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
900
15
2000
30
µA
µA
0.800
0.800
0.808
0.812
V
V
Main Control Loop
IQ
Input DC Supply Current
Normal
Shutdown Supply Current
VFB
Feedback Reference Voltage
ITH = 1.2V (Note 3) LTC1778E
ITH = 1.2V (Note 3) LTC1778I
∆VFB(LINEREG)
Feedback Voltage Line Regulation
VIN = 4V to 30V, ITH = 1.2V (Note 3)
∆VFB(LOADREG)
Feedback Voltage Load Regulation
ITH = 0.5V to 1.9V (Note 3)
IFB
Feedback Input Current
VFB = 0.8V
gm(EA)
Error Amplifier Transconductance
ITH = 1.2V (Note 3)
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.8V
tON
On-Time
ION = 30µA, VON = 0V (LTC1778-1)
ION = 15µA, VON = 0V (LTC1778-1)
tON(MIN)
Minimum On-Time
ION = 180µA
●
●
0.792
0.792
0.002
●
%/V
– 0.05
– 0.3
%
–5
±50
nA
●
1.4
1.7
2
mS
●
0.76
0.8
0.84
V
–1
–2
µA
198
396
233
466
268
536
ns
ns
50
100
ns
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LTC1778/LTC1778-1
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
tOFF(MIN)
Minimum Off-Time
ION = 30µA
VSENSE(MAX)
Maximum Current Sense Threshold
VPGND – VSW
VRNG = 1V, VFB = 0.76V
VRNG = 0V, VFB = 0.76V
VRNG = INTVCC, VFB = 0.76V
VSENSE(MIN)
Minimum Current Sense Threshold
VPGND – VSW
VRNG = 1V, VFB = 0.84V
VRNG = 0V, VFB = 0.84V
VRNG = INTVCC, VFB = 0.84V
∆VFB(OV)
Output Overvoltage Fault Threshold
VFB(UV)
Output Undervoltage Fault Threshold
VRUN/SS(ON)
RUN Pin Start Threshold
VRUN/SS(LE)
RUN Pin Latchoff Enable Threshold
VRUN/SS(LT)
RUN Pin Latchoff Threshold
RUN/SS Pin Falling
IRUN/SS(C)
Soft-Start Charge Current
VRUN/SS = 0V
– 0.5
IRUN/SS(D)
Soft-Start Discharge Current
VRUN/SS = 4.5V, VFB = 0V
0.8
VIN(UVLO)
Undervoltage Lockout
VIN Falling
VIN(UVLOR)
Undervoltage Lockout Release
VIN Rising
TG RUP
TG Driver Pull-Up On Resistance
TG RDOWN
BG RUP
●
●
●
113
79
158
TYP
MAX
UNITS
250
400
ns
133
93
186
153
107
214
mV
mV
mV
– 67
– 47
– 93
mV
mV
mV
5.5
7.5
9.5
%
520
600
680
mV
0.8
1.5
2
V
4
4.5
V
3.5
4.2
V
– 1.2
–3
µA
1.8
3
µA
●
3.4
3.9
V
●
3.5
4
V
TG High
2
3
Ω
TG Driver Pull-Down On Resistance
TG Low
2
3
Ω
BG Driver Pull-Up On Resistance
BG High
3
4
Ω
BG RDOWN
BG Driver Pull-Down On Resistance
BG Low
1
2
Ω
TG tr
TG Rise Time
CLOAD = 3300pF
20
ns
TG tf
TG Fall Time
CLOAD = 3300pF
20
ns
BG tr
BG Rise Time
CLOAD = 3300pF
20
ns
BG tf
BG Fall Time
CLOAD = 3300pF
20
ns
●
RUN/SS Pin Rising
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
∆VLDO(LOADREG)
Internal VCC Load Regulation
ICC = 0mA to 20mA, VEXTVCC = 4V
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, VEXTVCC Rising
∆VEXTVCC
EXTVCC Switch Drop Voltage
ICC = 20mA, VEXTVCC = 5V
∆VEXTVCC(HYS)
EXTVCC Switchover Hysteresis
●
4.7
●
4.5
5
5.3
V
– 0.1
±2
%
4.7
150
V
300
200
mV
mV
PGOOD Output (LTC1778 Only)
∆VFBH
PGOOD Upper Threshold
VFB Rising
5.5
7.5
9.5
∆VFBL
∆VFB(HYS)
VPGL
PGOOD Lower Threshold
VFB Falling
– 5.5
– 7.5
– 9.5
%
PGOOD Hysteresis
VFB Returning
1
2
%
PGOOD Low Voltage
IPGOOD = 5mA
0.15
0.4
V
Note 1: Absolute Maximum Ratings are those values beyond which the life of
a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD as follows:
LTC1778E: TJ = TA + (PD • 130°C/W)
Note 3: The LTC1778 is tested in a feedback loop that adjusts VFB to achieve
a specified error amplifier output voltage (ITH).
%
Note 4: The LTC1778E is guaranteed to meet performance specifications from
0°C to 70°C. Specifications over the –40°C to 85°C operating temperature
range are assured by design, characterization and correlation with statistical
process controls. The LTC1778I is guaranteed over the full – 40°C to 125°C
operating temperature range.
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LTC1778/LTC1778-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Transient Response
(Discontinuous Mode)
Transient Response
VOUT
50mV/DIV
Start-Up
RUN/SS
2V/DIV
VOUT
50mV/DIV
VOUT
1V/DIV
IL
5A/DIV
IL
5A/DIV
20µs/DIV
LOAD STEP 0A TO 10A
VIN = 15V
VOUT = 2.5V
FCB = 0V
FIGURE 9 CIRCUIT
20µs/DIV
LOAD STEP 1A TO 10A
VIN = 15V
VOUT = 2.5V
FCB = INTVCC
FIGURE 9 CIRCUIT
1778 G01
Efficiency vs Load Current
100
DISCONTINUOUS
MODE
CONTINUOUS
MODE
70
50
0.001
FCB = 5V
FIGURE 9 CIRCUIT
0.1
0.01
1
LOAD CURRENT (A)
280
ILOAD = 1A
90
ILOAD = 10A
80
0
5
25
10
15
20
INPUT VOLTAGE (V)
IOUT = 0A
240
200
30
5
Frequency vs Load Current
0
250
ITH VOLTAGE (V)
200
∆VOUT (%)
FIGURE 9 CIRCUIT
2.0
–0.1
–0.2
100
1.5
CONTINUOUS
MODE
1.0
–0.3
DISCONTINUOUS
MODE
0.5
50
2
4
6
LOAD CURRENT (A)
8
10
1778 G26
–0.4
25
ITH Voltage vs Load Current
2.5
FIGURE 9 CIRCUIT
CONTINUOUS MODE
0
20
1778 G05
Load Regulation
0
15
10
1778 G04
300
FREQUENCY (kHz)
260
INPUT VOLTAGE (V)
1778 G03
150
IOUT = 10A
220
10
DISCONTINUOUS
MODE
FCB = 0V
FIGURE 9 CIRCUIT
85
VIN = 10V
VOUT = 2.5V
EXTVCC = 5V
FIGURE 9 CIRCUIT
60
Frequency vs Input Voltage
300
FREQUENCY (kHz)
80
1778 G19
VIN = 15V
VOUT = 2.5V
RLOAD = 0.25Ω
95
EFFICIENCY (%)
90
50ms/DIV
1778 G02
Efficiency vs Input Voltage
100
EFFICIENCY (%)
IL
5A/DIV
0
2
6
4
LOAD CURRENT (A)
8
10
1778 G06
0
0
10
5
LOAD CURRENT (A)
15
1778 G07
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LTC1778/LTC1778-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Current Sense Threshold
vs ITH Voltage
VRNG =
200
On-Time vs ION Current
On-Time vs VON Voltage
10k
2V
VVON = 0V
1k
ON-TIME (ns)
0
100
400
200
10
0
1.0
1.5
2.0
ITH VOLTAGE (V)
0.5
2.5
10
ION CURRENT (µA)
1
3.0
200
150
100
50
50
25
75
0
TEMPERATURE (°C)
Maximum Current Sense
Threshold vs VRNG Voltage
100
150
MAXIMUM CURRENT SENSE THRESHOLD (mV)
MAXIMUM CURRENT SENSE THRESHOLD (mV)
250
VRNG = 1V
125
100
75
50
25
0
0
125
0.2
0.4
VFB (V)
0.6
Maximum Current Sense
Threshold vs RUN/SS Voltage
100
75
50
25
0
2.5
3
RUN/SS VOLTAGE (V)
200
150
100
50
0
3.5
1778 G23
0.5
0.75
1.0
1.25
1.5
VRNG VOLTAGE (V)
1.75
150
Feedback Reference Voltage
vs Temperature
0.82
VRNG = 1V
140
130
120
110
100
–50 –25
2.0
1778 G10
FEEDBACK REFERENCE VOLTAGE (V)
MAXIMUM CURRENT SENSE THRESHOLD (mV)
125
2
250
Maximum Current Sense
Threshold vs Temperature
VRNG = 1V
1.5
0.8
300
1778 G09
1778 G22
150
3
1778 G21
Current Limit Foldback
IION = 30µA
VVON = 0V
2
1
VON VOLTAGE (V)
0
1778 G20
On-Time vs Temperature
0
–50 –25
0
100
1778 G08
ON-TIME (ns)
600
–100
300
IION = 30µA
800
0.7V
0.5V
100
–200
MAXIMUM CURRENT SENSE THRESHOLD (mV)
1000
1.4V
1V
ON-TIME (ns)
CURRENT SENSE THRESHOLD (mV)
300
50
25
0
75
TEMPERATURE (°C)
100
125
1778 G11
0.81
0.80
0.79
0.78
–50 –25
75
0
25
50
TEMPERATURE (°C)
100
125
1778 G12
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LTC1778/LTC1778-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Input and Shutdown Currents
vs Input Voltage
2.0
1200
60
EXTVCC OPEN
INPUT CURRENT (µA)
1.6
1.4
1.2
50
800
40
SHUTDOWN
600
30
400
20
200
SHUTDOWN CURRENT (µA)
1000
1.8
gm (mS)
INTVCC Load Regulation
0
–0.1
∆INTVCC (%)
Error Amplifier gm vs Temperature
–0.2
–0.3
–0.4
10
EXTVCC = 5V
1.0
–50 –25
0
50
25
0
75
TEMPERATURE (°C)
100
125
20
15
25
10
INPUT VOLTAGE (V)
5
30
1778 G13
35
8
–0.25
2
0
–50
FCB PIN CURRENT (µA)
3
FCB PIN CURRENT (µA)
0
2
–0.50
–0.75
–1.00
50
25
0
75
TEMPERATURE (°C)
100
125
–1.50
–50 –25
50
25
75
0
TEMPERATURE (°C)
1778 G14
UNDERVOLTAGE LOCKOUT THRESHOLD (V)
RUN/SS THRESHOLD (V)
4.5
LATCHOFF ENABLE
4.0
3.5
LATCHOFF THRESHOLD
75
0
25
50
TEMPERATURE (°C)
PULL-UP CURRENT
100
125
–2
–50 –25
50
25
0
75
TEMPERATURE (°C)
100
125
1778 G16
Undervoltage Lockout Threshold
vs Temperature
5.0
–25
0
1778 G15
RUN/SS Latchoff Thresholds
vs Temperature
3.0
–50
PULL-DOWN CURRENT
1
–1
–1.25
–25
50
RUN/SS Pin Current
vs Temperature
10
4
10
30
40
20
INTVCC LOAD CURRENT (mA)
1778 G25
FCB Pin Current vs Temperature
6
0
1778 G24
EXTVCC Switch Resistance
vs Temperature
EXTVCC SWITCH RESISTANCE (Ω)
–0.5
0
0
100
125
1778 G17
4.0
3.5
3.0
2.5
2.0
–50 –25
75
0
25
50
TEMPERATURE (C)
100
125
1778 G18
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LTC1778/LTC1778-1
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PI FU CTIO S
RUN/SS (Pin 1): Run Control and Soft-Start Input. A
capacitor to ground at this pin sets the ramp time to full
output current (approximately 3s/µF) and the time delay
for overcurrent latchoff (see Applications Information).
Forcing this pin below 0.8V shuts down the device.
PGOOD (Pin 2, LTC1778): Power Good Output. Open
drain logic output that is pulled to ground when the output
voltage is not within ±7.5% of the regulation point.
ION (Pin 7): On-Time Current Input. Tie a resistor from VIN
to this pin to set the one-shot timer current and thereby set
the switching frequency.
VFB (Pin 8): Error Amplifier Feedback Input. This pin
connects the error amplifier input to an external resistive
divider from VOUT.
EXTVCC (Pin 9): External VCC Input. When EXTVCC exceeds 4.7V, an internal switch connects this pin to INTVCC
and shuts down the internal regulator so that controller
and gate drive power is drawn from EXTVCC. Do not exceed
7V at this pin and ensure that EXTVCC < VIN.
VON (Pin 2, LTC1778-1): On-Time Voltage Input. Voltage
trip point for the on-time comparator. Tying this pin to the
output voltage or an external resistive divider from the
output makes the on-time proportional to VOUT. The
comparator input defaults to 0.7V when the pin is grounded
or unavailable (LTC1778) and defaults to 2.4V when the
pin is tied to INTVCC. Tie this pin to INTVCC in high VOUT
applications to use a lower RON value.
VIN (Pin 10): Main Input Supply. Decouple this pin to
PGND with an RC filter (1Ω, 0.1µF).
VRNG (Pin 3): Sense Voltage Range Input. The voltage at
this pin is ten times the nominal sense voltage at maximum output current and can be set from 0.5V to 2V by a
resistive divider from INTVCC. The nominal sense voltage
defaults to 70mV when this pin is tied to ground, 140mV
when tied to INTVCC.
BG (Pin 12): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
FCB (Pin 4): Forced Continuous Input. Tie this pin to
ground to force continuous synchronous operation at low
load, to INTVCC to enable discontinuous mode operation
at low load or to a resistive divider from a secondary output
when using a secondary winding.
ITH (Pin 5): Current Control Threshold and Error Amplifier
Compensation Point. The current comparator threshold
increases with this control voltage. The voltage ranges
from 0V to 2.4V with 0.8V corresponding to zero sense
voltage (zero current).
SGND (Pin 6): Signal Ground. All small-signal components and compensation components should connect to
this ground, which in turn connects to PGND at one point.
INTVCC (Pin 11): Internal 5V Regulator Output. The driver
and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7µF
low ESR tantalum capacitor.
PGND (Pin 13): Power Ground. Connect this pin closely to
the source of the bottom N-channel MOSFET, the (–)
terminal of CVCC and the (–) terminal of CIN.
SW (Pin 14): Switch Node. The (–) terminal of the bootstrap capacitor CB connects here. This pin swings from a
diode voltage drop below ground up to VIN.
TG (Pin 15): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage swing equal to INTVCC superimposed on the switch node voltage SW.
BOOST (Pin 16): Boosted Floating Driver Supply. The (+)
terminal of the bootstrap capacitor CB connects here. This
pin swings from a diode voltage drop below INTVCC up to
VIN + INTVCC.
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LTC1778/LTC1778-1
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FU CTIO AL DIAGRA
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RON
VIN
2 VON**
7 ION
4 FCB
10 VIN
9 EXTVCC
+
4.7V
0.7V
CIN
2.4V
+
1µA
–
0.8V
REF
1
0.8V
5V
REG
+
–
F
tON =
16
VVON
(10pF)
IION
R
S
TG
Q
FCNT
14
L1
SWITCH
LOGIC
IREV
VOUT
DB
–
–
M1
SW
+
ICMP
CB
15
ON
20k
+
BOOST
INTVCC
11
SHDN
1.4V
BG
OV
12
+
COUT
CVCC
M2
VRNG
PGND
3
×
13
PGOOD*
0.7V
2
3.3µA
R2
1
240k
+
1V
Q2 Q4
–
Q6
ITHB
0.74V
UV
VFB
8
Q3 Q1
R1
+
Q5
SGND
6
OV
+
–
–
0.8V
–
SS
+
RUN
SHDN
1.2µA
EA
+
–
–
×4
6V
+
0.6V
0.8V
*LTC1778
**LTC1778-1
0.86V
5 ITH
RC
CC1
0.6V
1 RUN/SS CSS
1778 FD
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LTC1778/LTC1778-1
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OPERATIO
Main Control Loop
The LTC1778 is a current mode controller for DC/DC
step-down converters. In normal operation, the top
MOSFET is turned on for a fixed interval determined by a
one-shot timer OST. When the top MOSFET is turned off,
the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by
sensing the voltage between the PGND and SW pins using
the bottom MOSFET on-resistance . The voltage on the ITH
pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this
voltage by comparing the feedback signal VFB from the
output voltage with an internal 0.8V reference. If the load
current increases, it causes a drop in the feedback voltage
relative to the reference. The ITH voltage then rises until the
average inductor current again matches the load current.
At low load currents, the inductor current can drop to zero
and become negative. This is detected by current reversal
comparator IREV which then shuts off M2, resulting in
discontinuous operation. Both switches will remain off
with the output capacitor supplying the load current until
the ITH voltage rises above the zero current level (0.8V) to
initiate another cycle. Discontinuous mode operation is
disabled by comparator F when the FCB pin is brought
below 0.8V, forcing continuous synchronous operation.
The operating frequency is determined implicitly by the
top MOSFET on-time and the duty cycle required to
maintain regulation. The one-shot timer generates an ontime that is proportional to the ideal duty cycle, thus
holding frequency approximately constant with changes
in VIN. The nominal frequency can be adjusted with an
external resistor RON.
Overvoltage and undervoltage comparators OV and UV
pull the PGOOD output low if the output feedback voltage
exits a ±7.5% window around the regulation point.
Furthermore, in an overvoltage condition, M1 is turned off
and M2 is turned on and held on until the overvoltage
condition clears.
Foldback current limiting is provided if the output is
shorted to ground. As VFB drops, the buffered current
threshold voltage ITHB is pulled down by clamp Q3 to a 1V
level set by Q4 and Q6. This reduces the inductor valley
current level to one sixth of its maximum value as VFB
approaches 0V.
Pulling the RUN/SS pin low forces the controller into its
shutdown state, turning off both M1 and M2. Releasing
the pin allows an internal 1.2µA current source to charge
up an external soft-start capacitor CSS. When this voltage
reaches 1.5V, the controller turns on and begins switching, but with the ITH voltage clamped at approximately
0.6V below the RUN/SS voltage. As CSS continues to
charge, the soft-start current limit is removed.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the internal controller circuitry is derived from the
INTVCC pin. The top MOSFET driver is powered from a
floating bootstrap capacitor CB. This capacitor is recharged from INTVCC through an external Schottky diode
DB when the top MOSFET is turned off. When the EXTVCC
pin is grounded, an internal 5V low dropout regulator
supplies the INTVCC power from VIN. If EXTVCC rises
above 4.7V, the internal regulator is turned off, and an
internal switch connects EXTVCC to INTVCC. This allows
a high efficiency source connected to EXTVCC, such as an
external 5V supply or a secondary output from the
converter, to provide the INTVCC power. Voltages up to
7V can be applied to EXTVCC for additional gate drive. If
the input voltage is low and INTVCC drops below 3.5V,
undervoltage lockout circuitry prevents the power
switches from turning on.
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The basic LTC1778 application circuit is shown in
Figure 1. External component selection is primarily determined by the maximum load current and begins with
the selection of the sense resistance and power MOSFET
switches. The LTC1778 uses the on-resistance of the
synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and
operating frequency largely determines the inductor value.
Finally, CIN is selected for its ability to handle the large
RMS current into the converter and COUT is chosen with
low enough ESR to meet the output voltage ripple and
transient specification.
resulting in nominal sense voltages of 50mV to 200mV.
Additionally, the VRNG pin can be tied to SGND or INTVCC
in which case the nominal sense voltage defaults to 70mV
or 140mV, respectively. The maximum allowed sense
voltage is about 1.33 times this nominal value.
Choosing the LTC1778 or LTC1778-1
The gate drive voltage is set by the 5V INTVCC supply.
Consequently, logic-level threshold MOSFETs must be
used in LTC1778 applications. If the input voltage is
expected to drop below 5V, then sub-logic level threshold
MOSFETs should be considered.
Maximum Sense Voltage and VRNG Pin
Inductor current is determined by measuring the voltage
across a sense resistance that appears between the PGND
and SW pins. The maximum sense voltage is set by the
voltage applied to the VRNG pin and is equal to approximately (0.133)VRNG. The current mode control loop will
not allow the inductor current valleys to exceed
(0.133)VRNG/RSENSE. In practice, one should allow some
margin for variations in the LTC1778 and external component values and a good guide for selecting the sense
resistance is:
RSENSE =
VRNG
10 • IOUT(MAX)
An external resistive divider from INTVCC can be used to
set the voltage of the VRNG pin between 0.5V and 2V
The LTC1778 requires two external N-channel power
MOSFETs, one for the top (main) switch and one for the
bottom (synchronous) switch. Important parameters for
the power MOSFETs are the breakdown voltage V(BR)DSS,
threshold voltage V(GS)TH, on-resistance RDS(ON), reverse
transfer capacitance CRSS and maximum current IDS(MAX).
When the bottom MOSFET is used as the current sense
element, particular attention must be paid to its onresistance. MOSFET on-resistance is typically specified
with a maximum value RDS(ON)(MAX) at 25°C. In this case,
additional margin is required to accommodate the rise in
MOSFET on-resistance with temperature:
RDS(ON)(MAX) =
RSENSE
ρT
The ρT term is a normalization factor (unity at 25°C)
accounting for the significant variation in on-resistance
2.0
ρT NORMALIZED ON-RESISTANCE
The LTC1778 has an open-drain PGOOD output that
indicates when the output voltage is within ±7.5% of the
regulation point. The LTC1778-1 trades the PGOOD pin for
a VON pin that allows the on-time to be adjusted. Tying the
VON pin high results in lower values for RON which is useful
in high VOUT applications. The VON pin also provides a
means to adjust the on-time to maintain constant frequency operation in applications where VOUT changes and
to correct minor frequency shifts with changes in load
current. Finally, the VON pin can be used to provide
additional current limiting in positive-to-negative converters and as a control input to synchronize the switching
frequency with a phase locked loop.
Power MOSFET Selection
1.5
1.0
0.5
0
– 50
50
100
0
JUNCTION TEMPERATURE (°C)
150
1778 F02
Figure 2. RDS(ON) vs. Temperature
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The power dissipated by the top and bottom MOSFETs
strongly depends upon their respective duty cycles and
the load current. When the LTC1778 is operating in
continuous mode, the duty cycles for the MOSFETs are:
VOUT
VIN
V –V
= IN OUT
VIN
DTOP =
DBOT
The resulting power dissipation in the MOSFETs at maximum output current are:
PTOP = DTOP IOUT(MAX)2 ρT(TOP) RDS(ON)(MAX)
+ k VIN2 IOUT(MAX) CRSS f
PBOT = DBOT IOUT(MAX)2 ρT(BOT) RDS(ON)(MAX)
Both MOSFETs have I2R losses and the top MOSFET
includes an additional term for transition losses, which are
largest at high input voltages. The constant k = 1.7A–1 can
be used to estimate the amount of transition loss. The
bottom MOSFET losses are greatest when the bottom duty
cycle is near 100%, during a short-circuit or at high input
voltage.
Tying a resistor RON from VIN to the ION pin yields an ontime inversely proportional to VIN. For a step-down converter, this results in approximately constant frequency
operation as the input supply varies:
f=
VOUT
[ HZ ]
VVON RON(10pF)
To hold frequency constant during output voltage changes,
tie the VON pin to VOUT or to a resistive divider from VOUT
when VOUT > 2.4V. The VON pin has internal clamps that
limit its input to the one-shot timer. If the pin is tied below
0.7V, the input to the one-shot is clamped at 0.7V. Similarly, if the pin is tied above 2.4V, the input is clamped at
2.4V. In high VOUT applications, tying VON to INTVCC so
that the comparator input is 2.4V results in a lower value
for RON. Figures 3a and 3b show how RON relates to
switching frequency for several common output voltages.
1000
SWITCHING FREQUENCY (kHz)
with temperature, typically about 0.4%/°C as shown in
Figure 2. For a maximum junction temperature of 100°C,
using a value ρT = 1.3 is reasonable.
VOUT = 3.3V
VOUT = 1.5V
100
100
1000
RON (kΩ)
Operating Frequency
The operating frequency of LTC1778 applications is determined implicitly by the one-shot timer that controls the
on-time tON of the top MOSFET switch. The on-time is set
by the current into the ION pin and the voltage at the VON
pin (LTC1778-1) according to:
V
tON = VON (10pF )
IION
VON defaults to 0.7V in the LTC1778.
10000
1778 F03a
Figure 3a. Switching Frequency vs RON
for the LTC1778 and LTC1778-1 (VON = 0V)
1000
SWITCHING FREQUENCY (kHz)
The choice of operating frequency is a tradeoff between
efficiency and component size. Low frequency operation
improves efficiency by reducing MOSFET switching losses
but requires larger inductance and/or capacitance in order
to maintain low output ripple voltage.
VOUT = 2.5V
VOUT = 12V
VOUT = 5V
VOUT = 3.3V
100
100
1000
RON (kΩ)
10000
1778 F03b
Figure 3b. Switching Frequency vs RON
for the LTC1778-1 (VON = INTVCC)
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Because the voltage at the ION pin is about 0.7V, the
current into this pin is not exactly inversely proportional to
VIN, especially in applications with lower input voltages.
To correct for this error, an additional resistor RON2
connected from the ION pin to the 5V INTVCC supply will
further stabilize the frequency.
VIN(MIN) = VOUT
tON + tOFF(MIN)
tON
A plot of maximum duty cycle vs frequency is shown in
Figure 5.
5V
RON
0.7 V
Changes in the load current magnitude will also cause
frequency shift. Parasitic resistance in the MOSFET
switches and inductor reduce the effective voltage across
the inductance, resulting in increased duty cycle as the
load current increases. By lengthening the on-time slightly
as current increases, constant frequency operation can be
maintained. This is accomplished with a resistive divider
from the ITH pin to the VON pin and VOUT. The values
required will depend on the parasitic resistances in the
specific application. A good starting point is to feed about
25% of the voltage change at the ITH pin to the VON pin as
shown in Figure 4a. Place capacitance on the VON pin to
filter out the ITH variations at the switching frequency. The
resistor load on ITH reduces the DC gain of the error amp
and degrades load regulation, which can be avoided by
using the PNP emitter follower of Figure 4b.
Minimum Off-time and Dropout Operation
The minimum off-time tOFF(MIN) is the smallest amount of
time that the LTC1778 is capable of turning on the bottom
MOSFET, tripping the current comparator and turning the
MOSFET back off. This time is generally about 250ns. The
minimum off-time limit imposes a maximum duty cycle of
tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached,
Inductor Selection
Given the desired input and output voltages, the inductor
value and operating frequency determine the ripple
current:
⎛ V ⎞⎛ V ⎞
∆IL = ⎜ OUT ⎟ ⎜ 1 − OUT ⎟
VIN ⎠
⎝ f L ⎠⎝
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
2.0
SWITCHING FREQUENCY (MHz)
RON2 =
due to a dropping input voltage for example, then the
output will drop out of regulation. The minimum input
voltage to avoid dropout is:
1.5
DROPOUT
REGION
1.0
0.5
0
0
1.0
0.25
0.50
0.75
DUTY CYCLE (VOUT/VIN)
1778 F05
Figure 5. Maximum Switching Frequency vs Duty Cycle
RVON1
30k
RVON1
3k
VON
VOUT
CVON
0.01µF
RVON2
100k
LTC1778
RC
ITH
VOUT
10k
CVON
0.01µF
RVON2
10k
INTVCC
LTC1778
RC
Q1
2N5087
ITH
CC
CC
(4a)
VON
1778 F04
(4b)
Figure 4. Correcting Frequency Shift with Load Current Changes
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ripple. Highest efficiency operation is obtained at low
frequency with small ripple current. However, achieving
this requires a large inductor. There is a tradeoff between
component size, efficiency and operating frequency.
A reasonable starting point is to choose a ripple current
that is about 40% of IOUT(MAX). The largest ripple current
occurs at the highest VIN. To guarantee that ripple current
does not exceed a specified maximum, the inductance
should be chosen according to:
⎛ VOUT ⎞ ⎛
VOUT ⎞
L=⎜
1
−
⎟⎜
⎟
⎝ f ∆IL(MAX) ⎠ ⎝ VIN(MAX) ⎠
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron
cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. A variety of inductors designed
for high current, low voltage applications are available
from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko.
Schottky Diode D1 Selection
The Schottky diode D1 shown in Figure 1 conducts during
the dead time between the conduction of the power
MOSFET switches. It is intended to prevent the body diode
of the bottom MOSFET from turning on and storing charge
during the dead time, which can cause a modest (about
1%) efficiency loss. The diode can be rated for about one
half to one fifth of the full load current since it is on for only
a fraction of the duty cycle. In order for the diode to be
effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these
components be placed adjacently. The diode can be omitted if the efficiency loss is tolerable.
CIN and COUT Selection
The input capacitance CIN is required to filter the square
wave current at the drain of the top MOSFET. Use a low
ESR capacitor sized to handle the maximum RMS current.
IRMS ≅ IOUT(MAX)
VOUT
VIN
VIN
–1
VOUT
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT(MAX) / 2. This simple worst-case condition is
commonly used for design because even significant
deviations do not offer much relief. Note that ripple
current ratings from capacitor manufacturers are often
based on only 2000 hours of life which makes it advisable
to derate the capacitor.
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple and load step
transients. The output ripple ∆VOUT is approximately
bounded by:
⎛
1 ⎞
∆VOUT ≤ ∆IL ⎜ ESR +
⎟
8 fCOUT ⎠
⎝
Since ∆IL increases with input voltage, the output ripple is
highest at maximum input voltage. Typically, once the ESR
requirement is satisfied, the capacitance is adequate for
filtering and has the necessary RMS current rating.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Special polymer capacitors offer very low ESR
but have lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density
but it is important to only use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR, but
can be used in cost-sensitive applications providing that
consideration is given to ripple current ratings and long
term reliability. Ceramic capacitors have excellent low
ESR characteristics but can have a high voltage coefficient
and audible piezoelectric effects. The high Q of ceramic
capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be
taken to ensure that ringing from inrush currents and
switching does not pose an overvoltage hazard to the
power switches and controller. To dampen input voltage
transients, add a small 5µF to 50µF aluminum electrolytic
capacitor with an ESR in the range of 0.5Ω to 2Ω. High
performance through-hole capacitors may also be used,
Kool Mµ is a registered trademark of Magnetics, Inc.
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but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance.
Top MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected to the BOOST
pin supplies the gate drive voltage for the topside MOSFET.
This capacitor is charged through diode DB from INTVCC
when the switch node is low. When the top MOSFET turns
on, the switch node rises to VIN and the BOOST pin rises
to approximately VIN + INTVCC. The boost capacitor needs
to store about 100 times the gate charge required by the
top MOSFET. In most applications 0.1µF to 0.47µF, X5R or
X7R dielectric capacitor is adequate.
Discontinuous Mode Operation and FCB Pin
The FCB pin determines whether the bottom MOSFET
remains on when current reverses in the inductor. Tying
this pin above its 0.8V threshold enables discontinuous
operation where the bottom MOSFET turns off when
inductor current reverses. The load current at which
current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and
will vary with changes in VIN. Tying the FCB pin below the
0.8V threshold forces continuous synchronous operation,
allowing current to reverse at light loads and maintaining
high frequency operation.
In addition to providing a logic input to force continuous
operation, the FCB pin provides a means to maintain a
flyback winding output when the primary is operating in
discontinuous mode. The secondary output VOUT2 is normally set as shown in Figure 6 by the turns ratio N of the
+
VIN
CIN
VIN
1N4148
OPTIONAL
EXTVCC
CONNECTION
5V < VOUT2 < 7V
TG
•
+
LTC1778
EXTVCC
SW
R4
T1
1:N
FCB
R3
• +
VOUT2
COUT2
1µF
VOUT1
⎛ R4 ⎞
VOUT 2(MIN) = 0.8V⎜ 1 + ⎟
⎝ R3 ⎠
Fault Conditions: Current Limit and Foldback
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage. In
the LTC1778, the maximum sense voltage is controlled by
the voltage on the VRNG pin. With valley current control,
the maximum sense voltage and the sense resistance
determine the maximum allowed inductor valley current.
The corresponding output current limit is:
ILIMIT =
VSNS(MAX)
RDS(ON)
1
+ ∆IL
ρT 2
The current limit value should be checked to ensure that
ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit
generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power
loss in the converter. Note that it is important to check for
self-consistency between the assumed MOSFET junction
temperature and the resulting value of ILIMIT which heats
the MOSFET switches.
Caution should be used when setting the current limit
based upon the RDS(ON) of the MOSFETs. The maximum
current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and
maximum values for RDS(ON), but not a minimum. A
reasonable assumption is that the minimum RDS(ON) lies
the same amount below the typical value as the maximum
lies above it. Consult the MOSFET manufacturer for further
guidelines.
COUT
BG
SGND
transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary
load current, then VOUT2 will droop. An external resistor
divider from VOUT2 to the FCB pin sets a minimum voltage
VOUT2(MIN) below which continuous operation is forced
until VOUT2 has risen above its minimum.
PGND
1778 F06
Figure 6. Secondary Output Loop and EXTVCC Connection
To further limit current in the event of a short circuit to
ground, the LTC1778 includes foldback current limiting. If
the output falls by more than 25%, then the maximum
sense voltage is progressively lowered to about one sixth
of its full value.
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INTVCC Regulator
An internal P-channel low dropout regulator produces the
5V supply that powers the drivers and internal circuitry
within the LTC1778. The INTVCC pin can supply up to
50mA RMS and must be bypassed to ground with a
minimum of 4.7µF low ESR tantalum capacitor. Good
bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. Applications
using large MOSFETs with a high input voltage and high
frequency of operation may cause the LTC1778 to exceed
its maximum junction temperature rating or RMS current
rating. Most of the supply current drives the MOSFET
gates unless an external EXTVCC source is used. In continuous mode operation, this current is IGATECHG = f(Qg(TOP)
+ Qg(BOT)). The junction temperature can be estimated
from the equations given in Note 2 of the Electrical
Characteristics. For example, the LTC1778CGN is limited
to less than 14mA from a 30V supply:
will start-up using the internal linear regulator until the
boosted output supply is available.
External Gate Drive Buffers
The LTC1778 drivers are adequate for driving up to about
30nC into MOSFET switches with RMS currents of 50mA.
Applications with larger MOSFET switches or operating at
frequencies requiring greater RMS currents will benefit
from using external gate drive buffers such as the LTC1693.
Alternately, the external buffer circuit shown in Figure 7
can be used. Note that the bipolar devices reduce the
signal swing at the MOSFET gate, and benefit from an
increased EXTVCC voltage of about 6V.
10Ω
TG
TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C
For larger currents, consider using an external supply with
the EXTVCC pin.
INTVCC
BOOST
Q1
FMMT619
GATE
OF M1
Q2
FMMT720
10Ω
BG
Q3
FMMT619
GATE
OF M2
Q4
FMMT720
PGND
SW
1778 F07
Figure 7. Optional External Gate Driver
EXTVCC Connection
The EXTVCC pin can be used to provide MOSFET gate drive
and control power from the output or another external
source during normal operation. Whenever the EXTVCC
pin is above 4.7V the internal 5V regulator is shut off and
an internal 50mA P-channel switch connects the EXTVCC
pin to INTVCC. INTVCC power is supplied from EXTVCC until
this pin drops below 4.5V. Do not apply more than 7V to
the EXTVCC pin and ensure that EXTVCC ≤ VIN. The following list summarizes the possible connections for EXTVCC:
1. EXTVCC grounded. INTVCC is always powered from the
internal 5V regulator.
2. EXTVCC connected to an external supply. A high efficiency supply compatible with the MOSFET gate drive
requirements (typically 5V) can improve overall
efficiency.
3. EXTVCC connected to an output derived boost network.
The low voltage output can be boosted using a charge
pump or flyback winding to greater than 4.7V. The system
Soft-Start and Latchoff with the RUN/SS Pin
The RUN/SS pin provides a means to shut down the
LTC1778 as well as a timer for soft-start and overcurrent
latchoff. Pulling the RUN/SS pin below 0.8V puts the
LTC1778 into a low quiescent current shutdown (IQ <
30µA). Releasing the pin allows an internal 1.2µA current
source to charge up the external timing capacitor CSS. If
RUN/SS has been pulled all the way to ground, there is a
delay before starting of about:
tDELAY =
1.5V
CSS = 1.3s/µF CSS
1.2µA
(
)
When the voltage on RUN/SS reaches 1.5V, the LTC1778
begins operating with a clamp on ITH of approximately
0.9V. As the RUN/SS voltage rises to 3V, the clamp on ITH
is raised until its full 2.4V range is available. This takes an
additional 1.3s/µF, during which the load current is folded
back until the output reaches 75% of its final value. The pin
can be driven from logic as shown in Figure 7. Diode D1
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reduces the start delay while allowing CSS to charge up
slowly for the soft-start function.
After the controller has been started and given adequate
time to charge up the output capacitor, CSS is used as a
short-circuit timer. After the RUN/SS pin charges above
4V, if the output voltage falls below 75% of its regulated
value, then a short-circuit fault is assumed. A 1.8µA current then begins discharging CSS. If the fault condition
persists until the RUN/SS pin drops to 3.5V, then the controller turns off both power MOSFETs, shutting down the
converter permanently. The RUN/SS pin must be actively
pulled down to ground in order to restart operation.
The overcurrent protection timer requires that the soft-start
timing capacitor CSS be made large enough to guarantee
that the output is in regulation by the time CSS has reached
the 4V threshold. In general, this will depend upon the size
of the output capacitance, output voltage and load current
characteristic. A minimum soft-start capacitor can be
estimated from:
CSS > COUT VOUT RSENSE (10 – 4 [F/V s])
Generally 0.1µF is more than sufficient.
Overcurrent latchoff operation is not always needed or
desired. Load current is already limited during a shortcircuit by the current foldback circuitry and latchoff
operation can prove annoying during troubleshooting.
The feature can be overridden by adding a pull-up current
greater than 5µA to the RUN/SS pin. The additional
current prevents the discharge of CSS during a fault and
also shortens the soft-start period. Using a resistor to V IN
as shown in Figure 8a is simple, but slightly increases
shutdown current. Connecting a resistor to INTV CC as
INTVCC
RSS*
VIN
3.3V OR 5V
D1
RUN/SS
RSS*
D2*
RUN/SS
2N7002
CSS
CSS
1778 F08
*OPTIONAL TO OVERRIDE
OVERCURRENT LATCHOFF
(8a)
(8b)
Figure 8. RUN/SS Pin Interfacing with Latchoff Defeated
shown in Figure 8b eliminates the additional shutdown
current, but requires a diode to isolate CSS . Any pull-up
network must be able to pull RUN/SS above the 4.2V
maximum threshold of the latchoff circuit and overcome
the 4µA maximum discharge current.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in the circuit produce losses, four main sources
account for most of the losses in LTC1778 circuits:
1. DC I2R losses. These arise from the resistances of the
MOSFETs, inductor and PC board traces and cause the
efficiency to drop at high output currents. In continuous
mode the average output current flows through L, but is
chopped between the top and bottom MOSFETs. If the two
MOSFETs have approximately the same RDS(ON), then the
resistance of one MOSFET can simply be summed with the
resistances of L and the board traces to obtain the DC I2R
loss. For example, if RDS(ON) = 0.01Ω and RL = 0.005Ω, the
loss will range from 15mW to 1.5W as the output current
varies from 1A to 10A.
2. Transition loss. This loss arises from the brief amount
of time the top MOSFET spends in the saturated region
during switch node transitions. It depends upon the input
voltage, load current, driver strength and MOSFET
capacitance, among other factors. The loss is significant
at input voltages above 20V and can be estimated from:
Transition Loss ≅ (1.7A–1) VIN2 IOUT CRSS f
3. INTVCC current. This is the sum of the MOSFET driver
and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high
efficiency source, such as an output derived boost network or alternate supply if available.
4. CIN loss. The input capacitor has the difficult job of
filtering the large RMS input current to the regulator. It
must have a very low ESR to minimize the AC I2R loss and
sufficient capacitance to prevent the RMS current from
causing additional upstream losses in fuses or batteries.
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Other losses, including COUT ESR loss, Schottky diode D1
conduction loss during dead time and inductor core loss
generally account for less than 2% additional loss.
When making adjustments to improve efficiency, the
input current is the best indicator of changes in efficiency.
If you make a change and the input current decreases, then
the efficiency has increased. If there is no change in input
current, then there is no change in efficiency.
Checking Transient Response
Selecting a standard value of 1.8µH results in a maximum
ripple current of:
∆IL =
⎛ 2.5V ⎞
⎜1–
⎟ = 5.1A
28V ⎠
250kHz 1.8µH ⎝
2.5V
(
)(
)
Next, choose the synchronous MOSFET switch. Choosing
a Si4874 (RDS(ON) = 0.0083Ω (NOM) 0.010Ω (MAX),
θJA = 40°C/W) yields a nominal sense voltage of:
VSNS(NOM) = (10A)(1.3)(0.0083Ω) = 108mV
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ∆ILOAD (ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating a feedback error signal used by
the regulator to return VOUT to its steady-state value.
During this recovery time, VOUT can be monitored for
overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 9
will provide adequate compensation for most applications. For a detailed explanation of switching control loop
theory see Application Note 76.
Tying VRNG to 1.1V will set the current sense voltage range
for a nominal value of 110mV with current limit occurring
at 146mV. To check if the current limit is acceptable,
assume a junction temperature of about 80°C above a
70°C ambient with ρ150°C = 1.5:
Design Example
Because the top MOSFET is on for such a short time, an
Si4884 RDS(ON)(MAX) = 0.0165Ω, CRSS = 100pF, θJA =
40°C/W will be sufficient. Checking its power dissipation
at current limit with ρ100°C = 1.4:
As a design example, take a supply with the following
specifications: VIN = 7V to 28V (15V nominal), VOUT = 2.5V
±5%, IOUT(MAX) = 10A, f = 250kHz. First, calculate the
timing resistor with VON = VOUT:
RON =
2.5V
= 1.42MΩ
(0.7V)(250kHz)(10pF )
and choose the inductor for about 40% ripple current at
the maximum VIN:
⎛ 2.5V ⎞
L=
⎜ 1−
⎟ = 2.3µH
28V ⎠
250kHz 0.4 10A ⎝
2.5V
(
)( )( )
ILIMIT ≥
146mV
1
5.1A = 12A
2
(1.5)(0.010Ω) ( )
+
and double check the assumed TJ in the MOSFET:
PBOT =
28V – 2 .5V
12A
28V
2
( ) (1.5)(0.010Ω) = 1.97 W
TJ = 70°C + (1.97W)(40°C/W) = 149°C
2.5V
12A
28V
2
( ) (1.4)(0.0165Ω) +
2
(1.7)(28V) (12A)(100pF )(250kHz)
PTOP =
= 0.30W + 0.40W = 0.7 W
TJ = 70°C + (0.7W)(40°C/W) = 98°C
The junction temperatures will be significantly less at
nominal current, but this analysis shows that careful
attention to heat sinking will be necessary in this circuit.
1778fb
17
LTC1778/LTC1778-1
U
W
U U
APPLICATIO S I FOR ATIO
CIN is chosen for an RMS current rating of about 5A at
85°C. The output capacitors are chosen for a low ESR of
0.013Ω to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage will be
only:
∆VOUT(RIPPLE) = ∆IL(MAX) (ESR)
= (5.1A) (0.013Ω) = 66mV
However, a 0A to 10A load step will cause an output
change of up to:
∆VOUT(STEP) = ∆ILOAD (ESR) = (10A) (0.013Ω) = 130mV
An optional 22µF ceramic output capacitor is included to
minimize the effect of ESL in the output ripple. The
complete circuit is shown in Figure 9.
PC Board Layout Checklist
When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires
a dedicated ground plane layer. Also, for higher currents,
it is recommended to use a multilayer board to help with
heat sinking power components.
1
R3
11k
R4
39k
RPG
100k 2
3
CC1
500pF
4
RC
20k
CC2
100pF
5
6
7
R1
14.0k
R2
30.1k
8
C2
6.8nF
LTC1778
RUN/SS BOOST
PGOOD
TG
VRNG
SW
FCB
ITH
SGND
16
15
• Place CIN, COUT, MOSFETs, D1 and inductor all in one
compact area. It may help to have some components on
the bottom side of the board.
• Place LTC1778 chip with pins 9 to 16 facing the power
components. Keep the components connected to pins
1 to 8 close to LTC1778 (noise sensitive components).
• Use an immediate via to connect the components to
ground plane including SGND and PGND of LTC1778.
Use several bigger vias for power components.
• Use compact plane for switch node (SW) to improve
cooling of the MOSFETs and to keep EMI down.
• Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
• Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of
power component. You can connect the copper areas to
any DC net (VIN, VOUT, GND or to any other DC rail in
your system).
DB
CMDSH-3
CB
0.22µF
M1
Si4884
L1
1.8µH
+
PGND
BG
INTVCC
ION
VIN
VFB
EXTVCC
VIN
5V TO 28V
CIN
10µF
35V
×3
14
13
M2
Si4874
D1
B340A
COUT1-2
180µF
4V
×2
COUT3
22µF
6.3V
X7R
VOUT
2.5V
10A
12
11
+
CSS
0.1µF
• The ground plane layer should not have any traces and
it should be as close as possible to the layer with power
MOSFETs.
CVCC
4.7µF
10
RF
1Ω
9
CF
0.1µF
RON
1.4MΩ
1778 F09
CIN: UNITED CHEMICON THCR60EIHI06ZT
COUT1-2: CORNELL DUBILIER ESRE181E04B
L1: SUMIDA CEP125-1R8MC-H
Figure 9. Design Example: 2.5V/10A at 250kHz
1778fb
18
LTC1778/LTC1778-1
U
W
U U
APPLICATIO S I FOR ATIO
When laying out a printed circuit board, without a ground
plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in
Figure 10.
• Segregate the signal and power grounds. All small
signal components should return to the SGND pin at
one point which is then tied to the PGND pin close to the
source of M2.
• Place M2 as close to the controller as possible, keeping
the PGND, BG and SW traces short.
• Connect the input capacitor(s) CIN close to the power
MOSFETs. This capacitor carries the MOSFET AC
current.
• Keep the high dV/dT SW, BOOST and TG nodes away
from sensitive small-signal nodes.
• Connect the INTVCC decoupling capacitor CVCC closely
to the INTVCC and PGND pins.
• Connect the top driver boost capacitor CB closely to the
BOOST and SW pins.
• Connect the VIN pin decoupling capacitor CF closely to
the VIN and PGND pins.
CSS
2
3
4
RUN/SS
BOOST
PGOOD
TG
VRNG
SW
FCB
PGND
15
DB
14
5
BG
ITH
+
M1
13
D1
RC
CC2
12
CIN
VIN
M2
CVCC
6
7
SGND
INTVCC
ION
VIN
8
VFB
EXTVCC
–
11
10
R1
R2
L
16
+
CC1
CB
LTC1778
1
9
–
VOUT
COUT
CF
+
RF
RON
BOLD LINES INDICATE HIGH CURRENT PATHS
1778 F10
Figure 10. LTC1778 Layout Diagram
1778fb
19
LTC1778/LTC1778-1
U
TYPICAL APPLICATIO S
1.5V/10A at 300kHz from 3.3V Input
CSS
0.1µF
1
RR2
39k
RR1
11k
RPG
100k 2
3
CC1
680pF
4
RC
20k
5
CC2
100pF
6
7
R1
10k
8
LTC1778
RUN/SS BOOST
PGOOD
TG
VRNG
SW
DB
CMDSH-3
16
CB
0.22µF
15
CIN1-2
22µF
6.3V
×2
M1
IRF7811A
14
L1, 0.68µH
+
PGND
FCB
BG
ITH
SGND
INTVCC
ION
VIN
VFB
EXTVCC
13
M2
IRF7811A
D1
B320B
COUT
270µF
2V
×2
+
VIN
3.3V
CIN3
330µF
6.3V
VOUT
1.5V
10A
12
CVCC
4.7µF
11
10
5V
9
RON
576k
R2
8.87k
1778 TA01
CIN1-2: MURATA GRM42-2X5R226K6.3
COUT: CORNELL DUBILIER ESRE271M02B
1.2V/6A at 300kHz
CSS
0.1µF
1
RPG
100k 2
3
CC1
470pF
4
RC
20k
5
CC2
100pF
6
7
R1
20k
R2
10k
8
C2
2200pF
LTC1778
RUN/SS
BOOST
PGOOD
TG
VRNG
FCB
SW
PGND
BG
ITH
SGND
INTVCC
ION
VIN
VFB
EXTVCC
16
15
DB
CMDSH-3
CB
0.22µF
14
M1
1/2 FDS6982S
L1
1.8µH
13
CIN
10µF
25V
×2
M2
1/2 FDS6982S
+
COUT1
180µF
2V
VIN
5V TO 25V
VOUT
1.2V
6A
COUT2
10µF
6.3V
12
11
CVCC
4.7µF
10
RF
1Ω
9
CF
0.1µF
RON
510k
1778 TA02
CIN: TAIYO YUDEN TMK432BJ106MM
COUT1: CORNELL DUBILIER ESRD181M02B
COUT2: TAIYO YUDEN JMK316BJ106ML
L1: TOKO 919AS-1R8N
1778fb
20
LTC1778/LTC1778-1
U
TYPICAL APPLICATIO S
Single Inductor, Positive Output Buck/Boost
CSS
0.1µF
1
2
3
4
CC1 1nF RC
47k
5
CC2
220pF
6
7
C1
100pF
8
LTC1778-1
RUN/SS BOOST
VON
TG
VRNG
SW
DB
CMDSH-3
16
CB
0.22µF
15
CIN
22µF
D2
50V IR 12CWQ03FN
×2
M1
IRF7811A
14
PGND
BG
ITH
SGND
INTVCC
ION
VIN
VFB
EXTVCC
13
M2
IRF7811A
COUT
100µF
20V
×6
M3
Si4888
12
CVCC
4.7µF
11
D1
B340A
RF
1Ω
10
CF
0.1µF
9
PGND
CIN: MARCON THER70EIH226ZT
COUT: AVX TPSV107M020R0085
L1: SCHOTT 36835-1
RON1
1.5M
1%
RON2
1.5M
1%
R2
140k
1%
VIN
6V TO 18V
VOUT
12V
L1 4.8µH
+
FCB
R1
10k 1%
VIN IOUT
18V 6A
12V 5A
6V 3.3A
1778 TA04
12V/5A at 300kHz
LTC1778-1
16
1
RUN/SS BOOST
2
3
CC1
2.2nF
4
RC
20k
5
CC2
100pF
6
7
R1
10k
R2
140k
8
C2
2200pF
VON
TG
VRNG
SW
FCB
ITH
SGND
ION
VFB
15
DB
CMDSH-3
CB
0.22µF
14
CIN
22µF
50V
M1
L1 10µH
+
PGND
BG
INTVCC
VIN
EXTVCC
13
M2
D1
VIN
14V TO 28V
VOUT
12V
5A
COUT
220µF
16V
12
11
+
CSS
0.1µF
CVCC
4.7µF
10
RF
1Ω
9
CF
0.1µF
RON
1.6M
1778 TA05
CIN: UNITED CHEMICON THCR70E1H226ZT
COUT: SANYO 16SV220M
L1: SUMIDA CDRH127-100
M1, M2: FAIRCHILD FDS6680A
D1: DIODES, INC. B340A
(847) 696-2000
(619) 661-6835
(847) 956-0667
(408) 822-2126
(805) 446-4800
1778fb
21
LTC1778/LTC1778-1
U
TYPICAL APPLICATIO S
Positive-to-Negative Converter, –5V/5A at 300kHz
CSS
0.1µF
1
2
3
4
CC1
4700pF
RC
10k
5
CC2
100pF
6
7
R1
10k
R2
52.3k
8
LTC1778-1
16
RUN/SS BOOST
VON
TG
VRNG
SW
FCB
ITH
SGND
ION
VFB
15
VIN IOUT
20V 8A
10V 6.7A
5V 5A
DB
CMDSH-3
CB
0.22µF
14
CIN1
10µF
25V
×2
M1
IRF7811A
BG
INTVCC
VIN
EXTVCC
VIN
5V TO 20V
L1 2.7µH
+
PGND
CIN2
10µF
35V
13
M2
IRF7822
D1
B340A
12
11
CVCC
4.7µF
10
RF
1Ω
9
CF
0.1µF
COUT
100µF
6V
×3
VOUT
–5V
RON
698k
1778 TA06
CIN1: TAIYO YUDEN TMK432BJ106MM
CIN2: SANYO 35CV10GX
COUT: PANASONIC EEFUD0J101R
L1: PANASONIC ETQPAF2R7H
1778fb
22
LTC1778/LTC1778-1
U
PACKAGE DESCRIPTIO
GN Package
16-Lead Plastic SSOP (Narrow 0.150)
(LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.045 ±.005
16 15 14 13 12 11 10 9
.254 MIN
.009
(0.229)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
.015 ± .004
× 45°
(0.38 ± 0.10)
.007 – .0098
(0.178 – 0.249)
2 3
4
5 6
7
.0532 – .0688
(1.35 – 1.75)
8
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN16 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1778fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LTC1778/LTC1778-1
U
TYPICAL APPLICATIO
Typical Application 2.5V/3A at 1.4MHz
CSS
0.1µF
LTC1778
RUN/SS BOOST
1
RPG
100k 2
3
4
CC1
470pF
RC
33k
8
C2
2200pF
SW
SGND
7
R2
24.9k
VRNG
15
CB
0.22µF
14
CIN
10µF
25V
M1
1/2 Si9802
L1, 1µH
+
PGND
BG
ITH
6
R1
11.5k
TG
FCB
5
CC2
100pF
PGOOD
16
DB
CMDSH-3
INTVCC
ION
VIN
VFB
EXTVCC
13
M2
1/2 Si9802
VIN
9V TO 18V
VOUT
2.5V
3A
COUT
120µF
4V
12
11
CVCC
4.7µF
10
RF
1Ω
9
CF
0.1µF
RON
220k
1778 TA03
CIN: TAIYO YUDEN TMK432BJ106MM
COUT: CORNELL DUBILIER ESRD121M04B
L1: TOKO A921CY-1R0M
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DESCRIPTION
COMMENTS
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High Efficiency, Synchronous Step-Down Controller
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Mobile VID; 0.925V ≤ VOUT ≤ 2V; 3.5V ≤ VIN ≤ 36V
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Current Mode; 550kHz; Very Small Solution Size
LTC1773
Synchronous Step-Down Controller
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LTC1876
2-Phase, Dual Synchronous Step-Down Controller with
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3.5V ≤ VIN ≤ 36V, Power Good Output, 300kHz Operation
LTC3713
Low VIN High Current Synchronous Step-Down Controller
1.5V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)VIN, IOUT Up to 20A
LTC3778
Low VOUT, No RSENSE Synchronous Step-Down Controller
0.6V ≤ VOUT ≤ (0.9)VIN, 4V ≤ VIN ≤ 36V, IOUT Up to 20A
60V Synchronous Step-Down Controller
Current Mode, Output Slew Rate Control
®
LT 3800
Burst Mode is a registered trademark of Linear Technology Corporation.
1778fb
24
Linear Technology Corporation
LT/LT 0405 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2001