Maxim MAX1762EUB High-efficiency, 10-pin î¼max, step-down controllers for notebook Datasheet

19-1923; Rev 1; 10/05
KIT
ATION
EVALU
E
L
B
AVAILA
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
The MAX1762/MAX1791 PWM step-down controllers
provide high efficiency, excellent transient response,
and high DC output accuracy needed for stepping
down high-voltage batteries to generate low-voltage
CPU core, I/O, and chipset RAM supplies in notebook
computers and PDAs.
Maxim’s proprietary Quick-PWM™ pulse-width modulator is a free-running constant on-time type with input
feed-forward. Its high operating frequency (300kHz)
allows small external components to be utilized in PC
board area-critical applications such as subnotebook
computers and smart phones. PWM operation occurs
at heavy loads, and automatic switchover to pulse-skipping operation occurs at lighter loads. The external
high-side p-channel and low-side n-channel MOSFETs
require no bootstrap components. The MAX1762/
MAX1791 are simple, easy to compensate, and do not
have the noise sensitivity of conventional fixed-frequency current-mode PWMs.
These devices achieve high efficiency at a reduced
cost by eliminating the current-sense resistor found in
traditional current-mode PWMs. Efficiency is further
enhanced by their ability to drive synchronous-rectifier
MOSFETs. The MAX1762/MAX1791 come in a 10-pin
µMAX package and offer two fixed voltages (Dual
Mode™) for each device, 1.8V/2.5V/adj (MAX1762) and
3.3V/5.0V/adj (MAX1791).
________________________Applications
Features
♦ High Operating Frequency (300kHz)
♦ No Current-Sense Resistor
♦ Accurate Current Limit
♦ ±1% Total DC Error over Line and During
Continuous Conduction
♦ Dual Mode Fixed Output
1.8V/2.5V/adj (MAX1762)
3.3V/5.0V/adj (MAX1791)
♦ 0.5V to 5.5V Output Adjust Range
♦ 5V to 20V Input Range
♦ Automatic Light-Load Pulse Skipping Operation
♦ Free-Running On-Demand PWM
♦ Foldback Mode™ UVLO
♦ PFET/NFET Synchronous Buck
♦ 4.65V at 25mA Linear Regulator Output
♦ 5µA Shutdown Supply Current
♦ 230µA Quiescent Supply Current
♦ 10-Pin µMAX Package
Ordering Information
TEMP RANGE
PIN-PACKAGE
Notebooks
Handy-Terminals
MAX1762EUB
PART
-40°C to +85°C
10 µMAX
Subnotebooks
PDAs
MAX1791EUB
-40°C to +85°C
10 µMAX
Digital Cameras
Smart Phones
Pin Configuration
1.8V/2.5V Logic
and I/O Supplies
Typical Operating Circuit
VBATT
(5V TO 20V)
TOP VIEW
VL 1
VL
VP
MAX1762
MAX1791
REF
DH
FB
CS
OUT
DL
SHDN
GND
VOUT
1.8V/3.3V
10 VP
REF
2
FB
3
OUT
4
7
DL
SHDN
5
6
GND
MAX1762
MAX1791
9
DH
8
CS
µMAX
Quick-PWM, Dual Mode, and Foldback Mode are a trademarks of Maxim Integrated Products, Inc.
________________________________________________________________ Maxim Integrated Products
1
For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
MAX1762/MAX1791
General Description
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
ABSOLUTE MAXIMUM RATINGS
VP, SHDN to GND ..................................................-0.3V to +22V
VP to VL ..................................................................-0.3V to +22V
OUT, VL to GND .......................................................-0.3V to +6V
DL, FB, REF to GND ....................................-0.3V to (VL + 0.3V)
DH to GND....................................................-0.3V to (VP + 0.3V)
CS to GND ....................................................-2.0V to (VP + 0.3V)
REF Short Circuit to GND ...........................................Continuous
Continuous Power Dissipation (TA = +70°C)
10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VVP = 15V, VL enabled, CVL = 1µF, CREF = 0.1µF, TA = 0 to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
(Note 1)
PARAMETER
SYMBOL
VP Input Voltage Range
VVP
VL Input Voltage Range
VVL
CONDITIONS
MIN
MAX
UNITS
5
20
V
VL (overdriven)
4.75
5.25
V
1.773
1.8
1.827
V
OUT Output Voltage
(MAX1762, 1.8V Fixed)
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode
OUT Output Voltage
(MAX1762, 2.5V Fixed)
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode
2.463
2.5
2.538
V
OUT Output Voltage
(MAX1791, 3.3V Fixed)
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode
3.250
3.3
3.350
V
OUT Output Voltage
(MAX1791, 5V Fixed)
VOUT
VVP = 7V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode
4.925
5
5.075
V
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = OUT, continuous conduction mode
1.231
1.250
1.269
V
5.5
V
800
1700
kΩ
0.1
µA
OUT Output Voltage (Adj Mode)
Output Voltage Adjust Range
2
TYP
0.5
OUT Input Resistance
Adjustable-output mode
300
FB Input Bias Current
VFB = 1.3V
-0.1
Soft-Start Ramp Time
Zero to full ILIM
On-Time (Note 2)
tON
Minimum Off-Time
tOFF
1700
µs
VOUT = 1.25V, VVP = 6V
666
740
814
VOUT = 5V, VVP = 6V
2550
2830
3110
(Note 2)
300
400
500
ns
153
260
µA
VVL = float
227
410
VVL = 5V
93
200
VL Quiescent Supply Current
FB = GND, VVL = 5V, OUT forced above the
regulation point
VP Quiescent Supply Current
FB = GND, OUT forced
above the regulation point,
VVP = 20V
ns
µA
VL Shutdown Supply Current
VVL = 5V, SHDN = GND
2
15
µA
VP Shutdown Supply Current
SHDN = GND, measured at VP, VVL = 0 or 5V
4
12
µA
VL Output Voltage
ILOAD = 0 to 25mA, VVP = 5V to 20V
4.65
4.75
V
4.5
_______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
(VVP = 15V, VL enabled, CVL = 1µF, CREF = 0.1µF, TA = 0 to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
(Note 1)
CONDITIONS
MIN
TYP
MAX
UNITS
Reference Voltage
PARAMETER
VVL = 4.75V to 5.25V, no load
1.98
2
2.02
V
Reference Load Regulation
IREF = 0 to 50µA
0.01
V
REF Sink Current
REF in regulation
REF Fault Lockout Voltage
Falling edge
Output Undervoltage Threshold
(Foldback)
With respect to regulation point, no load
60
70
80
%
Output Undervoltage Lockout
Time (Foldback)
From SHDN signal going high VOUT < 0.6 x
regulation point
10
20
42
ms
-90
-100
-110
mV
Current-Limit Threshold
SYMBOL
VILIM
Thermal Shutdown Threshold
Hysteresis = 10oC
VL Undervoltage Lockout
Threshold
Rising edge, hysteresis = 20mV, PWM
disabled below this level
DH Gate Driver On-Resistance
VVP = 6V to 20V, measure at 50mA
DL Gate Driver On-Resistance
(Pullup)
10
µA
1.6
V
o
160
4.1
C
4.4
V
5
8
Ω
DL, high state, measure at 50mA
5
8
Ω
DL Gate Driver On-Resistance
(Pulldown)
DL, low state, measure at 50mA
1
5
Ω
DH Gate Driver Source/Sink
Current
VDH = 3V, VVP = 6V
0.6
A
DL Gate Driver Sink Current
VDL = 2.5V
0.9
A
DL Gate Driver Source Current
VDL = 2.5V
0.5
A
SHDN Logic Input High
Threshold Voltage
VIH
SHDN Logic Input Low
Threshold Voltage
VIL
1.6
MAX1762 VOUT = 1.8V fixed
Dual Mode Threshold Voltage
MAX1791 VOUT = 3.3V fixed
MAX1762 VOUT = 2.5V fixed
MAX1791 VOUT = 5V fixed
SHDN Logic Input Current
SHDN = 0 or 5V
V
0.6
V
50
100
150
mV
2.5
3.25
4
V
+2
µA
-2
_______________________________________________________________________________________
3
MAX1762/MAX1791
ELECTRICAL CHARACTERISTICS (continued)
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
ELECTRICAL CHARACTERISTICS
(VVP = 15V, VL enabled, CVL = 1µF, CREF = 0.1µF, TA = -40 to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
VVP
VL Input Voltage Range
VVL
OUT Output Voltage
(MAX1762, 1.8V Fixed)
CONDITIONS
MIN
TYP
MAX
UNITS
5
20
V
VL (overdriven)
4.75
5.25
V
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode
1.773
1.827
V
OUT Output Voltage
(MAX1762, 2.5V Fixed)
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode
2.463
2.538
V
OUT Output Voltage
(MAX1791, 3.3V Fixed)
VOUT
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode
3.250
3.350
V
OUT Output Voltage
(MAX1791, 5V Fixed)
VOUT
VVP = 7V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode
4.925
5.075
V
OUT Output Voltage (adj Mode)
VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = OUT, continuous conduction mode
1.231
1.269
V
FB Input Bias Current
VFB = 1.3V
-0.2
0.2
µA
On-Time (Note 2)
tON
Minimum Off-Time
tOFF
VOUT = 1.25V, VVP = 6V
666
814
VOUT = 5V, VVP = 6V
2550
3110
(Note 2)
250
550
ns
260
µA
ns
VL Quiescent Supply Current
FB = GND, VVL = 5V, OUT forced above the
regulation point
VP Quiescent Supply Current
FB = GND, OUT forced above
the regulation point VVP = 20V
VL Shutdown Supply Current
VVL = 5V, SHDN = GND
15
µA
VP Shutdown Supply Current
SHDN = GND, measured at VP, VVL = 0 or 5V
12
µA
VVL = float
410
VVL = 5V
200
µA
VL Output Voltage
ILOAD = 0 to 25mA, VVP = 5V to 20V
4.5
4.75
V
Reference Voltage
VVL = 4.75V to 5.25V, no load
1.98
2.02
V
Reference Load Regulation
IREF = 0 to 50µA
0.01
V
REF Sink Current
REF in regulation
10
Output Undervoltage Threshold
(Foldback)
With respect to regulation point, no load
60
80
%
Output Undervoltage Lockout
Time (Foldback)
From SHDN signal going high, VOUT < 0.6 x
regulation point
10
42
ms
-90
-110
mV
Current-Limit Threshold
4
SYMBOL
VP Input Voltage Range
VILIM
_______________________________________________________________________________________
µA
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
(VVP = 15V, VL enabled, CVL = 1µF, CREF = 0.1µF, TA = -40 to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
SYMBOL
VL Undervoltage Lockout
Threshold
CONDITIONS
Rising edge, hysteresis = 20mV, PWM
disabled below this level
SHDN Logic Input High
Threshold Voltage
VIH
SHDN Logic Input Low
Threshold Voltage
VIL
MIN
4.1
TYP
MAX
UNITS
4.4
V
1.6
MAX1762 VOUT = 1.8V fixed
MAX1791 VOUT = 3.3V fixed
Dual Mode Threshold Voltage
MAX1762 VOUT = 2.5V fixed
MAX1791 VOUT = 5V fixed
V
0.6
V
50
150
mV
2.5
4
V
Note 1: Specifications to -40°C are guaranteed by design, not production tested.
Note 2: One-shot times are measured at the DH pin (VP = 15V, CDH = 400pF, 90% point to 90% point; see drawing below for
measurement details).
tON
DH
90%
90%
_______________________________________________________________________________________
5
MAX1762/MAX1791
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(TA = +25°C, unless otherwise noted.)
MAX1762
EFFICIENCY vs. LOAD (1.8V)
EFFICIENCY (%)
VVP = 18V
60
VVP = 12V
50
40
VVP = 7V
VVP = 12V
60
VVP = 18V
50
90
40
70
50
30
30
20
20
10
10
10
0
0
10
100
1000
1
10
100
1000
10,000
0.1
10
100
MAX1791
EFFICIENCY vs. LOAD (5V)
MAX1791
EFFICIENCY vs. LOAD (3.3V)
MAX1791
EFFICIENCY vs. LOAD (3.0V)
40
70
VVP = 18V
60
VVP = 12V
50
100
40
VVP = 7V
70
40
20
20
10
10
10
30
0
0
100
1000
10,000
0.1
1
LOAD CURRENT (mA)
10
100
1000
0.1
10,000
1
10
FREQUENCY vs. SUPPLY VOLTAGE
VL VOLTAGE ERROR vs. OUTPUT CURRENT
350
VOUT = 5.0V
325
VOUT = 2.5V
300
0.1
VVP = 12V
VOUT = 2.5V
0
VL VOLTAGE ERROR (%)
MAX1762/91 toc07
VOUT = 3.3V
100
LOAD CURRENT (mA)
LOAD CURRENT (mA)
375
FREQUENCY (kHz)
VVP = 12V
50
20
10
VVP = 18V
60
30
1
VVP = 7V
80
30
0
VVP = 5V
90
EFFICIENCY (%)
VVP = 12V
50
80
EFFICIENCY (%)
60
VVP = 5V
90
MAX1762/91 toc08
MAX1762/91 toc04
VVP = 18V
70
100
-0.1
-0.2
-0.3
-0.4
275
VOUT = 1.8V
-0.5
LOAD = 1A
250
5
8
-0.6
11
14
SUPPLY VOLTAGE (V)
10,000
MAX1762/91 toc06
LOAD CURRENT (mA)
80
6
1000
LOAD CURRENT (mA)
VVP = 7V
0.1
1
LOAD CURRENT (mA)
100
90
VVP = 18V
0
0.1
10,000
VVP = 12V
40
20
1
VVP = 7V
60
30
0.1
VVP = 5V
80
MAX1762/91 toc05
EFFICIENCY (%)
70
80
70
100
MAX1762/91 toc03
VVP = 7V
80
VVP = 5V
90
EFFICIENCY (%)
VVP = 5V
90
100
MAX1762/91 toc01
100
MAX1762
EFFICIENCY vs. LOAD (1V)
MAX1762/91 toc02
MAX1762
EFFICIENCY vs. LOAD (2.5V)
EFFICIENCY (%)
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
17
0
5
10
15
20
25
VL CURRENT (mA)
_______________________________________________________________________________________
1000
10,000
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
SUPPLY CURRENT vs. INPUT VOLTAGE
(SHUTDOWN)
SUPPLY CURRENT vs. INPUT VOLTAGE
200
MAX1762/91 toc10
5.8
5.6
SUPPLY CURRENT (µA)
250
SUPPLY CURRENT (µA)
6.0
MAX1762/91 toc09
300
VOUT = 3.3V
NO LOAD
150
100
5.4
5.2
5.0
NO LOAD
4.8
4.6
4.4
50
4.2
4.0
0
5
8
11
14
5
17
8
MAX1762/91 toc13
MAX1762/91 toc12
MAX1762/91 toc11
7V
VOUT
AC-COUPLED
20mV/div
100µs/div
VVP = 12V, ILOAD = 0 TO 2A, VOUT = 1.8V
VVP = 12V, IL = 0 TO 2A, VOUT = 2.5V
SHUTDOWN AND STARTUP WAVEFORMS
(IL = 300mA)
OUTPUT OVERLOAD WAVEFORMS
MAX1762/91 toc14
ILOAD
2A/div
IL
2A/div
100µs/div
100µs/div
VVP = 7.5V TO 8V, ILOAD = 0, VOUT = 2.5V
VOUT
AC-COUPLED
100mV/div
VOUT
AC-COUPLED
100mV/div
VP
1V/div
SHUTDOWN AND STARTUP WAVEFORMS
(IL = 2.5A)
MAX1762/91 toc15
VOUT
AC-COUPLED
ILOAD
2A/div
100µs/div
17
LOAD-TRANSIENT RESPONSE
LOAD-TRANSIENT RESPONSE
LINE-TRANSIENT RESPONSE
14
VP (V)
VP (V)
VVP = 12V, ILOAD = 0 TO 3A, VOUT = 2.5V
11
MAX1762/MAX1791
Typical Operating Characteristics (continued)
(TA = +25°C, unless otherwise noted.)
2ms/div
VVP = 8V, ILOAD = 300mA, VOUT = 2.5V
MAX1762/91 toc16
SHDN
5V/div
SHDN
5V/div
VOUT
2V/div
VOUT
2V/div
ILX
1A/div
ILX
2A/div
2ms/div
VVP = 8V, ILOAD = 2.5A, VOUT = 2.5V
_______________________________________________________________________________________
7
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
MAX1762/MAX1791
Pin Description
PIN
NAME
FUNCTION
1
VL
+4.65V Linear Regulator Output. Serves as the supply input for the DL gate driver and supplies up to
25mA to external loads. VL can be overdriven using an external 5V supply. Bypass VL to GND with
at least a 1µF ceramic capacitor.
2
REF
3
FB
4
OUT
5
SHDN
Shutdown Input. Connect to a voltage less than VIL (<0.6V) to shut down the device. Connect to a
voltage greater than VIH (>1.6V) for normal operation.
6
GND
Analog and Power Ground
7
DL
Low-Side Gate Driver Output. DL swings between VL and GND.
8
CS
Current-Sense Connection. For lossless current sensing, connect CS to the junction of the MOSFETs
and inductor. For more accurate current sensing, connect CS to a current-sense resistor from the
source of the low-side switch to GND.
9
DH
High-Side Gate Driver Output. DH swings between VP and GND.
10
VP
Battery Voltage Supply Input. Used for PWM one-shot timing and as the input for the VL regulator
and DH gate drivers.
2V Reference Voltage Output. Bypass to GND with 0.1µF ceramic capacitor. REF can deliver up to
50µA for external loads.
Feedback Input. Connect to an external resistive divider from OUT to GND in adjustable version.
Regulates to 1.25V. FB also serves as Dual Mode select pin. Connect FB to GND for a fixed 1.8V
(MAX1762) or 3.3V (MAX1791) output, or to VL for a fixed 2.5V (MAX1762) or 5.0V (MAX1791) output.
Output Voltage Connection. OUT is used for sensing the output voltage to determine the on-time and
also serves as the feedback input in fixed-output modes.
Standard Application Circuit
Detailed Description
The standard application circuit (Figure 1) generates a
low-voltage output for general-purpose use in notebook
computers (I/O supply, fixed CPU, core supply, and
DRAM supply). This DC-DC converter steps down battery voltage from 5V to 20V with high efficiency and
accuracy to a fixed voltage of 1.8V/2.5V/adj (MAX1762)
or 3.3V/5.0V/adj (MAX1791). Both the MAX1762 and
MAX1791 can be configured for adjustable output voltages (VOUT > 1.25V), using a resistive voltage-divider
from VOUT to FB to adjust the output voltage (Figure 2).
Similarly, Figure 3 shows an application circuit for VOUT
< 1.25V, where a resistive voltage-divider from REF to
FB is used to set the output voltage. Figure 4 shows
how to set the regulator’s current limit with an external
sense resistor from CS to GND. Table 1 lists the components for each application circuit, and Table 2 contains contact information for the component
manufacturers.
The MAX1762/MAX1791 step-down controllers are targeted at low-voltage chipsets and RAM power supplies
for notebook and subnotebook computers, with additional applications in digital cameras, PDAs, and
handy-terminals. Maxim’s proprietary Quick-PWM
pulse-width modulator (Figure 5) is specifically
designed for handling fast load steps while maintaining
a relatively constant operating frequency (300kHz) over
a wide range of input voltages (5V to 20V). The
MAX1762 has fixed 1.8V or 2.5V outputs, while the
MAX1791 has fixed 3.3V or 5.0V output voltages. Using
an external resistive divider, VOUT can be set between
0.5V and 5.5V on either device. Quick-PWM architecture circumvents the poor load-transient response of
fixed-frequency current-mode PWMs. This type of
design avoids the problems commonly encountered
with conventional constant-on-time and constant-offtime PWM schemes.
8
_______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
MAX1762/MAX1791
VVP
C2
1µF
C1
10µF
10Ω
VL
VP
1µF
MAX1762
MAX1791
C3
0.1µF
REF
DH
FB
CS
OUT
DL
Q1
L1
7µH
Q2
VOUT
C4
220µF
GND
SHDN
Figure 1. Typical Application Circuit for Fixed Voltage
VVP
C2
1µF
C1
10µF
10Ω
VL
VP
1µF
MAX1762
MAX1791
C3
0.1µF
REF
DH
FB
CS
OUT
DL
Q1
L1
7µH
Q2
VOUT
C4
220µF
R1
GND
SHDN
R2
Figure 2. Typical Application Circuit for Adjustable Output VOUT > 1.25V
VVP
C1
10µF
C2
1µF
10Ω
VL
VP
1µF
MAX1762
MAX1791
C3
0.1µF
REF
DH
FB
CS
OUT
DL
Q1
R1
L1
7µH
R2
SHDN
Q2
VOUT
C4
220µF
GND
Figure 3. Typical Application Circuit for VOUT < 1.25V
_______________________________________________________________________________________
9
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
VVP
C1
10µF
C2
1µF
10Ω
VL
VP
1µF
MAX1762
MAX1791
C3
0.1µF
REF
DH
FB
CS
OUT
DL
Q1
L1
10µH
Q2
VOUT
C4
150µF
RS
GND
SHDN
Figure 4. Operation with External Current-Sense Resistor
Table 1. Component Selection for Standard Applications
10
COMPONENT
1.8V/2.5V/3.3V/5.0V AT 2A
1V AT 2A
Input Voltage Range
5V to 20V
5V to 20V
Inductor (µH)
7
5.2
L1 Inductor
CDRH104-7R0NC
Sumida
CDRH104-5R2NC
Sumida
Q1 MOSFETS
NDS8958A
Fairchild
SI4539ADY
Fairchild
C1 Input Capacitor
TMK432BJ106KM
Taiyo Yuden
TMK432BJ106
Taiyo Yuden
C2 VL Cap
EMK3160J105KL
Taiyo Yuden
LMK316BJ475
Taiyo Yuden
C3 REF Cap
UMK316BI104KH
Taiyo Yuden
UMK316BI104KH
Taiyo Yuden
C4 Output Cap
10TPB220M
Sanyo
6TPB150M
Sanyo
______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
MAX1762/MAX1791
Table 2. Component Manufacturers
MANUFACTURER
USA PHONE
WEBSITE INFO
Coiltronics
561-241-7876
www.coiltronics.com
Fairchild Semiconductor
408-822-2181
www.fairchildsemi.com
Sanyo
619-661-6835
www.secc.co.jp
USA
847-956-0666
Japan
81-3-3607-5111
Sumida
www.sumida.com
Taiyo Yuden
408-573-4150
www.t-yuden.com
10Ω
VP
VP
ON-TIME
COMPUTE
Q
OUT
TOFF
TON
DH
TRIG
Q 1-SHOT
DH
VIN
1µF
CIN
Q1
DRIVER
S
TON
TRIG
Q
1-SHOT
Q
R
S
Q
R
VP
CS
LINEAR
REG
VL
SHDN
VOS
-100mV
REF
-30%
OUT
ON/OFF
CONTROL
FEEDBACK
MUX
(FIGURE 9)
TIMER
VP
REF
ILIM
REF
CVL
2V
VREF
CREF
VL
FB
OUT
DL
UVP
LATCH
DL
COUT
Q2
DRIVER
GND
MAX1762
MAX1791
OUT
FB
Figure 5. Functional Diagram
VP Input and VL Logic Supply
An internal linear regulator supplied by VP produces
the +4.65V supply (VL) that powers the PWM controller,
logic, reference, and other blocks within the
MAX1762/MAX1791. This +4.65V low-dropout linear
regulator can supply up to 25mA for external loads.
Bypass VL to GND with at least a 1µF ceramic capacitor. VVP can range between 5V and 20V. VL is turned
off when the device is in shutdown and drops by
approximately 500mV during a fault condition, such as
when the output is short circuited to ground, and recovers when SHDN is cycled or power is reset. If VL is not
driven externally, then V VP should be at least 5V to
ensure operation. If VVP is running from a 5V (±10%)
supply, V VP should be externally connected to VL.
______________________________________________________________________________________
11
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
Overdriving the VL regulator with an external 5V supply
also increases the MAX1762/MAX1791s’ efficiency.
The MAX1762/MAX1791 include an input undervoltage
lockout (UVLO) circuit that prevents the device from
switching until VL > 4.4V (max). UVLO ensures there is
a sufficient drive for the external MOSFETs, prevents
the high-side MOSFET from being turned on for near
100% duty cycle, and keeps the output in regulation.
Voltage Reference (REF)
The 2V reference (REF) is accurate to ±1% over temperature, making REF useful as a precision system reference. Bypass REF to GND with a 0.1µF (min) ceramic
capacitor. REF can supply up to 50µA for external
loads. However, if tight-accuracy specs for either VOUT
or REF are essential, avoid loading REF. Loading slightly reduces the main output voltage by an amount that
tracks the reference-voltage load regulation error.
Free-Running Constant On-Time PWM
Controller with Input Feed-Forward
The PWM control architecture is a quasi-fixed-frequency constant on-time current-mode type with voltage
feed-forward. This architecture relies on the output ripple voltage to provide the PWM ramp signal; thus, the
output filter capacitor’s ESR acts as a feedback resistor. The control algorithm is very simple. The high-side
switch on-time is determined solely by a one-shot
whose period is inversely proportional to input voltage
and directly proportional to output voltage. There is
another one-shot that sets a minimum amount of offtime (500ns max). The on-time one-shot triggers when
all of the following conditions are met: the error comparator is low, the low-side switch current is below the
current-limit threshold, and the minimum off-time oneshot has timed out.
On-Time One-Shot
The on-time of the one-shot is inversely proportional to
the battery voltage as measured by the VP input, and
directly proportional to the output voltage sensed at
OUT:
t ON = K ×
(VOUT + 0.075V)
VBATT
where K is internally fixed at 3.349µs, and 0.075V is a
factor that accounts for the expected drop across the
synchronous switch. This arrangement maintains a
switching frequency that is nearly constant as V BATT,
ILOAD, and VOUT are changed. Table 3 shows the operating frequency range for the MAX1762/MAX1791.
Note that the output voltage adjust range for continuous-conduction operation is restricted by the non12
adjustable 0.5µs (max) minimum off-time. Worst-case
dropout performance is determined by the minimum
on-time spec. The worst-case duty factor limit is:
( )
t ON MIN
( )
( )
t ON MIN + t OFF MAX
=
2.55µs
2.55µs+0.5µs
= 84%
with VBATT = 6V and VOUT = 5V. Therefore, with IR voltage drops in the loop included, the minimum input voltage to achieve VOUT = 5V is about 6.1V, using the
step-down transfer function equation for duty cycle (DC
= VOUT/VIN). Typical units exhibit better performance.
Note that transient response is somewhat degraded
near dropout, and the circuit may need additional bulk
output capacitance to support fast load changes.
Automatic Pulse-Skipping Switchover
This PWM control algorithm automatically switches over
to pulse-skipping operation at light loads. The
MAX1762/MAX1791 truncates the low-side switch’s ontime when the inductor current drops to zero. The load
current level at which pulse-skipping/PWM crossover
occurs is equal to 1/2 the peak-to-peak ripple current,
which is a function of the inductor value (Figure 6):
ILOAD(SKIP) =
K × VOUT
2L
 VVP - VOUT 


VVP


The inductor current is never allowed to go negative. If
the output voltage is above its regulation point and the
inductor current reaches zero, the low-side driver is
switched off. Once the output voltage falls below its
regulation point, the high-side driver is switched on.
This causes a dead time in between when the highside and low-side drivers are on, skipping pulses and
resulting in the switching frequency slowing at light
loads, thereby improving efficiency.
MOSFET Gate Drivers
The DH and DL drivers are optimized for driving moderate-size power MOSFETs. This is consistent with the
low duty factor seen in the notebook CPU environment
where a large VBATT - VOUT differential exists. The highside driver (DH) is rated for 0.6A source/sink capability
and swings from VP to GND. The low-side driver (DL) is
Table 3. Operating Frequency
DEVICE
MAX1762/MAX1791
K
(µs)
MIN
(kHz)
TYP
(kHz)
MAX
(kHz)
3.349
268.7
298.5
328
______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
IPEAK
ILOAD = IPEAK/2
0
ON-TIME
TIME
INDUCTOR CURRENT
INDUCTOR CURRENT
ILOAD
ILIMIT
0
TIME
Figure 6. Pulse-Skipping/Discontinuous Crossover Point
Figure 7. “Valley” Current-Limit Threshold Point
rated for +0.5A, -0.9A source/sink capability and
swings from VL to GND.
The internal pulldown transistor that drives DL low is
robust, with a 1Ω typical on-resistance. This helps prevent DL from being pulled up during the fast rise time of
the inductor node, due to capacitive coupling from the
drain to the gate of the low-side synchronous-rectifier
MOSFET. However, for high-current applications, some
combinations of high-and low-side FETS may cause
excessive gate-drain coupling, which can lead to poor
efficiency, EMI, and shoot-through currents.
An adaptive dead-time circuit monitors the DL output
and prevents the high-side FET from turning on until DL
is fully turned off. The dead time at the other edge (DH
turning off) is determined by a fixed 35ns (typ) internal
delay.
sense voltage that appears at CS (Figure 8). Keep the
impedance at this mode low to avoid errors at CS.
Low-Side Current-Limit Sensing (ILIM)
The current-limit circuit employs a unique “valley” current-sensing algorithm that uses the on-state resistance
of the low-side MOSFET as a current-sensing element.
If the current-sense signal is below the current-limit
threshold (-100mV from CS to GND), the PWM is not
allowed to initiate a new cycle (Figure 7). The actual
peak current is greater than the current-limit threshold
by an amount equal to the inductor ripple current.
Therefore, the exact current-limit characteristic and
maximum load capability are a function of the MOSFET
on-resistance, inductor value, and battery voltage.
If greater current-limit accuracy is desired, CS must be
connected to the junction of the low-side switch source
and a current-sense resistor to GND. The current limit
will be 0.1V/RSENSE, and the accuracy will be ±10%.
A resistive voltage-divider from the inductor’s switching
mode to ground can be used to adjust the current-limit
POR and Soft-Start
Power-on reset (POR) occurs when VBATT rises above
approximately 2V, resetting the fault latch and soft-start
counter and preparing the PWM for operation. UVLO
circuitry inhibits switching until VVP rises above 4.1V,
whereupon an internal digital soft-start timer begins to
ramp up the maximum allowed current limit. The ramp
occurs in five steps: 20%, 40%, 60%, 80%, and 100%;
100% current is available after approximately 1.7ms.
Output Undervoltage Protection
The output UVLO function is similar to foldback current
limiting but employs a timer rather than a variable current limit. The output undervoltage protection is
enabled 20ms after POR or when coming out of shutdown. If the output is under 70% of the nominal value,
VP
DH
1.0kΩ
MAX1762
MAX1791
VOUT
CS
1.0kΩ
DL
Figure 8. Using a Resistive Voltage-Divider to Adjust CurrentLimit Sense Voltage to 200mV
______________________________________________________________________________________
13
MAX1762/MAX1791
IPEAK
∆i
VBATT - VOUT
=
∆t
L
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
then the PWM is latched off and does not restart until
VP power is cycled, or SHDN is toggled low then high.
Design Procedure
Begin by establishing the input voltage range and maximum load current before choosing an inductor and its
associated ripple-current ratio (LIR). The following four
factors dictate the rest of the design:
1) Input voltage range. The maximum value (V VP
(MAX) ) must accommodate the maximum AC
adapter voltage. The minimum value (VVP(MIN) )
must account for the lowest input voltage after
drops due to connectors, fuses, and battery selector switches. If there is a choice at all, lower input
voltages result in better efficiency.
2) Maximum load current. There are two values to
consider. The peak load current (ILOAD(MAX)) determines the instantaneous component stress and filtering requirements and thus drives output
capacitor selection, inductor saturation rating, and
the design of the current-limit circuit. The continuous load current (ILOAD) determines the thermal
stress and thus drives the selection of input capacitors, MOSFETs, and other critical heat-contributing
components. Modern notebook CPUs generally
exhibit, ILOAD = ILOAD(MAX) x 0.8.
3) Switching frequency. The MAX1762/MAX1791
have a nominal switching frequency of 300kHz.
4) Inductor ripple-current ratio (LIR). LIR is the ratio
of the peak-to-peak ripple current to the average
inductor current. Size and efficiency trade-offs must
be considered when setting the inductor ripple-current ratio. Low inductor values cause large ripple
currents, resulting in the smallest size but poor efficiency and high output noise. The minimum practical inductor value is one that causes the circuit to
operate at critical conduction (where the inductor
current just touches zero with every cycle). Inductor
values lower than this grant no further size-reduction benefit.
The MAX1762/MAX1791s’ pulse-skipping algorithm initiates skip mode at the critical conduction point. So, the
inductor operating point also determines the load-current value at which switchover occurs. The optimum
point is usually found between 20% and 50% ripple
current.
The inductor ripple current also impacts transientresponse performance, especially at low VVP - VOUT
difference. Low inductor values allow the inductor current to slew faster, replenishing charge removed from
the output filter capacitors by a sudden load step. The
14
peak amplitude of the output transient (VSAG) is also a
function of the maximum duty factor, which can be calculated from the on-time and minimum off-time:
VSAG =
 V

(∆ILOAD(MAX) )2 × L  K OUT + t OFF(MIN) 
 VVP

 V -V


2 × COUT × VOUT K  VP OUT  - t OFF(MIN) 
 

VVP



where minimum off-time = 0.5µs (max).
Inductor Selection
The switching frequency (on-time) and operating point
(% ripple or LIR) determine the inductor value as follows:
L=
VOUT (VVP - VOUT )
VVP × ƒ × LIR × ILOAD(MAX)
Example: ILOAD(MAX) = 2A, VVP = 7V, VOUT = 1.6V, f =
300kHz, 35% ripple current or LIR = 0.35:
L=
1.6V(7V -1.6V)
= 5.9µH
7 × 300kHz × 0.35 × 2A
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice. The core must be large
enough not to saturate at the peak inductor current
(IPEAK):
IPEAK = ILOAD(MAX) + [(LIR/2) ✕ ILOAD(MAX)]
Determining Current Limit
The minimum current-limit threshold must be great
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The valley of the inductor current occurs at ILOAD(MAX) minus
half of the ripple current; therefore:
IVALLEY > ILOAD(MAX) - [(LIR/2) ✕ ILOAD(MAX)]
where IVALLEY = minimum current-limit threshold voltage divided by the RDS(ON) of Q2. For the MAX1762/
MAX1791, the minimum current-limit threshold is 90mV.
Use the worst-case maximum value for RDS(ON) from
the MOSFET Q2 data sheet, and add some margin for
the rise in RDS(ON) with temperature. A good general
rule is to allow 0.5% additional resistance for each °C of
temperature rise.
______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
IVALLEY = 90mV / 52mΩ = 1.73A
Checking the corresponding ILOAD(MAX) reveals:
once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge
is no longer a problem (see the VSAG equation in the
Design Procedure section).
The amount of overshoot due to stored inductor energy
can be calculated as:
1.73A
I
ILOAD(MAX) = VALLEY =
= 2.1A
1 - 0.5 LIR 1 - 0.5 × 0.35
A current-sense resistor can be connected from CS to
GND to set the current limit for the device. The
MAX1762/MAX1791 use the sense resistor instead of
the RDS(ON) of Q2 to limit the current. The maximum
value of the sense resistor can be calculated with the
equation:
ILIMIT = 90mV / RSENSE
Output Capacitor Selection
The output filter capacitor must have low enough effective series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements. In CPU VCORE converters and other applications where the output is subject
to large load transients, the output capacitor’s size
depends on how much ESR is needed to prevent the
output from dipping too low under a load transient.
Ignoring the sag due to finite capacitance:
RESR ≤
VDIP
ƒILOAD(MAX)
where VDIP is the maximum tolerable transient voltage
drop. In non-CPU applications, the output capacitor’s
size depends on how much ESR is needed to maintain
an acceptable level of output voltage ripple:
RESR ≤
VP-P
LIR × ILOAD(MAX)
where VP-P is the peak-to-peak output voltage ripple.
The actual microfarad capacitance value required
relates to the physical size needed to achieve low ESR,
as well as to the chemistry of the capacitor technology.
Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value (this is true
of tantalum, SP, POS, and other electrolytic-type
capacitors).
When using low-capacity filter capacitors such as
ceramics, capacitor size is usually determined by the
capacity needed to prevent V SAG and V SOAR from
causing problems during load transients. Generally,
∆V ≤
LIPEAK 2
2CVOUT
where IPEAK is the peak inductor current.
Stability Considerations
Stability is determined by the value of the ESR zero
(fESR) relative to the switching frequency (f). The point
of instability is given by the following equation:
ƒ ESR ≤
ƒ
π
where:
ƒ ESR ≤
1
2 × π × RESR × COUT
For a typical 300kHz application, the ESR zero frequency must be well below 95kHz, preferably below 50kHz.
Tantalum, Sanyo POSCAP, and Panasonic SP capacitors in widespread use at the time of publication have
typical ESR zero frequencies of 20kHz. In the design
example used for inductor selection, the ESR needed
to support a specified ripple voltage is found by the
equation:
RESR =
VRIPPLE(p-p)
ƒLIR × ILOAD
where LIR is the inductor ripple current ratio, and ILOAD
is the average DC load. Using a LIR = 0.35 and an
average load current of 2A, the ESR needed to support
50mVP-P ripple is 71mΩ.
Do not use high-value ceramic capacitors directly
across the fast feedback inputs (FB to GND) without
taking precautions to ensure stability. Large ceramic
capacitors can have a high-ESR zero frequency and
cause erratic, unstable operation. However, it is easy to
add enough series resistance by placing the capacitors a couple of inches downstream from the junction of
the inductor and FB pin.
Unstable operation manifests itself in two related but distinctly different ways: double-pulsing and fast-feedback
loop instability. Double pulsing occurs due to noise on
the output or because the ESR is so low that there is not
enough voltage ramp in the output voltage signal. This
______________________________________________________________________________________
15
MAX1762/MAX1791
Examining the 2A circuit example with a maximum
RDS(ON) = 52mΩ at +85°C temperature reveals the following:
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
“fools” the error comparator into triggering a new cycle
immediately after the 500ns minimum off-time period has
expired. Double pulsing is more annoying than harmful,
resulting in nothing worse than increased output ripple.
However, it can indicate the possible presence of loop
instability, which is caused by insufficient ESR. Loop
instability can result in oscillations at the output after line
or load perturbations that can cause the output voltage
to fall below the tolerance limit.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient (refer to the
MAX1762/MAX1791 EV kit manual) and carefully
observe the output voltage ripple envelope for overshoot and ringing. It can help to simultaneously monitor
the inductor current with an AC current probe. Do not
allow more than one cycle of ringing after the initial
step-response under- or overshoot.
Input Capacitor Selection
The input capacitor must meet the ripple-current requirement (I RMS ) imposed by the switching currents.
Nontantalum chemistries (ceramic or OS-CON™) are preferred due to their resilience to power-up surge currents:
 VOUT (VVP - VOUT ) 
IRMS = ILOAD × 

VVP


Power MOSFET Selection
DC bias and output power considerations dominate the
selection of the power MOSFETs used with the
MAX1762/MAX1791. Take care not to exceed the
device’s maximum voltage ratings. In general, both
switches are exposed to the supply voltage, so select
MOSFETs with VDS (max) greater than VP (max). Gate
drives to the n-channel and p-channel MOSFETs are
not symmetrical. The n-channel device is driven from
ground to the logic supply VL, while the p-channel
device is driven from VP to ground. The maximum rating for VGS for the n-channel device is usually not an
issue; however, VGS (max) for the p-channel must be at
least VP (max). Since VGS (max) is usually lower than
V DS (max), gate drive constraints often dictate the
required p-channel breakdown rating.
For moderate input-to-output differentials, the high-side
MOSFET (Q1) can be sized smaller than the low-side
MOSFET (Q2) without compromising efficiency. The
high-side switch operates at a very low duty cycle
under these conditions, so most conduction losses
occur in Q2. For maximum efficiency, choose a highside MOSFET (Q1) that has conduction losses (I2R x
OS-CON is a trademark of Sanyo.
16
duty cycle) equal to the switching losses (CV VP 2f).
Make sure that the conduction losses at the minimum
input voltage do not exceed the package thermal limits
or violate the overall thermal budget. Conduction losses
plus switching losses at the maximum input voltage
should not exceed the package ratings or violate the
overall thermal budget (see MOSFET Power Dissipation).
In addition to efficiency considerations, the selection of
the RDS(ON) of the low-side MOSFET must account for
the regulator’s required current limit. Choose a MOSFET that has a low enough resistance over the operating temperature range such that the device does not
enter current limit during normal operation (see the
Determining Current Limit section). Conversely, ultralow RDS(ON) devices may set the current limit too high
and may result in only incremental improvements in efficiency. Some large n-channel FETs also have substantial interelectrode capacitance. Verify that the
MAX1762/ MAX1791 DL driver can hold the gate off
when the high side switch turns on. Cross-conduction
problems can occur when the high-side switch turns on
due to coupling through the n-channel’s parasitic drainto-gate capacitance.
The MAX1762/MAX1791 have adaptive dead-time circuitry that prevents the high-side and low-side
MOSFETs from conducting at the same time (see MOSFET Gate Drivers). Even with this protection, it is still
possible for delays internal to the MOSFET to prevent
one MOSFET from turning off while the other is turned
on. The maximum mismatch time that can be tolerated
is 60ns. Select devices that have low turn-off times, and
make sure that NFET(tD(off,max)) - PFET(tD(on,min)) <
60ns, and PFET(tD(off,max)) - NFET(tD(on,min)) < 60ns.
Failure to do so may result in efficiency-killing shootthrough currents.
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty factor
extremes. For the high-side MOSFET, the worst-case
power dissipation (PD) due to resistance occurs at minimum battery voltage:
 V

PD(Q1 resistance) =  OUT  × ILOAD2 × RDS(ON)
V

 VP(MIN) 
Generally, a small high-side MOSFET is desired to
reduce switching losses at high input voltage. However,
the RDS(ON) required to stay within package power-dissipation limits often limits how small the MOSFET can
be. Again, the optimum occurs when the switching (AC)
losses equal the conduction (RDS(ON)) losses. High-
______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum battery voltage is applied, due to the squared term in the CV2f
switching loss equation. If the high-side MOSFET chosen for adequate R DS(ON) at low battery voltages
becomes extraordinarily hot when subjected to
VVP(MAX), reconsider your choice of high-side MOSFET.
Calculating the power dissipation in Q1 due to switching losses is difficult since it must allow for difficult
quantifying factors that influence the turn-on and turnoff times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PC board layout characteristics. The following switching loss calculation provides only a very
rough estimate and is no substitute for breadboard
evaluation, preferably including a verification using a
thermocouple mounted on Q1:
C

RSS × VVP(MAX)2 × ƒ × ILOAD
PD (Q1 switching) = 



IGATE


unchanged, this voltage is approximately 0.7V (a diode
drop) at both transition edges while both switches are
off. In between the edges, the low-side switch conducts; the drop is IL ✕ RDS(ON). If a Schottky clamp is
connected across the low-side switch, the initial and
final voltage drops is reduced, improving efficiency
slightly.
Choose a Schottky diode (D1) having a forward voltage
low enough to prevent the Q2 MOSFET body diode
from turning on during the dead time. As a general rule,
a diode having a DC current rating equal to 1/3 of the
load current is sufficient. This diode is optional and can
be removed if efficiency isn’t critical.
Applications Issues
Dropout Performance
where CRSS is the reverse transfer capacitance of Q1,
and IGATE is the peak gate-drive source/sink current.
The output voltage adjust range for continuous-conduction operation is restricted by the nonadjustable 500ns
(max) minimum off-time one-shot. When working with
low input voltages, the duty-factor limit must be calculated using worst-case values for on- and off-times.
Manufacturing tolerances and internal propagation
delays introduce an error to the t ON K-factor. Also,
keep in mind that transient response performance of
buck regulators operating close to dropout is poor, and
bulk output capacitance must often be added.
For the low-side MOSFET, the worst-case power dissipation always occurs at maximum battery voltage:
Dropout design example: VIN = 7V (min), VOUT = 5V, f
= 300kHz. The required duty cycle is :

VOUT 
PD(Q2) = 1  × ILOAD2 × RDS


V
VP(MAX) 

V
+ VSW 5V + 0.1V
DCREQ = OUT
=
= 0.74
VVP - VSW
7V - 0.1V
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
ILOAD(MAX) but are not quite high enough to exceed
the current limit and cause the fault latch to trip. To protect against this possibility, the circuit must be overdesigned to tolerate:
ILOAD = ILIMIT(HIGH) + (LIR / 2 ) ✕ ILOAD(MAX)
where I LIMIT(HIGH) is the maximum valley current
allowed by the current-limit circuit, including threshold
tolerance and on-resistance variation. This means that
the MOSFET must be very well heatsinked. If short-circuit protection without overload protection is enough, a
normal ILOAD value can be used for calculating component stresses.
During the period when the high-side switch is off, current circulates from ground to the junction of both FETs
and the inductor. As a consequence, the polarity of the
switching node is negative with respect to ground. If
The worst-case on-time is:
V
+ 0.075
5V + 0.075
t ON(MIN) = OUT
×K=
×
VVP
7V
3.35µs × 90% = 2.18µs
The maximum IC duty factor based on timing constraints of the MAX1762/MAX1792 is:
Duty =
t ON(MIN)
2.18µs
=
= 0.82,
t ON(MIN) + t OFF(MAX) 2.18µs + 0.5µs
which meets the required duty cycle. Remember to
include inductor resistance and MOSFET on-state voltage drops (VSW) when doing worst-case dropout dutyfactor calculations.
Fixed Output Voltages
The MAX1762/MAX1791 Dual Mode operation allows
the selection of common voltages without requiring
external components (Figure 9). Connect FB to GND for
______________________________________________________________________________________
17
MAX1762/MAX1791
side switching losses do not usually become an issue
until the input is greater than approximately 15V.
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
OUT
FIXED
1.8V
TO ERROR
AMP
FIXED
3.3V
(Figure 10). Refer to the MAX1791 EV kit manual for a
specific layout example.
If possible, mount all of the power components on the
top side of the board, with their ground terminals flush
against one another. Follow these guidelines for good
PC board layout:
•
Isolate the power components on the top side from
the sensitive analog components on the bottom
side with a ground shield. Use a separate GND
plane under OUT. Avoid the introduction of AC currents into the GND ground planes. Run the power
plane ground currents on the top side only, if possible.
•
Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable, jitter-free operation.
•
Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PC boards (2oz vs. 1oz) can enhance
full-load efficiency by 1% or more. Correctly routing
PC board traces is a difficult task that must be
approached in terms of fractions of centimeters,
where a single milliohm of excess trace resistance
causes a measurable efficiency penalty.
•
Inductor and GND connections to the synchronous
rectifiers for current limiting must be made using
Kelvin sensed connections to guarantee the current-limit accuracy. With 8-pin SO MOSFETs, this is
best done by routing power to the MOSFETs from
outside using the top copper layer, while connecting GND and CS inside (underneath) the µMAX
package.
•
When trade-offs in trace lengths must be made, it’s
preferable to allow the inductor charging path to be
made longer than the discharge path. For example,
it’s better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
allow distance between the inductor and the lowside MOSFET or between the inductor and the output filter capacitor.
•
Ensure that the OUT connection to COUT is short
and direct. However, in some cases it may be desirable to deliberately introduce some trace length
between the OUT connector node and the output
filter capacitor (see Stability Considerations).
•
Route high-speed switching nodes (CS, DH, and
DL) away from sensitive analog areas (FB). Use
GND as an EMI shield to keep radiated switching
noise away from the IC’s feedback divider and analog bypass capacitors.
FB
0.150V
MAX1762
2.5V
Figure 9. Feedback MUX
a fixed +1.8V (MAX1762) or 3.3V (MAX1791) output.
Connect FB to VL for a fixed 2.5V (MAX1762) or 5.0V
(MAX1791) output. Otherwise, connect FB to a resistive
voltage-divider for an adjustable output.
Setting the Output Voltage
Select VOUT > 1.25V for the MAX1762/MAX1791 by
connecting FB to a resistive voltage-divider between
V OUT and GND (Figure 2). Choose R2 to be about
10kΩ, and solve for R1 using the equation:
 R1 
VOUT = V ƒFB × 1+

 R2 
where VFB = 1.25V. For a VOUT = 3.0V, R2 = 10kΩ and
R1 = 14kΩ.
For a desired VOUT < 1.25V, connect FB to a resistive
voltage-divider between REF and OUT (Figure 3).
Choose R1 to be about 50kΩ, and solve for R2 using
the equation:
V
-V 
R2 =  OUT FB  × R1
V
V
 FB REF 
where VFB = 1.25V and VREF = 2.0V. For a V OUT =
1.0V, R1 = 50kΩ and R2 = 16.5kΩ. Under these conditions, a minimum load of VREF - V FB / R1 >15µA is
required.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low
switching losses and clean, stable operation. This is
especially true when multiple converters are on the
same PC board where one circuit can affect the other.
The switching power stages require particular attention
18
______________________________________________________________________________________
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
MAX1762/MAX1791
USE AGND PLANE TO:
USE PGND PLANE TO:
- BYPASS VCC AND REF
- BYPASS VVP
- TERMINATE EXTERNAL FB
- CONNECT PGND TO THE TOPSIDE STAR GROUND
DIVIDER (IF USED)
- PIN-STRAP CONTROL
INPUTS
AGND
VOUT
L1
PGND
C2
D1
P1
VL
N1
VIA TO GROUND
GND
VBATT
C1
CONNECT PGND TO AGND
BENEATH THE MAX1762/MAX1791 AT
ONE POINT ONLY AS SHOWN.
NOTE: EXAMPLE SHOWN IS FOR DUAL n-CHANNEL MOSFET.
Figure 10. PC Board Layout Example
Layout Procedure
1) Place the power components first, with ground terminals adjacent (Q1 source, CIN, COUT). If possible, make all these connections on the top layer
with wide, copper-filled areas.
2) Mount the controller IC adjacent to the synchronous-rectifier MOSFETs, preferably on the back
side in order to keep CS, GND, and the DL gate
drive lines short and wide. The DL gate trace must
be short and wide (measuring 50mils to 100mils
wide if the MOSFET is 1in from the controller IC).
sides. The top-side star ground is a star connection
of the input capacitors, side 1 low-side MOSFET.
Keep the resistance low between the star ground
and the source of the low-side MOSFETs for accurate current limit. Connect the top-side star ground
(used for MOSFET, input, and output capacitors) to
the small island with a single short, wide connection
(preferably just a via).
6) Connect the output power planes directly to the output filter capacitor positive and negative terminals
with multiple vias.
3) Place the VL bypass capacitor near the controller
IC.
4) Make the DC-DC controller ground connections as
follows: Near the IC, create a small analog ground
plane. Connect this plane to GND, and use this
plane for the ground connection for the REF and
VVP bypass capacitors and FB dividers.
Chip Information
TRANSISTOR COUNT: 3520
PROCESS: S8E1FP
5) On the board’s top side (power planes), make a
star ground to minimize crosstalk between the two
______________________________________________________________________________________
19
Package Information
e
10LUMAX.EPS
MAX1762/MAX1791
High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks
4X S
10
10
INCHES
H
Ø0.50±0.1
0.6±0.1
1
1
0.6±0.1
BOTTOM VIEW
TOP VIEW
D2
MILLIMETERS
MAX
DIM MIN
0.043
A
0.006
A1
0.002
A2
0.030
0.037
0.120
D1
0.116
0.118
0.114
D2
0.116
0.120
E1
0.118
E2
0.114
0.199
H
0.187
L
0.0157 0.0275
L1
0.037 REF
b
0.007
0.0106
e
0.0197 BSC
c
0.0035 0.0078
0.0196 REF
S
α
0°
6°
MAX
MIN
1.10
0.15
0.05
0.75
0.95
3.05
2.95
3.00
2.89
3.05
2.95
2.89
3.00
4.75
5.05
0.40
0.70
0.940 REF
0.177
0.270
0.500 BSC
0.090
0.200
0.498 REF
0°
6°
E2
GAGE PLANE
A2
c
A
b
A1
α
E1
D1
L
L1
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL
DOCUMENT CONTROL NO.
21-0061
REV.
1
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
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© 2005 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products, Inc.