Micrel MIC4423ZN Dual 3a-peak low-side mosfet driver Datasheet

MIC4423/4424/4425
Micrel, Inc.
MIC4423/4424/4425
Dual 3A-Peak Low-Side MOSFET Driver
Bipolar/CMOS/DMOS Process
General Description
Features
The MIC4423/4424/4425 family are highly reliable BiCMOS/
DMOS buffer/driver/MOSFET drivers. They are higher output current versions of the MIC4426/4427/4428, which are
improved versions of the MIC426/427/428. All three families
are pin-compatible. The MIC4423/4424/4425 drivers are capable of giving reliable service in more demanding electrical
environments than their predecessors. They will not latch
under any conditions within their power and voltage ratings.
They can survive up to 5V of noise spiking, of either polarity,
on the ground pin. They can accept, without either damage
or logic upset, up to half an amp of reverse current (either
polarity) forced back into their outputs.
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The MIC4423/4424/4425 series drivers are easier to use, more
flexible in operation, and more forgiving than other CMOS
or bipolar drivers currently available. Their BiCMOS/DMOS
construction dissipates minimum power and provides rail-torail voltage swings.
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Reliable, low-power bipolar/CMOS/DMOS construction
Latch-up protected to >500mA reverse current
Logic input withstands swing to –5V
High 3A-peak output current
Wide 4.5V to 18V operating range
Drives 1800pF capacitance in 25ns
Short <40ns typical delay time
Delay times consistent with in supply voltage change
Matched rise and fall times
TTL logic input independent of supply voltage
Low equivalent 6pF input capacitance
Low supply current
3.5mA with logic-1 input
350µA with logic-0 input
Low 3.5Ω typical output impedance
Output voltage swings within 25mV of ground or VS.
‘426/7/8-, ‘1426/7/8-, ‘4426/7/8-compatible pinout
Inverting, noninverting, and differential configurations
Primarily intended for driving power MOSFETs, the
MIC4423/4424/4425 drivers are suitable for driving other loads
(capacitive, resistive, or inductive) which require low-impedance, high peak currents, and fast switching times. Heavily
loaded clock lines, coaxial cables, or piezoelectric transducers
are some examples. The only known limitation on loading is
that total power dissipated in the driver must be kept within
the maximum power dissipation limits of the package.
Note: See MIC4123/4124/4125 for high power and narrow
pulse applications.
Functional Diagram
VS
0.1mA
0.6mA
Integrated Component Count:
4 Resistors
4 Capacitors
52 Transistors
IN V E R T I N G
OUTA
INA
2kΩ
NONINVERTING
0.1mA
0.6mA
IN V E R T I N G
OUTB
INB
2kΩ
NONINVERTING
GND
Ground Unused Inputs
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
July 2005
1
MIC4423/4424/4425
MIC4423/4424/4425
Micrel, Inc.
Ordering Information
Part Number
Temperatre
Range
Package
Configuration
Standard
Pb-Free
MIC4423CWM
MIC4423ZWM
0°C to +70°C
16-pin Wide SOIC
Dual Inverting
MIC4423BWM
MIC4423YWM
–40°C to +85°C
16-pin Wide SOIC
Dual Inverting
MIC4423BM
MIC4423YM
–40°C to +85°C
8-pin SOIC
Dual Inverting
MIC4423CN
MIC4423ZN
0°C to +70°C
8-pin Plastic DIP
Dual Inverting
MIC4423BN
MIC4423YN
–40°C to +85°C
8-pin Plastic DIP
Dual Inverting
MIC4424CWM
MIC4424ZWM
0°C to +70°C
16-pin Wide SOIC
Dual Non-Inverting
MIC4424BWM
MIC4424YWM
–40°C to +85°C
16-pin Wide SOIC
Dual Non-Inverting
MIC4424BM
MIC4424YM
–40°C to +85°C
8-pin SOIC
Dual Non-Inverting
MIC4424CN
MIC4424ZN
0°C to +70°C
8-pin Plastic DIP
Dual Non-Inverting
MIC4424BN
MIC4424YN
–40°C to +85°C
8-pin Plastic DIP
Dual Non-Inverting
MIC4425CWM
MIC4425ZWM
0°C to +70°C
16-pin Wide SOIC
Inverting + Non-Inverting
MIC4425BWM
MIC4425YWM
–40°C to +85°C
16-pin Wide SOIC
Inverting + Non-Inverting
MIC4425BM
MIC4425YM
–40°C to +85°C
8-pin SOIC
Inverting + Non-Inverting
MIC4425CN
Contact Factory
0°C to +70°C
8-pin Plastic DIP
Inverting + Non-Inverting
MIC4425BN
MIC4425YN
–40°C to +85°C
8-pin Plastic DIP
Inverting + Non-Inverting
Pin Configuration
Driver Configuration
MIC4423xN/M
NC 1
MIC4423xWM
8 NC
INA 2
7 OUTA
GND 3
6 VS
INB 4
5 OUTB
INA 2
WM Package Note:
Duplicate GND, VS, OUTA,
INB 4
and OUTB pins must be
externally connected together.
8-pin DIP (N)
8-pin SOIC (M)
A
7 OUTA
INA 2
A
B
5 OUTB
INB 7
B
MIC4424xN/M
NC 1
16 N C
INA
2
15 OUTA
NC 3
14 OUTA
GND 4
13 V S
GND 5
12 V S
NC 6
11 O U T B
INB 7
10 O U T B
A
7 OUTA
INA 2
A
INB 4
B
5 OUTB
INB 7
B
10 OUTB
11 OUTB
14 OUTA
15 OUTA
10 OUTB
11 OUTB
MIC4423xWM
INA 2
A
7 OUTA
INA 2
A
INB 4
B
5 OUTB
INB 7
B
9 NC
NC 8
15 OUTA
MIC4423xWM
INA 2
MIC4425xN/M
14 OUTA
16-pin Wide SOIC (WM)
14 OUTA
15 OUTA
10 OUTB
11 OUTB
Pin Description
Pin Number
DIP, SOIC
Pin Number
Wide SOIC
Pin Name
Pin Function
2/4
2/7
INA/B
Control Input
3
4, 5
GND
Ground: Duplicate pins must be externally connected together.
6
12, 13
VS
7/5
14, 15 / 10, 11
OUTA/B
1, 8
1, 3, 6, 8, 9, 16
NC
MIC4423/4424/4425
Supply Input: Duplicate pins must be externally connected together.
Output: Duplicate pins must be externally connected together.
not connected
2
July 2005
MIC4423/4424/4425
Micrel, Inc.
Absolute Maximum Ratings (Note 1)
Operating Ratings (Note 2)
Supply Voltage ........................................................... +22V
Input Voltage .................................. VS + 0.3V to GND – 5V
Junction Temperature ................................................150°C
Storage Temperature Range ......................–65°C to 150°C
Lead Temperature (10 sec.) ......................................300°C
ESD Susceptability, Note 3 ..................................... 1000V
Supply Voltage (VS) ......................................+4.5V to +18V
Temperature Range
C Version ....................................................0°C to +70°C
B Version ................................................–40°C to +85°C
Package Thermal Resistance
DIP θJA ............................................................. 130°C/W
DIP θJC ............................................................... 42°C/W
Wide-SOIC θJA ................................................. 120°C/W
Wide-SOIC θJC ................................................... 75°C/W
SOIC θJA ........................................................... 120°C/W
SOIC θJC ............................................................ 75°C/W
MIC4423/4424/4425 Electrical Characteristics (Note 5)
4.5V ≤ VS ≤ 18V; TA = 25°C, bold values indicate –40°C ≤ TA ≤ +85°C; unless noted.
Symbol
Parameter
Conditions
Min
Typ
Max
Units
Input
VIH
Logic 1 Input Voltage
VIL
Logic 0 Input Voltage
IIN
Input Current
2.4
0V ≤ VIN ≤ VS
V
–1
–10
0.8
V
1
10
µA
µA
Output
VOH
High Output Voltage
VOL
Low Output Voltage
RO
Output Resistance HI State
Output Resistance LO State
IPK
Peak Output Current
I
Latch-Up Protection
Withstand Reverse Current
VS–0.025
V
0.025
V
IOUT = 10mA, VS = 18V
2.8
5
Ω
VIN = 0.8V, IOUT = 10mA, VS = 18V
3.7
8
Ω
IOUT = 10mA, VS = 18V
3.5
5
Ω
VIN = 2.4V, IOUT = 10mA, VS = 18V
4.3
8
Ω
3
A
>500
mA
Switching Time (Note 4)
tR
Rise Time
test Figure 1, CL = 1800pF
23
28
35
60
ns
ns
tF
Fall Time
test Figure 1, CL = 1800pF
25
32
35
60
ns
ns
tD1
Delay Tlme
test Ffigure 1, CL = 1800pF
33
32
75
100
ns
ns
tD2
Delay Time
test Figure 1, CL = 1800pF
38
38
75
100
ns
ns
tPW
Pulse Width
test Figure 1
IS
Power Supply Current
VIN = 3.0V (both inputs)
1.5
2
2.5
3.5
mA
mA
IS
Power Supply Current
VIN = 0.0V (both inputs)
0.15
0.2
0.25
0.3
mA
mA
400
ns
Power Supply
Note 1.
Note 2.
Note 3.
Note 4.
Note 5.
Exceeding the absolute maximum rating may damage the device.
The device is not guaranteed to function outside its operating rating.
Devices are ESD sensitive. Handling precautions recommended. ESD tested to human body model, 1.5k in series with 100pF.
Switching times guaranteed by design.
Specification for packaged product only.
July 2005
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MIC4423/4424/4425
MIC4423/4424/4425
Micrel, Inc.
Test Circuit
VS = 18V
VS = 18V
0.1µF
A
INA
MIC4423
B
INB
INPUT
5V
90%
0.1µF
OUTA
1800pF
INA
OUTB
1800pF
INB
tD1
tP W
tF
tD2
INPUT
tR
4.7µF
OUTA
1800pF
OUTB
1800pF
B
5V
90%
2.5V
tP W ≥ 0.5µs
10%
0V
VS
90%
tD1
tP W
tR
tD2
tF
O U TPU T
O U TPU T
10%
0V
10%
0V
Figure 1b. Noninverting Driver Switching Time
Figure 1a. Inverting Driver Switching Time
MIC4423/4424/4425
A
MIC4424
2.5V
tP W ≥ 0.5µs
10%
0V
VS
90%
4.7µF
4
July 2005
MIC4423/4424/4425
Micrel, Inc.
Typical Characteristic Curves
100
3300pF
60
80
1800pF
2200pF
40
20
10 12 14
VSUPPLY (V)
1000
CLOAD (pF)
500kHz
20kHz
100kHz
Supply Current
vs. Frequency
10000
ISUPPLY (mA)
100
VSUPPLY = 12V
90
80
10000pF
70
60
50
1000pF
40
100pF
3300pF
30
20
10
0
10
100
1000
FREQUENCY (kHz)
July 2005
Rise and Fall Time
vs. Temperature
VS = 18V
CLOAD = 1800pF
T
R
1000
CLOAD (pF)
10000
Propagation Delay vs.
Input Amplitude
50
VS = 18V
CLOAD = 1800pF
40
TF
TD2
30
TD1
20
10
Supply Current
vs. Frequency
VSUPPLY = 18V
1000
CLOAD (pF)
0
100
18
0
-75
-30
15
60
105 150
JUNCTION TEMPERATURE (˚C)
10000
Supply Current vs.
Capacitive Load
40
30
20
10
0
100
16
10
18V
50
10 12 14
VSUPPLY (V)
100
VSUPPLY = 18V
90
80
10000pF
70
60
50
1000pF
3300pF
40
100pF
30
20
10
0
10
100
1000
FREQUENCY (kHz)
ISUPPLY (mA)
90
80
70
60
8
20
40
0
100
6
TIME (ns)
30
12V
20
4
40
5V
60
100
0
18
Supply Current vs.
Capacitive Load
100
V
= 5V
90 SUPPLY
80
70
60
2MHz
50
40
30
100kHz
500kHz
20
10
0
100
1000
10000
CLOAD (pF)
ISUPPLY (mA)
TFALL (ns)
80
ISUPPLY (mA)
16
T (ns)
8
Fall Time vs.
Capacitive Load
100
18V
470pF
5
0
100
ISUPPLY (mA)
6
12V
40
90
80
70
60
0
2
4
6
8
INPUT (V)
10
12
Supply Current vs.
Capacitive Load
VSUPPLY = 12V
2MHz
500kHz
50
40
30
20
10
0
100
20kHz
100kHz
1000
CLOAD (pF)
10000
Supply Current
vs. Frequency
100
VSUPPLY = 5V
90
80
70
60
ISUPPLY (mA)
4
5V
60
20
470pF
0
1000pF
2200pF
40
20
80
1800pF
3300pF
60
1000pF
Rise Time
vs. Capacitive Load
100
4700pF
TFALL (ns)
TRISE (ns)
100
4700pF
80
Fall Time vs.
Supply Voltage
TRISE (ns)
Rise Time vs.
Supply Voltage
50
40
30
20
10
0
10
10000pF
4700pF
2200pF
1000pF
100pF
100
FREQUENCY (kHz)
1000
MIC4423/4424/4425
MIC4423/4424/4425
Micrel, Inc.
60
60
CLOAD = 2200 pF
50
30
TD1
20
Quiescent Sypply Current
vs. Voltage
TJ = 25˚C
BOTH INPUTS = 1
1
BOTH INPUTS = 0
0.1
10
4
6
8
10 12 14
VSUPPLY (V)
16
0
-55
18
Quiescent Current
vs. Temperature
6
VS = 10V
1.0
INPUTS = 1
0.8
0.6
0.4
INPUTS = 0
0.2
0
-55
-25 5
35 65 95
TEMPERATURE (˚C)
MIC4423/4424/4425
125
-25 5
35 65 95
TEMPERATURE (˚C)
125
0.01
Output Resistance (Output
High) vs. Supply Voltage
5
4
4
125˚C
25˚C
2
1
0
6
8
10 12 14
VSUPPLY (V)
16
18
6
5
3
4
Output Resistance (Output
Low) vs. Supply Voltage
RDS(ON) (Ω)
IQUIESCENT (mA)
1.2
TD1
20
10
1.4
TD2
T (ns)
TD2
RDS(ON) (Ω)
T (ns)
40
30
10
CLOAD = 2200 pF
50
40
0
Delay Time
vs. Temperature
IQUIESCENT (mA)
Delay Time vs.
Supply Voltage
125˚C
25˚C
3
2
1
4
6
8
10 12 14
VSUPPLY (V)
6
16
18
0
4
6
8
10 12 14
VSUPPLY (V)
16
18
July 2005
MIC4423/4424/4425
Micrel, Inc.
Application Information
requires attention to the ground path. Two things other than
the driver affect the rate at which it is possible to turn a load
off: The adequacy of the grounding available for the driver,
and the inductance of the leads from the driver to the load.
The latter will be discussed in a separate section.
Although the MIC4423/24/25 drivers have been specifically
constructed to operate reliably under any practical circumstances, there are nonetheless details of usage which will
provide better operation of the device.
Best practice for a ground path is obviously a well laid out
ground plane. However, this is not always practical, and a
poorly-laid out ground plane can be worse than none. Attention
to the paths taken by return currents even in a ground plane
is essential. In general, the leads from the driver to its load,
the driver to the power supply, and the driver to whatever is
driving it should all be as low in resistance and inductance
as possible. Of the three paths, the ground lead from the
driver to the logic driving it is most sensitive to resistance or
inductance, and ground current from the load are what is most
likely to cause disruption. Thus, these ground paths should
be arranged so that they never share a land, or do so for as
short a distance as is practical.
Supply Bypassing
Charging and discharging large capacitive loads quickly
requires large currents. For example, charging 2000pF from
0 to 15 volts in 20ns requires a constant current of 1.5A. In
practice, the charging current is not constant, and will usually
peak at around 3A. In order to charge the capacitor, the driver
must be capable of drawing this much current, this quickly,
from the system power supply. In turn, this means that as far
as the driver is concerned, the system power supply, as seen
by the driver, must have a VERY low impedance.
As a practical matter, this means that the power supply bus
must be capacitively bypassed at the driver with at least
100X the load capacitance in order to achieve optimum
driving speed. It also implies that the bypassing capacitor
must have very low internal inductance and resistance at
all frequencies of interest. Generally, this means using two
capacitors, one a high-performance low ESR film, the other
a low internal resistance ceramic, as together the valleys in
their two impedance curves allow adequate performance over
a broad enough band to get the job done. PLEASE NOTE
that many film capacitors can be sufficiently inductive as to
be useless for this service. Likewise, many multilayer ceramic
capacitors have unacceptably high internal resistance. Use
capacitors intended for high pulse current service (in-house
we use WIMA™ film capacitors and AVX Ramguard™ ceramics; several other manufacturers of equivalent devices also
exist). The high pulse current demands of capacitive drivers
also mean that the bypass capacitors must be mounted
very close to the driver in order to prevent the effects of lead
inductance or PCB land inductance from nullifying what you
are trying to accomplish. For optimum results the sum of the
lengths of the leads and the lands from the capacitor body to
the driver body should total 2.5cm or less.
To illustrate what can happen, consider the following: The
inductance of a 2cm long land, 1.59mm (0.062") wide on a
PCB with no ground plane is approximately 45nH. Assuming a dl/dt of 0.3A/ns (which will allow a current of 3A to flow
after 10ns, and is thus slightly slow for our purposes) a voltage of 13.5 Volts will develop along this land in response to
our postulated ∆Ι. For a 1cm land, (approximately 15nH) 4.5
Volts is developed. Either way, anyone using TTL-level input
signals to the driver will find that the response of their driver
has been seriously degraded by a common ground path for
input to and output from the driver of the given dimensions.
Note that this is before accounting for any resistive drops in
the circuit. The resistive drop in a 1.59mm (0.062") land of
2oz. Copper carrying 3A will be about 4mV/cm (10mV/in) at
DC, and the resistance will increase with frequency as skin
effect comes into play.
The problem is most obvious in inverting drivers where the
input and output currents are in phase so that any attempt
to raise the driver’s input voltage (in order to turn the driver’s
load off) is countered by the voltage developed on the common ground path as the driver attempts to do what it was
supposed to. It takes very little common ground path, under
these circumstances, to alter circuit operation drastically.
Bypass capacitance, and its close mounting to the driver serves
two purposes. Not only does it allow optimum performance
from the driver, it minimizes the amount of lead length radiating at high frequency during switching, (due to the large Δ I)
thus minimizing the amount of EMI later available for system
disruption and subsequent cleanup. It should also be noted
that the actual frequency of the EMI produced by a driver is
not the clock frequency at which it is driven, but is related to
the highest rate of change of current produced during switching, a frequency generally one or two orders of magnitude
higher, and thus more difficult to filter if you let it permeate your
system. Good bypassing practice is essential to proper
operation of high speed driver ICs.
Output Lead Inductance
The same descriptions just given for PCB land inductance
apply equally well for the output leads from a driver to its load,
except that commonly the load is located much further away
from the driver than the driver’s ground bus.
Generally, the best way to treat the output lead inductance
problem, when distances greater than 4cm (2") are involved,
requires treating the output leads as a transmission line. Unfortunately, as both the output impedance of the driver and the
input impedance of the MOSFET gate are at least an order of
magnitude lower than the impedance of common coax, using
coax is seldom a cost-effective solution. A twisted pair works
about as well, is generally lower in cost, and allows use of a
wider variety of connectors. The second wire of the twisted
pair should carry common from as close as possible to the
Grounding
Both proper bypassing and proper grounding are necessary
for optimum driver operation. Bypassing capacitance only
allows a driver to turn the load ON. Eventually (except in rare
circumstances) it is also necessary to turn the load OFF. This
July 2005
7
MIC4423/4424/4425
MIC4423/4424/4425
Micrel, Inc.
ible with TTL signals, or with CMOS powered from any supply
voltage between 3V and 15V.
ground pin of the driver directly to the ground terminal of the
load. Do not use a twisted pair where the second wire in the
pair is the output of the other driver, as this will not provide a
complete current path for either driver. Likewise, do not use
a twisted triad with two outputs and a common return unless
both of the loads to be driver are mounted extremely close
to each other, and you can guarantee that they will never be
switching at the same time.
The MIC4423/24/25 drivers can also be driven directly by the
SG1524/25/26/27, TL494/95, TL594/95, NE5560/61/62/68,
TSC170, MIC38C42, and similar switch mode power supply
ICs. By relocating the main switch drive function into the driver
rather than using the somewhat limited drive capabilities of a
PWM IC. The PWM IC runs cooler, which generally improves
its performance and longevity, and the main switches switch
faster, which reduces switching losses and increase system
efficiency.
For output leads on a printed circuit, the general rule is to make
them as short and as wide as possible. The lands should also
be treated as transmission lines: i.e. minimize sharp bends,
or narrowings in the land, as these will cause ringing. For a
rough estimate, on a 1.59mm (0.062") thick G-10 PCB a pair
of opposing lands each 2.36mm (0.093") wide translates to a
characteristic impedance of about 50Ω. Half that width suffices
on a 0.787mm (0.031") thick board. For accurate impedance
matching with a MIC4423/24/25 driver, on a 1.59mm (0.062")
board a land width of 42.75mm (1.683") would be required,
due to the low impedance of the driver and (usually) its load.
This is obviously impractical under most circumstances.
Generally the tradeoff point between lands and wires comes
when lands narrower than 3.18mm (0.125") would be required
on a 1.59mm (0.062") board.
The input protection circuitry of the MIC4423/24/25, in addition to providing 2kV or more of ESD protection, also works to
prevent latchup or logic upset due to ringing or voltage spiking
on the logic input terminal. In most CMOS devices when the
logic input rises above the power supply terminal, or descends
below the ground terminal, the device can be destroyed or
rendered inoperable until the power supply is cycled OFF
and ON. The MIC4423/24/25 drivers have been designed to
prevent this. Input voltages excursions as great as 5V below
ground will not alter the operation of the device. Input excursions above the power supply voltage will result in the excess
voltage being conducted to the power supply terminal of the
IC. Because the excess voltage is simply conducted to the
power terminal, if the input to the driver is left in a high state
when the power supply to the driver is turned off, currents as
high as 30mA can be conducted through the driver from the
input terminal to its power supply terminal. This may overload
the output of whatever is driving the driver, and may cause
other devices that share the driver’s power supply, as well as
the driver, to operate when they are assumed to be off, but
it will not harm the driver itself. Excessive input voltage will
also slow the driver down, and result in much longer internal
propagation delays within the drivers. TD2, for example, may
increase to several hundred nanoseconds. In general, while
the driver will accept this sort of misuse without damage,
proper termination of the line feeding the driver so that line
spiking and ringing are minimized, will always result in faster
and more reliable operation of the device, leave less EMI to
be filtered elsewhere, be less stressful to other components
in the circuit, and leave less chance of unintended modes of
operation.
To obtain minimum delay between the driver and the load, it
is considered best to locate the driver as close as possible to
the load (using adequate bypassing). Using matching transformers at both ends of a piece of coax, or several matched
lengths of coax between the driver and the load, works in
theory, but is not optimum.
Driving at Controlled Rates
Occasionally there are situations where a controlled rise or
fall time (which may be considerably longer than the normal
rise or fall time of the driver’s output) is desired for a load. In
such cases it is still prudent to employ best possible practice
in terms of bypassing, grounding and PCB layout, and then
reduce the switching speed of the load (NOT the driver) by
adding a noninductive series resistor of appropriate value
between the output of the driver and the load. For situations
where only rise or only fall should be slowed, the resistor can
be paralleled with a fast diode so that switching in the other
direction remains fast. Due to the Schmitt-trigger action of the
driver’s input it is not possible to slow the rate of rise (or fall)
of the driver’s input signal to achieve slowing of the output.
Power Dissipation
CMOS circuits usually permit the user to ignore power dissipation. Logic families such as 4000 series and 74Cxxx have
outputs which can only source or sink a few milliamps of current, and even shorting the output of the device to ground or
VCC may not damage the device. CMOS drivers, on the other
hand, are intended to source or sink several Amps of current.
This is necessary in order to drive large capacitive loads at
frequencies into the megahertz range. Package power dissipation of driver ICs can easily be exceeded when driving
large loads at high frequencies. Care must therefore be paid
to device dissipation when operating in this domain.
Input Stage
The input stage of the MIC4423/24/25 consists of a singleMOSFET class A stage with an input capacitance of ≤38pF.
This capacitance represents the maximum load from the
driver that will be seen by its controlling logic. The drain load
on the input MOSFET is a –2mA current source. Thus, the
quiescent current drawn by the driver varies, depending on
the logic state of the input.
Following the input stage is a buffer stage which provides
~400mV of hysteresis for the input, to prevent oscillations
when slowly-changing input signals are used or when noise
is present on the input. Input voltage switching threshold is
approximately 1.5V which makes the driver directly compatMIC4423/4424/4425
The Supply Current vs Frequency and Supply Current vs
Load characteristic curves furnished with this data sheet
aid in estimating power dissipation in the driver. Operating
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However, in this instance the RO required may be either the on
resistance of the driver when its output is in the high state, or
its on resistance when the driver is in the low state, depending
on how the inductor is connected, and this is still only half the
story. For the part of the cycle when the inductor is forcing
current through the driver, dissipation is best described as
frequency, power supply voltage, and load all affect power
dissipation.
Given the power dissipation in the device, and the thermal
resistance of the package, junction operating temperature
for any ambient is easy to calculate. For example, the thermal resistance of the 8-pin plastic DIP package, from the
datasheet, is 150°C/W. In a 25°C ambient, then, using a
maximum junction temperature of 150°C, this package will
dissipate 960mW.
PL2 = I VD (1 – D)
where VD is the forward drop of the clamp diode in the driver
(generally around 0.7V). The two parts of the load dissipation
must be summed in to produce PL
Accurate power dissipation numbers can be obtained by summing the three sources of power dissipation in the device:
PL = PL1 + PL2
• Load power dissipation (PL)
• Quiescent power dissipation (PQ)
• Transition power dissipation (PT)
Quiescent Power Dissipation
Quiescent power dissipation (PQ, as described in the input
section) depends on whether the input is high or low. A low
input will result in a maximum current drain (per driver) of
≤0.2mA; a logic high will result in a current drain of ≤2.0mA.
Quiescent power can therefore be found from:
Calculation of load power dissipation differs depending on
whether the load is capacitive, resistive or inductive.
Resistive Load Power Dissipation
Dissipation caused by a resistive load can be calculated as:
PQ = VS [D IH + (1 – D) IL]
PL = I2 RO D
where:
where:
IH =
IL =
D=
VS =
I = the current drawn by the load
RO = the output resistance of the driver when the
output is high, at the power supply voltage used
(See characteristic curves)
D = fraction of time the load is conducting (duty cycle)
quiescent current with input high
quiescent current with input low
fraction of time input is high (duty cycle)
power supply voltage
Transition Power Dissipation
Transition power is dissipated in the driver each time its
output changes state, because during the transition, for a
very brief interval, both the N- and P-channel MOSFETs in
the output totem-pole are ON simultaneously, and a current
is conducted through them from VS to ground. The transition
power dissipation is approximately:
Capacitive Load Power Dissipation
Dissipation caused by a capacitive load is simply the energy
placed in, or removed from, the load capacitance by the
driver. The energy stored in a capacitor is described by the
equation:
PT = f VS (A•s)
E = 1/2 C V2
where (A•s) is a time-current factor derived from Figure 2.
As this energy is lost in the driver each time the load is charged
or discharged, for power dissipation calculations the 1/2 is
removed. This equation also shows that it is good practice
not to place more voltage in the capacitor than is necessary,
as dissipation increases as the square of the voltage applied
to the capacitor. For a driver with a capacitive load:
Total power (PD) then, as previously described is just
PD = PL + PQ +PT
Examples show the relative magnitude for each term.
EXAMPLE 1: A MIC4423 operating on a 12V supply driving
two capacitive loads of 3000pF each, operating at 250kHz,
with a duty cycle of 50%, in a maximum ambient of 60°C.
PL = f C (VS)2
where:
First calculate load power loss:
f = Operating Frequency
C = Load Capacitance
VS = Driver Supply Voltage
PL = f x C x (VS)2
PL = 250,000 x (3 x 10–9 + 3 x 10–9) x 122
= 0.2160W
Inductive Load Power Dissipation
For inductive loads the situation is more complicated. For
the part of the cycle in which the driver is actively forcing
current into the inductor, the situation is the same as it is in
the resistive case:
Then transition power loss:
PT = f x VS x (A•s)
= 250,000 • 12 • 2.2 x 10–9 = 6.6mW
PL1 = I2 RO D
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Micrel, Inc.
Then quiescent power loss:
then:
PQ = VS x [D x IH + (1 – D) x IL]
= 12 x [(0.5 x 0.0035) + (0.5 x 0.0003)]
= 0.0228W
PD = 0.174 + 0.025 + 0.0150
= 0.213W
In a ceramic package with an θJA of 100°C/W, this amount of
power results in a junction temperature given the maximum
40°C ambient of:
Total power dissipation, then, is:
PD = 0.2160 + 0.0066 + 0.0228
= 0.2454W
(0.213 x 100) + 40 = 61.4°C
The actual junction temperature will be lower than calculated
both because duty cycle is less than 100% and because the
graph lists RDS(on) at a TJ of 125°C and the RDS(on) at 61°C TJ
will be somewhat lower.
Assuming an SOIC package, with an θJA of 120°C/W, this will
result in the junction running at:
0.2454 x 120 = 29.4°C
above ambient, which, given a maximum ambient temperature of 60°C, will result in a maximum junction temperature
of 89.4°C.
Definitions
CL = Load Capacitance in Farads.
D = Duty Cycle expressed as the fraction of time the input
to the driver is high.
EXAMPLE 2: A MIC4424 operating on a 15V input, with one
driver driving a 50Ω resistive load at 1MHz, with a duty cycle
of 67%, and the other driver quiescent, in a maximum ambient temperature of 40°C:
f = Operating Frequency of the driver in Hertz
IH = Power supply current drawn by a driver when both
inputs are high and neither output is loaded.
PL = I2 x RO x D
First, IO must be determined.
IL = Power supply current drawn by a driver when both
inputs are low and neither output is loaded.
IO = VS / (RO + RLOAD)
ID = Output current from a driver in Amps.
Given RO from the characteristic curves then,
PD = Total power dissipated in a driver in Watts.
IO = 15 / (3.3 + 50)
IO = 0.281A
PL = Power dissipated in the driver due to the driver’s
load in Watts.
PQ = Power dissipated in a quiescent driver in Watts.
PL = (0.281)2 x 3.3 x 0.67
= 0.174W
PT = F x VS x (A•s)/2
PT = Power dissipated in a driver when the output
changes states (“shoot-through current”) in Watts.
NOTE: The “shoot-through” current from a dual
transition (once up, once down) for both drivers is
stated in the graph on the following page in amperenanoseconds. This figure must be multiplied by the
number of repetitions per second (frequency to find
Watts).
and:
(because only one side is operating)
= (1,000,000 x 15 x 3.3 x 10–9) / 2
= 0.025 W
and:
RO= Output resistance of a driver in Ohms.
PQ = 15 x [(0.67 x 0.00125) + (0.33 x 0.000125) +
(1 x 0.000125)]
VS = Power supply voltage to the IC in Volts.
(this assumes that the unused side of the driver has its input
grounded, which is more efficient)
= 0.015W
MIC4423/4424/4425
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Crossover
Energy Loss
A•s (Ampere-seconds)
10-8
10-9
10-10
0
2
4
6
8 10 12 14 16 18
VIN
NOTE: THE VALUES ON THIS GRAPH REPRESENT THE LOSS SEEN BY BOTH
DRIVERS IN A PACKAGE DURING ONE COMPLETE CYCLE. FOR A SINGLE
DRIVER DIVIDE THE STATED VALUES BY 2. FOR A SINGLE TRANSITION OF A
SINGLE DRIVER, DIVIDE THE STATED VALUE BY 4.
Figure 2.
MAXIMUM PACKAGE
POWER DISSIPATION (mW)
1250
1000
SOIC
750
PDIP
500
250
0
25
50
75
100
125
150
AMBIENT TEMPERATURE (°C)
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Package Information
PIN 1
DIMENSIONS:
INCH (MM)
0.380 (9.65)
0.370 (9.40)
0.255 (6.48)
0.245 (6.22)
0.135 (3.43)
0.125 (3.18)
0.300 (7.62)
0.013 (0.330)
0.010 (0.254)
0.018 (0.57)
0.100 (2.54)
0.130 (3.30)
0.0375 (0.952)
0.380 (9.65)
0.320 (8.13)
8-Pin Plastic DIP (N)
8-Pin SOIC (M)
MIC4423/4424/4425
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MIC4423/4424/4425
Micrel, Inc.
PIN 1
DIMENSIONS:
INCHES (MM)
0.301 (7.645)
0.297 (7.544)
0.027 (0.686)
0.031 (0.787)
0.297 (7.544)
0.293 (7.442)
0.103 (2.616)
0.050 (1.270) 0.016 (0.046) 0.099 (2.515)
TYP
TYP
0.094 (2.388)
0.090 (2.286)
0.409 (10.389)
0.405 (10.287)
7
TYP
0.015
R
(0.381)
0.015
(0.381)
SEATING MIN
PLANE
0.330 (8.382)
0.326 (8.280)
0.032 (0.813) TYP
0.408 (10.363)
0.404 (10.262)
0.022 (0.559)
0.018 (0.457)
5
TYP
10 TYP
16-Pin Wide SOIC (WM)
MICREL INC.
TEL
2180 FORTUNE DRIVE
+ 1 (408) 944-0800
FAX
SAN JOSE, CA 95131
+ 1 (408) 474-1000
WEB
USA
http://www.micrel.com
This information furnished by Micrel in this data sheet is believed to be accurate and reliable. However no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 1999 Micrel, Inc.
July 2005
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MIC4423/4424/4425