TI MF10-N Mf10-n universal monolithic dual switched capacitor filter Datasheet

MF10-N
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MF10-N Universal Monolithic Dual Switched Capacitor Filter
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FEATURES
DESCRIPTION
•
•
The MF10-N consists of 2 independent and extremely
easy to use, general purpose CMOS active filter
building blocks. Each block, together with an external
clock and 3 to 4 resistors, can produce various 2nd
order functions. Each building block has 3 output
pins. One of the outputs can be configured to perform
either an allpass, highpass or a notch function; the
remaining 2 output pins perform lowpass and
bandpass functions. The center frequency of the
lowpass and bandpass 2nd order functions can be
either directly dependent on the clock frequency, or
they can depend on both clock frequency and
external resistor ratios. The center frequency of the
notch and allpass functions is directly dependent on
the clock frequency, while the highpass center
frequency depends on both resistor ratio and clock.
Up to 4th order functions can be performed by
cascading the two 2nd order building blocks of the
MF10-N; higher than 4th order functions can be
obtained by cascading MF10-N packages. Any of the
classical filter configurations (such as Butterworth,
Bessel, Cauer and Chebyshev) can be formed.
1
•
•
•
•
•
•
•
Easy to Use
Clock to Center Frequency Ratio Accuracy
±0.6%
Filter Cutoff Frequency Stability Directly
Dependent on External Clock Quality
Low Sensitivity to External Component
Variation
Separate Highpass (or Notch or Allpass),
Bandpass, Lowpass Outputs
fO × Q Range up to 200 kHz
Operation up to 30 kHz
20-pin 0.3″ Wide PDIP Package
20-pin Surface Mount (SOIC) Wide-Body
Package
For pin-compatible device with improved performance
refer to LMF100 datasheet.
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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System Block Diagram
Package in 20 pin molded wide body SOIC and 20 pin PDIP.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2)
Supply Voltage (V+ − V−)
14V
+
V + 0.3V
Voltage at Any Pin
V− − 0.3V
Input Current at Any Pin
Package Input Current
(3)
5 mA
(3)
20 mA
Power Dissipation (4)
500 mW
Storage Temperature
150°C
ESD Susceptability (5)
2000V
Soldering Information
SO Package
(1)
(2)
(3)
(4)
(5)
2
N Package: 10 sec
260°C
Vapor Phase (60 Sec.)
215°C
Infrared (15 Sec.)
220°C
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. DC and AC electrical specifications do not
apply when operating the device beyond its specified operating conditions.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
When the input voltage (VIN) at any pin exceeds the power supply rails (VIN < V− or VIN > V+) the absolute value of current at that pin
should be limited to 5 mA or less. The 20 mA package input current limits the number of pins that can exceed the power supply
boundaries with a 5 mA current limit to four.
The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature,
TA. The maximum allowable power dissipation at any temperature is PD = (TJMAX − TA)/θJA or the number given in the Absolute
Maximum Ratings, whichever is lower. For this device, TJMAX = 125°C, and the typical junction-to-ambient thermal resistance of the
MF10ACN/CCN when board mounted is 55°C/W. For the MF10AJ/CCJ, this number increases to 95°C/W and for the
MF10ACWM/CCWM this number is 66°C/W.
Human body model, 100 pF discharged through a 1.5 kΩ resistor.
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Operating Ratings (1)
Temperature Range (TMIN ≤ TA ≤ TMAX)
(1)
0°C ≤ TA ≤ 70°C
MF10ACN, MF10CCN, MF10CCWM
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. DC and AC electrical specifications do not
apply when operating the device beyond its specified operating conditions.
Electrical Characteristics
V+ = +5.00V and V− = −5.00V unless otherwise specified. Boldface limits apply for TMIN to TMAX; all other limits TA = TJ =
25°C.
Symbol
Parameter
Conditions
MF10ACN, MF10CCN,
MF10CCWM
Typical (1)
V+ − V−
IS
Supply Voltage
Center Frequency Range
fCLK
Clock Frequency Range
fCLK/fO
fCLK/fO
50:1 Clock to Center Frequency
Ratio Deviation
100:1 Clock to Center Frequency
Ratio Deviation
9
Max
14
Clock Applied to Pins 10 & 11
No Input Signal
Min
fO × Q < 200 kHz
Hz
20
kHz
Min
5.0
10
Hz
1.0
MHz
Max
1.5
MF10A
MF10C
MF10A
MF10C
Min
DC Offset Voltage (5)
Max
Min
Max
VOS3
±0.2
±0.6
±0.6
Q = 10, Mode 1
Vpin12 = 5V
fCLK = 250
KHz
±0.2
±1.5
±1.5
Q = 10, Mode 1
Vpin12 = 0V
fCLK = 500 kHz
±0.2
±0.6
±0.6
±0.2
±1.5
±1.5
DC Offset Voltage (5)
Min
Max
VOS2
DC Offset Voltage
10
(5)
±2
±6
±6
Vpin12 = 0V
fCLK = 500 kHz
±2
±6
±6
%
0
±0.2
±0.2
dB
±5.0
±20
±20
mV
−150
−185
−185
−85
−85
Vpin12 = +5V
(fCLK/fO = 50)
SA/B = V+
Vpin12 = +5V
(fCLK/fO = 50)
SA/B = V−
Vpin12 = +5V
(fCLK/fO = 50)
All Modes
Vpin12 = 0V
(fCLK/fO = 100)
SA/B = V+
−300
mV
Vpin12 = 0V
(fCLK/fO = 100)
SA/B = V−
−140
mV
Vpin12 = 0V
(fCLK/fO = 100)
All Modes
−140
mV
−70
−70
−100
−100
−20
−20
DC Offset Voltage (5)
VOUT
Minimum Output
BP, LP Pins
RL = 5k
±4.25
±3.8
±3.8
Voltage Swing
N/AP/HP Pin
RL = 3.5k
±4.25
±3.8
±3.8
Op Amp Gain BW Product
SR
Op Amp Slew Rate
(1)
(2)
(3)
(4)
(5)
mV
mV
VOS3
GBW
%
mV
Vpin12 = 5V
fCLK = 250 kHz
Mode 1 R1 = R2 = 10k
VOS2
mA
0.2
Q = 10, Mode 1
DC Offset Voltage (5)
12
30
Q Error (MAX) (4)
DC Lowpass Gain
12
V
0.1
Q = 10, Mode 1
VOS1
8
Units
Max
Clock Feedthrough
HOLP
Design
Limit (3)
Min
Maximum Supply Current
fO
Tested
Limit (2)
mV
V
V
2.5
MHz
7
V/μs
Typicals are at 25°C and represent most likely parametric norm.
Tested limits are ensured to AOQL (Average Outgoing Quality Level).
Design limits are specified but not 100% tested. These limits are not used to calculate outgoing quality levels.
The accuracy of the Q value is a function of the center frequency (fO). This is illustrated in the curves under the heading “Typical
Performance Characteristics”.
VOS1, VOS2, and VOS3 refer to the internal offsets as discussed in OFFSET VOLTAGE.
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Electrical Characteristics (continued)
V+ = +5.00V and V− = −5.00V unless otherwise specified. Boldface limits apply for TMIN to TMAX; all other limits TA = TJ =
25°C.
Symbol
Parameter
MF10ACN, MF10CCN,
MF10CCWM
Conditions
Typical (1)
(6)
(7)
Maximum Output Short Circuit
Current (7)
Design
Limit (3)
Units
Vpin12 = +5V, (fCLK/fO = 50)
83
dB
Vpin12 = 0V, (fCLK/fO = 100)
80
dB
Source
20
mA
Sink
3.0
mA
Dynamic Range (6)
ISC
Tested
Limit (2)
For ±5V supplies the dynamic range is referenced to 2.82V rms (4V peak) where the wideband noise over a 20 kHz bandwidth is
typically 200 μV rms for the MF10-N with a 50:1 CLK ratio and 280 μV rms for the MF10-N with a 100:1 CLK ratio.
The short circuit source current is measured by forcing the output that is being tested to its maximum positive voltage swing and then
shorting that output to the negative supply. The short circuit sink current is measured by forcing the output that is being tested to its
maximum negative voltage swing and then shorting that output to the positive supply. These are the worst case conditions.
Logic Input Characteristics
Boldface limits apply for TMIN to TMAX; all other limits TA = TJ = 25°C
MF10ACN, MF10CCN, MF10CCWM
Parameter
Min Logical “1”
CMOS Clock Input
Voltage
Max Logical “0”
Min Logical “1”
Max Logical “0”
Min Logical “1”
TTL Clock Input
Voltage
Max Logical “0”
Min Logical “1”
Max Logical “0”
(1)
(2)
(3)
4
Conditions
V+ = +5V, V− = −5V, VLSh = 0V
V+ = +10V, V− = 0V, VLSh = +5V
V+ = +5V, V− = −5V, VLSh = 0V
V+ = +10V, V− = 0V, VLSh = 0V
Typical (1)
Tested
Limit (2)
Design
Limit (3)
Units
+3.0
+3.0
V
−3.0
−3.0
V
+8.0
+8.0
V
+2.0
+2.0
V
+2.0
+2.0
V
+0.8
+0.8
V
+2.0
+2.0
V
+0.8
+0.8
V
Typicals are at 25°C and represent most likely parametric norm.
Tested limits are ensured to AOQL (Average Outgoing Quality Level).
Design limits are specified but not 100% tested. These limits are not used to calculate outgoing quality levels.
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Typical Performance Characteristics
Power Supply Current vs.
Power Supply Voltage
Positive Output Voltage Swing vs.
Load Resistance (N/AP/HP Output)
Figure 1.
Figure 2.
Negative Output Voltage Swing vs.
Load Resistance (N/AP/HP Output)
Negative Output Swing vs. Temperature
Figure 3.
Figure 4.
Positive Output Swing vs. Temperature
Crosstalk vs. Clock Frequency
Figure 5.
Figure 6.
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Typical Performance Characteristics (continued)
6
Q Deviation vs. Temperature
Q Deviation vs. Temperature
Figure 7.
Figure 8.
Q Deviation vs. Clock Frequency
Q Deviation vs. Clock Frequency
Figure 9.
Figure 10.
fCLK/fO Deviation vs. Temperature
fCLK/fO Deviation vs. Temperature
Figure 11.
Figure 12.
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Typical Performance Characteristics (continued)
fCLK/fO Deviation vs. lock Frequency
fCLK/fO Deviation vs. Clock Frequency
Figure 13.
Figure 14.
Deviation of fCLK/fO vs. Nominal Q
Deviation of fCLK/fO vs. Nominal Q
Figure 15.
Figure 16.
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PIN DESCRIPTIONS
LP(1,20), BP(2,19), N/AP/HP(3,18) The second order lowpass, bandpass and notch/allpass/highpass outputs.
These outputs can typically sink 1.5 mA and source 3 mA. Each output typically swings to within 1V of
each supply.
INV(4,17) The inverting input of the summing op-amp of each filter. These are high impedance inputs, but the
non-inverting input is internally tied to AGND, making INVA and INVB behave like summing junctions (low
impedance, current inputs).
S1(5,16) S1 is a signal input pin used in the allpass filter configurations (see modes 4 and 5). The pin should be
driven with a source impedance of less than 1 kΩ. If S1 is not driven with a signal it should be tied to
AGND (mid-supply).
SA/B(6) This pin activates a switch that connects one of the inputs of each filter's second summer to either AGND
(SA/B tied to V−) or to the lowpass (LP) output (SA/B tied to V+). This offers the flexibility needed for
configuring the filter in its various modes of operation.
VA+(7),VD+(8) Analog positive supply and digital positive supply. These pins are internally connected through the
IC substrate and therefore VA+ and VD+ should be derived from the same power supply source. They have
been brought out separately so they can be bypassed by separate capacitors, if desired. They can be
externally tied together and bypassed by a single capacitor.
VA−(14), VD−(13) Analog and digital negative supplies. The same comments as for VA+ and VD+ apply here.
LSh(9) Level shift pin; it accommodates various clock levels with dual or single supply operation. With dual ±5V
supplies, the MF10-N can be driven with CMOS clock levels (±5V) and the LSh pin should be tied to the
system ground. If the same supplies as above are used but only TTL clock levels, derived from 0V to +5V
supply, are available, the LSh pin should be tied to the system ground. For single supply operation (0V
and +10V) the VA−, VD−pins should be connected to the system ground, the AGND pin should be biased at
+5V and the LSh pin should also be tied to the system ground for TTL clock levels. LSh should be biased
at +5V for CMOS clock levels in 10V single-supply applications.
CLKA(10), CLKB(11) Clock inputs for each switched capacitor filter building block. They should both be of the
same level (TTL or CMOS). The level shift (LSh) pin description discusses how to accommodate their
levels. The duty cycle of the clock should be close to 50% especially when clock frequencies above 200
kHz are used. This allows the maximum time for the internal op-amps to settle, which yields optimum filter
operation.
50/100/CL(12) By tying this pin high a 50:1 clock-to-filter-center-frequency ratio is obtained. Tying this pin at midsupplies (i.e. analog ground with dual supplies) allows the filter to operate at a 100:1 clock-to-centerfrequency ratio. When the pin is tied low (i.e., negative supply with dual supplies), a simple current limiting
circuit is triggered to limit the overall supply current down to about 2.5 mA. The filtering action is then
aborted.
AGND(15) This is the analog ground pin. This pin should be connected to the system ground for dual supply
operation or biased to mid-supply for single supply operation. For a further discussion of mid-supply
biasing techniques see the Applications Information. For optimum filter performance a “clean” ground must
be provided.
Definition of Terms
fCLK: the frequency of the external clock signal applied to pin 10 or 11.
fO: center frequency of the second order function complex pole pair. fO is measured at the bandpass outputs of
the MF10-N, and is the frequency of maximum bandpass gain (Figure 17).
fnotch: the frequency of minimum (ideally zero) gain at the notch outputs.
fz: the center frequency of the second order complex zero pair, if any. If fz is different from fO and if QZ is high, it
can be observed as the frequency of a notch at the allpass output (Figure 26).
Q: “quality factor” of the 2nd order filter. Q is measured at the bandpass outputs of the MF10-N and is equal to fO
divided by the −3 dB bandwidth of the 2nd order bandpass filter (Figure 17). The value of Q determines the
shape of the 2nd order filter responses as shown in Figure 22.
8
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QZ: the quality factor of the second order complex zero pair, if any. QZ is related to the allpass characteristic,
which is written:
(1)
where QZ = Q for an all-pass response.
HOBP: the gain (in V/V) of the bandpass output at f = fO.
HOLP: the gain (in V/V) of the lowpass output as f → 0 Hz (Figure 18).
HOHP: the gain (in V/V) of the highpass output as f → fCLK/2 (Figure 19).
HON: the gain (in V/V) of the notch output as f → 0 Hz and as f → fCLK/2, when the notch filter has equal gain
above and below the center frequency (Figure 20). When the low-frequency gain differs from the high-frequency
gain, as in modes 2 and 3a (Figure 27 and Figure 24), the two quantities below are used in place of HON.
HON1: the gain (in V/V) of the notch output as f → 0 Hz.
HON2: the gain (in V/V) of the notch output as f → fCLK/2.
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(a)
(b)
Figure 17. 2nd-Order Bandpass Response
10
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(a)
(b)
Figure 18. 2nd-Order Low-Pass Response
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(a)
(b)
Figure 19. 2nd-Order High-Pass Response
12
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(a)
(b)
Figure 20. 2nd-Order Notch Response
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(a)
(b)
Figure 21. 2nd-Order All-Pass Response
(a) Bandpass
(b) Low Pass
(d) Notch
(c) High-Pass
(e) All-Pass
Figure 22. Response of various 2nd-order filters as a function of Q.
Gains and center frequencies are normalized to unity.
14
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Modes of Operation
The MF10-N is a switched capacitor (sampled data) filter. To fully describe its transfer functions, a time domain
approach is appropriate. Since this is cumbersome, and since the MF10-N closely approximates continuous
filters, the following discussion is based on the well known frequency domain. Each MF10-N can produce a full
2nd order function. See Table 1 for a summary of the characteristics of the various modes.
MODE 1: Notch 1, Bandpass, Lowpass Outputs:
fnotch = fO (See Figure 23)
(2)
fO= center frequency of the complex pole pair
(3)
fnotch= center frequency of the imaginary zero pair = fO.
(4)
(5)
= quality factor of the complex pole pair
BW
= the −3 dB bandwidth of the bandpass output.
Circuit dynamics:
(6)
MODE 1a: Non-Inverting BP, LP (See Figure 24)
(7)
Figure 23. MODE 1
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VIN should be driven from a low impedance (<1 kΩ) source.
Figure 24. MODE 1a
MODE 2: Notch 2, Bandpass, Lowpass: fnotch < fO (See Figure 25)
(8)
MODE 3: Highpass, Bandpass, Lowpass Outputs (See Figure 26)
(9)
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Figure 25. MODE 2
*In Mode 3, the feedback loop is closed around the input summing amplifier; the finite GBW product of this op amp
causes a slight Q enhancement. If this is a problem, connect a small capacitor (10 pF − 100 pF) across R4 to provide
some phase lead.
Figure 26. MODE 3
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MODE 3a: HP, BP, LP and Notch with External Op Amp (See Figure 27)
(10)
MODE 4: Allpass, Bandpass, Lowpass Outputs (See Figure 28)
(11)
*Due to the sampled data nature of the filter, a slight mismatch of fz and fO occurs causing a 0.4 dB peaking
around fO of the allpass filter amplitude response (which theoretically should be a straight line). If this is
unacceptable, Mode 5 is recommended.
18
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Figure 27. MODE 3a
Figure 28. MODE 4
MODE 5: Numerator Complex Zeros, BP, LP (See Figure 29)
(12)
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MODE 6a: Single Pole, HP, LP Filter (See Figure 30)
(13)
MODE 6b: Single Pole LP Filter (Inverting and Non-Inverting) (See Figure 31)
(14)
Figure 29. MODE 5
Figure 30. MODE 6a
Figure 31. MODE 6b
20
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Table 1. Summary of Modes. Realizable filter types (e.g. low-pass) denoted by asterisks.
Unless otherwise noted, gains of various filter outputs are inverting and adjustable by resistor ratios.
Mode
BP
LP
HP
N
AP
Adjustable fCLK/fO
3
No
2
No
3
Yes (above fCLK/50 or
fCLK/100)
4
Yes
Universal State-Variable Filter. Best
general-purpose mode.
7
Yes
As above, but also includes resistortuneable notch.
No
Gives Allpass response with HOAP =
−1 and HOLP = −2.
Notes
1
*
*
1a
HOBP1 = −Q
HOBP2 = +1
HOLP + 1
2
*
*
3
*
*
*
3a
*
*
*
*
*
*
3
*
*
*
4
Gives flatter allpass response than
above if R1 = R2 = 0.02R4.
3
Single pole.
2
Single pole.
4
5
6a
*
6b
HOLP1 = +1
HOLP2 = -R3/R2
*
Number of
Resistors
*
*
*
May need input buffer. Poor
dynamics for high Q.
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APPLICATIONS INFORMATION
The MF10-N is a general-purpose dual second-order state variable filter whose center frequency is proportional
to the frequency of the square wave applied to the clock input (fCLK). By connecting pin 12 to the appropriate DC
voltage, the filter center frequency fO can be made equal to either fCLK/100 or fCLK/50. fO can be very accurately
set (within ±6%) by using a crystal clock oscillator, or can be easily varied over a wide frequency range by
adjusting the clock frequency. If desired, the fCLK/fO ratio can be altered by external resistors as in Figure 25,
Figure 26, Figure 27, Figure 29, Figure 30, and Figure 31. The filter Q and gain are determined by external
resistors.
All of the five second-order filter types can be built using either section of the MF10-N. These are illustrated in
Figure 17 through Figure 21 along with their transfer functions and some related equations. Figure 22 shows the
effect of Q on the shapes of these curves. When filter orders greater than two are desired, two or more MF10-N
sections can be cascaded.
DESIGN EXAMPLE
In order to design a second-order filter section using the MF10-N, we must define the necessary values of three
parameters: f0, the filter section's center frequency; H0, the passband gain; and the filter's Q. These are
determined by the characteristics required of the filter being designed.
As an example, let's assume that a system requires a fourth-order Chebyshev low-pass filter with 1 dB ripple,
unity gain at DC, and 1000 Hz cutoff frequency. As the system order is four, it is realizable using both secondorder sections of an MF10-N. Many filter design texts include tables that list the characteristics (fO and Q) of each
of the second-order filter sections needed to synthesize a given higher-order filter. For the Chebyshev filter
defined above, such a table yields the following characteristics:
f0A = 529 Hz QA = 0.785
f0B = 993 Hz QB = 3.559
For unity gain at DC, we also specify:
H0A = 1
H0B = 1
The desired clock-to-cutoff-frequency ratio for the overall filter of this example is 100 and a 100 kHz clock signal
is available. Note that the required center frequencies for the two second-order sections will not be obtainable
with clock-to-center-frequency ratios of 50 or 100. It will be necessary to adjust
(15)
externally. From Table 1, we see that Mode 3 can be used to produce a low-pass filter with resistor-adjustable
center frequency.
In most filter designs involving multiple second-order stages, it is best to place the stages with lower Q values
ahead of stages with higher Q, especially when the higher Q is greater than 0.707. This is due to the higher
relative gain at the center frequency of a higher-Q stage. Placing a stage with lower Q ahead of a higher-Q stage
will provide some attenuation at the center frequency and thus help avoid clipping of signals near this frequency.
For this example, stage A has the lower Q (0.785) so it will be placed ahead of the other stage.
For the first section, we begin the design by choosing a convenient value for the input resistance: R1A = 20k. The
absolute value of the passband gain HOLPA is made equal to 1 by choosing R4A such that: R4A = −HOLPAR1A = R1A
= 20k. If the 50/100/CL pin is connected to mid-supply for nominal 100:1 clock-to-center-frequency ratio, we find
R2A by:
(16)
The resistors for the second section are found in a similar fashion:
22
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(17)
The complete circuit is shown in Figure 32 for split ±5V power supplies. Supply bypass capacitors are highly
recommended.
Figure 32. Fourth-Order Chebyshev Low-Pass Filter from Example in 3.1.
±5V Power Supply. 0V–5V TTL or −5V ±5V CMOS Logic Levels.
Figure 33. Fourth-Order Chebyshev Low-Pass Filter from Example in 3.1.
Single +10V Power Supply. 0V–5V TTL Logic Levels. Input Signals
Should be Referred to Half-Supply or Applied through a Coupling Capacitor.
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Resistive Divider with
Decoupling Capacitor
Operational Amplifier
with Divider
Voltage Regulator
Figure 34. Three Ways of Generating V+/2 for Single-Supply Operation
SINGLE SUPPLY OPERATION
The MF10-N can also operate with a single-ended power supply. Figure 33 shows the example filter with a
single-ended power supply. VA+ and VD+ are again connected to the positive power supply (8V to 14V), and VA−
and VD− are connected to ground. The AGND pin must be tied to V+/2 for single supply operation. This half-supply
point should be very “clean”, as any noise appearing on it will be treated as an input to the filter. It can be derived
from the supply voltage with a pair of resistors and a bypass capacitor (See Figure 34), or a low-impedance halfsupply voltage can be made using a three-terminal voltage regulator or an operational amplifier (See Figure 34
and Figure 34). The passive resistor divider with a bypass capacitor is sufficient for many applications, provided
that the time constant is long enough to reject any power supply noise. It is also important that the half-supply
reference present a low impedance to the clock frequency, so at very low clock frequencies the regulator or opamp approaches may be preferable because they will require smaller capacitors to filter the clock frequency. The
main power supply voltage should be clean (preferably regulated) and bypassed with 0.1 μF.
DYNAMIC CONSIDERATIONS
The maximum signal handling capability of the MF10-N, like that of any active filter, is limited by the power
supply voltages used. The amplifiers in the MF10-N are able to swing to within about 1V of the supplies, so the
input signals must be kept small enough that none of the outputs will exceed these limits. If the MF10-N is
operating on ±5V, for example, the outputs will clip at about 8 Vp–p. The maximum input voltage multiplied by the
filter gain should therefore be less than 8 Vp–p.
Note that if the filter Q is high, the gain at the lowpass or highpass outputs will be much greater than the nominal
filter gain (Figure 22). As an example, a lowpass filter with a Q of 10 will have a 20 dB peak in its amplitude
response at fO. If the nominal gain of the filter HOLP is equal to 1, the gain at fO will be 10. The maximum input
signal at fO must therefore be less than 800 mVp–p when the circuit is operated on ±5V supplies.
Also note that one output can have a reasonable small voltage on it while another is saturated. This is most likely
for a circuit such as the notch in Mode 1 (Figure 23). The notch output will be very small at fO, so it might appear
safe to apply a large signal to the input. However, the bandpass will have its maximum gain at fO and can clip if
overdriven. If one output clips, the performance at the other outputs will be degraded, so avoid overdriving any
filter section, even ones whose outputs are not being directly used. Accompanying Figure 23 through Figure 31
are equations labeled “circuit dynamics”, which relate the Q and the gains at the various outputs. These should
be consulted to determine peak circuit gains and maximum allowable signals for a given application.
OFFSET VOLTAGE
The MF10-N's switched capacitor integrators have a higher equivalent input offset voltage than would be found in
a typical continuous-time active filter integrator. Figure 35 shows an equivalent circuit of the MF10-N from which
the output DC offsets can be calculated. Typical values for these offsets with SA/B tied to V+ are:
Vos1 = opamp offset = ±5 mV
24
Vos2 = −150 mV @ 50:1
−300 mV @ 100:1
Vos3 = −70 mV @ 50:1
−140 mV @ 100:1
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When SA/B is tied to V−, Vos2 will approximately halve. The DC offset at the BP output is equal to the input offset
of the lowpass integrator (Vos3). The offsets at the other outputs depend on the mode of operation and the
resistor ratios, as described in the following expressions.
(18)
(19)
Figure 35. MF10-N Offset Voltage Sources
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Figure 36. Method for Trimming VOS
For most applications, the outputs are AC coupled and DC offsets are not bothersome unless large signals are
applied to the filter input. However, larger offset voltages will cause clipping to occur at lower AC signal levels,
and clipping at any of the outputs will cause gain nonlinearities and will change fO and Q. When operating in
Mode 3, offsets can become excessively large if R2 and R4 are used to make fCLK/fO significantly higher than the
nominal value, especially if Q is also high. An extreme example is a bandpass filter having unity gain, a Q of 20,
and fCLK/fO = 250 with pin 12 tied to ground (100:1 nominal). R4/R2 will therefore be equal to 6.25 and the offset
voltage at the lowpass output will be about +1V. Where necessary, the offset voltage can be adjusted by using
the circuit of Figure 36. This allows adjustment of VOS1, which will have varying effects on the different outputs as
described in the above equations. Some outputs cannot be adjusted this way in some modes, however (VOS(BP)
in modes 1a and 3, for example).
SAMPLED DATA SYSTEM CONSIDERATIONS
The MF10-N is a sampled data filter, and as such, differs in many ways from conventional continuous-time filters.
An important characteristic of sampled-data systems is their effect on signals at frequencies greater than onehalf the sampling frequency. (The MF10-N's sampling frequency is the same as its clock frequency.) If a signal
with a frequency greater than one-half the sampling frequency is applied to the input of a sampled data system, it
will be “reflected” to a frequency less than one-half the sampling frequency. Thus, an input signal whose
frequency is fs/2 + 100 Hz will cause the system to respond as though the input frequency was fs/2 − 100 Hz.
This phenomenon is known as “aliasing”, and can be reduced or eliminated by limiting the input signal spectrum
to less than fs/2. This may in some cases require the use of a bandwidth-limiting filter ahead of the MF10-N to
limit the input spectrum. However, since the clock frequency is much higher than the center frequency, this will
often not be necessary.
Another characteristic of sampled-data circuits is that the output signal changes amplitude once every sampling
period, resulting in “steps” in the output voltage which occur at the clock rate (Figure 37). If necessary, these can
be “smoothed” with a simple R–C low-pass filter at the MF10-N output.
The ratio of fCLK to fC (normally either 50:1 or 100:1) will also affect performance. A ratio of 100:1 will reduce any
aliasing problems and is usually recommended for wideband input signals. In noise sensitive applications,
however, a ratio of 50:1 may be better as it will result in 3 dB lower output noise. The 50:1 ratio also results in
lower DC offset voltages, as discussed in OFFSET VOLTAGE.
The accuracy of the fCLK/fO ratio is dependent on the value of Q. This is illustrated in Typical Performance
Characteristics. As Q is changed, the true value of the ratio changes as well. Unless the Q is low, the error in
fCLK/fO will be small. If the error is too large for a specific application, use a mode that allows adjustment of the
ratio with external resistors.
It should also be noted that the product of Q and fO should be limited to 300 kHz when fO < 5 kHz, and to 200
kHz for fO > 5 kHz.
26
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Figure 37. The Sampled-Data Output Waveform
Connection Diagram
Figure 38. SOIC and PDIP Packages (Top View)
See Package Numbers DW and NFH0020A
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REVISION HISTORY
Changes from Revision B (April 2013) to Revision C
•
28
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 27
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Sep-2015
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
MF100CCM
NRND
SOIC
DW
20
TBD
Call TI
Call TI
0 to 70
MF10CCWM
MF100CCWM
NRND
SOIC
DW
20
TBD
Call TI
Call TI
0 to 70
MF10CCWM
MF10CCN
NRND
PDIP
NFH
20
18
TBD
Call TI
Call TI
0 to 70
MF10CCN
MF10CCN/NOPB
ACTIVE
PDIP
NFH
20
18
Pb-Free
(RoHS)
CU SN
Level-1-NA-UNLIM
0 to 70
MF10CCN
MF10CCWM/NOPB
ACTIVE
SOIC
DW
20
36
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
0 to 70
MF10CCWM
MF10CCWMX
NRND
SOIC
DW
20
TBD
Call TI
Call TI
0 to 70
MF10CCWM
MF10CCWMX/NOPB
ACTIVE
SOIC
DW
20
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
0 to 70
MF10CCWM
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
10-Sep-2015
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
10-Sep-2015
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
MF10CCWMX/NOPB
Package Package Pins
Type Drawing
SOIC
DW
20
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
1000
330.0
24.4
Pack Materials-Page 1
10.9
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
13.3
3.25
12.0
24.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
10-Sep-2015
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
MF10CCWMX/NOPB
SOIC
DW
20
1000
367.0
367.0
45.0
Pack Materials-Page 2
MECHANICAL DATA
NFH0020A
N0020A
N20A (Rev G)
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