LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 LMR62421 SIMPLE SWITCHER® 24Vout, 2.1A Step-Up Voltage Regulator in SOT-23 Check for Samples: LMR62421 FEATURES DESCRIPTION • • • • • • • • • The LMR62421 is an easy-to-use, space-efficient 2.1A low-side switch regulator ideal for Boost and SEPIC DC-DC regulation. It provides all the active functions to provide local DC/DC conversion with fasttransient response and accurate regulation in the smallest PCB area. Switching frequency is internally set to 1.6 MHz, allowing the use of extremely small surface mount inductor and chip capacitors while providing efficiencies near 90%. Current-mode control and internal compensation provide ease-of-use, minimal component count, and high-performance regulation over a wide range of operating conditions. External shutdown features an ultra-low standby current of 80 nA ideal for portable applications. Tiny 5-pin SOT-23 and 6-pin WSON packages provide space-savings. Additional features include internal soft-start, circuitry to reduce inrush current, pulse-bypulse current limit, and thermal shutdown. 1 2 • Input Voltage Range of 2.7V to 5.5V Output Voltage up to 24V Switch Current up to 2.1A 1.6 MHz Switching Frequency Low Shutdown Iq, 80 nA Cycle-by-Cycle Current Limiting Internally Compensated Internal Soft-Start 5-Pin SOT-23 (2.92 x 2.84 x 1mm) and 6-Pin WSON (3 x 3 x 0.8 mm) Packaging Fully Enabled for WEBENCH® Power Designer PERFORMANCE BENEFITS • • Extremely Easy to Use Tiny Overall Solution Reduces System Cost APPLICATIONS • • • • • Boost / SEPIC Conversions from 3.3V, 5V Rails Space Constrained Applications Embedded Systems LCD Displays LED Applications System Performance Efficiency vs Load Current VOUT = 20V Efficiency vs Load Current VOUT = 12V 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2011–2013, Texas Instruments Incorporated LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com Typical Application VOUT VIN L1 D1 R2 R3 4 C3 3 C2 2 5 R1 1 C1 GND Connection Diagrams SW 1 5 VIN 4 EN PGND 1 6 SW VIN 2 5 AGND EN 3 4 FB GND 2 FB 3 Figure 1. 5-Pin SOT-23 (Top View) See DBV Package 2 Figure 2. 6-Pin WSON (Top View) See NGG0006A Package Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 PIN DESCRIPTIONS - 5-Pin SOT-23 Pin Name Function 1 SW 2 GND Switch node. Connect to the inductor, output diode. 3 FB Feedback pin. Connect FB to external resistor divider to set output voltage. 4 EN Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V. 5 VIN Supply voltage for power stage, and input supply voltage. Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin. PIN DESCRIPTIONS - 6-Pin WSON Pin Name Function 1 PGND Power ground pin. Place PGND and output capacitor GND close together. 2 VIN Supply voltage for power stage, and input supply voltage. 3 EN Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V. 4 FB Feedback pin. Connect FB to external resistor divider to set output voltage. 5 AGND 6 SW DAP GND Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 4. Switch node. Connect to the inductor, output diode. Signal & Power ground. Connect to pin 1 & pin 5 on top layer. Place 4-6 vias from DAP to bottom layer GND plane. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 3 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1) (2) VIN -0.5V to 7V SW Voltage -0.5V to 26.5V FB Voltage -0.5V to 3.0V EN Voltage -0.5V to VIN + 0.3V ESD Susceptibility (3) 2kV Junction Temperature (4) 150°C Storage Temp. Range -65°C to 150°C For soldering specifications: SNOA549 (1) (2) (3) (4) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For specified specifications and the test conditions, see Electrical Characteristics. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device. Operating Ratings (1) VIN 2.7V to 5.5V VEN (2) 0V to VIN −40°C to +125°C Junction Temperature Range (1) (2) 4 Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not ensured. For specified specifications and the test conditions, see Electrical Characteristics. Do not allow this pin to float or be greater than VIN +0.3V. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 Electrical Characteristics (1) (2) Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature range of (TJ = -40°C to 125°C). Minimum and Maximum limits are ensured through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. VIN = 5V unless otherwise indicated under the Conditions column. Symbol VFB ΔVFB/VIN Feedback Voltage Line Regulation Feedback Input Bias Current Conditions Min Typ Max −40°C ≤ to TJ ≤ +125°C (SOT-23) 1.230 1.255 1.280 0°C ≤ to TJ ≤ +125°C (SOT-23) 1.236 1.255 1.274 −40°C ≤ to TJ ≤ +125°C (WSON) 1.225 1.255 1.285 −0°C ≤ to TJ ≤ +125°C (WSON) 1.229 1.255 1.281 VIN = 2.7V to 5.5V 0.06 2000 kHz 1200 1600 Maximum Duty Cycle 88 96 DMIN Minimum Duty Cycle ICL Switch Current Limit Soft Start UVLO %/V µA Switching Frequency SS % 5 % SOT-23 170 330 WSON 190 350 2.1 Quiescent Current (switching) A 4 ms 7.0 VEN = 0V 80 Undervoltage Lockout VIN Rising 2.3 1.7 mΩ 3 Quiescent Current (shutdown) VIN Falling V 1 FSW Switch On Resistance Units 0.1 DMAX IQ 11 mA nA 2.65 V 1.9 Shutdown Threshold Voltage See (3) Enable Threshold Voltage See (3) I-SW Switch Leakage VSW = 24V 1.0 µA I-EN Enable Pin Current Sink/Source 100 nA θJA Junction to Ambient 0 LFPM Air Flow (4) WSON 80 SOT-23 118 θJC Junction to Case WSON 18 SOT-23 60 TSD Thermal Shutdown Temperature (5) 160 Thermal Shutdown Hysteresis 10 VEN_TH (2) (3) (4) (5) Feedback Voltage IFB RDS(ON) (1) Parameter 0.4 1.8 V °C/W °C/W °C Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL). Typical numbers are at 25°C and represent the most likely parametric norm. Do not allow this pin to float or be greater than VIN +0.3V. Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air. Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 5 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com Typical Performance Characteristics 6 Current Limit vs Temperature FB Pin Voltage vs Temperature Figure 3. Figure 4. Oscillator Frequency vs Temperature Typical Maximum Output Current vs VIN Figure 5. Figure 6. RDSON vs Temperature Efficiency vs Load Current, Vo = 20V Figure 7. Figure 8. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 Typical Performance Characteristics (continued) Efficiency vs Load Current, Vo = 12V Output Voltage Load Regulation Figure 9. Figure 10. Output Voltage Line Regulation Figure 11. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 7 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com Simplified Internal Block Diagram EN VIN ThermalSHDN Control Logic + UVLO = 2.3V Oscillator Corrective - Ramp SW cv S R R Q 1.6 MHz + + - FB VREF = 1.255V NMOS Internal Compensation ILIMIT ISENSE-AMP Soft-Start + - GND Figure 12. Simplified Block Diagram 8 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 APPLICATION INFORMATION THEORY OF OPERATION The following operating description of the LMR62421 will refer to the Simplified Block Diagram (Figure 12) the simplified schematic (Figure 13), and its associated waveforms (Figure 14). The LMR62421 supplies a regulated output voltage by switching the internal NMOS control switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the internal NMOS control switch. During this on-time, the SW pin voltage (VSW) decreases to approximately GND, and the inductor current (IL) increases with a linear slope. IL is measured by the current sense amplifier, which generates an output proportional to the switch current. The sensed signal is summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the feedback voltage and VREF. When the PWM comparator output goes high, the output switch turns off until the next switching cycle begins. During the switch off-time, inductor current discharges through diode D1, which forces the SW pin to swing to the output voltage plus the forward voltage (VD) of the diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output voltage . I L (t) + VL (t) - D1 L1 I C (t) Control + VIN + Q1 VSW( t ) C1 VO(t) - Figure 13. Simplified Schematic Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 9 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com VO + VD Vsw (t) t VIN VL(t) t VIN - VOUT - VD I L (t) iL t I DIODE (t) t ( iL - - i OUT ) I Capacitor (t) t - i OUT 'v VOUT (t) DTS TS Figure 14. Typical Waveforms CURRENT LIMIT The LMR62421 uses cycle-by-cycle current limiting to protect the internal NMOS switch. It is important to note that this current limit will not protect the output from excessive current during an output short circuit. The input supply is connected to the output by the series connection of an inductor and a diode. If a short circuit is placed on the output, excessive current can damage both the inductor and diode. Design Guide ENABLE PIN / SHUTDOWN MODE The LMR62421 has a shutdown mode that is controlled by the Enable pin (EN). When a logic low voltage is applied to EN, the part is in shutdown mode and its quiescent current drops to typically 80 nA. Switch leakage adds up to another 1 µA from the input supply. The voltage at this pin should never exceed VIN + 0.3V. THERMAL SHUTDOWN Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature exceeds 160°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature drops to approximately 150°C. 10 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 SOFT-START This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s reference voltage ramps to its nominal value of 1.255V in approximately 4.0ms. This forces the regulator output to ramp up in a more linear and controlled fashion, which helps reduce inrush current. INDUCTOR SELECTION The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN): VOUT VIN §1 · 1 =¨ ¸= c ©1 - D¹ D (1) Therefore: D= VOUT - VIN VOUT (2) Power losses due to the diode (D1) forward voltage drop, the voltage drop across the internal NMOS switch, the voltage drop across the inductor resistance (RDCR) and switching losses must be included to calculate a more accurate duty cycle (See Calculating Efficiency and Junction Temperature for a detailed explanation). A more accurate formula for calculating the conversion ratio is: VOUT K = c D VIN where • η equals the efficiency of the LMR62421 application. (3) The inductor value determines the input ripple current. Lower inductor values decrease the size of the inductor, but increase the input ripple current. An increase in the inductor value will decrease the input ripple current. 'i L I L (t) iL VIN - VOUT VIN L DTS L TS t Figure 15. Inductor Current Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 11 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com 2'iL § VIN · =¨ ¸ DTS © L ¹ § VIN · ¸¸ x DTS ÂiL = ¨¨ © 2L ¹ (4) A good design practice is to design the inductor to produce 10% to 30% ripple of maximum load. From the previous equations, the inductor value is then obtained. § VIN · ¸ x DTS L =¨ ©2 x 'iL¹ where • 1/TS = FSW = switching frequency (5) One must also ensure that the minimum current limit (2.1A) is not exceeded, so the peak current in the inductor must be calculated. The peak current (ILPK ) in the inductor is calculated by: ILpk = IIN + ΔIL (6) ILpk = IOUT / D' + ΔIL (7) or When selecting an inductor, make sure that it is capable of supporting the peak input current without saturating. Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating correctly. Because of the speed of the internal current limit, the peak current of the inductor need only be specified for the required maximum input current. For example, if the designed maximum input current is 1.5A and the peak current is 1.75A, then the inductor should be specified with a saturation current limit of >1.75A. There is no need to specify the saturation or peak current of the inductor at the 3A typical switch current limit. Because of the operating frequency of the LMR62421, ferrite based inductors are preferred to minimize core losses. This presents little restriction since the variety of ferrite-based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended inductors see Example Circuits. INPUT CAPACITOR An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent Series Inductance). The recommended input capacitance is 10 µF to 44 µF depending on the application. The capacitor manufacturer specifically states the input voltage rating. Make sure to check any recommended deratings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature. The ESL of an input capacitor is usually determined by the effective cross sectional area of the current path. At the operating frequencies of the LMR62421, certain capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than that required to provide stable operation. As a result, surface mount capacitors are strongly recommended. Multilayer ceramic capacitors (MLCC) are good choices for both input and output capacitors and have very low ESL. For MLCCs it is recommended to use X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions. 12 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 OUTPUT CAPACITOR The LMR62421 operates at frequencies allowing the use of ceramic output capacitors without compromising transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple. The output capacitor is selected based upon the desired output ripple and transient response. The initial current of a load transient is provided mainly by the output capacitor. The output impedance will therefore determine the maximum voltage perturbation. The output ripple of the converter is a function of the capacitor’s reactance and its equivalent series resistance (ESR): § · VOUT x D ¸ ÂVOUT = ÂIL x R ESR + ¨¨ ¸ © 2 x FSW x RLoad x C OUT ¹ (8) When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the output ripple will be approximately sinusoidal and 90° phase shifted from the switching action . Given the availability and quality of MLCCs and the expected output voltage of designs using the LMR62421, there is really no need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic capacitances in the inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not. Since the output capacitor is one of the two external components that control the stability of the regulator control loop, most applications will require a minimum at 4.7 µF of output capacitance. Like the input capacitor, recommended multilayer ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired operating voltage and temperature. SETTING THE OUTPUT VOLTAGE The output voltage is set using the following equation where R1 is connected between the FB pin and GND, and R2 is connected between VOUT and the FB pin. VO R2 C3 VFB R LOAD R1 Figure 16. Setting Vout A good value for R1 is 10kΩ. § VOUT · - 1¸¸ x R1 R 2 = ¨¨ © VREF ¹ (9) Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 13 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com COMPENSATION The LMR62421 uses constant frequency peak current mode control. This mode of control allows for a simple external compensation scheme that can be optimized for each application. A complicated mathematical analysis can be completed to fully explain the LMR62421’s internal & external compensation, but for simplicity, a graphical approach with simple equations will be used. Below is a Gain & Phase plot of a LMR62421 that produces a 12V output from a 5V input voltage. The Bode plot shows the total loop Gain & Phase without external compensation. 80 180 gm-Pole 60 RC-Pole 90 40 dB 20 0 -20 Vi = 5V Vo = 12V Io = 500 mA Co = 10 PF Lo = 5 PH 0 gm-Zero -90 -40 RHP-Zero -60 -80 10 100 1k 10k 100k -180 1M FREQUENCY Figure 17. LMR62421 Without External Compensation One can see that the Crossover frequency is fine, but the phase margin at 0dB is very low (22°). A zero can be placed just above the crossover frequency so that the phase margin will be bumped up to a minimum of 45°. Below is the same application with a zero added at 8 kHz. 80 60 40 gm-Pole RC-Pole Vi = 5V Vo = 12V Io = 500 mA Co = 10 mF Lo = 5 mH 180 90 D = 0.625 Cf = 220 pF gm-zero 0 0 Fz-cf = 8 kHz RHP-Zero = 107 kHz -20 Fp-cf = 77 kHz Fp-rc = 660 Hz -90 -40 Ext (Cf) -Zero -60 Ext (Cf)-Pole RHP-Zero -180 -80 10 100 1k 10k 100k 1M dB 20 FREQUENCY Figure 18. LMR62421 With External Compensation 14 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 The simplest method to determine the compensation component value is as follows. Set the output voltage with the following equation. § VOUT · - 1¸¸ x R1 R 2 = ¨¨ © VREF ¹ where • R1 is the bottom resistor • R2 is the resistor tied to the output voltage. and (10) The next step is to calculate the value of C3. The internal compensation has been designed so that when a zero is added between 5 kHz & 10 kHz the converter will have good transient response with plenty of phase margin for all input & output voltage combinations. FZERO - CF = 1 = 5 kHz o 10 kHz 2S(R2 x Cf) (11) Lower output voltages will have the zero set closer to 10 kHz, and higher output voltages will usually have the zero set closer to 5 kHz. It is always recommended to obtain a Gain/Phase plot for your actual application. One could refer to the Typical Appplication section to obtain examples of working applications and the associated component values. Pole @ origin due to internal gm amplifier: FP-ORIGIN (12) Pole due to output load and capacitor: FP- RC = 1 2S(R Load COUT) (13) This equation only determines the frequency of the pole for perfect current mode control (CMC). Therefore, it doesn’t take into account the additional internal artificial ramp that is added to the current signal for stability reasons. By adding artificial ramp, you begin to move away from CMC to voltage mode control (VMC). The artifact is that the pole due to the output load and output capacitor will actually be slightly higher in frequency than calculated. In this example it is calculated at 650 Hz, but in reality it is around 1 kHz. The zero created with capacitor C3 & resistor R2: VO R2 VFB C3 R LOAD R1 Figure 19. Setting External Pole-Zero Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 15 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 FZERO - CF = www.ti.com 1 2S(R2 x C3) (14) There is an associated pole with the zero that was created in the above equation. FPOLE - CF = 1 2S((R1 R2) x C3) (15) It is always higher in frequency than the zero. A right-half plane zero (RHPZ) is inherent to all boost converters. One must remember that the gain associated with a right-half plane zero increases at 20dB per decade, but the phase decreases by 45° per decade. For most applications there is little concern with the RHPZ due to the fact that the frequency at which it shows up is well beyond crossover, and has little to no effect on loop stability. One must be concerned with this condition for large inductor values and high output currents. 2 RHPZERO = (D') RLoad 2S x L (16) There are miscellaneous poles and zeros associated with parasitics internal to the LMR62421, external components, and the PCB. They are located well over the crossover frequency, and for simplicity are not discussed. PCB Layout Considerations When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The most important consideration when completing a Boost Converter layout is the close coupling of the GND connections of the COUT capacitor and the LMR62421 PGND pin. The GND ends should be close to one another and be connected to the GND plane with at least two through-holes. There should be a continuous ground plane on the bottom layer of a two-layer board. The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup and inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with the AGND of R1 placed as close as possible to the GND (pin 5 for the WSON) of the IC. The VOUT trace to R2 should be routed away from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible. However, making the traces wide increases radiated noise, so the designer must make this trade-off. Radiated noise can be decreased by choosing a shielded inductor. The remaining components should also be placed as close as possible to the IC. Please see Application Note AN-1229 SNVA054 for further considerations and the LMR62421 demo board as an example of a good layout. SEPIC Converter The LMR62421 can easily be converted into a SEPIC converter. A SEPIC converter has the ability to regulate an output voltage that is either larger or smaller in magnitude than the input voltage. Other converters have this ability as well (CUK and Buck-Boost), but usually create an output voltage that is opposite in polarity to the input voltage. This topology is a perfect fit for Lithium Ion battery applications where the input voltage for a single cell Li-Ion battery will vary between 3V & 4.5V and the output voltage is somewhere in between. Most of the analysis of the LMR62421 Boost Converter is applicable to the LMR62421 SEPIC Converter. SEPIC Design Guide: SEPIC Conversion ratio without loss elements: Vo D = VIN D' (17) Therefore: D= 16 VO VO + VIN (18) Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 Small ripple approximation: In a well-designed SEPIC converter, the output voltage, and input voltage ripple, the inductor ripple and is small in comparison to the DC magnitude. Therefore it is a safe approximation to assume a DC value for these components. The main objective of the Steady State Analysis is to determine the steady state duty-cycle, voltage and current stresses on all components, and proper values for all components. In a steady-state converter, the net volt-seconds across an inductor after one cycle will equal zero. Also, the charge into a capacitor will equal the charge out of a capacitor in one cycle. Therefore: I L2 § D· = ¨ ' ¸ x I L1 ©D¹ and IL1 = § D · x § VO · ¨ D' ¸ ¨ R ¸ © ¹ © ¹ (19) Substituting IL1 into IL2 VO IL2 = R (20) The average inductor current of L2 is the average output load. VL(t) AREA 1 t (s) AREA 2 DTS TS Figure 20. Inductor Volt-Sec Balance Waveform Applying Charge balance on C1: VC1 = D' (Vo ) D (21) Since there are no DC voltages across either inductor, and capacitor C6 is connected to Vin through L1 at one end, or to ground through L2 on the other end, we can say that VC1 = VIN (22) Therefore: VIN = D' (Vo ) D (23) This verifies the original conversion ratio equation. It is important to remember that the internal switch current is equal to IL1 and IL2. During the D interval. Design the converter so that the minimum ensured peak switch current limit (2.1A) is not exceeded. Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 17 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com VIN VO L1 D1 C6 LMR62421 C1 1 6 2 5 3 4 L2 R2 R3 C2 C5 C4 C3 R1 Figure 21. SEPIC CONVERTER Schematic Steady State Analysis with Loss Elements i L1( t ) i sw iC1( t ) vC1( t ) + i D1( t ) vD1( t ) i L 2( t ) VIN i C2( t ) vL2( t ) + - + R L1 vL1( t ) + vC2( t ) vO ( t ) - + R on R L2 18 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 Using inductor volt-second balance & capacitor charge balance, the following equations are derived: I L2 § VO · =¨R¸ © ¹ and § VO · § D · ¨ R ¸ x ¨ D' ¸ © ¹ © ¹ IL1 = Vo § =¨ VIN ¨© (24) § · ¨ ¸ ¨ ¸ 1 D·¨ ¸ ¸ ¸ D' ¹ ¨ § VD R L2 · §¨ D ·¸ § R ON · §¨ D 2 ·¸ § R L1 · ¸ ¸+ ¨ ¨1+ + ¨ ¸+ ¸¸ ¨ ¨ ¨© VO R ¸¹ ¨© D' 2 ¸¹ © R ¹ ¨© D' 2 ¸¹ © R ¹ ¸ © ¹ (25) Therefore: § · ¨ ¸ ¨ ¸ 1 ¸ K= ¨ ¨§ VD R L2 · §¨ D ·¸ § R ON · §¨ D 2 ·¸ § R L1 · ¸ ¸+ ¨ ¨1+ + ¸+ ¨ ¸¸ ¨ ¨ ¨© VO R ¸¹ ¨© D' 2 ¸¹ © R ¹ ¨© D' 2 ¸¹ © R ¹ ¸ © ¹ (26) One can see that all variables are known except for the duty cycle (D). A quadratic equation is needed to solve for D. A less accurate method of determining the duty cycle is to assume efficiency, and calculate the duty cycle. § D ·xK ¨1 - D¸ © ¹ (27) VO · § ¨(V x K) +V ¸ O¹ © IN (28) VO = VIN D= Table 1. Efficiencies for Typical SEPIC Application Vin 2.7V Vin 3.3V Vin Vo 3.1V lin 770 mA lo η 5V Vo 3.1V Vo 3.1V lin 600mA lin 375 mA 500 mA lo 500mA lo 500 mA 75% η 80% η 83% Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 19 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com SEPIC Converter PCB Layout The layout guidelines described for the LMR62421 Boost-Converter are applicable to the SEPIC Converter. Below is a proper PCB layout for a SEPIC Converter. CIN PCB VIN PGND L1 FB EN 4 3 AGND VIN 5 2 PGND SW 6 1 CIN COUT D1 VO C6 L2 Figure 22. SEPIC PCB Layout WSON Package The LMR62421 packaged in the 6–pin WSON: Figure 23. Internal WSON Connection 20 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 For certain high power applications, the PCB land may be modified to a "dog bone" shape (see Figure 24). Increasing the size of ground plane, and adding thermal vias can reduce the RθJA for the application. COPPER PGND 1 6 SW Vin 2 5 AGND EN 3 4 FB COPPER Figure 24. PCB Dog Bone Layout LMR62421 Design Example 1 L1 4 VIN R3 1 M: EN 6.8 PH 2.9A 3 FB 2 2A 20V GND 5 Vin C1 10 PF 10V 12V 1 SW D1 C2 10 PF 25V R2 86.6k C3 220 pF 25V R LOAD R1 10.2k Figure 25. Vin = 3V - 5V, Vout = 12V @ 500 mA Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 21 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com LMR62421 Design Example 2 L1 10 PH 1.2A 1 M: 3 4 FB SHDN R3 2 1A 20V GND VIN 5V 1 5 SW Vin D1 C1 10 PF 6.3V R2 30.1k C2 10 PF 10V C3 1 nF R LOAD R1 10k Figure 26. Vin = 3V, Vout = 5V @ 500 mA LMR62421 Design Example 3 L1 10 PH 1.2A 1 M: 4 R3 SHDN 3 FB 2 GND VIN 5 Vin C1 22 PF 6.3V 500 mA 30V 20V 1 SW D1 R2 150k C2 4.7 PF 50V C3 470 pF 50V R LOAD R1 10k Figure 27. Vin = 3.3V, Vout = 20V @ 100 mA 22 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 LMR62421 www.ti.com SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 LMR62421 SEPIC Design Example 4 VIN L1 6.8 PH 1.2A 2.2 PF 16V 1A 20V D1 VO C6 LMR62421 1 6 2 5 L2 6.8 PH 1.2A R2 16.5k 3 C1 22 PF 10V C2 4 R3 100k C5 2.2 nF C3 10 PF 10V C4 (opt) R1 10.2k (opt) Figure 28. Vin = 2.7V - 5V, Vout = 3.3V @ 500mA Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 23 LMR62421 SNVS734B – OCTOBER 2011 – REVISED APRIL 2013 www.ti.com REVISION HISTORY Changes from Revision A (April 2013) to Revision B • 24 Page Changed layout of National Data Sheet to TI format .......................................................................................................... 23 Submit Documentation Feedback Copyright © 2011–2013, Texas Instruments Incorporated Product Folder Links: LMR62421 PACKAGE OPTION ADDENDUM www.ti.com 11-Apr-2013 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish (2) MSL Peak Temp Op Temp (°C) Top-Side Markings (3) (4) LMR62421XMF/NOPB ACTIVE SOT-23 DBV 5 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SH8B LMR62421XMFE/NOPB ACTIVE SOT-23 DBV 5 250 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SH8B LMR62421XMFX/NOPB ACTIVE SOT-23 DBV 5 3000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 SH8B LMR62421XSD/NOPB ACTIVE WSON NGG 6 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L270B LMR62421XSDE/NOPB ACTIVE WSON NGG 6 250 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L270B LMR62421XSDX/NOPB ACTIVE WSON NGG 6 4500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L270B (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Top-Side Marking for that device. Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 11-Apr-2013 Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. 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Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 8-Apr-2013 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) LMR62421XMF/NOPB SOT-23 DBV 5 1000 178.0 8.4 LMR62421XMFE/NOPB SOT-23 DBV 5 250 178.0 LMR62421XMFX/NOPB SOT-23 DBV 5 3000 178.0 LMR62421XSD/NOPB WSON NGG 6 1000 LMR62421XSDE/NOPB WSON NGG 6 LMR62421XSDX/NOPB WSON NGG 6 3.2 3.2 1.4 4.0 8.0 Q3 8.4 3.2 3.2 1.4 4.0 8.0 Q3 8.4 3.2 3.2 1.4 4.0 8.0 Q3 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 250 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 4500 330.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 Pack Materials-Page 1 W Pin1 (mm) Quadrant PACKAGE MATERIALS INFORMATION www.ti.com 8-Apr-2013 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LMR62421XMF/NOPB SOT-23 DBV 5 1000 210.0 185.0 35.0 LMR62421XMFE/NOPB SOT-23 DBV 5 250 210.0 185.0 35.0 LMR62421XMFX/NOPB SOT-23 DBV 5 3000 210.0 185.0 35.0 LMR62421XSD/NOPB WSON NGG 6 1000 210.0 185.0 35.0 LMR62421XSDE/NOPB WSON NGG 6 250 210.0 185.0 35.0 LMR62421XSDX/NOPB WSON NGG 6 4500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA NGG0006A SDE06A (Rev A) www.ti.com IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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