TI1 LM20143QMHE/NOPB 3-a powerwise adjustable frequency synchronous buck regulator Datasheet

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LM20143, LM20143-Q1
SNVS528H – OCTOBER 2007 – REVISED JANUARY 2016
LM20143/LM20143-Q1 3-A PowerWise™ Adjustable Frequency Synchronous Buck
Regulator
1 Features
3 Description
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The LM20143 devices are full featured PowerWise™
adjustable frequency synchronous buck regulators
capable of delivering up to 3 A of continuous output
current. The current mode control loop can be
compensated to be stable with virtually any type of
output capacitor. For most cases, compensating the
device only requires two external components,
providing maximum flexibility and ease of use. The
device is optimized to work over the input voltage
range of 2.95 V to 5.5 V, making it suited for a wide
variety of low voltage systems.
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Available in AEC-Q100 Temperature Grade 1
Input Voltage Range: 2.95 V to 5.50 V
Accurate Current Limit Minimizes Inductor Size
97% Peak Efficiency
Adjustable Output Voltage Down to 0.80 V
Adjustable Switching Frequency (500 kHz to 1.5
MHz)
32-mΩ Integrated FET Switches
Starts into Prebiased Loads
Output Voltage Tracking
Peak Current Mode Control
Adjustable Soft-Start with External Capacitor
Precision Enable Pin with Hysteresis
Integrated OVP, UVLO, Power Good and Thermal
Shutdown
The device features internal over voltage protection
(OVP) and over current protection (OCP) circuits for
increased system reliability. A precision enable pin
and integrated UVLO allows the turn-on of the device
to be tightly controlled and sequenced. Start-up
inrush currents are limited by both an internally fixed
and externally adjustable Soft-Start circuit. Fault
detection and supply sequencing is possible with the
integrated power good circuit.
2 Applications
•
•
•
The frequency of this device can be adjusted from
500 kHz to 1.5 MHz by connecting an external
resistor from the RT pin to ground.
Simple to Design, High Efficiency Point of Load
Regulation from a 5-V or 3.3-V Bus
High Performance DSPs, FPGAs, ASICs, and
Microprocessors
Broadband, Networking, and Optical
Communications Infrastructure
The LM20143 is designed to work well in multi-rail
power supply architectures. The output voltage of the
device can be configured to track a higher voltage rail
using the SS/TRK pin. If the output of the LM20143 is
pre-biased at startup it will not sink current to pull the
output low until the internal soft-start ramp exceeds
the voltage at the feedback pin.
Device Information(1)
PART NUMBER
LM20143
PACKAGE
HTSSOP (16)
LM20143-Q1
BODY SIZE (NOM)
4.40 mm × 5.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Circuit
L
LM20143
PVIN
VIN
CIN
EN
RF
VOUT
SW
RFB1
FB
COUT
AVIN
CF
PGOOD
RT
RT
COMP
RC1
CC1
RFB2
VCC
SS/TRK
PGND AGND
CVCC
CSS
(optional)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM20143, LM20143-Q1
SNVS528H – OCTOBER 2007 – REVISED JANUARY 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
11
11
12
13
8
Application and Implementation ........................ 17
8.1 Application Information............................................ 17
8.2 Typical Applications ................................................ 17
9 Power Supply Recommendations...................... 27
10 Layout................................................................... 27
10.1 Layout Guidelines ................................................. 27
10.2 Layout Example .................................................... 28
10.3 Thermal Considerations ........................................ 29
11 Device and Documentation Support ................. 30
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Documentation Support ........................................
Related Links ........................................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
30
30
30
30
30
30
12 Mechanical, Packaging, and Orderable
Information ........................................................... 30
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (March 2013) to Revision H
•
Page
Added ESD Ratings table, Feature Descriptionsection, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................. 1
Changes from Revision F (March 2013) to Revision G
•
2
Page
Changed layout of National Data Sheet to TI format ............................................................................................................. 1
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5 Pin Configuration and Functions
PWP Package
16-Pin HTSSOP With Exposed Thermal Pad
Top View
SS/TRK
1
16
RT
FB
2
15
AGND
PGOOD
3
14
AVIN
COMP
4
13
VCC
Exposed Pad
On Bottom
(EP)
NC
5
12
EN
PVIN
6
11
PGND
PVIN
7
10
PGND
SW
8
9
SW
Pin Functions
PIN
NO.
NAME
TYPE (1)
DESCRIPTION
1
SS/TRK
I
Soft-Start or Tracking control input. An internal 5-µA current source charges an external capacitor to set
the Soft-Start ramp rate. If driven by a external source less than 800 mV, this pin overrides the internal
reference that sets the output voltage. If left open, an internal 1ms Soft-Start ramp is activated.
2
FB
I
Feedback input to the error amplifier from the regulated output. This pin is connected to the inverting
input of the internal transconductance error amplifier. An 800-mV reference connected to the noninverting input of the error amplifier sets the closed loop regulation voltage at the FB pin.
3
PGOOD
O
Power good output signal. Open drain output indicating the output voltage is regulating within tolerance.
is recommend for most applications.
4
COMP
O
External compensation pin. Connect the compensation network to resistor and capacitor to this pin to
compensate the device.
5
NC
-
This pin has no internal connection. While this pin may be left open, it is strongly recommended that this
pin be connected to Ground.
PVIN
P
Input voltage to the power switches inside the device. These pins should be connected together at the
device. A low ESR capacitor should be placed near these pins to stabilize the input voltage.
SW
P
Switch pin. The PWM output of the internal power switches.
PGND
G
Power ground pin for the internal power switches.
12
EN
I
Precision enable input for the device. An external voltage divider can be used to set the device turn-on
threshold. If not used the EN pin should be connected to PVIN.
13
VCC
P
Internal 2.7-V sub-regulator. This pin should be bypassed with a 1-µF ceramic capacitor.
14
AVIN
P
Analog input supply that generates the internal bias. Must be connected to VIN through a low pass RC
filter.
15
AGND
G
Quiet analog ground for the internal bias circuitry.
16
RT
I
Frequency adjust pin. Connecting a resistor on this pin to ground will set the oscillator frequency.
EP
Exposed Pad
-
Exposed metal thermal pad on the underside of the package with a weak electrical connection to
ground. It is recommended to connect this pad to the PC board ground plane copper in order to improve
heat dissipation and reduce the package θJA. Do not connect to any potential other than Ground.
6
7
8
9
10
11
(1)
P: Power, I: Input, O: Output, G: Ground
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6 Specifications
6.1 Absolute Maximum Ratings
(1)
See
AVIN, PVIN, EN, PGOOD, SS/TRK, COMP, FB, RT
Power Dissipation
Voltages from indicated pins to GND
MIN
MAX
UNIT
–0.3
6
V
(2)
2.6
W
Junction Temperature
150
°C
Lead Temperature (Soldering, 10 sec)
260
°C
150
°C
Storage Temperature
(1)
(2)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junctions-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX – TA) / θJA. The maximum power dissipations of 2.6 W is determined using TA = 25°C, θJA = 38°C/W, and
TJ_MAX = 125°C.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
MIN
MAX
UNIT
PVIN, AVIN to GND
2.95
5.5
V
Junction Temperature
−40
125
°C
6.4 Thermal Information
LM20143
THERMAL METRIC (1)
PWP (HTSSOP)
UNIT
16 PINS
(2)
RθJA
Junction-to-ambient thermal resistance
RθJC(top)
Junction-to-case (top) thermal resistance
39.3
°C/W
20.3
RθJB
°C/W
Junction-to-board thermal resistance
9.9
°C/W
ψJT
Junction-to-top characterization parameter
0.6
°C/W
ψJB
Junction-to-board characterization parameter
9.9
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
12.1
°C/W
(1)
(2)
(3)
4
(3)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
On JEDEC 4-Layer test board (JESD 51-7) with eight (8) thermal vias.
θJC refers to center of the Exposed Pad on the bottom of the package as the case.
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6.5 Electrical Characteristics
Unless otherwise stated, the following conditions apply: AVIN = PVIN = VIN = 5 V. All Typical limits are for TJ = 25°C only, all
Minimum and Maximum limits apply over the junction temperature (TJ) range of –40°C to 125°C (1). Typical values represent
the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
0.788
0.8
0.812
UNIT
VFB
Feedback pin voltage
VIN = 2.95 V to 5.5 V
ΔVOUT/ΔIOUT
Load Regulation
IOUT = 100 mA to 3 A
V
ICL
Switch Current Limit Threshold
VIN = 3.3 V
4.8
5.3
A
RDS_ON
High-Side Switch On Resistance
ISW = 3.5 A
36
55
mΩ
RDS_ON
Low-Side Switch On Resistance
ISW = 3.5 A
32
52
mΩ
IQ
Operating Quiescent Current
Non-switching, VFB = VCOMP
3.5
6
mA
ISD
Shutdown Quiescent current
VEN = 0 V
90
180
µA
VUVLO
VIN Under Voltage Lockout
Rising VIN
2.7
2.95
V
VUVLO_HYS
VIN Under Voltage Lockout Hysteresis
Falling VIN
45
100
mV
VVCC
VCC Voltage
IVCC = 0 µA
2.45
2.7
2.95
V
ISS
Soft-Start Pin Source Current
VSS/TRK = 0 V
2
4.5
7
µA
VTRACK
SS/TRK Accuracy, VSS - VFB
VSS/TRK = 0.4 V
–10
3
15
mV
FOSCH
Oscillator Frequency
RT = 49.9 kΩ
1350
1500
1650
kHz
FOSCL
Oscillator Frequency
RT = 249 kΩ
450
510
570
kHz
DCMAX
Maximum Duty Cycle
ILOAD = 0 A
TON_TIME
Minimum On Time
TCL_BLANK
Current Sense Blanking Time
0.08
4.3
2.45
%/A
OSCILLATOR
85%
After Rising VSW
100
ns
80
ns
ERROR AMPLIFIER AND MODULATOR
IFB
Feedback pin bias current
VFB = 0.8 V
ICOMP_SRC
COMP Output Source Current
VFB = VCOMP = 0.6 V
80
100
1
µA
ICOMP_SNK
COMP Output Sink Current
VFB = 1.0 V, VCOMP = 0.6 V
80
100
µA
Gm
Error Amplifier Transconductance
ICOMP = ± 50 µA
AVOL
Error Amplifier Voltage Gain
450
510
100
600
nA
µmho
2000
V/V
POWER GOOD
VOVP
Over Voltage Protection Rising Threshold
With respect to VFB
VOVP_HYS
Over Voltage Protection Hysteresis
With respect to VFB
VPGTH
PGOOD Rising Threshold
With respect to VFB
VPGHYS
PGOOD Falling Hysteresis
With respect to VFB
TPGOOD
PGOOD deglitch time
IOL
PGOOD Low Sink Current
VPGOOD = 0.4 V
IOH
PGOOD High Leakage Current
VPGOOD = 5 V
VIH_EN
EN Pin turn-on Threshold
VEN Rising
VEN_HYS
EN Pin Hysteresis
105%
92%
0.6
108%
111%
2%
3%
94%
96%
2%
3%
16
µs
1
mA
5
100
1.18
1.28
nA
ENABLE
1.08
V
66
mV
160
°C
10
°C
THERMAL SHUTDOWN
TSD
Thermal Shutdown
TSD_HYS
Thermal Shutdown Hysteresis
(1)
Minimum and Maximum limits are specified by test, design, or statistical correlation.
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6.6 Typical Characteristics
Unless otherwise specified: CIN = COUT = 100 µF, L = 1.0 µH, VIN = 5 V, VOUT = 1.2 V, RLOAD = 1.2 Ω, fSW = 1 MHz, TA = 25°C
for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
VIN = 5 V
fSW = 1.5 MHz
VIN = 3.3 V
Figure 1. Efficiency vs Load Current
VIN = 5 V
Figure 3. Efficiency vs Load Current
VIN = 5.0 V
fSW 500 kHz
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fSW = 1.0 MHz
Figure 4. Efficiency vs Load Current
VIN = 3.3 V
Figure 5. Efficiency vs Load Current
6
Figure 2. Efficiency vs Load Current
VIN = 3.3 V
fSW = 1.0 MHz
fSW = 1.5 MHz
fSW = 500 kHz
Figure 6. Efficiency vs Load Current
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Typical Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100 µF, L = 1.0 µH, VIN = 5 V, VOUT = 1.2 V, RLOAD = 1.2 Ω, fSW = 1 MHz, TA = 25°C
for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Figure 7. High-Side FET Resistance vs Temperature
Figure 8. Low-Side FET Resistance vs Temperature
Figure 9. Error Amplifier Gain vs Frequency
Figure 10. Line Regulation
Figure 11. Load Regulation
Figure 12. Feedback Pin Voltage vs Temperature
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Typical Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100 µF, L = 1.0 µH, VIN = 5 V, VOUT = 1.2 V, RLOAD = 1.2 Ω, fSW = 1 MHz, TA = 25°C
for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
8
Figure 13. Switching Frequency vs Temperature
Figure 14. Switching Frequency vs RT
Figure 15. Quiescent Current vs VIN (Not Switching)
Figure 16. Shutdown Current vs Temperature
Figure 17. Enable Threshold vs Temperature
Figure 18. UVLO Threshold vs Temperature
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Typical Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100 µF, L = 1.0 µH, VIN = 5 V, VOUT = 1.2 V, RLOAD = 1.2 Ω, fSW = 1 MHz, TA = 25°C
for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Figure 19. Peak Current Limit vs Temperature
Figure 20. Peak Current Limit vs VOUT
Figure 21. Peak Current Limit vs VIN
Figure 22. Load Transient Response
Figure 23. Line Transient Response
Figure 24. Start Up (Soft-Start)
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Typical Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100 µF, L = 1.0 µH, VIN = 5 V, VOUT = 1.2 V, RLOAD = 1.2 Ω, fSW = 1 MHz, TA = 25°C
for efficiency curves, loop gain plots and waveforms, and TJ = 25°C for all others.
Figure 25. Start Up (Tracking)
Figure 26. Short Circuit Input Current vs VIN
Figure 27. Power Down
10
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7 Detailed Description
7.1 Overview
The LM20143 switching regulator features all of the functions necessary to implement an efficient low voltage
buck regulator using a minimum number of external components. This easy to use regulator features two
integrated switches and is capable of supplying up to 3 A of continuous output current. The regulator utilizes
peak current mode control with nonlinear slope compensation to optimize stability and transient response over
the entire output voltage range. Peak current mode control also provides inherent line feed-forward, cycle-bycycle current limiting and easy loop compensation. The switching frequency can be varied from 500 kHz to 1.5
MHz with an external resistor to ground. The device can operate at high switching frequency allowing use of a
small inductor while still achieving efficiencies as high as 96%. The precision internal voltage reference allows
the output to be set as low as 0.8 V. Fault protection features include: current limiting, thermal shutdown, over
voltage protection, and shutdown capability. The device is available in the HTSSOP 16-pin package featuring an
exposed pad to aid thermal dissipation. The LM20143 can be used in numerous applications to efficiently stepdown from a 5 V or 3.3 V bus. The typical application circuit for the LM20143 is shown in Figure 32 in the design
guide.
7.2 Functional Block Diagram
+2.7V
REGULATOR
AVIN
2.7V
VCC
UVLO
+
-
SLOPE COMP
PVIN
COMP
2.7V
CURRENT SENSE
+
5 PA
DISCHARGE (50 Ps)
SS/TRK
ERROR AMP
gm = 510 Pmho
+
+
FB
4.8A
VREF
+
- 800 mV
DISCHARGE
CURRENT
LIMIT
+
-
PVIN
+
-
864 mV
PWM COMPARATOR
OVERVOLTAGE
+
-
PG-L
752 mV
UNDERVOLTAGE
+
-
DIODE
EMULATION
CONTROL
LOGIC
+
-
SW
PVIN
THERMAL
PROTECTION
EN
1.18V
+
-
PGND
OSCILLATOR
RT
PG-L
PGOOD
AGND
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7.3 Feature Description
7.3.1 Peak Current Mode Control
In most cases, the peak current mode control architecture used in the LM20143 only requires two external
components to achieve a stable design. The compensation can be selected to accommodate any capacitor type
or value. The external compensation also allows the user to set the crossover frequency and optimize the
transient performance of the device.
For duty cycles above 50% all current mode control buck converters require the addition of an artificial ramp to
avoid sub-harmonic oscillation. This artificial linear ramp is commonly referred to as slope compensation. What
makes the LM20143 unique is the amount of slope compensation will change depending on the output voltage.
When operating at high output voltages the device will have more slope compensation than when operating at
lower output voltages. This is accomplished in the LM20143 by using a non-linear parabolic ramp for the slope
compensation. The parabolic slope compensation of the LM20143 is much better than the traditional linear slope
compensation because it optimizes the stability of the device over the entire output voltage range.
7.3.2 Precision Enable
The enable (EN) pin allows the output of the device to be enabled or disabled with an external control signal.
This pin is a precision analog input that enables the device when the voltage exceeds 1.2 V (typical). The EN pin
has 100 mV of hysteresis and will disable the output when the enable voltage falls below 1.1 V (typical). If the EN
pin is not used, it should be connected to VIN. Since the enable pin has a precise turn-on threshold it can be
used along with an external resistor divider network from VIN to configure the device to turn-on at a precise input
voltage. The precision enable circuitry will remain active even when the device is disabled.
7.3.3 Current Limit
The precise current limit of the LM20143 is set at the factory to be within 10% over the entire operating
temperature range. This enables the device to operate with smaller inductors that have lower saturation currents.
When the peak inductor current reaches the current limit threshold, an over current event is triggered and the
internal high-side FET turns off and the low-side FET turns on allowing the inductor current to ramp down until
the next switching cycle. For each sequential over-current event, the reference voltage is decremented and PWM
pulses are skipped resulting in a current limit that does not aggressively fold back for brief over-current events,
while at the same time providing frequency and voltage foldback protection during hard short circuit conditions.
7.3.4 Pre-Bias Start Up Capability
The LM20143 is in a pre-biased state when the device starts up with an output voltage greater than zero. This
often occurs in many multi-rail applications such as when powering an FPGA, ASIC, or DSP. In these
applications the output can be pre-biased through parasitic conduction paths from one supply rail to another.
Even though the LM20143 is a synchronous converter it will not pull the output low when a pre-bias condition
exists. During start up the LM20143 will not sink current until the Soft-Start voltage exceeds the voltage on the
FB pin. Since the device can not sink current it protects the load from damage that might otherwise occur if
current is conducted through the parasitic paths of the load.
7.3.5 Soft-Start and Voltage Tracking
The SS/TRK pin is a dual function pin that can be used to set the start up time or track an external voltage
source. The start up or Soft-Start time can be adjusted by connecting a capacitor from the SS/TRK pin to ground.
The Soft-Start feature allows the regulator output to gradually reach the steady state operating point, thus
reducing stresses on the input supply and controlling start up current. If no Soft-Start capacitor is used the device
defaults to the internal Soft-Start circuitry resulting in a start up time of approximately 1 ms. For applications that
require a monotonic start up or utilize the PGOOD pin, an external Soft-Start capacitor is recommended. The
SS/TRK pin can also be set to track an external voltage source. The tracking behavior can be adjusted by two
external resistors connected to the SS/TRK pin as shown in Figure 29.
12
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Feature Description (continued)
7.3.6 Power Good and Overvoltage Fault Handling
The LM20143 has built in under and over voltage comparators that control the power switches. Whenever there
is an excursion in output voltage above the set OVP threshold, the part will terminate the present on-pulse, turnon the low-side FET, and pull the PGOOD pin low. The low-side FET will remain on until either the FB voltage
falls back into regulation or the zero cross detection is triggered which in turn tri-states the FETs. If the output
reaches the UVP threshold the part will continue switching and the PGOOD pin will be asserted and go low.
Typical values for the PGOOD resistor are on the order of 100 kΩ or less. To avoid false tripping during transient
glitches the PGOOD pin has 16 µs of built in deglitch time to both rising and falling edges.
7.3.7 UVLO
The LM20143 has a built-in under-voltage lockout protection circuit that keeps the device from switching until the
input voltage reaches 2.7 V (typical). The UVLO threshold has 45 mV of hysteresis that keeps the device from
responding to power-on glitches during start up. If desired the turn-on point of the supply can be changed by
using the precision enable pin and a resistor divider network connected to VIN as shown in Figure 31. in the
design guide.
7.3.8 Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event that the maximum
junction temperature is exceeded. When activated, typically at 160°C, the LM20143 tri-states the power FETs
and resets soft start. After the junction cools to approximately 150°C, the part starts up using the normal start up
routine. This feature is provided to prevent catastrophic failures from accidental device overheating.
7.4 Device Functional Modes
7.4.1 Light Load Operation
The LM20143 offers increased efficiency when operating at light loads. Whenever the load current is reduced to
a point where the peak to peak inductor ripple current is greater than two times the load current, the part will
enter the diode emulation mode preventing significant negative inductor current. The point at which this occurs is
the critical conduction boundary and can be calculated by Equation 1
IBOUNDARY =
(VIN ± VOUT) x D
2 x L x fSW
(1)
Several diagrams are shown in Figure 28 illustrating continuous conduction mode (CCM), discontinuous
conduction mode, and the boundary condition.
It can be seen that in diode emulation mode, whenever the inductor current reaches zero the SW node will
become high impedance. Ringing will occur on this pin as a result of the LC tank circuit formed by the inductor
and the parasitic capacitance at the node. If this ringing is of concern an additional RC snubber circuit can be
added from the switch node to ground.
At very light loads, usually below 100 mA, several pulses may be skipped in between switching cycles, effectively
reducing the switching frequency and further improving light-load efficiency.
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Switchnode Voltage
Device Functional Modes (continued)
Continuous Conduction Mode (CCM)
VIN
Time (s)
Inductor Current
Continuous Conduction Mode (CCM)
IAVERAGE
Inductor Current
Time (s)
DCM - CCM Boundary
IAVERAGE
Switchnode Voltage
Time (s)
Discontinuous Conduction Mode (DCM)
VIN
Inductor Current
Time (s)
Discontinuous Conduction Mode (DCM)
IPeak
Time (s)
Figure 28. Modes of Operation for LM20143
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Device Functional Modes (continued)
7.4.2 Tracking an External Supply
By using a properly chosen resistor divider network connected to the SS/TRK pin, as shown in Figure 29, the
output of the LM20143 can be configured to track an external voltage source to obtain a simultaneous or
ratiometric start up.
External
Power Supply
VOUT1
LM20143
R1
EN
VOUT2
SS/TRK
R2
Figure 29. Tracking an External Supply
Since the Soft-Start charging current ISS is always present on the SS/TRK pin, the size of R2 should be less than
10 kΩ to minimize the errors in the tracking output. Once a value for R2 is selected the value for R1 can be
calculated using appropriate equation in Figure 30, to give the desired start up. Figure 30 shows two common
start up sequences; the top waveform shows a simultaneous start up while the waveform at the bottom illustrates
a ratiometric start up.
SIMULTANEOUS START UP
VOLTAGE
VOUT1
VOUT2
§VOUT2 ·
-1¸¸ x R2
R1 = ¨¨
© 0.8V ¹
VEN
VOUT2 < 0.8 x VOUT1
TIME
RATIOMETRIC START UP
VOUT1
VOLTAGE
VOUT2
R1 = ( VOUT1 -1) x R2
VEN
TIME
Figure 30. Common Start Up Sequences
A simultaneous start up is preferred when powering most FPGAs, DSPs, or other microprocessors. In these
systems the higher voltage, VOUT1, usually powers the I/O, and the lower voltage, VOUT2, powers the core. A
simultaneous start up provides a more robust power up for these applications since it avoids turning on any
parasitic conduction paths that may exist between the core and the I/O pins of the processor.
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Device Functional Modes (continued)
The second most common power on behavior is known as a ratiometric start up. This start up is preferred in
applications where both supplies need to be at the final value at the same time.
Similar to the Soft-Start function, the fastest start up possible is 1ms regardless of the rise time of the tracking
voltage. When using the track feature the final voltage seen by the SS/TRACK pin should exceed 1V to provide
sufficient overdrive and transient immunity.
7.4.3 Using Precision Enable and Power Good
The precision enable (EN) and power good (PGOOD) pins of the LM20143 can be used to address many
sequencing requirements. The turn-on of the LM20143 can be controlled with the precision enable pin by using
two external resistors as shown in Figure 31.
External
Power Supply
VOUT1
RA
LM20143
VOUT2
EN
RB
Figure 31. Sequencing LM20143 with Precision Enable
The value for resistor RB can be selected by the user to control the current through the divider. Typically this
resistor will be selected to be between 10 kΩ and 1 MΩ. Once the value for RB is chosen the resistor RA can be
solved using Equation 2 to set the desired turn-on voltage.
RA =
VTO
VIH_EN
- 1 x RB
(2)
When designing for a specific turn-on threshold (VTO) the tolerance on the input supply, enable threshold
(VIH_EN), and external resistors needs to be considered to insure proper turn-on of the device.
The LM20143 features an open drain power good (PGOOD) pin to sequence external supplies or loads and to
provide fault detection. This pin requires an external resistor (RPG) to pull PGOOD high while when the output is
within the PGOOD tolerance window. Typical values for this resistor range from 10 kΩ to 100 kΩ.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM20143 is a step down DC/DC converter, typically used for to convert 3.3 V or 5 V rail to lower DC
voltages with a maximum output current of 3 A. The following design procedure can be used to select all the
external components for the LM20143. Alternately, the WEBENCH® Design Tool may be used to complete the
design. By using this tool the user can optimize the design by using an iterative design procedure and selecting
different components from the extensive component database. Along with this datasheet and WEBENCH, the
user can reference the LM20143 Quickstart Calculator tool as a free download. The Quickstart tool is an excel
sheet summary of the Typical Applications section.
8.2 Typical Applications
Several circuit designs can be created for applications with differing requirements. Three examples
demonstrating this are detailed in this section.
8.2.1 3.3-V or 5-V Supply Rail Design
This section walks the designer through the steps necessary to select the external components to build a fully
functional power supply. As with any DC-DC converter numerous trade-offs are possible to optimize the design
for efficiency, size, or performance. These will be taken into account and highlighted throughout this section. The
circuit shown in Figure 32 may be used as a reference to facilitate component selection. Unless otherwise
indicated all formulas assume units of amps (A) for current, farads (F) for capacitance, henries (H) for inductance
and volts (V) for voltages.
LM20143
PVIN
SW
VIN
CIN
L
VOUT
RFB1
EN
RF
AVIN
FB
CF
VIN
COUT
RFB2
RPG
RT
RT
COMP
RC1
CC1
PGOOD
VCC
SS/TRK
PGND GND
VPG
CVCC
CSS
Figure 32. Typical Application Circuit
8.2.1.1 Design Requirements
For this design example, use the parameters listed in Table 1 as the input parameters.
Table 1. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
3.3 V to 5 V
Output voltage
1.2 V
Operating frequency
1.5 MHz
Output current rating
3A
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8.2.1.2 Detailed Design Procedure
Table 2 lists components chosen for a circuit design with the listed specifications. The example calculations in
this section will use values representative of this circuit design.
Table 2. Bill of Materials
Designator
Description
Part Number
Manufacturer
Qty
U1
Synchronous Buck Regulator
LM20143
Texas Instruments
1
CIN
47 µF, 1210, X5R, 6.3 V
GRM32ER60J476ME20
Murata
1
COUT
47 µF, 1210, X5R, 6.3 V
GRM32ER60J476ME20
Murata
1
L
1.2 µH, 17 mΩ
DO1813H-122ML
Coilcraft
1
RPG
10 kΩ, 0603
CRCW06031002F-e3
Vishay-Dale
1
RF
1 Ω, 0603
CRCW06031R0J-e3
Vishay-Dale
1
CF
100 nF, 0603, X7R, 16 V
GRM188R71C104KA01
Murata
1
CVCC
1 µF, 0603, X5R, 6.3 V
GRM188R60J105KA01
Murata
1
RC1
1 kΩ, 0603
CRCW06031001F-e3
Vishay-Dale
1
CC1
4.7 nF, 0603, X7R, 25 V
VJ0603Y472KXXA
Vishay-Vitramon
1
CSS
33 nF, 0603, X7R, 25 V
VJ0603Y333KXXA
Vishay-Vitramon
1
RT
49.9 kΩ, 0603
CRCW06034992F-e3
Vishay-Dale
1
RFB1
4.99 kΩ, 0603
CRCW06034991F-e3
Vishay-Dale
1
RFB2
10 kΩ, 0603
CRCW06031002F-e3
Vishay-Dale
1
8.2.1.2.1 Duty Cycle Calculation
The first equation to calculate for any buck converter is duty-cycle. Ignoring conduction losses associated with
the FETs and parasitic resistances it can be approximated with Equation 3.
D=
VOUT
VIN
(3)
The example design the typical duty cycle is calculated to be 66%.
8.2.1.2.2 Inductor Selection (L)
The inductor value is determined based on the operating frequency, load current, ripple current, and duty cycle.
To optimize the performance and prevent the device from entering current limit at maximum load, the inductance
is typically selected such that the ripple current, ΔiL, is between 25% and 50% of the rated output current. In
general, the inductor ripple current, ΔiL, should be greater than 10% of the rated output current to provide
adequate current sense information for the current mode control loop. Figure 33 illustrates the switch and
inductor ripple current waveforms. Once the input voltage, output voltage, operating frequency, and desired ripple
current are known, the minimum value for the inductor can be calculated with Equation 4.
VOUT ˜ (1 D)
VOUT ˜ (1 D)
dL d
fSW ˜ 0.5 ˜ IOUT(max)
fSW ˜ 0.25 ˜ IOUT(max)
(4)
For the example design the calculated inductance is between .405 μH and 0.810 μH.
The inductor selected should have a saturation current rating greater than the peak current limit of the device
listed in the Electrical Characteristics table. Keep in mind the specified current limit does not account for delay of
the current limit comparator, therefore the current limit in the application may be higher than the specified value.
Peak current can be calculated using Equation 5. This value needs to be less than the part current limit
threshold. If needed, slightly smaller value inductors can be used, however, the peak inductor current should be
kept below the peak current limit of the device.
(VIN VOUT ) ˜ D
IL(max) IOUT
L ˜ fSW
(5)
For the example design the peak inductor current is between 4.563 A and 6.125 A for the selected inductor ripple
current range.
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Due to exceeding current limit and adding flexibility to accommodate the entire output voltage range a inductance
of 1.2 μH was selected, resulting in a peak inductor current of 4.056 A.
VSW
VIN
Time
IL
IL AVG = IOUT
'IL
Time
Figure 33. Switch and Inductor Current Waveforms
8.2.1.2.3 Output Capacitor Selection (COUT)
The output capacitor, COUT, filters the inductor ripple current and provides a source of charge for transient load
conditions. A wide range of output capacitors may be used with the LM20143 that provide excellent performance.
The best performance is typically obtained using ceramic, SP, or OSCON type chemistries. Typical trade-offs are
that the ceramic capacitor provides extremely low ESR to reduce the output ripple voltage and noise spikes,
while the SP and OSCON capacitors provide a large bulk capacitance in a small volume for transient loading
conditions.
When selecting the value for the output capacitor the two performance characteristics to consider are the output
voltage ripple and transient response. The output voltage ripple can be approximated by using Equation 6.
'VOUT = 'iL x RESR +
1
8 x fSW x COUT
where
•
•
•
•
ΔVOUT (V) is the amount of peak to peak voltage ripple at the power supply output
RESR (Ω) is the series resistance of the output capacitor
fSW(Hz) is the switching frequency
COUT (F) is the output capacitance used in the design
(6)
The amount of output ripple that can be tolerated is application specific; however a general recommendation is to
keep the output ripple less than 1% of the rated output voltage. Keep in mind ceramic capacitors are sometimes
preferred because they have very low ESR; however, depending on package and voltage rating of the capacitor
the value of the capacitance can drop significantly with applied voltage. The output capacitor selection will also
affect the output voltage droop during a load transient. The peak droop on the output voltage during a load
transient is dependent on many factors; however, an approximation of the transient droop ignoring loop
bandwidth can be obtained using Equation 7. Both the tolerance and voltage coefficient of the capacitor needs to
be examined when designing for a specific output ripple or transient drop target.
VDROOP = 'IOUTSTEP x RESR +
L x 'IOUTSTEP2
COUT x (VIN - VOUT)
where
•
•
•
•
•
•
•
COUT (F) is the minimum required output capacitance
L (H) is the value of the inductor
VDROOP (V) is the output voltage drop ignoring loop bandwidth considerations
ΔIOUTSTEP (A) is the load step change
RESR (Ω) is the output capacitor ESR
VIN (V) is the input voltage
VOUT (V) is the set regulator output voltage
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For the example design, a 47-µF ceramic capacitor is selected for the output capacitor to provide good transient
and DC performance in a relatively small package. From the technical specifications of this capacitor, the ESR is
roughly 3 mΩ, and the effective in-circuit capacitance is approximately 32 µF (reduced from 47 µF due to the 1.2
V DC bias). With these values, the peak-to-peak voltage ripple on the output when operating from a 5-V input
can be calculated to be 3 mV. A load transient from 1.5 A to 3 A is calculated to create a 27-mV droop.
8.2.1.2.4 Input Capacitor Selection (CIN)
Good quality input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the
switch current during the on-time. In general it is recommended to use a ceramic capacitor for the input as they
provide both a low impedance and small footprint. One important note is to use a good dielectric for the ceramic
capacitor such as X5R or X7R. These provide better over temperature performance and also minimize the DC
voltage derating that occurs on Y5V capacitors. For most applications, a 22-µF, X5R, 6.3-V input capacitor is
sufficient; however, additional capacitance may be required if the connection to the input supply is far from the
PVIN pins. The input capacitor should be placed as close as possible PVIN and PGND pins of the device.
Non-ceramic input capacitors should be selected for RMS current rating and minimum ripple voltage. A good
approximation for the required ripple current rating is given by Equation 8.
IIN-RMS = IOUT D(1 - D)
(8)
As indicated by the RMS ripple current equation, highest requirement for RMS current rating occurs at 50% duty
cycle. For this case, the RMS ripple current rating of the input capacitor should be greater than half the output
current. For best performance, low ESR ceramic capacitors should be placed in parallel with higher capacitance
capacitors to provide the best input filtering for the device.
For the example design the rated RMS current must be at least 1.5 A. A 47-µF ceramic capacitor provides the
necessary input capacitance for the evaluation board. For improved bypassing, a small 1-µF high frequency
capacitor is placed in parallel with the 47-µF bulk capacitor to filter high frequency noise pulses on the supply.
8.2.1.2.5 Setting the Output Voltage (RFB1, RFB2)
The resistors RFB1 and RFB2 are selected to set the output voltage for the device. For most applications, RFB1
should be between 4.99 kΩ to 49.9 kΩ.
RFB1 =
VOUT
0.8
- 1 x RFB2
(9)
For the example design RFB1 = 4 kΩ and RFB2 = 10 kΩ to set the output voltage at 1.2 V.
Table 3 contains common output voltage values for RFB1 and RFB2.
Table 3. Suggested Values for RFB1 and RFB2
20
RFB1(kΩ)
RFB2(kΩ)
VOUT
short
open
0.8
4.99
10
1.2
8.87
10.2
1.5
12.7
10.2
1.8
21.5
10.2
2.5
31.6
10.2
3.3
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8.2.1.2.6 Adjusting the Operating Frequency (RT)
The LM20143 supports a wide range of switching frequencies from 500 kHz to 1.5 MHz. The operating frequency
of the LM20143 can be adjusted by connecting a resistor from the RT pin to ground. The choice of switching
frequency is usually a compromise between efficiency and the size of the circuit. Lower switching frequency
implies reduced switching losses, usually resulting in higher overall efficiency. However, higher switching
frequency allows use of smaller inductor and output capacitor components, resulting in a smaller design. The
optimal switching frequency is usually a tradeoff in a given application and thus should be determined on a caseby-case basis. For the design example a relatively high switching frequency was selected to minimize circuit size.
Equation 10 can be used to calculate the value of RT for a given operating frequency.
RT =
154750
fSW
- 55
where
•
•
fSW is the switching frequency in kHz
RT is the frequency adjust resistor in kΩ
(10)
Please refer to the curve Oscillator Frequency verses RT in the typical performance characteristics section. If the
RT resistor is omitted the device will not operate.
For the example design RT was calculated to be 49.9 kΩ to give a switching frequency of 1.5 MHz.
8.2.1.2.7 AVIN Filtering Components (CF and RF)
To prevent high frequency noise spikes from disturbing the sensitive analog circuitry connected to the AVIN and
AGND pins, a high frequency RC filter is required between PVIN and AVIN. These components are shown in
Figure 32 as CF and RF. The required value for RF is 1 Ω. CF must be used. Recommended value of CF is 1.0
µF. The filter capacitor, CF should be placed as close to the IC as possible with a direct connection from AVIN to
AGND. A good quality X5R or X7R ceramic capacitor should be used for CF.
For the example design, RF 1 Ω and CF is 1.0 µF, providing greater than 16 dB of attenuation at the set 1.5-MHz
switching frequency.
8.2.1.2.8 Sub-Regulator Bypass Capacitor (CVCC)
The capacitor at the VCC pin provides noise filtering and stability for the internal sub-regulator. The
recommended value of CVCC should be no smaller than 1 µF and no greater than 10 µF. The capacitor should be
a good quality ceramic X5R or X7R capacitor. In general, a 1-µF ceramic capacitor is recommended for most
applications. In the example design, CVCC is 1 µF.
8.2.1.2.9 Setting the Start Up Time (CSS)
The addition of a capacitor connected from the SS pin to ground sets the time at which the output voltage will
reach the final regulated value. Larger values for CSS will result in longer start up times. While the Soft-Start
capacitor can be sized to meet many start up requirements, there are limitations to its size. The Soft-Start time
can never be faster than 1 ms due to the internal default 1-ms start up time. When the device is enabled there is
an approximate time interval of 50 μs when the Soft-Start capacitor will be discharged just prior to the Soft-Start
ramp. If the enable pin is rapidly pulsed or the Soft-Start capacitor is large there may not be enough time for CSS
to completely discharge resulting in start up times less than predicted. To aid in discharging of Soft-Start
capacitor during long disable periods an external 1-MΩ resistor from SS/TRK to ground can be used without
greatly affecting the start up time. Equation 11 can be used to calculate the desired soft start time. For the
LM20143, ISS is nominally 5 µA, that may be found in the electrical characteristics table.
0.8V ˜ CSS
tSS
ISS
(11)
For the example design the start-up time was set to be around 5ms resulting in a 33 nF capacitor.
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Table 4 shows values of CSS for some common start up times.
Table 4. Start Up Times for Different Soft-Start Capacitors
Start Up Time (ms)
CSS (nF)
1
none
5
33
10
68
15
100
20
120
8.2.1.2.10 Loop Compensation (RC1, CC1)
The purpose of loop compensation is to meet static and dynamic performance requirements while maintaining
adequate stability. Optimal loop compensation depends on the output capacitor, inductor, load, and the device
itself. Table 5 gives values for the compensation network that will result in a stable system when using a 100-µF,
6.3-V ceramic X5R output capacitor and 1-µH inductor.
Table 5. Recommended Compensation for COUT = 100 µF, L = 1 µH & fSW = 1 MHz
VIN
VOUT
CC1 (nF)
RC1 (kΩ)
5.00
3.30
4.7
17.8
5.00
2.50
4.7
12.1
5.00
1.80
4.7
7.68
5.00
1.50
4.7
5.9
5.00
1.20
4.7
3.57
5.00
0.80
4.7
1.58
3.30
2.50
4.7
13
3.30
1.80
4.7
9.76
3.30
1.50
4.7
6.49
3.30
1.20
4.7
4.64
3.30
0.80
4.7
1.58
If the desired solution differs from those provided, the loop transfer function should be analyzed to optimize the
loop compensation. The overall loop transfer function is the product of the power stage and the feedback network
transfer functions. For stability purposes, the objective is to have a loop gain slope that is –20 db/decade from a
very low frequency to beyond the crossover frequency. Figure 34 shows the transfer functions for power stage,
feedback/compensation network, and the resulting closed loop system for the LM20143.
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Output Filter Pole, fP(FIL)
AM
0 dB
Output Filter Zero, fZ(FIL)
Complex Double Pole, fP(MOD)
Modulator and Output Filter
Transfer Function
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Pole, fP2(EA)
0 dB
Error Amp Zero, fZ(EA)
AEA + AM
Error Amp Pole, fP(EA)
0 dB
Complex Double Pole, fP(MOD)
fC
Error Amplifier
Transfer Function
Optional Error Amp
Compensated Closed
Loop Transfer Function
GAIN (dB)
Error Amp Pole, fP1(EA)
AEA
fSW/2
FREQUENCY (Hz)
Figure 34. LM20143 Loop Compensation
The power stage transfer function is dictated by the modulator, output LC filter, and load; while the feedback
transfer function is set by the feedback resistor ratio, error amp gain, and external compensation network.
To achieve a –20 dB/decade slope, the error amplifier zero, located at fZ(EA), should positioned to cancel the
output filter pole (fP(FIL)). An additional error amp pole, located at fP2(EA), can be added to cancel the output filter
zero at fZ(FIL). Cancellation of the output filter zero is recommended if larger value, non-ceramic output capacitors
are used.
Compensation of the LM20143 is achieved by adding an RC network as shown in Figure 35.
LM20143
COMP
RC1
CC2
(optional)
CC1
Figure 35. Compensation Network for LM20143
A good starting value for CC1 for most applications is 4.7 nF. Once the value of CC1 is chosen the value of RC
should be calculated using Equation 12 to cancel the output filter pole (fP(FIL)) as shown in Figure 34.
RC1 =
-1
15 x D
CC1
IOUT
1-D
x
+
+
VIN
COUT
VOUT fSW x L
(12)
A higher crossover frequency can be obtained, usually at the expense of phase margin, by lowering the value of
CC1 and recalculating the value of RC1. Likewise, increasing CC1 and recalculating RC1 will provide additional
phase margin at a lower crossover frequency. As with any attempt to compensate the LM20143 the stability of
the system should be verified for desired transient droop and settling time.
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If the output filter zero, fZ(FIL) approaches the crossover frequency (FC), an additional capacitor (CC2) should be
placed at the COMP pin to ground. This capacitor adds a pole to cancel the output filter zero assuring the
crossover frequency will occur before the double pole at fSW / 2 degrades the phase margin. The output filter
zero is set by the output capacitor value and ESR as shown in Equation 13.
fZ(FIL) =
1
2 x S x COUT x RESR
(13)
If needed, the value for CC2 should be calculated using Equation 14.
CC2 =
COUT x RESR
RC1
where
•
•
RESR is the output capacitor series resistance
RC1 is the calculated compensation resistance
(14)
For the example design RC1 is 1 kΩ, CC1 is 4.7 nF and CC2is left open.
8.2.1.3 Application Curves
VIN = 5 V
VOUT = 1.2 V
IOUT = 3 A
VIN = 5 V
VOUT = 1.2 V
Time Scale = 200 µs/div
Figure 37. Load Transient
Figure 36. Startup
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8.2.2 5-V Supply Rail Design
L
LM20143
PVIN
SW
VIN
CIN
VOUT
RFB1
EN
RF
AVIN
FB
CF
VIN
COUT
RFB2
RPG
RT
RT
COMP
RC1
CC1
PGOOD
VCC
SS/TRK
PGND GND
VPG
CVCC
CSS
Figure 38. Typical Application Circuit
8.2.2.1 Design Requirements
For this design example, use the parameters listed in Table 6 as the input parameters.
Table 6. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
5V
Output voltage
3.3 V
Operating frequency
750 kHz
Output current rating
3A
8.2.2.2 Detailed Design Procedure
Table 7 lists a selection of components using the design procedure explained in 3.3-V or 5-V Supply Rail Design
for a circuit design with the listed specifications.
Table 7. Bill of Materials
Designator
Description
Part Number
Manufacturer
Qty
U1
Synchronous Buck Regulator
LM20143
Texas Instruments
1
CIN
47 µF, 1210, X5R, 6.3 V
GRM32ER60J476ME20
Murata
1
COUT
100 µF, 1210, X5R, 6.3 V
GRM32ER60J107ME20
Murata
1
L
1.5 µH, 8.1 mΩ
MSS1038-152NL
Coilcraft
1
RF
1 Ω, 0603
CRCW06031R0J-e3
Vishay-Dale
1
CF
100 nF, 0603, X7R, 16 V
GRM188R71C104KA01
Murata
1
CVCC
1 µF, 0603, X5R, 6.3 V
GRM188R60J105KA01
Murata
1
RC1
10 kΩ, 0603
CRCW06031002F-e3
Vishay-Dale
1
CC1
3.3 nF, 0603, X7R, 25 V
VJ0603Y332KXXA
Vishay-Vitramon
1
CSS
33 nF, 0603, X7R, 25 V
VJ0603Y333KXXA
Vishay-Vitramon
1
RT
150 kΩ, 0603
CRCW06031503F-e3
Vishay-Dale
1
RFB1
31.6 kΩ, 0603
CRCW06033162F-e3
Vishay-Dale
1
RFB2
10.2 kΩ, 0603
CRCW06031022F-e3
Vishay-Dale
1
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8.2.3 3-V Supply Rail Design
L
LM20143
PVIN
SW
VIN
CIN
VOUT
RFB1
EN
RF
AVIN
FB
CF
VIN
COUT
RFB2
RPG
RT
RT
COMP
RC1
CC1
PGOOD
VCC
SS/TRK
PGND GND
VPG
CVCC
CSS
Figure 39. Typical Application Circuit
8.2.3.1 Design Requirements
For this design example, use the parameters listed in Table 8 as the input parameters.
Table 8. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
3.3 V
Output voltage
1.2 V
Operating frequency
750 kHz
Output current rating
3A
8.2.3.2 Detailed Design Procedure
Table 9 lists a selection of components using the design procedure explained in 3.3-V or 5-V Supply Rail Design
for a circuit design with the listed specifications.
Table 9. Bill of Materials
Designator
Description
Part Number
Manufacturer
Qty
U1
Synchronous Buck Regulator
LM20143
Texas Instruments
1
CIN
47 µF, 1210, X5R, 6.3 V
GRM32ER60J476ME20
Murata
1
COUT
100 µF, 1210, X5R, 6.3 V
GRM32ER60J107ME20
Murata
1
L
1.2 µH, 17 mΩ
DO1813H-122ML
Coilcraft
1
RF
1 Ω, 0603
CRCW06031R0J-e3
Vishay-Dale
1
CF
100 nF, 0603, X7R, 16 V
GRM188R71C104KA01
Murata
1
CVCC
1 µF, 0603, X5R, 6.3 V
GRM188R60J105KA01
Murata
1
RC1
2 kΩ, 0603
CRCW06032001F-e3
Vishay-Dale
1
CC1
4.7 nF, 0603, X7R, 25 V
VJ0603Y472KXXA
Vishay-Vitramon
1
CSS
33 nF, 0603, X7R, 25 V
VJ0603Y333KXXA
Vishay-Vitramon
1
RT
150 kΩ, 0603
CRCW06031503F-e3
Vishay-Dale
1
RFB1
4.99 kΩ, 0603
CRCW06034991F-e3
Vishay-Dale
1
RFB2
10 kΩ, 0603
CRCW06031002F-e3
Vishay-Dale
1
26
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SNVS528H – OCTOBER 2007 – REVISED JANUARY 2016
9 Power Supply Recommendations
The LM20143 converter is designed to operate from either a 3.3-V or 5-V rail. The input supply must be
compatible with the Absolute Maximum Ratings and Recommended Operating Conditions. Further more the
input supply must be able to supply the required average input current to the regulated load. The average input
current can be estimated with Equation 15.
CC2 =
COUT x RESR
RC1
where
•
η is efficiency
(15)
If the regulator is connected to the input supply through long wires or PCB traces with large impedance, special
care is required to achieve good performance. The parasitic inductance and resistance of the input cables may
have an adverse effect on the operation of the regulator. The parasitic inductance, in combination with the low
ESR ceramic input capacitors, can form an under-damped resonant circuit. This circuit may cause overvoltage
transients at the PVIN pin each time the input supply is cycled on and off. The parasitic resistance causes the
PVIN voltage to dip when the load on the regulator is switched on or exhibits a transient. If the regulator is
operating close to the minimum input voltage, this dip may cause false UVLO fault triggering and system reset.
The best way to solve these types of issues is to reduce the distance from the input supply to the regulator
and/or use an aluminum or tantalum input capacitor in parallel with the ceramics. The moderate ESR of the
electrolytic capacitors helps to damp the input resonant circuit and reduce any voltage overshoots. A value in the
range of 20 μF to 100 μF is usually sufficient to provide input damping and help to hold the input voltage steady
during large load transients.
Sometimes, an EMI input filter is used in front of the regulator. This can lead to instability, as well as some of the
effects mentioned above, unless it is carefully designed. The user guide Simple Success with Conducted EMI for
DC-DC Converters, SNVA489, provides helpful suggestions when designing an input filter for any switching
regulator.
10 Layout
10.1 Layout Guidelines
PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance
of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability.
Good layout can be implemented by following a few simple design rules.
1. Minimize area of switched current loops. In a buck regulator there are two loops where currents are switched
very fast. The first loop starts from the input capacitor, to the regulator VIN pin, to the regulator SW pin, to
the inductor then out to the output capacitor and load. The second loop starts from the output capacitor
ground, to the regulator PGND pins, to the inductor and then out to the load (see Figure 40). To minimize
both loop areas the input capacitor should be placed as close as possible to the PVIN pin. Grounding for
both the input and output capacitor should consist of a small localized top side plane that connects to PGND
and the die attach pad (DAP). The inductor should be placed as close as possible to the SW pin and output
capacitor.
2. Minimize the copper area of the switch node. Since the LM20143 has the SW pins on opposite sides of the
package it is recommended to via these pins down to the bottom or internal layer with 2 to 4 vias on each
SW pin. The SW pins should be directly connected with a trace that runs across the bottom of the package.
To minimize IR losses this trace should be no smaller that 50 mils wide, but no larger than 100 mils wide to
keep the copper area to a minimum. In general the SW pins should not be connected on the top layer since it
could block the ground return path for the power ground. The inductor should be placed as close as possible
to one of the SW pins to further minimize the copper area of the switch node.
3. Have a single point ground for all device analog grounds located under the DAP. The ground connections for
the compensation, feedback, and Soft-Start components should be connected together then routed to the
AGND pin of the device. The AGND pin should connect to PGND under the DAP. This prevents any
switched or load currents from flowing in the analog ground plane. If not properly handled poor grounding
can result in degraded load regulation or erratic switching behavior.
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Layout Guidelines (continued)
4. Minimize trace length to the FB pin. Since the feedback node can be high impedance the trace from the
output resistor divider to FB pin should be as short as possible. This is most important when high value
resistors are used to set the output voltage. The feedback trace should be routed away from the SW pin and
inductor to avoid contaminating the feedback signal with switch noise.
5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or
output of the converter and can improve efficiency. If voltage accuracy at the load is important make sure
feedback voltage sense is made at the load. Doing so will correct for voltage drops at the load and provide
the best output accuracy.
6. Provide adequate device heatsinking. Use as many vias as is possible to connect the DAP to the power
plane heatsink. For best results use a 4x4 via array with a minimum via diameter of 12 mils. See Thermal
Considerations section to insure enough copper heatsinking area is used to keep the junction temperature
below 125°C.
LM20143
PVIN
L
SW
VOUT
CIN
COUT
PGND
LOOP1
LOOP2
Figure 40. Schematic of LM20143 Highlighting Layout Sensitive Nodes
10.2 Layout Example
AGND
RT
CSS
CC1
CC2
RFB1
RFB2
RT
SS/TRK
AGND
FB
RC1
RF
RPG
PGOOD
COMP
NC
LM20143
CF
AVIN
VPGOOD
VCC
CVCC
EN
VIN
PVIN
PGND
PVIN
PGND
SW
VEN
SW
CIN
GND
SW
Via to GND
Via to VOUT
Via to PGOOD
Via to SW
L
COUT
GND
VOUT
Figure 41. Example Layout
28
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SNVS528H – OCTOBER 2007 – REVISED JANUARY 2016
10.3 Thermal Considerations
The thermal characteristics of the LM20143 are specified using the parameter θJA, which relates the junction
temperature to the ambient temperature. Although the value of θJA is dependant on many variables, it still can be
used to approximate the operating junction temperature of the device.
To obtain an estimate of the device junction temperature, one may use Equation 16 and Equation 17.
TJ = PDθJA + TA
(16)
PD = PIN x (1 - Efficiency) - 1.1 x IOUT2 x DCR
where
•
•
•
•
•
•
TJ is the junction temperature in °C
PIN is the input power in Watts (PIN = VIN x IIN)
θJA is the junction to ambient thermal resistance for the LM20143
TA is the ambient temperature in °C
IOUT is the output load current
DCR is the inductor series resistance
(17)
It is important to always keep the operating junction temperature (TJ) below 125°C for reliable operation. If the
junction temperature exceeds 160°C the device will cycle in and out of thermal shutdown. If thermal shutdown
occurs it is a sign of inadequate heatsinking or excessive power dissipation in the device.
Figure 42 provides a better approximation of the θJA for a given PCB copper area. The PCB heatsink area
consists of 2 oz. copper located on the bottom layer of the PCB directly under the HTSSOP exposed pad. The
bottom copper area is connected to the HTSSOP exposed pad by means of a 4 x 4 array of 12 mil thermal vias.
Figure 42. Thermal Resistance vs PCB Area
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SNVS528H – OCTOBER 2007 – REVISED JANUARY 2016
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
LM20143 Quickstart Design Tool, http://www.ti.com/tool/lm20143design-calc
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• AN-1691 LM20143 Evaluation Board, SNVA277
• AN-2162: Simple Success with Conducted EMI from DC-DC Converters, SNVA489
11.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 10. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
LM20143
Click here
Click here
Click here
Click here
Click here
LM20143-Q1
Click here
Click here
Click here
Click here
Click here
11.4 Trademarks
PowerWise is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
30
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PACKAGE OPTION ADDENDUM
www.ti.com
3-Nov-2015
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LM20143MH/NOPB
ACTIVE
HTSSOP
PWP
16
92
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
MH
LM20143MHE/NOPB
ACTIVE
HTSSOP
PWP
16
250
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
MH
LM20143MHX/NOPB
ACTIVE
HTSSOP
PWP
16
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
MH
LM20143QMH/NOPB
ACTIVE
HTSSOP
PWP
16
92
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
QMH
LM20143QMHE/NOPB
ACTIVE
HTSSOP
PWP
16
250
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
QMH
LM20143QMHX/NOPB
ACTIVE
HTSSOP
PWP
16
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
20143
QMH
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
3-Nov-2015
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM20143, LM20143-Q1 :
• Catalog: LM20143
• Automotive: LM20143-Q1
NOTE: Qualified Version Definitions:
• Catalog - TI's standard catalog product
• Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
LM20143MHE/NOPB
HTSSOP
PWP
16
250
178.0
12.4
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
6.95
5.6
1.6
8.0
12.0
Q1
LM20143MHX/NOPB
HTSSOP
PWP
16
2500
330.0
12.4
6.95
5.6
1.6
8.0
12.0
Q1
LM20143QMHE/NOPB
HTSSOP
PWP
16
250
178.0
12.4
6.95
5.6
1.6
8.0
12.0
Q1
LM20143QMHX/NOPB
HTSSOP
PWP
16
2500
330.0
12.4
6.95
5.6
1.6
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2015
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM20143MHE/NOPB
HTSSOP
PWP
LM20143MHX/NOPB
HTSSOP
PWP
16
250
210.0
185.0
35.0
16
2500
367.0
367.0
35.0
LM20143QMHE/NOPB
HTSSOP
PWP
LM20143QMHX/NOPB
HTSSOP
PWP
16
250
210.0
185.0
35.0
16
2500
367.0
367.0
35.0
Pack Materials-Page 2
MECHANICAL DATA
PWP0016A
MXA16A (Rev A)
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