19-2795; Rev 2; 6/07 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit The MAX8545/MAX8546/MAX8548 are voltage-mode pulse-width-modulated (PWM), step-down DC-DC controllers ideal for a variety of cost-sensitive applications. They drive low-cost n-channel MOSFETs for both the high-side switch and synchronous rectifier, and require no external current-sense resistor. These devices can supply output voltages as low as 0.8V. The MAX8545/MAX8546/MAX8548 have a wide 2.7V to 28V input range, and do not need any additional bias voltage. The output voltage can be precisely regulated from 0.8V to 0.83 x VIN. These devices can provide efficiency up to 95%. Lossless short-circuit and current-limit protection is provided by monitoring the RDS(ON) of the low-side MOSFET. The MAX8545 and MAX8548 have a current-limit threshold of 320mV, while the MAX8546 has a current-limit threshold of 165mV. All devices feature foldback-current capability to minimize power dissipation under short-circuit condition. Pulling the COMP/EN pin low with an open-collector or low-capacitance, opendrain device can shut down all devices. The MAX8545/MAX8546 operate at 300kHz and the MAX8548 operates at 100kHz. The MAX8545/MAX8546/ MAX8548 are compatible with low-cost aluminum electrolytic capacitors. Input undervoltage lockout prevents proper operation under power-sag operations to prevent external MOSFETs from overheating. Internal soft-start is included to reduce inrush current. These devices are offered in space-saving 10-pin µMAX® packages. Applications Set-Top Boxes Graphic Card and Video Supplies Desktops and Desknotes PCI Express Power Supplies Telecom Power Supplies Notebook Docking Station Supplies Cable Modems and Routers Networking Power Supplies Features ♦ 2.7V to 28V Input Range ♦ Foldback Short-Circuit Protection ♦ No Additional Bias Supply Needed ♦ 0.8V to 0.83 x VIN Output ♦ Up to 95% Efficiency ♦ Low-Cost External Components ♦ No Current-Sense Resistor ♦ All n-Channel MOSFET Design ♦ Adaptive Gate Drivers Eliminate Shoot-Through ♦ Lossless Overcurrent and Short-Circuit Protection ♦ 300kHz Switching Frequency (MAX8545/MAX8546) ♦ 100kHz Switching Frequency (MAX8548) ♦ Pin-Compatible with the MAX1967 ♦ Thermal Shutdown Selector Guide SWITCHING FREQUENCY (kHz) CURRENT-LIMIT THRESHOLD (mV) MAX8545 300 -320 MAX8546 300 -165 MAX8548 100 -320 PART Typical Operating Circuit INPUT 2.7V TO 28V VCC VL VIN Ordering Information TEMP RANGE PINPACKAGE PKG CODE MAX8545EUB -40°C to +85°C 10 µMAX U10-2 MAX8545EUB+ -40°C to +85°C 10 µMAX U10-2 MAX8546EUB -40°C to +85°C 10 µMAX U10-2 MAX8546EUB+ -40°C to +85°C 10 µMAX U10-2 MAX8548EUB -40°C to +85°C 10 µMAX U10-2 MAX8548EUB+ -40°C to +85°C 10 µMAX U10-2 PART +Denotes a lead-free package. µMAX is a registered trademark of Maxim Integrated Products, Inc. BST DH MAX8545 MAX8546 MAX8548 COMP/ EN OFF LX OUTPUT 0.8V TO 0.9 x VIN UP TO 6A DL GND FB ON OPTIONAL Pin Configuration appears at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX8545/MAX8546/MAX8548 General Description MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit ABSOLUTE MAXIMUM RATINGS (All voltages referenced to GND unless otherwise noted.) VIN to GND ............................................................-0.3V to +30V VCC to GND .............................-0.3V, lower of 6V or (VL + 0.3V) FB to GND ................................................................-0.3V to +6V BST to GND ............................................................-0.3V to +36V VL, DL, COMP to GND ..............................-0.3V to (VCC + 0.3V) BST to LX..................................................................-0.3V to +6V DH to LX ....................................................-0.3V to (VBST + 0.3V) VL Short to GND ......................................................................5s LX to GND ................................................................-1V to +30V Input Current (any pin) .....................................................±50mA Continuous Power Dissipation (TA = +70°C) 10-Pin µMAX (derate 5.6mW/°C above +70°C) ..........444mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VL = VCC = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER SYMBOL CONDITIONS MIN TYP MAX VCC = VL, VIN separate from VCC 4.9 28.0 VIN = VL = VCC 2.7 5.5 VIN Undervoltage Lockout (UVLO) Trip Level Rising and falling edge, hysteresis = 2% 2.35 VIN Operating Supply Current VFB = 0.88V (no switching) VL Output Voltage 5.5V < VIN < 28V, VCC = VL, 1mA < ILOAD < 25mA Thermal Shutdown Rising temperature, typical hysteresis = 10°C (Note 1) VIN Operating Range VIN 4.7 UNITS V 2.50 2.66 V 0.7 1.2 mA 5 5.3 V °C +160 OSCILLATOR Frequency fOSC Minimum Duty Cycle DCMIN Maximum Duty Cycle DCMAX MAX8545, MAX8546 250 300 360 MAX8548 80 100 120 DH output, MAX8545, MAX8546 5 MAX8548 10 DH output, MAX8545, MAX8546 83 86 MAX8548 90 95 kHz % % SOFT-START Digital Ramp Period Soft-Start Levels 2 MAX8545, MAX8546 MAX8548 MAX8545, MAX8546 MAX8548 6.6 10.2 ms VOUT / 64 VOUT / 32 V _______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit MAX8545/MAX8546/MAX8548 ELECTRICAL CHARACTERISTICS (continued) (VIN = VL = VCC = 5V, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER SYMBOL CONDITIONS MIN TYP MAX 2.7V < VCC < 5.5V, 0°C to +85°C 0.787 0.800 0.815 2.7V < VCC < 5.5V, -40°C to +85°C 0.782 0.800 0.815 UNITS ERROR AMPLIFIER FB Regulation Voltage FB to COMP/EN Gain 4000 FB to COMP/EN Transconductance -5µA < ICOMP/EN < +5µA FB Input Bias Current VFB = 0.88V COMP/EN Source Current VCOMP/EN = 0V Current-Limit Threshold Voltage (Across Low-Side MOSFET) Foldback Current-Limit Threshold Voltage (Across Low-Side MOSFET) When Output is Short V V /V 70 108 160 µS 15 1 2 µA 46 100 µA LX to GND, MAX8545, MAX8548, VFB = 0.8V -355 -320 -280 LX to GND MAX8546, VFB = 0.8V -185 -165 -140 LX to GND, VFB = 0V, MAX8545, MAX8548 -105 -75 -45 MAX8546, LX to GND, VFB = 0 -53 -38 -22 mV mV MOSFET DRIVERS Break-Before-Make Time Rising edge, DH going low to DL going high 96 Falling edge, DL going low to DH going high 28 ns DH On-Resistance in Low State 1.6 4 Ω DH On-Resistance in High State 2.5 5.5 Ω DL On-Resistance in Low State 1.1 2.5 Ω DL On-Resistance in High State 2.5 5.5 Ω BST Leakage Current VBST = 33V, VLX = 28V, VFB = 0.88V 0 50 µA LX Leakage Current VBST = 33V, VLX = 28V, VFB = 0.88V 33 100 µA Note 1: Thermal shutdown disables the buck regulator when the die reaches this temperature. Soft-start is reset but the VL regulator remains on. _______________________________________________________________________________________ 3 Typical Operating Characteristics (VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.) EFFICIENCY vs. LOAD CURRENT 50 40 30 VOUT = 1.8V 70 60 50 40 0.1 0 10 1 0.01 10 1 0.01 VOUT = 1.8V 90 80 EFFICIENCY (%) VOUT = 3.3V 60 50 40 VOUT = 1.8V 70 60 VOUT = 1.2V 50 40 0.1 1 10 40 10 1 EFFICIENCY vs. LOAD CURRENT EFFICIENCY vs. LOAD CURRENT CHANGE IN OUTPUT VOLTAGE vs. LOAD CURRENT 50 80 VOUT = 1.2V 40 50 20 20 1 LOAD CURRENT (A) 2.510 2.505 2.500 2.495 2.485 VIN = 17V 0 10 VOUT = 2.5V VIN = 12V 2.515 2.490 10 VIN = 12V 0.1 VOUT = 1.2V 40 30 0 VOUT = 1.8V 60 30 10 VOUT = 2.5V 70 2.520 OUTPUT VOLTAGE (V) 60 VOUT = 3.3V 90 MAX8545/46/48 toc08 VOUT = 1.8V (CIRCUIT OF FIGURE 2; TABLE 2b) 100 EFFICIENCY (%) VOUT = 2.5V 70 0.01 0.1 LOAD CURRENT (A) MAX8545/46/48 toc07 80 0.01 10 1 LOAD CURRENT (A) VOUT = 3.3V 90 0.1 VIN = 5V 0 LOAD CURRENT (A) (CIRCUIT OF FIGURE 2; TABLE 2b) 100 10 VIN = 3.3V 0.01 VOUT = 1.2V 50 20 0.01 VOUT = 1.8V 60 30 0 VOUT = 2.5V 70 20 0 VOUT = 3.3V 80 20 10 (CIRCUIT OF FIGURE 1; TABLE 1b) 90 30 VIN = 17V 10 1 EFFICIENCY vs. LOAD CURRENT 100 30 10 0.1 LOAD CURRENT (A) (CIRCUIT OF FIGURE 1; TABLE 1b) 100 MAX8545/46/48 toc04 90 EFFICIENCY (%) 0.1 EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 2; TABLE 2a) 70 VIN = 12V LOAD CURRENT (A) EFFICIENCY vs. LOAD CURRENT 80 40 0 LOAD CURRENT (A) 100 50 10 VIN = 5V EFFICIENCY (%) 0.01 60 20 10 MAX8545/46/48 toc05 0 VOUT = 1.8V 70 30 20 VIN = 3.3V VOUT = 1.2V 10 VOUT = 3.3V 80 30 20 4 90 MAX8545/46/48 toc06 60 VOUT = 3.3V 80 EFFICIENCY (%) 70 90 EFFICIENCY (%) EFFICIENCY (%) 80 (CIRCUIT OF FIGURE 2; TABLE 2a) 100 MAX8545/46/48 toc09 MAX8545/46/48 toc01 90 EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 1; TABLE 1a) 100 MAX8545/46/48 toc02 (CIRCUIT OF FIGURE 1; TABLE 1a) MAX8545/46/48 toc03 EFFICIENCY vs. LOAD CURRENT 100 EFFICIENCY (%) MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit 0.01 0.1 1 LOAD CURRENT (A) 2.480 10 1 2 3 4 LOAD CURRENT (A) _______________________________________________________________________________________ 5 6 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit CHANGE IN OUTPUT VOLTAGE vs. INPUT VOLTAGE CHANGE IN OUTPUT VOLTAGE vs. INPUT VOLTAGE MAX8545/46/48 toc11 1.83 ILOAD = 6V OUTPUT VOLTAGE (V) ILOAD = 6V OUTPUT VOLTAGE (V) 2.52 MAX8545/46/48 toc10 1.84 1.82 1.81 1.80 2.51 2.50 2.49 1.79 2.48 1.78 3.0 3.5 4.0 4.5 10 12 14 INPUT VOLTAGE (V) FREQUENCY vs. INPUT VOLTAGE 20 22 24 MAX8545/46/48 toc12 VOUT = 2.5V NO LOAD MAX8545/ MAX8546 310 VIN = 12V VOUT = 2.5V NO LOAD MAX8545/ MAX8546 306 FREQUENCY (kHz) FREQUENCY (kHz) 18 FREQUENCY vs. TEMPERATURE 310 306 16 INPUT VOLTAGE (V) 302 298 294 MAX8545/46/48 toc13 2.5 302 298 294 290 2.70 7.76 12.82 17.88 22.94 290 -40.00 28.00 -15.00 INPUT VOLTAGE (V) 10.00 35.00 60.00 85.00 TEMPERATURE (°C) LOAD TRANSIENT RESPONSE MAX8545 toc14 VIN = 17V VOUT = 2.5V VOUT AC COUPLED 100mV/div IOUT 5A/div 0 40µs/div _______________________________________________________________________________________ 5 MAX8545/MAX8546/MAX8548 Typical Operating Characteristics (continued) (VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.) Typical Operating Characteristics (continued) (VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.) SHUTDOWN WAVEFORM STARTUP WAVEFORM MAX8545 toc16 MAX8545 toc15 VIN 5V/div VIN 5V/div VOUT 1V/div VOUT 1V/div INDUCTOR CURRENT 2A/div INDUCTOR CURRENT 5A/div ILOAD = 3A ILOAD = 3A 2ms/div 1ms/div SHORT-CIRCUIT WAVEFORM SHORT-CIRCUIT WAVEFORM MAX8545 toc17 0 0 MAX8545 toc18 VIN 20V/div VIN 5V/div IIN 2A/div IIN 10A/div VOUT 2V/div VOUT 2V/div 0 INDUCTOR CURRENT 5A/div IOUT 5A/div 0 1ms/div GAIN AND PHASE vs. FREQUENCY MAX8545/46/48 toc19 60 40 GAIN 0 150 -20 120 -40 -60 PHASE 90 PHASE (DEGREES) 180 20 GAIN (dB) MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit -80 -100 -120 -140 60 VIN = 17V, VOUT = 2.5V ILOAD = 6A 0.1 1 10 30 100 FREQUENCY (kHz) 6 _______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit PIN NAME FUNCTION 1 COMP/EN 2 FB 3 VCC Internal Chip Supply. Connect VCC to VL through a 10Ω resistor. Bypass VCC to GND with at least a 0.1µF ceramic capacitor. 4 VIN Power Supply for LDO Regulator for VIN > 5.5V, and Chip Supply for VIN < 5.5V. Bypass VIN with at least a 1µF ceramic capacitor to GND. 5 VL Output of Internal 5V LDO. Connect VL to VIN for VIN < 5.5V. Bypass VL with at least a 1µF ceramic capacitor to GND. 6 DL Low-Side External MOSFET Gate-Driver Output. DL swings from VL to GND. 7 GND 8 LX Inductor Switching Node. LX is used for both current limit and the return supply of the DH driver. 9 DH High-Side External MOSFET Gate-Driver Output. DH swings from BST to LX. 10 BST Positive Supply of DH Driver. Connect a 0.1µF ceramic capacitor between BST and LX. Compensation Input. Pull COMP/EN low with an open-collector or open-drain device to turn off the output. Feedback Input. Connect a resistive-divider network to set VOUT. FB threshold is 0.8V. Ground and Negative Current-Sense Input Functional Diagram VIN 5V LINEAR REGULATOR VL TEMPERATURE SHUTDOWN 1V RAMP GENERATOR BST PWM COMP DH COMP/EN ERROR AMPLIFIER CONTROL LOGIC LX FB DL INTERNAL CHIP SUPPLY GND VCC CURRENT-LIMIT COMPARATOR 800mV REF SOFT-START 100kHz/ 300kHz* CLOCK GENERATOR MAX8545 MAX8546 MAX8548 FB *SEE SELECTOR GUIDE FOLDBACK _______________________________________________________________________________________ 7 MAX8545/MAX8546/MAX8548 Pin Description MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit Detailed Description The MAX8545/MAX8546/MAX8548 are BiCMOS switchmode power-supply controllers designed to implement simple, buck-topology regulators in cost-sensitive applications. The main power-switching circuit consists of two n-channel MOSFETs, an inductor, and input/output filter capacitors. An all n-channel synchronous-rectified design provides high efficiency at reduced cost. These devices have an internal 5V linear regulator that steps down the input voltage to supply the IC and the gate drivers. The low-side-switch gate driver is directly powered from the 5V regulator (VL), while the highside-switch gate driver is indirectly powered from VL plus an external diode-capacitor boost circuit. Current-Limit and Short-Circuit Protection The MAX8545/MAX8546/MAX8548 employ a valley current-sensing algorithm that uses the RDS(ON) of the lowside n-channel MOSFET to sense the current. This eliminates the need for an external sense resistor usually placed in series with the output. The voltage measured across the low-side MOSFET’s RDS(ON) is compared to a fixed -320mV reference for the MAX8545/MAX8548 and a fixed -165mV reference for the MAX8546. The current limit is given by the equations below: ILIMIT = 320mV (MAX8545 / MAX8548) RDS(ON) ILIMIT = 165mV (MAX8546) RDS(ON) Aside from current limiting, these devices feature foldback short-circuit protection. This feature is designed to reduce the current limit by 80% as the output voltage drops to 0V. MOSFET Gate Drivers The DH and DL drivers are optimized for driving n-channel MOSFETs with low gate charge. An adaptive deadtime circuit monitors the DL output and prevents the high-side MOSFET from turning on until the low-side MOSFET is fully off. There must be a low-resistance, low-inductance connection from the DL driver to the MOSFET gate for the adaptive dead-time circuit to work properly. Otherwise, the sense circuitry in the MAX8545/ MAX8546/MAX8548 may detect the MOSFET gate as off while there is actually charge left on the gate. Use very short, wide traces measuring no less than 50mils to 100 8 mils wide if the MOSFET is 1in away from the MAX8545/ MAX8546/MAX8548. The same type of adaptive deadtime circuit monitors the DH off edge. The same recommendations apply for the gate connection of the high-side MOSFET. The internal pulldown transistor that drives DL low is robust, with a 1.1Ω (typ) on-resistance. This helps prevent DL from being pulled up due to capacitive coupling from the drain to the gate of the low-side synchronous-rectifier MOSFET during the fast rise time of the LX node. Soft-Start The MAX8545/MAX8546/MAX8548 feature an internally set soft-start function that limits inrush current. It accomplishes this by ramping the internal reference input to the controller’s transconductance error amplifier from 0 to the 0.8V reference voltage. The ramp time is 1024 oscillator cycles for the MAX8548 and 2048 oscillator cycles for the MAX8545/MAX8546. At the nominal 100kHz and 300kHz switching rate, the soft-start ramp is approximately 10.2ms and 6.8ms, respectively. High-Side Gate-Drive Supply (BST) A flying-capacitor boost circuit generates gate-drive voltage for the high-side n-channel MOSFET. The flying capacitor is connected between the BST and LX nodes. On startup, the synchronous rectifier (low-side MOSFET) forces LX to ground and charges the boost capacitor to VL. On the second half-cycle, the MAX8545/MAX8546/ MAX8548 turn on the high-side MOSFET by closing an internal switch between BST and DH. This provides the necessary gate-to-source voltage to drive the high-side MOSFET gate above its source at the input voltage. Internal 5V Linear Regulator All MAX8545/MAX8546/MAX8548 functions are internally powered from an on-chip, low-dropout 5V regulator (VL). These devices have a maximum input voltage (VIN) of 28V. Connect VCC to VL through a 10Ω resistor and bypass VCC to GND with a 0.1µF ceramic capacitor. The VIN-to-VL dropout voltage is typically 140mV, so when VIN is less than 5.5V, VL is typically VIN - 140mV. The internal linear regulator can source a minimum of 25mA and a maximum of approximately 40mA to supply power to the IC low-side and high-side MOSFET drivers. Duty-Cycle Limitations for Low VOUT/VIN Ratios The MAX8545/MAX8546/MAX8548s’ output voltage is adjustable down to 0.8V. However, the minimum duty cycle can limit the ability to supply low-voltage outputs _______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit VOUT + (RDS(ON) × ILOAD ) VIN Setting the Output Voltage An output voltage between 0.8V and (0.83 x VIN) can be configured by connecting FB to a resistive divider between the output and GND (see Figures 1 and 2). Select resistor R4 in the 1kΩ to 10kΩ range. R3 is then given by: where RDS(ON) x ILOAD is the voltage drop across the synchronous rectifier. Therefore, the maximum input voltage (VIN(DFMAX)) that can supply a given output voltage is: VIN(DFMAX ) ≤ 1 VOUT + RDS(ON) × ILOAD DCMIN ( ( )) If the circuit cannot attain the required duty cycle dictated by the input and output voltages, the output voltage still remains in regulation. However, there may be intermittent or continuous half-frequency operation as the controller attempts to lower the average duty cycle by deleting pulses. This can increase output voltage ripple and inductor current ripple, which increases noise and reduces efficiency. Furthermore, circuit stability is not guaranteed. Applications Information Design Procedures 1) Input Voltage Range. The maximum value (VIN(MAX)) must accommodate the worst-case high input voltage. The minimum value (VIN(MIN)) must account for the lowest input voltage after drops due to connectors, fuses, and switches are considered. In general, lower input voltages provide the best efficiency. 2) Maximum Load Current. There are two current values to consider. Peak load current (ILOAD(MAX)) determines the instantaneous component stresses and filtering requirements and is key in determining output capacitor requirements. ILOAD(MAX) also determines the required inductor saturation rating. Continuous load current (ILOAD) determines the thermal stresses, input capacitor, and MOSFETs, as well as the RMS ratings of other heat-contributing components such as the inductor. 3) Inductor Value. This choice provides tradeoffs between size, transient response, and efficiency. Higher inductance value results in lower inductor ripple current, lower peak current, lower switching losses, and, therefore, higher efficiency at the cost of slower transient response and larger size. Lower inductance values result in large ripple currents, smaller size, and poor efficiency, while also providing faster transient response. ⎡V ⎤ R3 = R4 ⎢ OUT − 1⎥ ⎣ VFB ⎦ where VFB = +0.8V. Inductor Selection Determine an appropriate inductor value with the following equation: L = VOUT × (VIN − VOUT ) VIN × fOSC × LIR × ILOAD(MAX ) where LIR is the ratio of inductor ripple current to average continuous maximum load current. Choosing LIR between 20% to 40% results in a good compromise between efficiency and economy. Choose a low-coreloss inductor with the lowest possible DC resistance. Ferrite-core-type inductors are often the best choice for performance; however, the MAX8548’s low switching frequency also allows the use of powdered iron core inductors in ultra-low-cost applications where efficiency is not critical. With any core material, the core must be large enough not to saturate at the peak inductor current (IPEAK): ⎛ LIR ⎞ IPEAK = ILOAD(MAX ) + ⎜ ⎟ ×I ⎝ 2 ⎠ LOAD(MAX ) Setting the Current Limit The MAX8545/MAX8546/MAX8548 provide valley current limit by sensing the voltage across the external low-side MOSFET. The minimum current-limit threshold voltage is -280mV for the MAX8545/MAX8548 and -140mV for the MAX8546. The MOSFET on-resistance required to allow a given peak inductor current is: RDS(ON)MAX ≤ RDS(ON)MAX ≤ 0.28V (for the MAX8545 / MAX8548) IVALLEY 0.14V IVALLEY (for the MAX8546) where I VALLEY = I LOAD(MAX) x (1 - LIR / 2), and RDS(ON)MAX is the maximum on-resistance of the lowside MOSFET at the maximum operating junction temperature. _______________________________________________________________________________________ 9 MAX8545/MAX8546/MAX8548 from high-voltage inputs. With high input voltages, the required duty factor is approximately: MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit +2.7V TO +5.5V INPUT C4 C3 C2 C1 R1 VIN D1 VL VCC C9 C5 MAX8545* MAX8546 MAX8548* OUT 1.8V/3A OR 6A L1 LX C6 DL COMP/ EN D2 Q1 BST DH C7 C8 GND R2 R3 C10 FB C11 R4 *FOLLOW THE DATA SHEET DESIGN PROCEDURE TO SELECT THE EXTERNAL COMPONENTS FOR THE MAX8545/MAX8548. Figure 1. Typical Application Circuit (2.7V to 5V) Input (see Tables 1a, 1b) +10V TO +24V INPUT R1 C4 C2 C1 C9 VCC Q1 BST DH MAX8545* MAX8546 MAX8548* COMP/ EN D1 VL VIN C12 D2 C3 C5 OUT 2.5V/3A OR 6A L1 LX C6 DL C7 C8 GND R2 R3 C10 C11 FB R4 *FOLLOW THE DATA SHEET DESIGN PROCEDURE TO SELECT THE EXTERNAL COMPONENTS FOR THE MAX8545/MAX8548. Figure 2. Typical Application Circuit (10V to 24V) Input (see Tables 2a, 2b) 10 ______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit Power MOSFET Selection The MAX8545/MAX8546/MAX8548 drive two external, logic-level, n-channel MOSFETs as the circuit switching elements. The key selection parameters are: 1) On-resistance (RDS(ON)): the lower, the better. 2) Maximum drain-to-source voltage (VDSS) should be at least 10% higher than the input supply rail at the high-side MOSFET’s drain. 3) Gate charges (Qg,Qgd,Qgs): the lower, the better. Choose the MOSFETs with rated RDS(ON) at VGS = 4.5V for an input voltage greater than 5V, and at VGS = 2.5V for an input voltage less than 5.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET (N1) that has conduction losses equal to the switching losses at nominal input voltage and maximum output current. For N2, make sure it does not spuriously turn on due to a dV/dt caused by N1 turning on as this would result in shoot-through current degrading the efficiency. MOSFETs with a lower Qgd/Qgs ratio have higher immunity to dV/dt. MOSFET Power Dissipation For proper thermal-management design, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage (for the low-side MOSFET (N2) the worst case is at VIN(MAX), for the highside MOSFET (N1) the worst case can be either at VIN(MIN) or VIN(MAX)). N1 and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore, the major losses are: the channel conduction loss (PN2CC), the body-diode conduction loss (P N2DC ), and the gate-drive loss (PN2DR): ⎛ V ⎞ PN2CC = ⎜1− OUT ⎟ × I2LOAD × RDS(ON) VIN ⎠ ⎝ Use RDS(ON) at TJ(MAX): PN2DC = 2 × ILOAD × VF × t dt × fS where VF is the body-diode forward voltage drop, tdt is the dead time between N1 and N2 switching transitions (which is 30ns), and fS is the switching frequency. Because of zero-voltage switch operation, the N2 gatedrive losses are due to charging and discharging the input capacitor, C ISS . These losses are distributed between the average DL gate driver’s pullup and pulldown resistors and the internal gate resistance. The RDL is typically 1.8Ω, and the internal gate resistance (R GATE ) of the MOSFET is typically 2Ω. The drive power dissipated in N2 is given by: 2 PN2DR = CISS × (VGS ) × fS × RGATE RGATE + RDL N1 operates as a duty-cycle control switch and has the following major losses: the channel conduction loss (PN1CC), the voltage and current overlapping switching loss (PN1SW), and the drive loss (PN1DR). N1 does not have a body-diode conduction loss because the diode never conducts current: ⎛V ⎞ 2 PN1CC = ⎜ OUT ⎟ × (ILOAD ) × RDS(ON) V ⎝ IN ⎠ Use RDS(ON) at TJ(MAX): PN1SW = VIN × ILOAD × fS × QGS + QGD IGATE where IGATE is the average DH high driver output-current capability determined by: IGATE(ON) = 1 VL × 2 RDH + RGATE where RDH is the high-side MOSFET driver’s average on-resistance (2.05Ω typ) and RGATE is the internal gate resistance of the MOSFET (2Ω typ): PN1DR = QGS × VGS × fS × RGATE RDH + RGATE where VGS ~ VL. In addition to the losses above, allow about 20% more for additional losses due to MOSFET output capacitance and N2 body-diode reverse recovery charge dissipated in N1. Refer to the MOSFET data sheet for thermal resistance specifications to calculate the PC board area needed. This information is essential to maintain the desired maximum operating junction temperature with the above calculated power dissipation. To reduce EMI caused by switching noise, add a 0.1µF ceramic capacitor from the high-side MOSFET drain to the low-side MOSFET source or add resistors in series ______________________________________________________________________________________ 11 MAX8545/MAX8546/MAX8548 A limitation of sensing current across a MOSFET’s onresistance is that the current-limit threshold is not accurate since MOSFET R DS(ON) specifications are not precise. This type of current limit provides a coarse level of fault protection. It is especially suited when the input source is already current-limited or otherwise protected. MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit with DH and DL to slow down the switching transitions. However, adding series resistors increases the power dissipation of the MOSFET, so ensure temperature ratings of the MOSFET are not exceeded. Input-Capacitor Selection The input capacitors (C2 and C3 in Figure 1) reduce noise injection and current peaks drawn from the input supply. The input capacitor must meet the ripple-current requirement (IRMS) imposed by the switching currents. The RMS input ripple current is given by: IRMS = ILOAD × VOUT × (VIN − VOUT ) VIN For optimal circuit reliability, choose a capacitor that has less than 10°C temperature rise at the RMS current. IRMS is maximum when the input voltage equals 2 x VOUT, where IRMS = 1/2 ILOAD. zero close to the LC double pole when possible to negate the sharp phase shift of the typically high-Q double LC pole (see the Compensation Design section). Aluminum electrolytic or POS capacitors are recommended. Higher output current requires multiple capacitors to meet the output ripple voltage. The MAX8545/MAX8546/MAX8548s’ response to a load transient depends on the selected output capacitor. After a load transient, the output instantly changes by (ESR x ∆I LOAD ) + (ESL x dI/dt). Before the controller can respond, the output deviates further depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on the closed-loop bandwidth. Higher bandwidth results in faster response time, preventing the output voltage from further deviation. Do not exceed the capacitor’s voltage or ripple-current ratings. Output Capacitor Selection Boost Diode and Capacitor Selection The key parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements. All these parameters affect the overall stability, output ripple voltage, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the ESR, and the voltage drop across the ESL. VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) A low-current Schottky diode, such as the CMPSH-3 from Central Semiconductor, works well for most applications. Do not use large power diodes since higher junction capacitance can charge up BST to LX voltage that could exceed the device rating of 6V. The boost capacitor should be in the range of 0.1µF to 0.47µF, depending on the specific input and output voltages and the external components and PCB layout. The boost capacitance needs to be as large as possible to prevent it from charging to excessive voltage, but small enough to adequately charge during the minimum lowside MOSFET conduction time, which happens at the maximum operating duty cycle (this occurs at the minimum input voltage). In addition, ensure the boost capacitor does not discharge to below the minimum gate-to-source voltage required to keep the high-side MOSFET fully enhanced for lowest on-resistance. This minimum gate-to-source voltage VGS(MIN) is determined by: The output voltage ripple as a consequence of the ESR and output capacitance is: VRIPPLE(ESR) = IP−P × ESR VRIPPLE(C) = IP−P 8 × COUT × fSW V × ESL VRIPPLE(ESL) = IN L + ESL ⎛ VIN − VOUT ⎞ ⎛ VOUT ⎞ IP−P = ⎜ ⎟⎜ ⎟ ⎝ fSW × L ⎠ ⎝ VIN ⎠ where IP-P is the peak-to-peak inductor current (see the Inductor Selection section). While these equations are suitable for initial capacitor selection to meet the ripple requirement, final values may also depend on the relationship between the LC double-pole frequency and the capacitor ESR-zero frequency. Generally, the ESR zero is higher than the LC double pole; however, it is preferable to keep the ESR 12 VGS(MIN) = VL − QG CBOOST where Q G is the total gate charge of the high-side MOSFET and CBOOST is the boost capacitor value. Compensation Design The MAX8545/MAX8546/MAX8548 use a voltage-mode control scheme that regulates the output voltage. This is done by comparing the error amplifier’s output (COMP) to a fixed internal ramp. The inductor and output capacitor create a double pole at the resonant frequency, which ______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit GMOD(DC) = VIN VRAMP where VRAMP = 1V. The pole frequency due to the inductor and output capacitor is: fPMOD = 1 2π LCOUT The zero frequency due to the output capacitor’s ESR is: 1 fZESR = 2π × ESR × COUT The output capacitor is usually comprised of several same capacitors connected in parallel. With n capacitors in parallel, the output capacitance is: COUT = n X CEACH The total ESR is: ESR = ESREACH n The ESR zero (f ZESR ) for a parallel combination of capacitors is the same as for an individual capacitor. The feedback divider has a gain of GFB = VFB/VOUT, where VFB is 0.8V. The transconductance error amplifier has DC gain GEA(dc) of 72dB. A dominant pole (fDPEA) is set by the compensation capacitor (C C ), the amplifier output resistance (RO) equals 37MΩ, and the compensation resistor (RC): fDPEA = 1 2π × CC × (RO + RC ) The compensation resistor and the compensation capacitor set a zero: fZEA = 1 2π × CC × RC The total closed-loop gain must equal unity at the crossover frequency. The crossover frequency should be higher than fZESR, so that the -1 slope is used to cross over at unity gain. Also, the crossover frequency should be less than or equal to 1/5 the switching frequency (fSW) of the controller: f fZESR < fC ≤ SW 5 The loop-gain equation at the crossover frequency is: VFB/VOUT x GEA(fC) x GMOD(fC) = 1 where G EA(fC) = g mEA × R C , and G MOD(fC) = GMOD(DC) × (fPMOD)2 / (fZESR × fC). The compensation resistor, RC, is calculated from: RC = VOUT/ gmEA x VFB x GMOD(fC) where gmEA = 108µS. Due to the underdamped (Q > 1) nature of the output LC double pole, the error-amplifier compensation zero should be approximately 0.2 fPMOD to provide good phase boost. CC is calculated from: CC = 5 2π × RC × fPMOD A small capacitor, CF, can also be added from COMP to GND to provide high-frequency decoupling. C F adds another high-frequency pole, fPHF, to the error-amplifier response. This pole should be greater than 100 times the error-amplifier zero frequency to have negligible impact on the phase margin. This pole should also be less than 1/2 the switching frequency for effective decoupling: 100 fZEA < fPHF < 0.5 fsw Select a value for fPHF in the range given above, then solve for CF using the following equation: CF = 1 2π × RC × fPHF PCB Layout Guidelines Careful PCB layout is critical to achieve low switching losses and stable operation. If possible, mount all the power components on the top side of the board with their ______________________________________________________________________________________ 13 MAX8545/MAX8546/MAX8548 has a gain drop of 40dB per decade, and a phase shift of 180°. The error amplifier must compensate for this gain drop and phase shift to achieve a stable highbandwidth, closed-loop system. The basic regulator loop consists of a power modulator (Figure 3), an output feedback divider, and an error amplifier. The power modulator has DC gain set by VIN/VRAMP, with a double pole set by the inductor and output capacitor, and a single zero set by the output capacitor (COUT) and its equivalent series resistance (ESR). Below are equations that define the power modulator: The DC gain of the power modulator is: MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit VIN DH RAMP GENERATOR N VOUT L PWM LX DL COUT N R3 FB COMP/EN R4 ERROR AMPLIFIER R2 C10 MAX8545 MAX8546 MAX8548 0.8V Figure 3. Compensation Scheme ground terminals flush against one another. Follow these guidelines for good PCB layout: 1) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. 2) Connect the power and analog grounds close to the IC pin 7. 3) Keep the power traces and load connections short. This practice is essential for high efficiency. Using thick copper PCBs (2oz vs. 1oz) can enhance fullload efficiency by 1% or more. Correctly routing PCB traces is a difficult task that must be approached in terms of fractions of centimeters, where a few milohms of excess trace resistance cause a measurable efficiency penalty. 4) LX and GND connections to the low-side MOSFET for current sensing must be made using Kelvin sense connections to guarantee the current-limit accuracy. With 8-pin MOSFETs, this is best done by routing power to the MOSFETs from outside using the top copper layer, while connecting LX and GND inside (underneath) the 8-pin package. 14 5) When tradeoffs in trace lengths must be made, it is preferable to allow the inductor charging current path to be longer than the discharge path. For example, it is better to allow some extra distance between the inductor and the low-side MOSFET or between the inductor and the output filter capacitor. 6) Ensure that the connection between the inductor and C3 is short and direct. 7) Route switching nodes (BST, LX, DH, and DL) away from sensitive analog areas (COMP and FB). Ensure the C1 ceramic bypass capacitor is immediately adjacent to the pins and as close to the device as possible. Furthermore, the VIN and GND pins of MAX8545/ MAX8546/MAX8548 must terminate at the two ends of C1 before connecting to the power switches and C2. ______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit COMPONENT QTY DESCRIPTION C1, C4 2 1µF, 10V X7R ceramic capacitors Taiyo Yuden LMK212BJ105MG C2 0 Not installed 1 1200µF, 10V, 44mΩ, 1.25A aluminum electrolytic capacitor SANYO 10MV1200AX (10 x 16 case size) 3 0.1µF, 10V X7R ceramic capacitors Kemet C0603C104M8RAC 2 1000µF, 6.3V, 69mΩ, 0.8A aluminum electrolytic capacitors SANYO 6.3MV1000AX (8 x 20 case size) C10 1 1.5nF, 10V X7R ceramic capacitor Kemet C0603C152M8RAC C11 0 Not installed 2 30V, 100mA Schottky diodes Central Semiconductor CMPSH-3 C3 C5, C8, C9 C6, C7 D1, D2 L1 Q1 1 1 Table 1b. Component Selection for Standard Applications for VIN = 2.7V to 5.5V, VOUT = 1.8V / 6A (Figure 1) (MAX8546 Only) COMPONENT QTY 2 C2, C3 2 1200µF, 10V, 44mΩ, 1.25A aluminum electrolytic capacitors SANYO 10MV1200AX (10 x 16 case size) C5, C8, C9 3 0.1µF, 10V X7R ceramic capacitors Kemet C0603C104M8RAC C6, C7 2 1500µF, 6.3V, 44mΩ, 1.25A aluminum electrolytic capacitors SANYO 6.3MV1500AX (10 x 20 case size) C10 1 1.5nF, 10V X7R ceramic capacitor Kemet C0603C152M8RAC C11 0 Not installed D1, D2 2 30V, 100mA Schottky diodes Central Semiconductor CMPSH-3 L1 1 2.1µH, 8A, 11.6mΩ inductor Sumida CEP122-2R1 Q1 1 20V, 18mΩ dual n-channel, 8-pin SO Fairchild FDS6898A (for 2.7V to 3.6VIN) Fairchild FDS6890A (for 4.5V to 5.5VIN) C1, C4 4.7µH, 5.7A, 18mΩ inductor Sumida CDRH124-4R7 20V/30V, 35mΩ dual n-channel, 8-pin SO Vishay Si4966DY (for 2.7V to 3.6VIN) Fairchild FDS6912A (for 4.5V to 5.5VIN) DESCRIPTION 1µF, 10V X7R ceramic capacitors Taiyo Yuden LMK212BJ105MG R1 1 10Ω ±5% resistor R2 1 110kΩ ±5% resistor R1 1 10Ω ±5% resistor R3 1 5.11kΩ ±1% resistor R2 1 150kΩ ±5% resistor R4 1 4.02kΩ ±1% resistor R3 1 5.11kΩ ±1% resistor R4 1 4.02kΩ ±1% resistor ______________________________________________________________________________________ 15 MAX8545/MAX8546/MAX8548 Table 1a. Component Selection for Standard Applications for VIN = 2.7V to 5.5V, VOUT = 1.8V / 3A (Figure 1) (MAX8546 Only) MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit Table 2a. Component Selection for Standard Applications for VIN = 10V to 24V, VOUT = 2.5V / 3A (Figure 2) (MAX8546 Only) COMPONENT QTY C1 1 1µF, 10V X7R ceramic capacitor Taiyo Yuden LMK212BJ105MG C2 0 Not installed 1 470µF, 35V, 39mΩ, 1.45A aluminum electrolytic capacitor SANYO 35MV470AX (10 x 22 case size) C3 2 1µF, 35V X7R ceramic capacitors Taiyo Yuden GMK316BJ105ML 3 0.1µF, 10V X7R ceramic capacitors Kemet C0603C104M8RAC 2 1000µF, 6.3V, 69mΩ, 0.8A aluminum electrolytic capacitors SANYO 6.3MV1000AX (8 x 20 case size) C10 1 6.8nF, 10V X7R ceramic capacitor Kemet C0603C6822M8RAC C11 0 Not installed 2 30V, 100mA Schottky diodes Central Semiconductor CMPSH-3 C4, C12 C5, C8, C9 C6, C7 D1, D2 COMPONENT QTY DESCRIPTION 1 1µF, 10V X7R ceramic capacitor Taiyo Yuden LMK212BJ105MG C2, C3 2 470µF, 35V, 39mΩ, 1.45A aluminum electrolytic capacitors SANYO 35MV470AX (10 x 22 case size) C4, C12 2 1µF, 35V X7R ceramic capacitors Taiyo Yuden GMK316BJ105ML C5, C8, C9 3 0.1µF, 10V X7R ceramic capacitors Kemet C0603C104M8RAC C6, C7 2 1500µF, 6.3V, 44mΩ, 1.25A aluminum electrolytic capacitors SANYO 6.3MV1500AX (10 x 20 case size) C10 1 6.8nF, 10V X7R ceramic capacitor Kemet C0603C682M8RAC C11 0 Not installed D1, D2 2 30V, 100mA Schottky diodes Central Semiconductor CMPSH-3 L1 1 4µH, 8.3A, 6.6mΩ inductor Sumida CEP125-4R0 1 30V, 18mΩ (LSFET)/35mΩ (HSFET), dual n-channel, 8-pin SO Fairchild FDS6982 C1 1 8.2µH, 5.8A, 9.5mΩ inductor Sumida CEP125-8R2 Q1 1 30V, 35mΩ, dual n-channel, 8-pin SO Fairchild FDS6912A Q1 R1 1 10Ω ±5% resistor R1 1 10Ω ±5% resistor R2 1 82kΩ ±5% resistor R2 1 68kΩ ±5% resistor R3 1 8.66kΩ ±1% resistor R3 1 8.66kΩ ±1% resistor R4 1 4.02kΩ ±1% resistor R4 1 4.02kΩ ±1% resistor L1 16 DESCRIPTION Table 2b. Component Selection for Standard Applications for VIN = 10V to 24V, VOUT = 2.5V / 6A (Figure 2) (MAX8546 Only) ______________________________________________________________________________________ Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit TRANSISTOR COUNT: 3351 PROCESS: BiCMOS TOP VIEW COMP/EN 1 FB 10 BST 9 DH 8 LX VIN 4 7 GND VL 5 6 DL VCC Chip Information 2 3 MAX8545 MAX8546 MAX8548 µMAX ______________________________________________________________________________________ 17 MAX8545/MAX8546/MAX8548 Pin Configuration Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.) e 10LUMAX.EPS MAX8545/MAX8546/MAX8548 Low-Cost, Wide Input Range, Step-Down Controllers with Foldback Current Limit 4X S 10 10 INCHES H Ø0.50±0.1 0.6±0.1 1 1 0.6±0.1 BOTTOM VIEW TOP VIEW D2 MILLIMETERS MAX DIM MIN 0.043 A 0.006 A1 0.002 A2 0.030 0.037 0.120 D1 0.116 0.118 0.114 D2 0.116 0.120 E1 0.118 E2 0.114 0.199 H 0.187 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S α 0° 6° MAX MIN 1.10 0.15 0.05 0.75 0.95 3.05 2.95 3.00 2.89 3.05 2.95 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0° 6° E2 GAGE PLANE A2 c A b A1 α E1 L D1 L1 FRONT VIEW SIDE VIEW PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE, 10L uMAX/uSOP APPROVAL DOCUMENT CONTROL NO. 21-0061 REV. 1 1 Revision History Pages changed at Rev 2: 1, 2, 8, 11, 15, 16, 18 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.