LTC3632 High Efficiency, High Voltage 20mA Synchronous Step-Down Converter FEATURES DESCRIPTION n The LTC®3632 is a high efficiency, high voltage step-down DC/DC converter with internal high side and synchronous power switches that draws only 12μA typical DC supply current at no load while maintaining output voltage regulation. n n n n n n n n n n n n n Wide Input Voltage Range: Operation from 4.5V to 50V Overvoltage Lockout Provides Protection Up to 60V Internal High Side and Low Side Power Switches No Compensation Required 20mA Output Current Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 12μA Wide Output Voltage Range: 0.8V to VIN 0.8V ±1% Feedback Voltage Reference Adjustable Peak Current Limit Internal and External Soft-Start Precise RUN Pin Threshold with Adjustable Hysteresis Few External Components Required Low Profile (0.75mm) 3mm × 3mm DFN and Thermally Enhanced MS8E Packages APPLICATIONS n n n n n n 4mA to 20mA Current Loops Industrial Control Supplies Distributed Power Systems Portable Instruments Battery-Operated Devices Automotive Power Systems The LTC3632 can supply up to 20mA load current and features a programmable peak current limit that provides a simple method for optimizing efficiency in lower current applications. The LTC3632’s combination of Burst Mode® operation, integrated power switches, low quiescent current, and programmable peak current limit provides high efficiency over a broad range of load currents. With its wide 4.5V to 50V input range and internal overvoltage monitor capable of protecting the part through 60V surges, the LTC3632 is a robust converter suited for regulating a wide variety of power sources. Additionally, the LTC3632 includes a precise run threshold and soft-start feature to guarantee that the power system start-up is well-controlled in any environment. The LTC3632 is available in the thermally enhanced 3mm × 3mm DFN and MS8E packages. L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION Efficiency and Power Loss vs Load Current 100 5V, 20mA Step-Down Converter SW LTC3632 RUN VFB HYST ISET SS GND 1.47M 280k 3632 TA01a VOUT 5V 10μF 20mA 80 EFFICIENCY (%) 1mH VIN EFFICIENCY 100 70 60 10 50 POWER LOSS POWER LOSS (mW) VIN 5V TO 50V 1μF 1000 90 40 30 20 0.1 VIN = 10V 1 VIN = 48V 1 LOAD CURRENT (mA) 10 3632 TA01b 3632fb 1 LTC3632 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN Supply Voltage ..................................... –0.3V to 60V SW Voltage (DC) ............................–0.3V to (VIN + 0.3V) RUN Voltage .............................................. –0.3V to 60V VFB, HYST, ISET, SS Voltages......................... –0.3V to 6V Operating Junction Temperature Range (Note 2).................................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) MS8E ................................................................ 300°C PIN CONFIGURATION TOP VIEW TOP VIEW SW VIN ISET SS 1 2 3 4 9 GND 8 7 6 5 SW 1 GND HYST VFB RUN VIN 2 ISET 3 9 GND SS 4 8 GND 7 HYST 6 VFB 5 RUN MS8E PACKAGE 8-LEAD PLASTIC MSOP DD PACKAGE 8-LEAD (3mm × 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 40°C/W, θJC = 5°-10°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 43°C/W, θJC = 3°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3632EMS8E#PBF LTC3632EMS8E#TRPBF LTFFZ 8-Lead Plastic MSOP –40°C to 125°C LTC3632IMS8E#PBF LTC3632IMS8E#TRPBF LTFFZ 8-Lead Plastic MSOP –40°C to 125°C LTC3632EDD#PBF LTC3632EDD#TRPBF LFGB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3632IDD#PBF LTC3632IDD#TRPBF LFGB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3632fb 2 LTC3632 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are for TA = 25°C (Note 2). VIN = 10V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Input Supply (VIN) VIN Input Voltage Operating Range 4.5 UVLO VIN Undervoltage Lockout VIN Rising VIN Falling Hysteresis OVLO VIN Overvoltage Lockout VIN Rising VIN Falling Hysteresis IQ DC Supply Current (Note 3) Active Mode Sleep Mode Shutdown Mode l l 50 V 3.80 3.75 4.15 4.00 150 4.50 4.35 54 52 56 54 2 59 57 V V V 125 12 3 220 22 6 μA μA μA V VRUN = 0V V V mV Output Supply (VFB) VFB Rising l 0.792 0.800 0.808 l 3 5 7 0 10 VFB Feedback Comparator Trip Voltage VHYST Feedback Comparator Hysteresis Voltage IFB Feedback Pin Current VFB = 1V ΔVLINEREG Feedback Voltage Line Regulation VIN = 4.5V to 50V VRUN RUN Pin Threshold Voltage RUN Rising RUN Falling Hysteresis 1.17 1.06 1.21 1.10 110 1.25 1.14 V V mV IRUN RUN Pin Leakage Current RUN = 1.3V –10 0 10 nA VHYSTL Hysteresis Pin Voltage Low RUN < 1V, IHYST = 1mA 0.07 0.1 V –10 0.001 mV nA %/V Operation IHYST Hysteresis Pin Leakage Current VHYST = 1.3V –10 0 10 nA ISS Soft-Start Pin Pull-Up Current VSS < 1.5V 4.5 5.5 6.5 μA tINTSS Internal Soft-Start Time SS Pin Floating IPEAK Peak Current Trip Threshold ISET Floating 500k Resistor from ISET to GND ISET Shorted to GND 60 mA mA mA RON Power Switch On-Resistance Top Switch Bottom Switch ISW = –10mA ISW = 10mA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3632 is tested under pulsed load conditions such that TJ ≈ TA. LTC3632E is guaranteed to meet specifications from 0°C to 85°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3632I is guaranteed over the full –40°C to 125°C operating junction temperature range. Note that the 0.75 l 40 8 50 25 10 5.0 2.5 ms 13 Ω Ω maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. The junction temperature (TJ, in °C) is calculated from the ambient temperature (TA, in °C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD • θJA), where θJA (in °C/W) is the package thermal impedance. Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. 3632fb 3 LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Load Current, VOUT = 5V Efficiency vs Load Current, VOUT = 3.3V 100 95 VIN = 24V 90 VIN = 48V 75 VIN = 36V 70 80 75 VIN = 48V 70 65 60 60 VOUT = 5V FIGURE 11 CIRCUIT 1 LOAD CURRENT (mA) 90 VIN = 24V 85 65 50 0.1 95 VIN = 12V 90 EFFICIENCY (%) EFFICIENCY (%) 80 100 95 VIN = 12V 85 55 VIN = 36V 50 0.1 VOUT = 5V FIGURE 11 CIRCUIT 1 LOAD CURRENT (mA) 80 ILOAD = 5mA 75 10 15 20 25 30 35 40 INPUT VOLTAGE (V) 5.03 0.10 0 –0.10 –0.20 4.98 –0.40 4.96 4.95 10 5 15 20 25 30 35 40 INPUT VOLTAGE (V) 45 110 3632 G07 3632 G06 Peak Current Trip Threshold vs Temperature and ISET 60 VIN = 10V PEAK CURRENT TRIP THRESHOLD (mA) 5.6 5.4 5.2 5.0 4.8 4.6 4.4 –40 –10 20 50 80 TEMPERATURE (°C) 110 3632 G08 20 10 5 15 LOAD CURRENT (mA) 0 50 3632 G05 FEEDBACK COMPARATOR HYSTERESIS (mV) 0.799 20 50 80 TEMPERATURE (°C) 4.99 Feedback Comparator Hysteresis Voltage vs Temperature 0.800 –10 5.00 4.97 3632 G04 0.798 –40 5.02 5.01 –0.30 Feedback Comparator Trip Voltage vs Temperature 10 VIN = 10V VOUT = 5V FIGURE 11 CIRCUIT 5.04 0.20 50 VIN = 10V 1 LOAD CURRENT (mA) Load Regulation ILOAD = 20mA FIGURE 11 CIRCUIT –0.50 45 VOUT = 2.5V FIGURE 11 CIRCUIT 3632 G03 OUTPUT VOLTAGE (V) ΔVOUT/VOUT (%) EFFICIENCY (%) ILOAD = 20mA VIN = 36V 5.05 0.40 90 VIN = 48V 50 0.1 10 0.30 70 70 Line Regulation 95 ILOAD = 1mA 75 55 0.50 85 80 3632 G02 Efficiency vs Input Voltage 100 VIN = 24V 65 VOUT = 3.3V FIGURE 11 CIRCUIT 55 10 VIN = 12V 85 60 3632 G01 FEEDBACK COMPARATOR TRIP VOLTAGE (V) Efficiency vs Load Current, VOUT = 2.5V EFFICIENCY (%) 100 0.801 TA = 25°C, unless otherwise noted. VIN = 10V ISET OPEN 50 40 30 RISET = 500k 20 ISET = GND 10 0 –40 –10 20 50 80 TEMPERATURE (°C) 110 3632 G09 3632fb 4 LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS Peak Current Trip Threshold vs RISET 60 40 30 20 10 0 200 400 600 800 RISET (kΩ) 1000 14 VIN = 10V ISET OPEN 50 12 VIN SUPPLY CURRENT (μA) VIN = 10V 50 0 40 30 RSET = 500k 20 ISET = GND 10 0 5 SLEEP 8 6 SHUTDOWN 8 7 7 6 TOP 5 4 3 BOTTOM 10 20 30 50 40 6 TOP 5 4 BOTTOM 3 2 0 –40 INPUT VOLTAGE (V) –10 20 50 80 TEMPERATURE (°C) 3632 G14 3632 G13 Switch Leakage Current vs Temperature 110 3632 G15 RUN Comparator Threshold Voltage vs Temperature Operating Waveforms 1.30 0.30 RUN COMPARATOR THRESHOLD (V) VIN = 50V SWITCH LEAKAGE CURRENT (μA) VIN = 10V 1 0 110 45 3632 G12 8 1 20 50 80 TEMPERATURE (°C) 25 35 INPUT VOLTAGE (V) Switch On-Resistance vs Temperature 2 2 –10 15 5 10 15 20 25 30 35 40 45 50 INPUT VOLTAGE (V) SWITCH ON-RESISTANCE (Ω) SWITCH ON-RESISTANCE (Ω) VIN SUPPLY CURRENT (μA) 12 10 SHUTDOWN 4 Switch On-Resistance vs Input Voltage VIN = 10V 0 –40 6 3632 G11 Quiescent Supply Current vs Temperature 4 8 0 0 1200 SLEEP 10 2 3632 G10 14 Quiescent Supply Current vs Input Voltage Peak Current Trip Threshold vs Input Voltage PEAK CURRENT TRIP THRESHOLD (mA) PEAK CURRENT TRIP THRESHOLD (mA) 60 TA = 25°C, unless otherwise noted. 0.25 0.20 0.15 0.10 SW = 50V 0.05 0 –40 SW = 0V –10 20 50 80 TEMPERATURE (°C) 110 3632 G16 1.25 SWITCH VOLTAGE 20V/DIV RISING OUTPUT VOLTAGE 100mV/DIV INDUCTOR CURRENT 50mA/DIV 1.20 1.15 FALLING 1.10 VIN = 48V VOUT = 5V ILOAD = 10mA 1.05 1.00 –40 –10 20 50 80 TEMPERATURE (°C) 20μs/DIV 3632 G18 110 3632 G17 3632fb 5 LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS Soft-Start Waveforms Load Step Transient Response OUTPUT VOLTAGE 2V/DIV OUTPUT VOLTAGE 100mV/DIV OUTPUT VOLTAGE 1V/DIV LOAD CURRENT 20mA/DIV CSS = 0.047μF 5ms/DIV 3632 G19 Short-Circuit Response INDUCTOR CURRENT 20mA/DIV VIN = 10V VOUT = 5V 1ms/DIV 3632 G20 VIN = 10V VOUT = 5V 200μs/DIV 3632 G21 PIN FUNCTIONS SW (Pin 1): Switch Node Connection to Inductor. This pin connects to the drains of the internal power MOSFET switches. VIN (Pin 2): Main Supply Pin. A ceramic bypass capacitor should be tied between this pin and GND (Pin 8). ISET (Pin 3): Peak Current Set Input. A resistor from this pin to ground sets the peak current trip threshold. Leave floating for the maximum peak current (50mA). Short this pin to ground for the minimum peak current (10mA). A 1μA current is sourced out of this pin. SS (Pin 4): Soft-Start Control Input. A capacitor to ground at this pin sets the ramp time to full current output during start-up. A 5.5μA current is sourced out of this pin. If left floating, the ramp time defaults to an internal 0.75ms soft-start. RUN (Pin 5): Run Control Input. A voltage on this pin above 1.2V enables normal operation. Forcing this pin below 0.7V shuts down the LTC3632, reducing quiescent current to approximately 3μA. VFB (Pin 6): Output Voltage Feedback. Connect to an external resistive divider to divide the output voltage down for comparison to the 0.8V reference. HYST (Pin 7): Run Hysteresis Open-Drain Logic Output. This pin is pulled to ground when RUN (Pin 5) is below 1.2V. This pin can be used to adjust the RUN pin hysteresis. See Applications Information. GND (Pin 8, Exposed Pad Pin 9): Ground. The exposed pad must be soldered to the printed circuit board ground plane for optimal electrical and thermal performance. 3632fb 6 LTC3632 BLOCK DIAGRAM VIN 1μA ISET 2 C2 3 – PEAK CURRENT COMPARATOR + RUN LOGIC AND SHOOTTHROUGH PREVENTION + 5 1.2V – SW L1 VOUT 1 C1 HYST 7 + REVERSE-CURRENT COMPARATOR FEEDBACK COMPARATOR GND GND 8 9 – VOLTAGE REFERENCE + + – 0.800V 5.5μA SS 4 VFB R1 3632 BD 6 R2 3632fb 7 LTC3632 OPERATION (Refer to Block Diagram) The LTC3632 is a step-down DC/DC converter with internal power switches that uses Burst Mode control, combining low quiescent current with high switching frequency, which results in high efficiency across a wide range of load currents. Burst Mode operation functions by using short “burst” cycles to ramp the inductor current through the internal power switches, followed by a sleep cycle where the power switches are off and the load current is supplied by the output capacitor. During the sleep cycle, the LTC3632 draws only 12μA of supply current. At light loads, the burst cycles are a small percentage of the total cycle time which minimizes the average supply current, greatly improving efficiency. Main Control Loop The feedback comparator monitors the voltage on the VFB pin and compares it to an internal 800mV reference. If this voltage is greater than the reference, the comparator activates a sleep mode in which the power switches and current comparators are disabled, reducing the VIN pin supply current to only 12μA. As the load current discharges the output capacitor, the voltage on the VFB pin decreases. When this voltage falls 5mV below the 800mV reference, the feedback comparator trips and enables burst cycles. At the beginning of the burst cycle, the internal high side power switch (P-channel MOSFET) is turned on and the inductor current begins to ramp up. The inductor current increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin exceeds 800mV, at which time the high side power switch is turned off and the low side power switch (N-channel MOSFET) turns on. The inductor current ramps down until the reverse-current comparator trips, signaling that the current is close to zero. If the voltage on the VFB pin is still less than the 800mV reference, the high side power switch is turned on again and another cycle commences. The average current during a burst cycle will normally be greater than the average load current. For this architecture, the maximum average output current is equal to half of the peak current. The hysteretic nature of this control architecture results in a switching frequency that is a function of the input voltage, output voltage and inductor value. This behavior provides inherent short-circuit protection. If the output is shorted to ground, the inductor current will decay very slowly during a single switching cycle. Since the high side switch turns on only when the inductor current is near zero, the LTC3632 inherently switches at a lower frequency during start-up or short-circuit conditions. Start-Up and Shutdown If the voltage on the RUN pin is less than 0.7V, the LTC3632 enters a shutdown mode in which all internal circuitry is disabled, reducing the DC supply current to 3μA. When the voltage on the RUN pin exceeds 1.2V, normal operation of the main control loop is enabled. The RUN pin comparator has 110mV of internal hysteresis, and therefore must fall below 1.1V to disable the main control loop. The HYST pin provides an added degree of flexibility for the RUN pin operation. This open-drain output is pulled to ground whenever the RUN comparator is not tripped, signaling that the LTC3632 is not in normal operation. In applications where the RUN pin is used to monitor the VIN voltage through an external resistive divider, the HYST pin can be used to increase the effective RUN comparator hysteresis. An internal 0.75ms soft-start function limits the ramp rate of the output voltage on start-up to prevent excessive input supply droop. If a longer ramp time and consequently less supply droop is desired, a capacitor can be placed from the SS pin to ground. The 5μA current that is sourced out of this pin will create a smooth voltage ramp on the capacitor. If this ramp rate is slower than the internal 0.75ms soft-start, then the output voltage will be limited by the ramp rate 3632fb 8 LTC3632 OPERATION (Refer to Block Diagram) on the SS pin instead. The internal and external soft-start functions are reset on start-up and after an undervoltage or overvoltage event on the input supply. In order to ensure a smooth start-up transition in any application, the internal soft-start also ramps the peak inductor current from 10mA during its 0.75ms ramp time to the set peak current threshold. The external ramp on the SS pin does not limit the peak inductor current during start-up; however, placing a capacitor from the ISET pin to ground does provide this capability. Peak Inductor Current Programming The offset of the peak current comparator nominally provides a peak inductor current of 50mA. This peak inductor current can be adjusted by placing a resistor from the ISET pin to ground. The 1μA current sourced out of this pin through the resistor generates a voltage that is translated into an offset in the peak current comparator, which limits the peak inductor current. Input Undervoltage and Overvoltage Lockout The LTC3632 implements a protection feature which disables switching when the input voltage is not within the 4.5V to 50V operating range. If VIN falls below 4V typical (4.35V maximum), an undervoltage detector disables switching. Similarly, if VIN rises above 55V typical (53V minimum), an overvoltage detector disables switching. When switching is disabled, the LTC3632 can safely sustain input voltages up to the absolute maximum rating of 60V. Switching is enabled when the input voltage returns to the 4.5V to 50V operating range. 3632fb 9 LTC3632 APPLICATIONS INFORMATION The basic LTC3632 application circuit is shown on the front page of this data sheet. External component selection is determined by the maximum load current requirement and begins with the selection of the peak current programming resistor, RISET. The inductor value L can then be determined, followed by capacitors CIN and COUT. Peak Current Resistor Selection The peak current comparator has a maximum current limit of 50mA nominally, which results in a maximum average current of 25mA. For applications that demand less current, the peak current threshold can be reduced to as little as 10mA. This lower peak current allows the use of lower value, smaller components (input capacitor, output capacitor and inductor), resulting in lower input supply ripple and a smaller overall DC/DC converter. The threshold can be easily programmed with an appropriately chosen resistor (RISET) between the ISET pin and ground. The value of resistor for a particular peak current can be computed by using Figure 1 or the following equation: RISET = IPEAK • 21 • 106 where 10mA < IPEAK < 50mA. The peak current is internally limited to be within the range of 10mA to 50mA. Shorting the ISET pin to ground programs the current limit to 10mA, and leaving it floating sets the current limit to the maximum value of 50mA. When selecting this resistor value, be aware that the maximum 1100 1000 900 800 RISET (k) 700 600 500 400 300 200 100 0 4 6 10 12 14 16 18 8 MAXIMUM LOAD CURRENT (mA) 20 3632 F01 Figure 1. RISET Selection average output current for this architecture is limited to half of the peak current. Therefore, be sure to select a value that sets the peak current with enough margin to provide adequate load current under all foreseeable operating conditions. Inductor Selection The inductor, input voltage, output voltage and peak current determine the switching frequency of the LTC3632. For a given input voltage, output voltage and peak current, the inductor value sets the switching frequency when the output is in regulation. A good first choice for the inductor value can be determined by the following equation: V V L = OUT • 1– OUT VIN f • IPEAK The variation in switching frequency with input voltage and inductance is shown in the following two figures for typical values of VOUT. For lower values of IPEAK, multiply the frequency in Figure 2 and Figure 3 by 50mA/IPEAK. An additional constraint on the inductor value is the LTC3632’s 100ns minimum on-time of the high side switch. Therefore, in order to keep the current in the inductor well controlled, the inductor value must be chosen so that it is larger than LMIN, which can be computed as follows: L MIN = VIN(MAX ) • tON(MIN) IPEAK(MAX ) where VIN(MAX) is the maximum input supply voltage for the application, tON(MIN) is 100ns, and IPEAK(MAX) is the maximum allowed peak inductor current. Although the above equation provides the minimum inductor value, higher efficiency is generally achieved with a larger inductor value, which produces a lower switching frequency. For a given inductor type, however, as inductance is increased DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating, both of which place an upper limit on the inductance. The recommended range of inductor values for small surface mount inductors as a function of peak current is shown in Figure 4. The values in this range are a good compromise between the trade-offs discussed above. For applications 3632fb 10 LTC3632 APPLICATIONS INFORMATION SWITCHING FREQUENCY (kHz) 350 VOUT = 5V ISET OPEN 300 where board area is not a limiting factor, inductors with larger cores can be used, which extends the recommended range of Figure 4 to larger values. L = 220μH 250 200 Inductor Core Selection L = 470μH 150 100 L = 1000μH 50 L = 2200μH 0 5 10 15 20 25 30 35 40 VIN INPUT VOLTAGE (V) 45 50 3632 F02 Figure 2. Switching Frequency for VOUT = 5V SWITCHING FREQUENCY (kHz) 250 L = 220μH VOUT = 3.3V ISET OPEN 200 150 L = 470μH 100 L = 1000μH 50 L = 2200μH 0 5 10 15 20 25 30 35 40 VIN INPUT VOLTAGE (V) 45 50 3632 F03 Figure 3. Switching Frequency for VOUT = 3.3V INDUCTOR VALUE (μH) 10000 Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of the more expensive ferrite cores. Actual core loss is independent of core size for a fixed inductor value but is very dependent of the inductance selected. As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequently output voltage ripple. Do not allow the core to saturate! Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate energy but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price vs size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Coiltronics, Coilcraft, TDK, Toko, Sumida and Vishay. CIN and COUT Selection 1000 100 10 50 The input capacitor, CIN, is needed to filter the trapezoidal current at the source of the top high side MOSFET. To prevent large ripple voltage, a low ESR input capacitor sized for the maximum RMS current should be used. Approximate RMS current is given by: PEAK INDUCTOR CURRENT (mA) 3632 F04 Figure 4. Recommended Inductor Values for Maximum Efficiency IRMS = IOUT(MAX ) • VOUT VIN • −1 VIN VOUT 3632fb 11 LTC3632 APPLICATIONS INFORMATION This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based only on 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. electrolytic capacitors have significantly higher ESR but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR characteristics but can have high voltage coefficient and audible piezoelectric effects. The high quality factor (Q) of ceramic capacitors in series with trace inductance can also lead to significant ringing. The output capacitor, COUT, filters the inductor’s ripple current and stores energy to satisfy the load current when the LTC3632 is in sleep. The output voltage ripple during a burst cycle is dominated by the output capacitor equivalent series resistance (ESR) and can be estimated by the following equation: Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. VOUT < ΔVOUT ≤ IPEAK • ESR 160 where the lower limit of VOUT/160 is due to the 5mV feedback comparator hysteresis. The value of the output capacitor must be large enough to accept the energy stored in the inductor without a large change in output voltage. Setting this voltage step equal to 1% of the output voltage, the output capacitor must be: I 2 PEAK COUT > 50 • L • VOUT Typically, a capacitor that satisfies the ESR requirement is adequate to filter the inductor ripple. To avoid overheating, the output capacitor must also be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by IRMS = IPEAK/2. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic, and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important only to use types that have been surge tested for use in switching power supplies. Aluminum Using Ceramic Input and Output Capacitors For applications with inductive source impedance, such as a long wire, a series RC network may be required in parallel with CIN to dampen the ringing of the input supply. Figure 5 shows this circuit and the typical values required to dampen the ringing. LTC3632 LIN VIN R= LIN CIN 4 • CIN 3632 F05 CIN Figure 5. Series RC to Reduce VIN Ringing Output Voltage Programming The output voltage is set by an external resistive divider according to the following equation: R1 VOUT = 0.8V • 1+ R2 3632fb 12 LTC3632 APPLICATIONS INFORMATION The resistive divider allows the VFB pin to sense a fraction of the output voltage as shown in Figure 6. Output voltage adjustment range is from 0.8V to VIN. VOUT R1 VFB LTC3632 R2 GND 3632 F06 The RUN pin can alternatively be configured as a precise undervoltage lockout (UVLO) on the VIN supply with a resistive divider from VIN to ground. The RUN pin comparator nominally provides 10% hysteresis when used in this method; however, additional hysteresis may be added with the use of the HYST pin. The HYST pin is an opendrain output that is pulled to ground whenever the RUN comparator is not tripped. A simple resistive divider can be used as shown in Figure 8 to meet specific VIN voltage requirements. Figure 6. Setting the Output Voltage VIN To minimize the no-load supply current, resistor values in the megohm range should be used; however, large resistor values should be used with caution. The feedback divider is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds the load current, the output voltage will be pulled up. In normal operation, this is generally a minor concern since the load current is much greater than the leakage. The increase in supply current due to the feedback resistors can be calculated from: V V IVIN = OUT • OUT R1+R2 VIN The LTC3632 has a low power shutdown mode controlled by the RUN pin. Pulling the RUN pin below 0.7V puts the LTC3632 into a low quiescent current shutdown mode (IQ ~ 3μA). When the RUN pin is greater than 1.2V, the controller is enabled. Figure 7 shows examples of configurations for driving the RUN pin from logic. VIN LTC3632 RUN RUN R2 LTC3632 HYST R3 3632 F08 Figure 8. Adjustable Undervoltage Lockout Specific values for these UVLO thresholds can be computed from the following equations: R1 Rising VIN UVLO Threshold = 1.21V • 1+ R2 Run Pin with Programmable Hysteresis VSUPPLY R1 4.7M LTC3632 RUN R1 Falling VIN UVLO Threshold = 1.10V • 1+ R2 +R3 The minimum value of these thresholds is limited to the internal VIN UVLO thresholds that are shown in the Electrical Characteristics table. The current that flows through this divider will directly add to the shutdown, sleep and active current of the LTC3632, and care should be taken to minimize the impact of this current on the overall efficiency of the application circuit. Resistor values in the megohm range may be required to keep the impact on quiescent shutdown and sleep currents low. Be aware that the HYST pin cannot be allowed to exceed its absolute maximum 3632 F07 Figure 7. RUN Pin Interface to Logic 3632fb 13 LTC3632 APPLICATIONS INFORMATION rating of 6V. To keep the voltage on the HYST pin from exceeding 6V, the following relation should be satisfied: R3 VIN(MAX) • < 6V R1+R2 +R3 The RUN pin may also be directly tied to the VIN supply for applications that do not require the programmable undervoltage lockout feature. In this configuration, switching is enabled when VIN surpasses the internal undervoltage lockout threshold. Soft-Start The internal 0.75ms soft-start is implemented by ramping both the effective reference voltage from 0V to 0.8V and the peak current limit set by the ISET pin (10mA to 50mA). To increase the duration of the reference voltage soft-start, place a capacitor from the SS pin to ground. An internal 5μA pull-up current will charge this capacitor, resulting in a soft-start ramp time given by: tSS = CSS • 0.8 V 5μA When the LTC3632 detects a fault condition (input supply undervoltage or overvoltage) or when the RUN pin falls below 1.1V, the SS pin is quickly pulled to ground and the internal soft-start timer is reset. This ensures an orderly restart when using an external soft-start capacitor. The duration of the 0.75ms internal peak current softstart may be increased by placing a capacitor from the ISET pin to ground. The peak current soft-start will ramp from 10mA to the final peak current value determined by a resistor from ISET to ground. A 1μA current is sourced out of the ISET pin. With only a capacitor connected between ISET and ground, the peak current ramps linearly from 10mA to 50mA, and the peak current soft-start time can be expressed as: tSS(ISET) = CISET • 0.8 V 1μA A linear ramp of peak current appears as a quadratic waveform on the output voltage. For the case where the peak current is reduced by placing a resistor from ISET to ground, the peak current offset ramps as a decaying exponential with a time constant of RISET • CISET. For this case, the peak current soft-start time is approximately 3 • RISET • CISET. Unlike the SS pin, the ISET pin does not get pulled to ground during an abnormal event; however, if the ISET pin is floating (programmed to 50mA peak current), the SS and ISET pins may be tied together and connected to a capacitor to ground. For this special case, both the peak current and the reference voltage will soft-start on power-up and after fault conditions. The ramp time for this combination is CSS(ISET) • (0.8V/6μA). Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses: VIN operating current and I2R losses. The VIN operating current dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. 1. The VIN operating current comprises two components: The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. 3632fb 14 LTC3632 APPLICATIONS INFORMATION 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. When switching, the average output current flowing through the inductor is “chopped” between the high side PMOS switch and the low side NMOS switch. Thus, the series resistance looking back into the switch pin is a function of the top and bottom switch RDS(ON) values and the duty cycle (DC = VOUT/VIN) as follows: Next, verify that this value meets the LMIN requirement. For this input voltage and peak current, the minimum inductor value is: RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 – DC) Next, CIN and COUT are selected. For this design, CIN should be size for a current rating of at least: The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add RSW to RL and multiply the result by the square of the average output current: I2R Loss = IO2(RSW + RL) Other losses, including CIN and COUT ESR dissipative losses and inductor core losses, generally account for less than 2% of the total power loss. Thermal Considerations The LTC3632 does not dissipate much heat due to its high efficiency and low peak current level. Even in worst-case conditions (high ambient temperature, maximum peak current and high duty cycle), the junction temperature will exceed ambient temperature by only a few degrees. Design Example As a design example, consider using the LTC3632 in an application with the following specifications: VIN = 24V, VOUT = 3.3V, IOUT = 20mA, f = 250kHz. Furthermore, assume for this example that switching should start when VIN is greater than 12V and should stop when VIN is less than 8V. First, calculate the inductor value that gives the required switching frequency: 3.3V 3.3V L = • 1– 220μH 250kHz • 50mA 24V L MIN = 24V • 100ns ≅ 48μH 50mA Therefore, the minimum inductor requirement is satisfied, and the 220μH inductor value may be used. IRMS = 20mA • 3.3V 24V • – 1 ≅ 7mA RMS 24V 3.3V Due to the low peak current of the LTC3632, decoupling the VIN supply with a 1μF capacitor is adequate for most applications. COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement. For a 50mV output ripple, the value of the output capacitor ESR can be calculated from: ΔVOUT = 50mV ≤ 50mA • ESR A capacitor with a 1Ω ESR satisfies this requirement. A 10μF ceramic capacitor has significantly less ESR than 1Ω. The output voltage can now be programmed by choosing the values of R1 and R2. Choose R2 = 240k and calculate R1 as: V R1= OUT – 1 • R2 = 750k 0.8V The undervoltage lockout requirement on VIN can be satisfied with a resistive divider from VIN to the RUN and HYST pins. Choose R1 = 2M and calculate R2 and R3 as follows: 1.21V • R1= 224k R2 = – 1.21V V IN(RISING) 1.1V • R1– R2 = 90.8k R3 = – 1.1V V IN(FALLING) 3632fb 15 LTC3632 APPLICATIONS INFORMATION Choose standard values for R2 = 226k and R3 = 91k. The ISET pin should be left open in this example to select maximum peak current (50mA). Figure 9 shows a complete schematic for this design example. 220μH VIN 24V VIN 1μF 2M SW LTC3632 RUN SS 226k 91k ISET VFB HYST GND 4. Flood all unused area on all layers with copper. Flooding with copper will reduce the temperature rise of power components. You can connect the copper areas to any DC net (VIN, VOUT, GND or any other DC rail in your system). VOUT 3.3V 20mA 10μF 2 VIN VIN SW 1 L1 VOUT LTC3632 5 750k CIN 240k 7 4 3632 F09 CSS RUN VFB R1 6 COUT HYST SS ISET 3 GND 8, 9 R2 RSET 3632 F10a Figure 9. 24V to 3.3V, 20mA Regulator at 250kHz PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3632. Check the following in your layout: L1 VIN CIN VOUT COUT 1. Large switched currents flow in the power switches and input capacitor. The loop formed by these components should be as small as possible. A ground plane is recommended to minimize ground impedance. R1 R2 2. Connect the (+) terminal of the input capacitor, CIN, as close as possible to the VIN pin. This capacitor provides the AC current into the internal power MOSFETs. 3. Keep the switching node, SW, away from all sensitive small-signal nodes. The rapid transitions on the switching node can couple to high impedance nodes, in particular VFB, and create increased output ripple. RSET CSS GND 3632 F10b VIAS TO GROUND PLANE VIAS TO INPUT SUPPLY (VIN) OUTLINE OF LOCAL GROUND PLANE Figure 10. Layout Example 3632fb 16 LTC3632 TYPICAL APPLICATIONS L1 1mH VIN 5V TO 50V VIN CIN 4.7μF SW LTC3632 RUN HYST ISET R1 4.2M VFB COUT 100μF VOUT 5V 20mA R2 800k SS CSS 470nF GND 3632 F11 CIN: TDK C5750X7R2A475MT COUT: AVX 1812D107MAT L1: COILCRAFT LPS6235-105ML Figure 11. High Efficiency 5V Regulator 3.3V, 20mA Regulator with Peak Current Soft-Start, Small Size VIN 4.5V TO 24V L1 470μH VIN CIN 1μF SW R1 294k LTC3632 RUN ISET SS CSS 0.1μF VFB COUT 10μF VOUT 3.3V 20mA R2 93.1k HYST GND Soft-Start Waveforms OUTPUT VOLTAGE 1V/DIV INDUCTOR CURRENT 20mA/DIV 3632 TA03a 3632 TA03b 2ms/DIV CIN: TDK C3216X7R1E105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43CN471K03 Positive-to-Negative Converter L1 1mH VIN CIN 1μF 20 SW LTC3632 RUN ISET R1 1M COUT 10μF VFB SS HYST GND R2 71.5k CIN: TDK C3225X7R1H105KT COUT: MURATA GRM32DR71C106KA01 L1: TYCO/COEV DQ6545-102M VOUT –12V 3632 TA04a MAXIMUM LOAD CURRENT (mA) VIN 4.5V TO 38V Maximum Load Current vs Input Voltage ISET OPEN VOUT = –3V VOUT = –5V 15 VOUT = –12V 10 5 5 10 15 20 25 30 35 40 VIN INPUT VOLTAGE (V) 45 50 3632 TA04b 3632fb 17 LTC3632 TYPICAL APPLICATIONS Small Size, Limited Peak Current, 4mA Regulator VIN 7V TO 50V L1 2.2mH VIN CIN 1μF R3 470k SW RUN R4 100k R5 33k R1 470k LTC3632 VFB COUT 10μF VOUT 5V 4mA R2 88.7k HYST SS ISET GND 3632 TA05a CIN: TDK C3225X7R1H105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43NN222K03 High Efficiency 15V, 4mA Regulator 95 L1 10mH VIN CIN 1μF SW LTC3632 RUN ISET SS VFB HYST GND CIN: AVX 18125C105KAT2A COUT: TDK C3216X7R1E475KT L1: COILCRAFT LPS6235-106ML R1 3M R2 169k 3642 TA07a COUT 4.7μF VOUT 15V 4mA 90 VIN = 24V 85 EFFICIENCY (%) VIN 15V TO 50V Efficiency vs Load Current 80 75 VIN = 36V VIN = 48V 70 65 60 55 50 0.1 1 LOAD CURRENT (mA) 4 3632 TA07b 3632fb 18 LTC3632 PACKAGE DESCRIPTION DD Package 8-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1698 Rev C) 0.70 p0.05 3.5 p0.05 1.65 p0.05 2.10 p0.05 (2 SIDES) PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC 2.38 p0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 p0.10 (4 SIDES) R = 0.125 TYP 5 0.40 p 0.10 8 1.65 p 0.10 (2 SIDES) PIN 1 TOP MARK (NOTE 6) (DD8) DFN 0509 REV C 0.200 REF 0.75 p0.05 4 0.25 p 0.05 1 0.50 BSC 2.38 p0.10 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE 3632fb 19 LTC3632 PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1662 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 1.88 (.074) 1 0.889 p 0.127 (.035 p .005) 1.88 p 0.102 (.074 p .004) 0.29 REF 1.68 (.066) 0.05 REF 5.23 (.206) MIN DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 1.68 p 0.102 3.20 – 3.45 (.066 p .004) (.126 – .136) 8 0.42 p 0.038 (.0165 p .0015) TYP 3.00 p 0.102 (.118 p .004) (NOTE 3) 0.65 (.0256) BSC 8 7 6 5 0.52 (.0205) REF RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 3.00 p 0.102 (.118 p .004) (NOTE 4) 4.90 p 0.152 (.193 p .006) DETAIL “A” 0o – 6o TYP GAUGE PLANE 1 0.53 p 0.152 (.021 p .006) DETAIL “A” 2 3 4 1.10 (.043) MAX 0.86 (.034) REF 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 0.65 (.0256) BSC 0.1016 p 0.0508 (.004 p .002) MSOP (MS8E) 0210 REV F NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 3632fb 20 LTC3632 REVISION HISTORY (Revision history begins at Rev B) REV DATE DESCRIPTION B 6/10 Text updates in Description PAGE NUMBER Updates to Electrical Characteristics Updates to graphs G08, G09, G17, G18, G19 Updated description for Pins 8 and 9 in Pin Functions 1 3 4, 5, 6 6 Text updates in Operation section 8, 9 Text updates in Applications Information section 13 Figure 10 graphic added 16 Asterisk and related text added to Typical Application 22 Related Parts updated 22 3632fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 21 LTC3632 TYPICAL APPLICATION 5V, 20mA Regulator for Automotive Applications VBATT 4.5V TO 50V TRANSIENTS UP TO 60V L1 1mH VIN CIN 1μF SW LTC3632 RUN ISET SS VFB R1 470k R2 88.7k HYST GND CIN: TDK C3225X7R2A105M COUT: KEMET C1210C106K4RAC L1: COILTRONICS DRA73-102-R COUT 10μF VOUT* 5V 20mA 3632 TA06a *VOUT = VBATT FOR VBATT < 5V RELATED PARTS PART NUMBER LTC3631/LTC3631-3.3/ LTC3631-5 LTC3642/LTC3642-3.3/ LTC3642-5 LTC1474 LT1934/LT1934-1 LT1939 LT3437 LT3470 LT3685 DESCRIPTION COMMENTS 45V, 100mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA, 3mm × 3mm DFN8, MSOP8E 45V, 50mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA, 3mm × 3mm DFN8, MSOP8E VIN: 3V to 18V, VOUT(MIN) = 1.2V, IQ = 10μA, ISD = 6μA, 18V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter MSOP8 VIN: 3.2V to 34V, VOUT(MIN) = 1.25V, IQ = 12μA, ISD < 1μA, 36V, 250mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation ThinSOT™ Package 25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO Controller 60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation 40V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter with Burst Mode Operation 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10μA, 3mm × 3mm DFN10 VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100μA, ISD < 1μA, 3mm × 3mm DFN10, TSSOP16E VIN: 4V to 40V, VOUT(MIN) = 1.2V, IQ = 26μA, ISD < 1μA, 2mm × 3mm DFN8, ThinSOT VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, 3mm × 3mm DFN10, MSOP10E 3632fb 22 Linear Technology Corporation LT 0610 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2009