2 Switch-Forward Current Mode Converter

AND8373/D
2 Switch-Forward Current
Mode Converter
Prepared by: Thierry Sutto
ON Semiconductor
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APPLICATION NOTE
Introduction
This application note describes the design of 120−W,
125 kHz, two−switch forward current mode converter with
the NCP1252 controller. It can viewed the practical
implementation of the 2−switch forward converter example
described in Ref. [1].
The NCP1252 controller offers everything to build
cost−effective and reliable ac−dc switching power supplies
implementing the forward converter: NCP1252 detects an
output overload without relying on the auxiliary Vcc, a
Brown−Out input offers protection against low input
voltages and improves the converter safety. Finally a SOIC8
package saves PCB space and represents a solution of choice
in cost sensitive projects.
The power supply described here operates from a dc input
voltage, as the forward converter is usually connected after
a Power Factor Correction (PFC) stage. It generates a 12−V
output at 10 A. The efficiency at full load is close to 90% at
the nominal output of the PFC.
A major advantage of the two−switch forward converter
is that the power switches only block the supply voltage
instead of twice the supply voltage as in the flyback or
single−switch forward converter.
Here after, the complete specification, of the two
switch−forward converter is described:
Table 1. Specification
Description
Input voltage Range
Value
Units
350−410
Vdc
Output Voltage
12
Vdc
Output Power
96
W
Output Peak Power during 5 sec
per 1 min
120
W
Minimum Output Load Current(s)
0
Adc
Number of Outputs
1
Nominal Output Voltage
12
±5%
Vdc
Maximum Output Current
8
Adc
Maximum Output Peak Current
10
Adc
Output ripple
50
mV
Maximum startup time
<1
s
< 100
mW
90
%
20, 50
& 100
%
> 50
%
Maximum Transient load step of
the maximum output current
50
%
Maximum Output drop voltage from
Iout = 5 to 10 A in 5 ms
250
mV
Standby Power
Target Efficiency at full load
@ Vin = 390 V dc
Load Conditions for Efficiency
Measurements (10%, 20%,..)
Min Load Efficiency (Pout = 1.2 W)
Power Supply Components Calculation
Transformer
The following equation extracted from the buck converter
running in Continuous Current Mode (CCM), turns ratio
will determine the turns ratio of the transformer:
V out + h @ V bulk
March, 2010 − Rev. 1
(eq. 1)
Where:
• Vout is the output voltage
• h is the targeted efficiency
• Vbulkmin is the minimum operating input voltage of the
forward
• DCmax is the maximum duty cycle that the NCP1252
can deliver
• N is the turns ratio of the transformer
Extracting the turns ratio from the previous equation, we
obtain:
N+
© Semiconductor Components Industries, LLC, 2010
min @ DC max @ N
1
V out
+
0.9
hV bulk minDC max
12
350
0.45
+ 0.085
(eq. 2)
Publication Order Number:
AND8373/D
AND8373/D
Given a DIout of 5 A, the above room temperature ESR
components would, generate an output voltage undershoot/
overshoot of:
Using this value in Equation 1, we can estimate the
minimum duty cycle at high line by changing the bulk
voltage parameter:
DC min +
V out
hV bulk
maxN
+
0.9
12
410
(eq. 3)
0.085
DV out + DI outR ESR,max + 5
+ 38.2%
max
sw
Ǔ
+ 0.45 , is
125 k
V bulkmin
L mag +
+ 350 + 13.4 mH
0.1 0.94
10%I p_pk
0.45
DC max
125 k
Fsw
DI L v
(eq. 4)
0.25
w 318 mF
1
10 k
318 m
50 m
v 2.27 A
22 m
(eq. 8)
V out
(1 * DC min)T sw
L
(eq. 9)
(eq. 10)
V
L + out (1 * DC min)T sw w 12 (1 * 0.38) 1 w 26 mH
2.27
DI L
125 k
If we consider a 10% drop in the inductor value at high
temperature and current, let us adopt a 29 mH output
inductor. But as this value is not standard part we will stick
to a 27−mH normalized value.
With the selected inductor value, we can calculate the rms
current in the output capacitor:
(eq. 5)
IC
The above case assumes an ESR much lower than the
capacitor impedance at the crossover frequency:
1
R ESR v
v
2pf cC out
2p
v
Using Equation 8, we can derive a minimum inductor
value for L:
The crossover frequency fc will arbitrarily be selected at
10 kHz. Beyond this value, the converter would pick−up
switching noise and would require a more carefull layout.
Below, the stringent dropout specification would lead to the
selection of a larger output capacitor. Considering a voltage
drop mostly dictated by fc, the output capacitance and the
step load current, we can derive a first capacitor value by
using a formula already encountered:
5
10 k
R ESR,max
DI L +
LC Output Filter:
DI out
w
2pf cDV out
2p
V ripple
To obtain the output inductor value, we can write the buck
ripple expression based on the off−time duration:
Based on this assumption the transformer manufacturer
offered the following transformer core: E30/15/7.
C out w
(eq. 7)
which is acceptable given a specification of 250 mV.
There is a rule of thumb to select an ESR capacitor equal
to the half of the calculated value with Equation 6. This rule
will take into account the process variation of the capacitor
plus some margin for a startup operation of the power supply
at very low ambient temperature.
The final check will include the circulating rms current.
However, given the nonpulsating nature of the buck output,
we do not expect this current to be that high.
Considering the output power level and the selected
capacitor, we can consider the total ripple voltage
contributed by the ESR term alone. Thus, if we adopt an ESR
of 22 mW (approximate value at 0°C), the maximum peak to
peak output ripple current must be lower than:
To ensure enough primary magnetizing current to
properly reset the core (drive the stray capacitance and allow
the voltage across the winding to reverse), one must usually
reduce the primary inductance from the core’s ungapped
value to one that will cause an adequate magnetizing current.
A popular rule of thumb as to make the magnetizing current
around 10% of the primary current. Since the primary
current is 0.94 A peak (the calculation of this peak current
is given on the following paragraph), we will let the
magnetizing current rise to 0.1 A. The desired primary
inductance, then, with a primary voltage of 350 Vdc and a
pulse duration of 3.6 ms
ǒDCF
28.5 m + 142 mV
out
,rms + I out
1 * DC min
Ǹ12t L
(eq. 11)
+ 10
1 * 0.38 + 1.06 A
Ǹ12 2.813
Where:
v 50 mW (eq. 6)
tL +
L out
Vout
Iout
We must also select a capacitor whose worst case ESR
remains below the capacitor impedance at the crossover
frequency, in order to limits its contribution to the transient
output drop. We are going to parallel two 1000 mF FM
capacitors from Panasonic.
@ 1
27 m
+
12
10
Fsw
1
125 k
+ 2.813 (eq. 12)
Given the equivalent capacitor current capability
(5.36 A), there is no problem here.
The secondary side peak current will be:
I s_pk + I out )
DI L
+ 10 ) 2.27 + 11.13 A
2
2
C = 2000 mF, FM series @ 16 V
IC,rms = 5.36 A (2*2.38 A) @ TA = +105°C
On the primary side, this current reflects to:
RESR,low = 8.5 mW (19 mW/2) @ TA = +20°C
I p_pk + I s_pkN ratio + 11.13
RESR,high = 28.5 mW (57 mW/2) @ TA = −10°C
And the valley reaches
ǒ
I p_valley + I out *
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2
Ǔ
ǒ
Ǔ
DI L
N ratio + 10 * 2.27
2
2
(eq. 13)
0.085 + 0.946 A (eq. 14)
0.085 + 0.75 A (eq. 15)
AND8373/D
Based on the following Equation 16, we are able to
calculate the rms current of a pulsating waveform with linear
current (see Figure 1):
I rms + I ǸDC
Ǹ1 ) 13 ǒDI2IǓ
(eq. 16)
I(t)
DIL
I
t
T
DCT
0
Figure 1. Pulsating Waveform with Linear Ripple Current
This waveform exactly despits the current we have with
a forward converter on the primary or secondary side of the
tranformer.
When this current is measured on the primary side, DI
represents the reflected secondary−side ripple summed with
the magnetizing current. Thus if we would like to accurately
I p,rms,10% +
I p,rms,10% +
Ǹ
ǒ
0.45 (1.1
Ǹ
calculate the primary rms current, the magnetizing current
should be added to the Ip_pk calculated with Equation 14.
The magnetizing inductance has been previously calculated
(Equation 4) with 10% of the primary peak current.
Therefore the primary rms current can be written has
followed:
ǒ
DC max (1.1 @ I p_pk) 2 * 1.1 @ I p_pkDI LN )
0.946) 2 * 1.1
0.946
Where:
• DCmax is the maximum duty cycle that the NCP1252
can deliver
• Ip_pk is the peak current calculated by Equation 14.
• ΔIL is the maximum output peak to peak current ripple
• N is the turns ratio of the transformer
The mosfets are selected based on the maximum input
voltage and a derating factor kM of 0.85. If we choose 500 V
devices (in a two−switch forward converter, the transistor
stress is limited to the input voltage), the maximum
high−voltage rail must be limited to
0.85 + 425 V
0.085 )
(2.27
Ǔ
(eq. 17)
Ǔ
0.085) 2
+ 0.63 A
3
If the PFC does not include skip cycle in light−load
operation, chances are that its output voltage will reach the
overvoltage protection (OVP) level. The converter thus
enters a kind of autorecovery hiccup mode. It is therefore
important to check that one respects Equation 18 despite the
OVP detection.
The FDP16N50 has been selected for this application. Its
specification are as follows:
• Package TO220
• BVDSS = 500 V
• RDS(on) = 0.434 W at Tj = 110°C (RDS(on) = 0.31 W @
Tj = 25°C multiplied by 1.4: RDS(on) derating factor for
110°C)
• QG = 45 nC
• QGD = 14 nC
Thanks to Equation 17, we can estimate its conduction
losses as
Mosfet Selection
V bulk,max + BV DSSk M + 500
2.27
(DI LN) 2
3
(eq. 18)
P cond + I p,rms,10% 2 R DS(on) @ T J + 110 oC + 0.632 2
As we are running a 2−switch forward application, the
voltage presents on each power switch at the turn−on is equal
to the half of the bulk voltage: the two following figures
0.434 + 173 mW (eq. 19)
illustrate the 2−switch forward arrangement and the
simulated voltage present on both power switches.
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AND8373/D
Bulk
M1
DRV_hi
VM1(t)
D3
Lout
D1
D2
DRV_lo
C1
D4
M2
Vout
VM2(t)
Figure 2. 2−switch Forward Arrangement
400V
Turn
ON
VDS of the
power mosfets
300V
200V
Turn
OFF
100V
0V
V(VBULK,Q5:c)
750mA
Power mosfets
current
500mA
250mA
0A
SEL>>
860.0us
−I(RSENSE2)
870.0us
Time
878.1us
Figure 3. Power Mosfet Curves: VDS(t) and ID(t) of the Both Power Mosfets
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AND8373/D
At the turn−on the power losses can be expressed as follow:
Dt
V bulk
VDS(t)
2
ID(t)
Ip_valley
PSW,on Losses
t
Figure 4. Turn−on Losses (PSW,on)
The average power losses at the switch on is a triangle
area, the exact calculation can be done via the following
integral calculation:
Where the overlap (Δt) is estimated via the following
equation:
Dt +
Dt
P SW,on + F sw
ŕ I (t)V
D
0
+
P SW,on +
I p_valley
Vbulk
6
2
DS(t)dt
Dt
I p_valleyV bulkDt
12
P SW,on +
I p_valleyV bulk,maxDt
12
0.75
410
12
F sw
46.7 n
(eq. 22)
This overlap estimation does not take into account that the
driver of the NCP1252 is a CMOS type, in that case the
output driver will not deliver a constant current.
Nevertheless the estimation is not so wrong. This overlap is
true for a bipolar driver stage that it delivers a constant
current.
As we have 2 power mosfets with our application the total
switch−on losses will the double of the losses from
Equation 21: 358 mW.
Experimental measurement:
(eq. 20)
F sw
F sw
Based on the previous equation we are able to estimate the
losses at each power mosfet switch on:
P SW,on +
Q GD
14 n
+
+ 46.7 ns
0.300
I DRV_pk
(eq. 21)
125 k + 149 mW
ID(t)
500 mA/div
VDS(t)
100 V/div
P(t) = ID(t) VDS(t)
30 W/div
Time
20 ns/div
Figure 5. Switching Losses During the Turn On of the LOW Side Mosfet
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AND8373/D
ID(t)
500 mA/div
VDS(t)
100 V/div
P(t) = ID(t) VDS(t)
30 W/div
Time
20 ns/div
Figure 6. Switching losses during the turn on of the HIGH side mosfet
losses have not been well estimated. This is probably due to
the wrong estimation of the driver current capability
(IDRV_pk): we took the hypothesis that the driver is able to
deliver a constant current as we have with a bipolar output
stage (UC384X like). But as the NCP1252’s output driver
stage is based on the CMOS technology, the current is
varying with the voltage on the power mosfet gate, thus it is
really difficult to estimate accurately the overlap.
As the losses of the power MOSFET in a 2−switch
forward are very low, the error introduce in these estimations
does not impact to much the heat sink calculation.
The losses at the turn off can be calculated using the
similar method: but now the peak current is at its max value.
The drain−to−source voltage of the power switch is close to
zero and switches to Vbulk.
Figures 5 and 6 represent the power losses of the power
mosfets (high and low side mosfet). From these figures we
can note that the drain to source voltage on the low and high
side mosfet is not at the half of bulk voltage as expected from
the theory and the simulation result.
Drain to source power mosfet voltage is not equal to the half
of the bulk voltage due to the parasitic element from the
transformer and the power mosfet. The low side power mosfet
voltage is equal to 150 V and 240 V for the high side one.
Thus the measured switch−on losses are the following:
• High side switch on losses: 386 mW
• Low side switch on losses: 155 mW
If we compare these experimental results with the theory
from Equation 21 where the switch on losses has been
estimated to 179 mW per switch, we can conclude that the
Dt
VDS(t)
Ip_pk
Vbulk
PSW,off Losses
ID(t)
t
Figure 7. Turn−off Losses (PSW,off)
Based on the equation used for the switch on losses, we are able to estimate the losses at each power mosfet:
P SW,off +
I p_pkV bulk,maxDt
6
F sw +
0.95
410
6
40 n
125 k + 324 mW
(eq. 23)
The overlap (Δt) is estimated via the following equation:
Dt +
Q GD
14 n
+
+ 40 ns
0.350
I DRV_pk
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(eq. 24)
AND8373/D
We are now able to estimate the overall losses on each
power mosfet:
P losses + P SW,on ) P cond ) P SW,off
+ 0.149 ) 0.173 ) 0.324
P losses + 646 mW
Tj
Rqj−c
Where:
• Switch−on losses: PSW,on = 149 mW
• Conduction losses: Pcond = 173 mW
• Switch−off losses: PSW,off = 324 mW
Once we have the total dissipation budget per MOSFET,
a heatsink can be calculated.
(eq. 25)
Tc
Rqc−hs
Rqhs−a
Ths
Ta
Plosses
Figure 8. Thermal path between the power mosfet and the heat sink
Where:
• Tj is the junction temperature of the power mosfet
• Tc is the case temperature of the power mosfet
• Ths is the heat sink temperature
• Ta is the ambient temperature
• Rθj−c is the thermal resistance between the junction and
the case of the power mosfet
• Rθc−hs is the thermal resistance between the case of the
power mosfet and the heat sink
• Rθhs−a is the thermal resistance between the heat sink
and the ambient temperature.
The following condition has to be checked to prevent any
over heating of the power mosfet during worst case
operation:
T jmax * T ambmax u P losses
ȍ Rq
As the magnetizing and demagnetizing voltage are similar
(Vbulk, thanks to the 2−switch forward structure); both on
and reset times are equal.
t reset + I mag,pk
The average current can now be derived in a snapshot:
I mag_avg +
+
+
(eq. 26)
Or it can be written as follow:
R qhsa t
T jmax * T ambmax
P losses
(eq. 27)
Thus the thermal resistance of the heat sink should be
lower than 67.4°C/W. A KL194/25.4/SW from Seifert
(ref.[2]) has been selected (14°C/W).
Diodes Selection
PIV +
The choice of the primary freewheeling diodes depends
on the transformer magnetizing inductor. The magnetizing
peak current can be calculated via the following equation:
V bulk,min DC max
+ 350
L mag
F sw
13.4 m
ǒ
ǒ
2
Fsw
Ǔ
DCmax
) t reset I mag_pk
Fsw
2
Fsw
0.45
) 3.6 m
125 k
Ǔ
(eq. 30)
94 m
2
125 k
Diode such as the MUR160 accommodates the
demagnetization task easily. Usually, in off−line application
as the magnetizing current remains low any 1 A high voltage
diode (500 or 600 V) can do the job.
Let us now take care of the secondary diodes. In the
forward converter, both secondary−side diodes sustain a
similar peak inverse voltage (PIV). Given a turns ratio of
0.085 and the diode’s derating factor kD, the diodes have to
sustain the following PIV:
R qhsa t 67.4 oCńW
I mag_pk +
(t on ) t reset)I mag_pk
I mag_avg + 42.3 mA
* ǒR qjc ) R qchsǓ
t 110 * 65 * (1 ) 1.2)
0.646
L mag
13.4 m
+ 94 m
+ 3.6 ms (eq. 29)
350
V bulkmin
NV bulkmax
+ 0.085 410 + 58 V (eq. 31)
0.60
kD
Thanks to the low PIV value, we are able to select the
following Schottky diode reference: MBRB30H60CT.
This diode (30 A, 60 V in a TO−220) case features a
maximum drop of 0.5 V at 125°C (see Figure 9)
(eq. 28)
45% + 94 mA
125 k
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AND8373/D
On average these diodes would dissipate around 5.3 W or
4.4% of the total output power. In order to improve the
efficiency it can be interesting to implement a synchronous
rectification to replace them.
For these diodes, we can re−use Equation 27 to calculate
the required heat sink.
IF, INSTANTANEOUS FORWARD CURRENT (A)
100
TJ = 125°C
10
TJ = 25°C
1
0.1
R qhs−a t
t 125 * 65 * (2 ) 1.2)
5.33
Vf = 0.5 V @ 10 A
& TJ = 125°C
0
0.2
0.4
0.6
Thus the thermal resistance of the heat sink should be
lower than 8°C/W.
For the demonstration board, the following heat sink has
been selected KL195/25.4/SW from Seifert (ref.[2]). It
provides a low thermal resistance of 6.2°C/W.
1.2
VF, INSTANTANEOUS FORWARD VOLTAGE (V)
Figure 9. MBRB30H60CT, Maximum Forward
Voltage versus Instaneous Current
NCP1252 Component Selection
The series diode would then dissipate the following power
in worst case conditions (Low line and maximum duty
cycle).
Switching Frequency Selection
A resistor connected between the Rt pin and the ground
precisely sets the switching frequency between 50 kHz and
a maximum of 500 kHz. The following curve helps to select
the resistor according the selected switching frequency.
(eq. 32)
P d_on + V fI outDC max + 0.5
10
0.45 + 2.25 W
The freewheeling diode would dissipate slightly more as
it conducts during the off time:
P d_off + V fI out(1 * DC min)
10
(eq. 33)
(1 * 0.39) + 3.05 W
Switching frequency, Fsw (kHz)
+ 0.5
(eq. 34)
R qhs−a t 8.06 oCńW
1.0
0.8
T Jmax * T AMBmax
* ǒR qj−c ) R qc−hsǓ
P losses
Switching frequency versus Rt resistor
500
450
400
350
300
250
200
150
100
50
0
0
20
40
60
Rt resistor (kOhm)
80
100
Figure 10. Switching Frequency Selection
The following equation could also be used to calculate the
resistor value according to the switching frequency
selection:
Rt +
1.95
10 9V R
F sw
t
If we select a 33 kW resistor, this will yield:
F sw +
10 9
125 k
10 9V R
Rt
t
+ 1.95
10 9
33 k
(eq. 36)
2.2 + 130 kHz
The measurement on the final board with a resistor equal
to 33 kW gives us 130 kHz for the switching frequency.
This oscillator resistor will be laid out as close as possible
to the Rt pin (pin #4) of the NCP1252 and its ground (pin #5).
As these pins are really close together, it will be not so
difficult to take into account this requirement. The
robustness of the controller against electrical noise will be
improved.
(eq. 35)
Where:
• VRt is the internal voltage reference present on the Rt
pin and equal to 2.2 V.
If we assume a switching frequency of 125 kHz,
R t + 1.95
1.95
2.2 + 34.3 kW
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AND8373/D
Sense Resistor
measurement and it will improve the robustness of the
power supply. Nevertheless this time constant should not be
too large compared to the switching period of the controller.
It is usually recommended to select a 150−300−ns time
constant for the current sense filter network.
The NCP1252 provides an internal ramp compensation
appearing on the CS pin. The resistor of the RC filter will
play a double function: ramp compensation and time
constant for filtering. Thus the ramp compensation will fix
the resistor value of the RC filter and then the capacitor will
be adjusted to respect the time constant previously defined.
One of the following chapters describes how to calculate
the ramp compensation resistor.
The NCP1252 featuring a maximum peak current to 1 V,
the sense resistor is computed via the following expression,
where a 20% margin appears on the primary peak current
(10% for the magnetizing current and 10% for general
margin):
R sense +
F CS
+
1.2
1.2I p_pk
1
+ 884 mW (eq. 37)
0.946
The power dissipation of the sense resistor with a 20%
margin on the primary peak current amounts to:
PR
sense
+ I p,rms,20% 2R sense + 0.695 2
(eq. 38)
0.884 + 427 mW
Where:
• Ip,rms,20% is the rms current of the primary peak current
with 20% margin on the peak current
As we are using 1206 resistor type sizes with a limited
power dissipation of 250 mW, we have to place 2 resistors
in parallel in order to fit the authorized power capability.
Thus we select 2 resistors of 1.5 W. The new power
dissipation will be 362 mW for both resistors and 180 mW
for each one.
Despite the presence of a Leading Edge Blanking (LEB =
130 ns), it is recommended to insert between the sense
resistor and the CS pin of the controller a small RC filter in
order to remove any parasitic noise from the application.
This small RC network will “clean” the current sense
Brown−out
By monitoring the level on BO pin, the NCP1252 protects
the forward converter against low input voltage conditions.
When the BO pin level falls below the VBO level, the
controllers stops pulsing until the input level goes back to
normal and resumes the operation via a new soft start
sequence.
The brown−out comparator features a fixed voltage
reference level (VBO). The hysteresis is implemented by
using the internal current connected between the BO pin and
the ground when the BO pin is below the internal voltage
reference (VBO).
S
Q
Q
Vbulk
RB O u p
R
BO
−
Grand
Re se t
+
shutdown
BOK
RB Olo
VBO
UVLO reset
IBO
Figure 11. BO Pin Setup
When BO pin voltage is below VBO (internal voltage
reference), the internal current source (IBO) is activated. The
following equation can be written:
The following equations show how to calculate the
resistors for BO pin.
First of all, select the bulk voltage value at which the
controller must start switching (Vbulkon) and the bulk
voltage for shutdown (Vbulkoff) the controller.
Where:
• Vbulkon = 370 V
• Vbulkoff = 350 V
• VBO = 1 V (fixed internal voltage reference)
• IBO = 10 mA (fixed internal current source)
ǒ
V bulkON + R BOup I BO )
Ǔ
V BO
) V BO
R BOlo
(eq. 39)
When BO pin voltage is higher than VBO, the internal
current source is now disabled. The following equation can
be written:
V BO +
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V bulkoffR BOlo
R BOlo ) R BOup
(eq. 40)
AND8373/D
From Equation 40 RBOup can be extracted:
R BOup +
ǒ
Ǔ
V bulkoff * V BO
R BOlo
V BO
We selected the following values for the brown out
resistor divider:
• RBOlo = 5.1 KW + 680 W
• RBOup = 1 MW + 1 MW
(eq. 41)
Equation 41 is substituted in Equation 39 and solved for
RBOlo, yields:
R BOlo +
ǒ
Ǔ
V BO V bulkon * V BO
*1
I BO V bulkoff * V BO
Soft Start
The soft start of the NCP1252 controls the peak current of
the forward converter during the startup sequence: this
prevent any over stress on the power components (primary
mosfet, secondary diode and magnetic component like
transformer and inductor) during this critical phase and it
reduces the output overshoot.
The soft start pin provides a current source connected to
an internal voltage reference. Thus a capacitor connected to
this current source generates a linear voltage slope that
controlling the peak current of the power supply via the
current sense resistor. The SS pin voltage is divided by 4 to
scale down the SS pin voltage to a compatible CS pin
voltage.
(eq. 42)
RBOup can be also written independently of RBOlo by
substituting Equation 42 into Equation 41 as follow:
R BOup +
V bulkon * V bulkoff
I BO
(eq. 43)
From Equation 42 and Equation 43, the resistor divider
value can be calculated:
ǒ
Ǔ
R BOlo + 1 370 * 1 * 1 + 5731 W
10 m 350 * 1
R BOup + 370 * 350 + 2.0 MW
10 m
Clock
Rcomp
S
CS
LEB
Rsense
Iss
DRV
Q
Soft Start
Status
Vdd
Q
R
Fixe d
De lay
120 ms
UVLO
+
−
SS
Soft start
Grand Reset
Figure 12. Soft Start Principle
Based on the following well known equation:
I SS + C SS
V SS
T SS
C SS + I SS
(eq. 44)
T SS
15 m
+ 10 m
+ 37.5 nF (eq. 45)
4.0
V SS
If we select Css = 33 nF, the soft start duration measured
(see Figure 13) on the demoboard is equal to 13 ms.
By extracting from the previous equation the capacitor
value we are able to calculate the soft start duration: if we
select Tss = 15 ms,
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AND8373/D
Soft Start pin
(2 V/div)
T ss = 13 ms
V ss = 4 V
CS pin
(0.5 V/div)
Time
(4 ms/div)
Figure 13. Soft Start Duration Illustration
half of the switching frequency and occur only during
Continuous Conduction Mode (CCM) with a duty−cycle
close or above 50%. To lower the current loop gain, one
usually injects between 50 and 100% of the inductor
downslope. Figure 14 depicts how internally the ramp is
generated:
The ramp compensation applied on CS pin is buffered
from the internal oscillator ramp. A switch placed between
the buffered internal oscillator ramp and Rramp disconnects
the ramp compensation during the off−time DRV signal.
Figure 13 illustrates that the max voltage on the soft start
pin is equal to 6.6 V, but the peak current of the forward
transformer linearly ramps from zero to 4.0 V. Above 4.0 V
on the SS pin, the controller will clamp to the max peak
current.
At the beginning of the soft start the peak current variation
is not linear due to the Discontinuous Mode Current (DCM)
operation of the forward at low peak current and low voltage
on the output.
Ramp Compensation Selection
Ramp compensation is a known means to cure
subharmonic oscillations. These oscillations take place at
Vdd
FB
2R
Clock
R
S
Rcomp
DRV
path
R
Buffered
Ramp
Rramp
Q
Q
−
CS
LEB
+
Rsense
Figure 14. Ramp Compensation Setup
In the NCP1252, the internal ramp swings with a slope of:
V ramp
S int +
F
DC max sw
In a forward application the secondary−side downslope
viewed on a primary side requires a projection over the sense
resistor Rsense. Thus:
(eq. 46)
S sense +
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(V out ) V f) N S
R
L out
N P sense
(eq. 47)
AND8373/D
where:
• Vout is output voltage level
• Vf the freewheel diode forward drop
• Lout, the secondary inductor value
• Ns/Np the transformer turn ratio
• Rsense: the sense resistor on the primary side
Assuming the selected amount of ramp compensation to
be applied is δcomp, then we must calculate the division ratio
to scale down Sint accordingly:
Ratio +
S sensed comp
S int
transformer magnetizing inductance. In some case illustrate
here after the power supply does not need additional ramp
compensation due to the high level of the natural primary
ramp.
The natural primary ramp is extracted from the following
formula:
S natural +
(eq. 51)
If the natural ramp compensation (δnatural_comp) is higher
than the ramp compensation needed (δcomp), the power
supply does not need additional ramp compensation. If not,
only the difference (δcomp−δnatural_comp) should be used to
calculate the accurate compensation value.
Thus the new division ratio is:
(eq. 49)
The previous ramp compensation calculation does not
take into account the natural primary ramp created by the
if d natural_comp t d comp å Ratio +
S sense(d comp * d natural_comp)
S int
(eq. 52)
• Vbulk = 350 V, minimum input voltage at which the
Then Rcomp can be calculated with the same equation used
when the natural ramp is neglected.
If we assume that our forward is based on the following
information:
2 switch−Forward Power supply specification:
• Regulated output: 12 V
• Lout = 27 mH
• Vf = 0.5 V (drop voltage on the regulated output)
• Current sense resistor : 0.75 W
• Switching frequency : 125 kHz
S int +
S natural
S sense
d natural_comp +
(eq. 48)
Ratio
1 * Ratio
(eq. 50)
Then the natural ramp compensation will be:
A few line of algebra determined Rcomp:
R comp + R ramp
V bulk
R
L mag sense
•
•
•
•
•
power supply works.
Duty cycle max : DCmax = 50%
Vramp = 3.5 V, Internal ramp level.
Rramp = 26.5 kW, Internal pull−up resistance
Targeted ramp compensation level: 100%
Transformer specification:
− Lmag = 13 mH
− Ns/Np = 0.087
Internal ramp compensation level
V ramp
F å S int + 3.5 125 kHz + 875 mVńms
0.50
DC max sw
(eq. 53)
Secondary−side downslope projected over the sense resistor is:
S sense +
(V out ) V f) N S
(12 ) 0.5)
R
å S sense
0.087
L out
N P sense
27 @ 10 −6
0.75 + 30.21 mVńms
(eq. 54)
V bulk
R
å S natural + 350 −3 0.75 + 20.19 mVńms
L mag sense
13 @ 10
(eq. 55)
S natural
å d natural_comp + 20.19 + 66.8%
30.21
S sense
(eq. 56)
Natural primary ramp:
S natural +
Thus the natural ramp compensation is:
d natural_comp +
Here the natural ramp compensation is lower than the desired ramp compensation, so an external compensation should be
added to prevent sub−harmonics oscillation.
Ratio +
S sense(d comp * d natural_comp)
S int
å Ratio +
30.21(1.00 * 0.67)
+ 0.0114
875
(eq. 57)
We can know calculate external resistor (Rcomp) to reach the correct compensation level.
R comp + R ramp
Ratio å R
0.0114 + 305 W
3
comp + 26.5 @ 10
1 * 0.0114
1 * Ratio
Thus with Rcomp = 330 W, 100% compensation ramp is applied on the CS pin.
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(eq. 58)
AND8373/D
As the ramp compensation resistor is now calculated, we
are able to calculate the capacitor value of the RC network
connected to the CS pin.
If we assume the time constant of the RC network is equal
to 220 ns, the capacitor value will be:
C CS +
t RC
220 n
+
+ 666 pF
330
R Comp
If we select a 680−pF normalized value for CCS we are
really close to the targeted time constant of 220 ns.
The following figure illustrates the behavior of the RC
filtering network.
(eq. 59)
V Rsense
(0.5 V/div)
CS pin
(0.5 V/div)
Time
(2 ms/div)
Figure 15. Comparison of the Voltage on the Current Sense
Resistor and After the RC Filter
wire: if the original ground wire is used, the current
measurement will be worse than in reality. The following
figure shows a comparison of the wrong and correct current
measurements over the sense resistor.
After filtering the current information of the forward
converter provided to the controller is free of noise.
Note: The measurements done in Figure 15 have to be
done by respecting a true clean ground probe connection.
Usually the scope probe is delivered with a long ground
Current
measurement
on Sense
:resistor with
standard probe
(0.5 V/div)
Short gnd
connection
(0.5 V/div)
Time
(4 ms/div)
Figure 16. Current Sense Measurement on Sense
Resistor with Standard and Short Ground Connection
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AND8373/D
Both following figures illustrate the different probe
connection for measuring the current sense information.
Figure 17 illustrates the standard probe connection: but as
the probe’s ground wire is quite long, the measurement
generates noise (see Figure 16 probe measurement
comparison).
GND
connection
Rsense
Rsense
Current
sense
information
Figure 18. Current Sense Measurement on Sense
Resistor with Short Ground Connection of the Probe
Secondary diode snubber calculation:
Figure 17. Current Sense Measurement on Sense
Resistor with Standard Probe Connection
Without snubbing elements (R2&C2, R4&C6) in parallel
with the secondary diodes (D5): some oscillations appear
across the secondary diode. These oscillations are the result
of the leakage inductance of the secondary side of the
transformer with the capacitor behavior of the diode when
it blocks. Thus in the worst case condition (max input
voltage) it is possible that the maximum reverse voltage of
the diode has been reached; with all the consequence.
Figure 18 illustrates the correct connection for measuring
on a power supply the current sense information. This
connection has been done just by removing the plastic tips
protection of the standard probe and by soldering two short
wires directly to the sense resistor pads.
Figure 19. Secondary Diode Snubbing
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AND8373/D
As depicted by the following figure without snubbing element the oscillations at the nominal input voltage reach the
maximum reverse voltage of the diode (60 V).
62 V with max
VRRM = 60 V for
the MBRB30H60!!!
Voltage
accross the
forward diode
of D5
(10 V/div)
Time
(2 ms/div)
Figure 20. Voltage Applied to the Forward Diode of D5
will be adjusted to damp completely all the oscillations
implying a quality coefficient (Q) of 1:
The principle of the snubber placed in parallel of each
diode is damp the oscillations. The oscillations take place at
the end of the conduction of the diode and they are the
consequences of the leakage inductance of the secondary
winding and the parasitic diode behavior of the diode.
Knowing the leakage inductance of the secondary
winding of the transformer and the oscillation frequency we
are able to determine the resistor to be placed in parallel of
the diode to damp the oscillations. In that case the resistor
R damp + L leakw r + 118 n
2p
22 M + 16 W (eq. 60)
After selecting a 22−W resistor for both secondary diodes,
the oscillation voltage is now limited to 36 V compared to
62 V without snubber at similar input voltage (373 Vdc). A
2.2−nF capacitor is placed in series with the resistor in order
to limit the losses due to the resistor presence.
36 V with snubber
on each diode
Voltage
accross the
freewheeling
diode of D5
(10 V/div)
Voltage
accross the
forward diode
of D5
(10 V/div)
Time
(2 ms/div)
Figure 21. Voltage Applied to the Secondary Diodes (D5) with Snubber
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AND8373/D
Board Performances
The following figures illustrate the general performances of this demoboard.
Startup Delay
As depicted by Figure 22, when the VCC voltage is rising from zero and crossing VCC(ON) level, the NCP1252 sends the first
pulses on the DRV pin only when the 120−ms startup delay is elapsed.
Delay: 120 ms
Vcc pin
(5 V/div)
SS pin
(5 V/div)
DRV pin
(10 V/div)
Time
(40 ms/div)
Figure 22. Startup Delay
Soft Start
compare to the middle and the end of the soft start: this non
linearity is related due to the DCM (Discontinuous Current
Mode) mode of operation of the forward during the first 2 or
3 ms of the soft start when the output voltage is low (< 1 V).
Figure 23 depicts a soft start sequence. The CS pin voltage
is following the shape of the SS pin voltage. At the beginning
of the soft start period, the peak current variation is not linear
SS pin
(2 V/div)
CS pin
(500 mV/div)
12 Vout
(5 V/div)
Time
(4 ms/div)
Figure 23. Soft Start at Full Load (10 A)
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AND8373/D
Jittering Frequency
switching frequency selected by the resistor connected to Rt
pin with a frequency modulation of 330 Hz. The jittering
modulation can be also observed by measuring the Rt pin
voltage.
The Jittering frequency featured by the NCP1252 helps to
spread out the switching noise and eases the filtering of the
power supply. The following figure illustrates the digital
jittering frequency of the NCP1252: ±5% of the centered
Max: 137 kHz
Center:131 kHz
Min: 125 kHz
Switching
frequency
(5 kHz/div)
DRV pin
(10 V/div)
Time
(400 ms/div)
Figure 24. Jittering Frequency Measurement
No Load Regulation
demonstration board does not have any dummy load and
ensure a correct no load regulation. This regulation is
achieved by skipping some driving cycles and by forcing the
NCP1252 in burst mode of operation.
Thanks to the skip cycle feature implemented on the
NCP1252, it is possible to achieve a real no load regulation
without triggering any over voltage protection. The
FB pin
(200 mV/div)
DRV pin
(10 V/div)
Time
(400 ms/div)
Figure 25. No Load Regulation (Real No Load to
the Output) Vout = 12.096 V
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AND8373/D
Step Load Stability
figures show the fast transient response without any
oscillations and exhibit a low drop voltage 165 mV (1.3% of
the nominal output).
In order to test the close loop stability, a maximum step
load of 5 A have been applied on the output. The following
FB pin
(2 V/div)
165 mV
Ac coupling on
12−V output
(0.1 V/div)
Time
(1 ms/div)
Figure 26. Step Load Response from 5 A to 10 A
FB pin
(2 V/div)
165 mV
Ac coupling on
12−V output
(0.1 V/div)
Time
(1 ms/div)
Figure 27. Step Load Response from 0.5 A to 5.5 A
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AND8373/D
Efficiency
The efficiency measurements have been done at room temp at different load conditions and at the nominal load with different
input voltage.
Efficiency versus Output load at ambient temperature
95%
90%
85%
Efficiency (%)
80%
75%
70%
65%
60%
55%
50%
0.0
2.0
4.0
6.0
8.0
10.0
Iout (A)
Figure 28. Efficiency Measurement at Room Temperature and Nominal
Input Voltage (390 V dc) versus Output Load Variation
Efficiency versus Input voltage at ambient temperature
91.0%
90.5%
Efficiency (%)
90.0%
89.5%
89.0%
88.5%
88.0%
360
370
380
390
400
410
Vin (V dc)
Figure 29. Efficiency Measurement at Room Temperature and
Nominal Output Load (10 A dc) versus Intput Voltage
One possible way to improve the efficiency of the demoboard is to implement a synchronous rectification, it will improve
by some percent the overall efficiency.
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Figure 30. Main Demoboard Schematic
1k
1
0
C15
220pF
0 TP 1 1
Rt
R20
39k
U4
NCP1252
1
2 FB SS 8
Vc c
3 BO
7
CS DRV
6
4 RT
GN D 5
1
0
1nF
R19 CS2
TP 8
1
C11
TP 6
SS
0
R3
47k
TP 1
M1
D6
MUR160
0 TP 9
DRV
100nF
DRV
0
VC C C12
NC
35V
C13
33nF C10
CS
D8
1SMA5931
0
D9
VC C
R13
1R 5
Vc c
0
1R 5
R12
D4
BZ X84C 18/ Z TX
0
R10
47k
TP 3
D R V_H _ref 0
TP 4
DRV_L
M2
DRV_LO
DRV_HI_ref
1
R18
10
100 1% TP
CS1
6200 1%
CS
R21
1
1nF
C14
0
FB
1
TP 7
BO
1
TP 5
FB
GN D
J5
0
DRV_H
D3
1SMA5931
DRV_HI
1
200k 1%
R16
1M 1%
0
2
2
2
2
1
1
R14
1M 1%
KL195
2
1
1
KL194
1
1
F D P16N 50
R17
0
HS2 Mosfet
1
HS3 Diode
2
2
1
FDP16N50
Vbulk
KL194
1
HS1 Mosfet
Vbulk
V12V
1
J202
Vc c
1
2 DRV
3
GN D
J204
DRV_HI
1
DRV_HI_ref
2
3
DRV_LO
4
CS
GN D 5
0
0
H EAD ER 3
H EAD ER 5
0
Daughter board connectors
U3
TL 4 3 1
R15
4. 7k
TP 2
12V SLB4−I/ 90
J3
T1 10
D5
R4 27uH
TP 1 6
22R
MB RB30H60
1
C5
C4
out put _power
2W
1000uF / F M 1000uF / F M
TP 1 5
C6 2. 2nF
16V
16V
out put _power
6
5
100V
SLB4−I/ 90
XFM R1
J4
R6
2. 2nF
V12V
10R
C7
R9a
R8
0
15k
180R
J6
FB
2
1
R11
R9b
1k
U2
3k
GN D
C9
10nF
SF H 615A_4
D1N4937 SLB4−I/ 90
J1
C1 47uF D1
SLB4−I/ 90
450V
J2
C2
R2 22R
2. 2nF
0
2W
100V 2306−H−RC
MU R 160
1 L1
2
AND8373/D
Demoboard Schematics
Main Board with its Auxiliary Supply
AND8373/D
• Self Vcc supply: in that case the NCP1010 regulator
As depicted by Figure 30, the NCP1252 feedback is
implemented via a TL431 arranged in a Type 2 corrector.
The power supply can be connected directly to a dc source,
where the minimum startup voltage is 370 Vdc. There are 2
options for supplying the Vcc to the NCP1252:
• External Vcc supply: in that case the dedicated
connector for the Vcc can be used to supply the 15 Vdc
to the demoboard. The NCP1010 controller for the
auxiliairy supply must be removed from the board.
U104
1
GN D 8
2 VC C GN D 7
NC
3
4 GND DRAIN 5
FB
(see Figure 31) is used to build the 15 V output via a
buck converter for stepping down the main bulk voltage
to the Vcc level. This auxiliary self supply is
implemented on the main board close to the bulk
voltage.
Vbulk
NCP1010P60
+
C102
47uF / 25V
D101
BZ X84C 12
R101
1k
C101
1n
R102
1k
0
L101
1
2. 2m H
2
Vc c
+ C103
47uF / 25V
D102
MUR1 6 0
0
Figure 31. Auxiliary Supply Based on the NCP1010 @ 65 kHz
Daughter Board: Driver Stage
We selected to drive also the low side power mosfet with the
pulse transformer to prevent any difference on the way to
drive the low and high side mosfet.
The daughter board or the drivers for the high side and low
side mosfet is done with a pulse transformer from PREMO.
D 301
MMSD 4148
MMBT489LT1G
Q301
XFM R 2
6
4
R 302
47
1
2
3
Vc c
DRV
GN D
J 202
H EAD ER 3
Q302
MMBT589LT1G
C 301
10n
C 302
220nF
3
R 304 1k
Q303
MMBT589LT1G
5
2
1
2
D 302
MMS D4 1 4 8
R301
47R
U 301
1
D 303
R 305
47R MMS D4 1 4 8
DRV_HI 1
DRV_HI_ref 2
D R V_LO 3
DRV_LO_ref 4
5
0
J 301
GN D
R 306
1k
Figure 32. Daughter Board: Driver Stage
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Q304
MMBT589LT1G
J 302
H EAD ER 5
AND8373/D
Demo Board Picture
DCin+
DCin GND
Daughter
Board Drivers
Out GND
12 V Out
Figure 33. Top View of the Demo Board
NCP1252 with its
Surrounding Components
Figure 34. Bottom View of the Demo Board
References
1. C. Basso, “Switch Mode Power Supplies: SPICE Simulations and Practical Designs”, McGraw−Hill, 2008.
2. Heatsink manufacturer link: http://www.seifert−electronic.de/en/produkt.php?id=50 or direct link to the datasheet
http://www.digtion−medien.de/seifert/Uploads/seifert_punkte/pdfs/50%5B0%5D.pdf
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arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
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AND8373/D