APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators 1. Introduction _________________________________________________ Power supply circuitry is always limited by cost, PCB area, height and by the desire to reduce complexity. The majority of systems today are mixed-signal, and in all but the simplest cases, the various analog and digital circuits need several different supply voltages to operate. One way to add additional outputs to a power supply without adding additional control ICs is to replace the standard inductor of a buck regulator with a multi-winding inductor. One winding is energized by the buck regulator, and there are many ways to create a second output by rectifying and filtering the voltage induced in the second winding. Buck regulators are found in nearly every power supply architecture, and the multi-winding inductor can be custom-wound to provide various turns ratios and even multiple outputs, but in the interest of controlling both complexity and cost, this application note will focus on off-the-shelf, 1:1 coupled inductors. Würth Elektronik eiSos offers several families of 1:1 coupled inductors with varying power levels and pinouts, many of which are suitable for adding a secondary output to a buck regulator. 2. The Three Most Common Used Topologies _______________________ Figures 1a, 1b and 1c show the three topologies that are most often used. Figure 1a shows how the secondary voltage, VOUT2 can be stacked on the main output voltage, VOUT1, creating a total voltage with respect to system ground of (2 x VOUT1). Figure 1b shows how the return of the secondary output can be connected to the common of an isolated secondary, creating an isolated voltage that could be used to power sensors or isolated communications equipment. (Note: most 1:1 inductors are not rated to withstand the voltages in isolated AC mains or telecom circuits. Isolation with 1:1 inductors is used to insulate against electrical noise.) In this case, the average value of VOUT2 is equal to VOUT1. Figure 1c shows how the system ground can be shifted to the other side of the secondary output capacitor to create a negative voltage. In this case, the absolute value of V OUT2 is equal to VOUT1. Figure 1a: Stacked for 2x VOUT Figure 1b: Isolated VOUT2 Figure 1c: Negative VOUT2 In all three cases the control loop of the buck regulator only regulates VOUT1, and hence the actual value of VOUT2 will vary with input voltage, with the voltage drops across the output diode in the secondary, with the load currents in both the main output and the secondary output and with duty cycle. For this reason a linear regulator is often used on the secondary to provide a truly regulated secondary output voltage. 2013-09-10, ChrR Page 1 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators 3. Design Example ______________________________________________ The following equations and design philosophies will guide the user through the steps needed to add a secondary output to a non-synchronous buck regulator using a control IC with a fixed current limit. Most buck regulator ICs with internal power MOSFETs have a comparator that monitors the current flowing through the control or “high-side” MOSFET (Q1 in Figure 1), which is equal to the main inductor current when this MOSFET is on. Good quality ICs will specify a minimum threshold for this current limit over the full range of operating temperature, and in most cases when this limit is reached the IC will immediately turn the MOSFET off. As will be shown, this represents a maximum output power that must be shared between the main output and the secondary output. For the design example the IC used will be the TPS54160 from Texas Instruments. The Electrical Characteristics table of the datasheet states that the minimum current limit threshold over the full temperature range is 1,8A. The design specifications are as follows: VIN = 10 V to 14 V, nominal 12 V. Input ripple voltage ∆vIN = 0,2 VP-P VOUT1 = 5,0 V, IO1 = 400 mA to 500 mA, continuous. Output ripple ∆vO1 = 60 mVP-P VOUT2 = 5 V (by definition), IO2-MAX = 200 mA. Output ripple ∆vO2 = 60 mVP-P Switching Frequency, fSW = 500 kHz Estimated Efficiency, η = 90 % POUT = 3,5 W The selection of the external components is the same regardless of how the secondary is connected; hence these calculations are valid for the cases shown in Figure 1a, 1b and 1c. 4. The 1:1 Inductor ______________________________________________ Magnetics form the heart of any switching converter, from this example delivering 3,5 W to a multiphase fullbridge converter delivering kilowatts of power. For the 1:1 coupled buck the principal difference from a standard buck is the increase in peak-to-peak ripple current induced in the main (primary) winding. This is due to the additional energy stored and delivered to the secondary output, and the induced current changes the waveshape from a triangle wave to a trapezoid wave. Figure 2 shows the currents in the primary winding in pink and the secondary winding in green. The dot convention of the two windings is arranged so that the secondary winding conducts while the control MOSFET is off and the recirculating diode (D1) is on. In this way the average voltage over a switching cycle applied to the secondary is approximately equal to V OUT1. Figure 2: Primary (pink) and Secondary (green) Currents in the 1:1 Coupled Inductor VIN = 12,0 V, VO1 = 5,0 V, VO2 ≈ VO1, IO1 = 500 mA, IO2 = 100 mA 2013-09-10, ChrR Page 2 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators It is important to distinguish between the average current and the DC current for the secondary. “Average Current”, IS-AVG, refers to the average height of the trapezoid waveform. “DC Current”, I O2-MAX, refers to the average current delivered to the load on the secondary. These two currents are not the same value, and much confusion can result from mixing them up. For clarity, the Average Current of a trapezoid waveform is equal to the DC Current divided by one minus the duty cycle. For the case of the secondary in a 1:1 coupled buck the maximum values are of interest for worst case calculations: DMAX VOUT 1 VD1 5 0,5 0,54 EQ.1 VIN MIN VD1 10 0,5 I A AVG I O 2 MAX 0,2 A 0,43 A EQ.2 1 DMAX 0,46 VD1 is the typical forward voltage of the recirculating diode, D1 The inductance per winding is selected in the same way as with a standard buck, and is based upon the control of the peak-peak ripple current in the primary winding, denoted Δip-TRI: DMIN VOUT 1 VD1 5 0,5 0,39 EQ.3 VIN MAX VD1 14 0,5 LMIN DMIN VIN MAX VOUT 1 EQ.4 i P TRI f SW 5. Inductor Ripple Current _______________________________________ One of the most important decisions to be made in designing the converter is how much ripple to allow. In normal buck converters the peak-peak ripple is usually set at 20 % to 40 % of the maximum DC output current. The range of 20 % to 40 % represents a good compromise, tested and confirmed by the design of countless switching converters, and gives a balance between size (larger ripple requires less inductance and therefore smaller inductors) and efficiency/noise (smaller ripple leads to lower RMS currents and lower EMI). EQ.4 selects the inductance for the purely triangular portion of the primary ripple current, but the total ripple current, ∆iP, is equal to the sums of primary triangular and the total secondary ripple currents. Due to this additional ripple, the recommended range for ∆iP-TRI is lower - from 10 % to 30 % of the maximum output current of the primary, IO1-MAX. For this example ∆iP-TRI = 30 % = 0,15 A: LMIN 0.39 14V 5V 45,5H 0.15 A 500kHz The next highest standard value of inductance is 47 µH, and this value, denoted L1, will be used henceforth. The triangular portion of primary ripple current is then: i P TRI DMIN VIN MAX VOUT 1 14V 5V 0.39 0,146 AP P L1 f SW 47 H 500kHz Calculating the ripple current in the secondary is unfortunately much less straightforward. This quantity is highly dependent upon the leakage inductance of the 1:1 inductor, the load current, the forward voltage drop of the output diode, and the DC resistance (DCR) of the inductor windings (Figure 3). RDC + LLEAK/2 LLEAK/2 RDC +VRDC- VOUT1 - +VD2- IOUT2 Figure 3: 1:1 Inductor with 1st Order Parasitic Components and Voltage Drops 2013-09-10, ChrR Page 3 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators Closer inspection, especially at higher secondary output currents reveals that Δi S is not purely triangular, but is in fact a parabola due to the trapezoid waveform of the voltage applied across the leakage inductance. An approximate equation is given here, based upon the assumption that the average voltage across the leakage inductance is constant and is equal to the forward voltage of the output diode: iS 2 VD 2 2 0,5V (1 DMIN ) iS (0,62) 0,40 AP P EQ.5 LLEAK f SW 3,1H 500kHz The difficulty in using this expression stems from accurately measuring the leakage inductance and from accurately predicting (or measuring) the diode voltage VD2. Leakage inductance is not always specified by inductor manufacturers, but it can be measured by short-circuiting one set of terminals of the inductor and measuring the inductance of the other terminals. For this example LLEAK was measured as 3,1 µH. Diode voltage must be assumed, hence a value of 0,5 V will be used. With ΔiS estimated the total primary ripple current can be calculated: iP iPTRI iS 0,146 A 0,40 0,55 AP P EQ.6 Peak current can now be calculated for the primary: I P PEAK I O1 MAX i P 0,55 A 0.5 A 0,77 A EQ.7 2 2 Peak current can be calculated for the secondary after the peak-peak secondary ripple current has been calculated: I S PEAK I S AVG iS 0,40 A 0,42 A 0,62 A EQ.8 2 2 The final step required before an actual inductor can be selected is to calculate the RMS currents for both the primary and the secondary. For the primary, using the DC output current is quick and very close to the true RMS value. For the secondary, the RMS value of a trapezoid wave is: I S RMS I S AVG 1 DMAX 1 iS 1 3 I S AVG 2 0,33 ARMS EQ.9 6. The Right Inductor ____________________________________________ In conclusion, the desired inductor should have an inductance of 47 µH per winding, be able to support a peak current of 0,77 A without saturating, and able to carry RMS currents of 0,5 A in one winding and 0,33 A in the other without overheating. With 1:1 coupled inductors it is especially important to read the datasheets carefully and to understand the conditions under which the saturation and RMS current ratings are valid. Good quality manufacturers will state these conditions explicitly. The Würth Elektronik WE-DD series of coupled inductors includes the 744 878 470, whose electrical characteristics are repeated below: Order L Code (µH) 744878470 47 IR (A) 0,9 ISAT (A) 1 RDC-TYP (Ω) 0,6 IR 40ºK over ambient temperature when both windings in series are energized by rated current mentioned. ISAT inductance drop of 10% typical when one winding by saturation current mentioned 2013-09-10, ChrR Page 4 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators The ISAT rating is for the combination of the currents in both windings, however one fortunate aspect of 1:1 coupled bucks is that when current in one winding is at its peak, current in the other winding is at its valley, and hence the primary peak current IP-PEAK represents the worst case. 7. Limits for Maximum Secondary Current __________________________ Two conditions limit the maximum current that the secondary output of a 1:1 coupled buck can deliver. The first is derived from the current limit threshold of the control IC. For this example where I LIM = 1,8 A: I O 2 LIMIT (1 DMIN )(2 I LIM 2 I O1MAX iPTRI ) 0.62 (3,6 A 1,0 A 0,146 A) 1,52 A EQ.10 This condition is most likely to occur when both outputs are delivering maximum load current and when input voltage is at a maximum. The second limitation comes from the boundary between Continuous Conduction Mode (CCM), where the current in the primary winding never reaches zero, and Discontinuous Conduction Mode (DCM), where primary current falls to zero before the end of the switching cycle. This limitation applies to non-synchronous buck converters and some synchronous buck converters if their control ICs do not allow inductor current to reverse polarity. Once the converter enters DCM the average voltage across the primary winding is less than VOUT1. VOUT2 tends to droop or even collapse, depending upon the load current it supplies. For all switching regulators the CCM/DCM boundary can be defined as the point when average inductor current equals onehalf of the ripple current. Referring back to Figure 2, it is clear that this condition is most likely to occur when primary load current is low and secondary load current is high. Writing a useful equation is difficult because of the high variability of the secondary ripple current and its dependence upon the non-linear relationship of the secondary output diode forward voltage to forward current. Lab testing is the best way to determine when the converter will enter DCM, and a current probe will show the exact point where primary current hits zero. If a current probe is not available, then a voltage probe and a multimeter will work. Figure 4 shows the switch-node voltage of the primary (where the control FET, inductor primary and diode all connect) just in DCM. The voltage probe will reveal the onset of a damped oscillation before the end of each switching cycle, and a multimeter will show the rapid drop in V O2 once DCM is reached. Figure 4: Primary switch node voltage (yellow). The section circled in red indicates DCM. VIN = 10,0 V, VO1 = 5,0 V, VO2 ≈ 3,0 V, IO1 = 500 mA, IO2 = 200 mA 2013-09-10, ChrR Page 5 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators If the desired secondary load current IO2-MAX is higher than the lower of these two limits, there are several remedies: 1. Current ripple can be reduced by increasing the inductance, increasing the switching frequency, or both. This helps with both types of limit. 2. For current limit: a different control IC with a higher current limit threshold or a controller with external power MOSFETs and an adjustable threshold can be used. 3. For DCM limit: switching to a synchronous buck regulator that allows current in the inductor to reverse polarity and flow below zero allows the converter to maintain an average voltage of V OUT1 across the windings of the 1:1 inductor even when IO1 = zero. Several semiconductor companies have released small, cost-effective synchronous buck regulators targeted towards this application. There is also a minimum current that the secondary output must supply in order to prevent V OUT2 from increasing at light or no-load conditions. The cause of this increase is a gradual transfer of energy to the secondary output capacitors from the energy in the leakage inductance of the inductor. Measuring leakage inductance and attempting to calculate the minimum load is generally impractical, and for this example lab testing with a potentiometer proved quick and effective. To keep the absolute value of V OUT2 below 5,5 V a maximum resistance of 1,1 kΩ is needed, and in the BOM a 1 kΩ resistor is used. 8. Accurac y of the Secondary Output Voltage _______________________ One disadvantage of using coupled inductors to generate secondary outputs is that only the primary output voltage is regulated by the feedback loop. Depending upon the tolerance of the reference voltage and of the feedback divider resistors, the main output V OUT1 enjoys a typical tolerance of ±1% to ±4% of the average VOUT1 value. VOUT2 is unregulated, however, and will shift with changes in input voltage and duty cycle. V OUT2 also shows a greater shift than VOUT1 with respect to the load currents of both the primary and the secondary. This is due to the voltage drops across the diodes and across the DC resistance (DCR) of the inductor windings and the PCB traces. Figure 5 shows a 1:1 inductor treated as an ideal transformer with a coupling efficient of 1, driven by a voltage equal to VOUT1 on the primary. By reflecting the voltage drops from the primary to the secondary it can be seen that VOUT2 is directly proportional to IO1 and is inversely proportional to IO2. VD2 + - RDC VOUT1 + + IO1 IO2 VD1 VOUT2 - + - RDC Figure 5: Primary and secondary voltage drops affect V OUT2 The first-order equation governing VOUT2 is then: VOUT 2 VOUT 1 I O1 DCR VD1 I O 2 DCR VD 2 EQ.11 In most design guides the forward voltage of a Schottky diode is assumed to be 0,5 V, however if VOUT2 is to be predicted with any accuracy the datasheets of the Schottkys used must be consulted or, better yet, the actual voltage drop measured. In practice the tolerance is also affected by the coupling coefficient between the two windings, and the effort of predicting VOUT2 to an accuracy of less than ±10 % is likely wasted, especially when the effect of load transients on either V OUT1 or VOUT2 are considered. For this example, even if the converter were sitting at just the right operating point to make the positive and negative terms of EQ.11 cancel, any load transient ∆i would cause a shift of ∆I * DCR. In conclusion, using V OUT2 without a linear 2013-09-10, ChrR Page 6 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators regulator is only practical for loads that have few or no load transients and that can tolerate a fairly wide range. Table 1 shows the tolerance of VOUT2 for this example over various load and line conditions: VIN (V) IO1 (mA) IO2 (mA) VOUT2 (V) VIN (V) IO1 (mA) IO2 (mA) VOUT2 (V) VIN (V) IO1 (mA) IO2 (mA) VOUT2 (V) 10.0 10.0 10.0 10.0 10.0 50 100 200 500 50 25 25 25 25 50 4.65 4.95 5.03 5.25 3.55 12.0 12.0 12.0 12.0 12.0 50 100 200 500 50 25 25 25 25 50 4.84 5.02 5.13 5.37 4.14 14.0 14.0 14.0 14.0 14.0 50 100 200 500 50 25 25 25 25 50 4.89 5.05 5.16 5.41 4.41 10.0 10.0 10.0 10.0 10.0 10.0 100 200 500 50 100 200 50 50 50 100 100 100 4.51 4.75 4.93 1.67 3.07 4.00 12.0 12.0 12.0 12.0 12.0 12.0 100 200 500 50 100 200 50 50 50 100 100 100 4.74 4.89 5.12 2.38 3.79 4.40 14.0 14.0 14.0 14.0 14.0 14.0 100 200 500 50 100 200 50 50 50 100 100 100 4.85 4.98 5.23 2.93 4.15 4.58 10.0 10.0 500 200 100 200 4.23 2.22 12.0 12.0 500 200 100 200 4.62 3.02 14.0 14.0 500 200 100 200 4.81 3.50 10.0 500 200 3.28 12.0 500 200 3.69 14.0 500 200 4.02 Table 1: Secondary Voltage vs. Input Voltage, Primary Load and Secondary Load 9. Output Capacitors ____________________________________________ Capacitors for smoothing the voltages at the converter outputs are selected with a similar philosophy as the inductor, but with a focus on maintaining a given maximum peak-to-peak ripple voltage instead of a ripple current. For the main output the ripple voltage is a function of the output capacitor´s impedance at the switching frequency and the peak-peak ripple current. The following equations assume that half of the impedance is due to the capacitive reactance, and half comes from the equivalent series resistance (ESR): CO1 MIN i P 0,55 A 4,55F EQ.12 vO1 f SW 4 60mV 500kHz 4 ESR1MAX vO1 60mV 55m EQ.13 2 i P 2 0,55 A For this example, a solid tantalum capacitor with 220 µF and ESR of 40 mΩ meets both requirements and provides much extra capacitance for responding to load transients. As a general practice, any time tantalum, aluminum or other high-ESR type capacitors are used at a switching converter´s input or output a 100 nF multi-layer ceramic capacitor (MLCC) placed in parallel will help reduce high frequency noise. The secondary output requires that CO2 both filter and hold up VOUT2 while the control FET is on. This requires higher capacitance and a higher RMS current rating for an output of the same power. Fortunately VOUT2 only delivers a maximum of 200 mA. Minimum capacitance and maximum ESR are determined as follows: 2013-09-10, ChrR Page 7 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators CO 2 MIN I S AVG DMAX 0,42 A 0,52 7,33F EQ.14 vO 2 f SW 60mV 500kHz ESR 2 MAX vO 2 60mV 143m EQ.15 I A AVG 0,42 A Here it is important to calculate the RMS current as well: I CO2 RMS I O 2 MAX DMAX 0,52 0,2 A 0,18 ARMS EQ.16 1 DMAX 0,38 Tantalum and aluminum capacitors are possibilities, but in the face of heavy RMS current, MLCCs are the best choice. For this example a 22 µF device rated at 10V with X5R dielectric and 1210 case size will be used. The ESR is approximately 3 mΩ and the RMS current rating is about 3 A. One important consideration for MLCCs is their loss of capacitance under a DC voltage bias. This particular capacitor drops to about 16 µF when used at 5 VDC. 10. Input Capacitors _____________________________________________ Input ripple voltage is often overlooked, but plays a vital role in conducted EMI. Input capacitors to buck regulators must withstand a discontinuous, heavy RMS current because they supply most of the AC current to the converter when the control FET is on. As with the secondary output capacitors, after minimum capacitance and maximum ESR have been selected, the capacitor or capacitors chosen must be evaluated for their RMS current ratings. C IN MIN ( I O1 MAX I O 2 MAX ) I IN PK DMAX (1 DMAX ) 0,52 0,48 0,7 A 1,75F EQ.17 v IN MAX f SW 0,2V 500kHz PO MAX i 3,5W 0,55 A P 1,02 A EQ.18 VIN MIN 2 10V 0,9 2 ESR IN MAX v IN 200mV 197m I IN PK 1,02 A EQ.19 The RMS current through the input capacitor is calculated as: I CIN RMS ( I O1MAX I O 2MAX ) DMAX (1 DMAX ) 0,7 A 0,52 0,46 0,35 ARMS EQ.20 MLCCs are again the best choice due to their low ESR and high RMS current ratings. Here, a 25 V rated 10 µF device with X5R dielectric will be used. When biased at 14 V, the actual capacitance is approximately 7 µF. 11. Diodes _____________________________________________________ The recirculating diode for the main output (D1) and the output diode for the secondary (D2) are selected by calculating the average current carried and multiplying by the forward voltage to determine the power dissipation. The diode package is then selected based upon control of the temperature. Schottky diodes are 2013-09-10, ChrR Page 8 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators preferred whenever they are available for their low forward voltage and near-zero reverse recovery charge, two features which combine to make them more efficient and less electrically noisy than PN diodes. For safety and margin when dealing with noise and transients, the reverse blocking voltage of the diodes should be selected at least 20% above the highest voltage in the system, which is VIN-MAX for bucks and coupled bucks. 20% above 14 V is 16,8 V, hence 20 V or higher rated diodes will be used. Power dissipation is determined by the following equations: PD1MAX I O1MAX VD1 (1 DMIN ) 0,5 A 0,5V (1 0,38) 0,16W EQ.21 PD 2MAX I O 2MAX VD 2 0,2 A 0,5V 0,1W EQ.22 The industry standard SMA package with a typical thermal resistance of 95ºC/W is a cost-effective choice, and 20V rated devices are available from many different manufacturers. 12. Control Loop Compensation ___________________________________ Peak Current Mode control (PCM) is a good choice for controlling a buck using coupled inductors for several reasons. When properly designed this control method simplifies the design of the compensation of the error amplifier. As the name implies, PCM also controls peak current in the inductor (in this case in the primary winding) naturally and on a cycle-by-cycle basis. Adding a second winding and a second load to a buck converter changes the control-to-output transfer function of the power stage, comprised of the duty cycle modulator and the output filter. The basic goal of compensating the control loop stays the same, however. This is provide a high gain at DC (for best tolerance of the main output voltage), a high bandwidth (for fast response to load transients) and a high phase margin (for stable response to load and line transients.) There are many design philosophies for compensating control loops, and this application note will focus on a basic method that provides a conservative design that is very stable. 13. Defining the Pow er Stage ____________________________________ The inputs that are needed to develop a linear, small-signal model of the power stage of a 1:1 coupled buck regulator are as listed along with their values for this design example: VIN-MIN = 10 V VOUT = 5,0 V IO = IO1-MAX + IO2-MAX = 0,5 A + 0,2 A = 0,7 A fSW = 500 kHz L1 = 47 µH, Power Stage Resistance, RL = RDSON of control MOSFET + DCR per winding of L1 = 200 + 600 = 800 mΩ Total Output Capacitance, CO = CO1 + CO2 = 220 µF + 16 µF = 236 µF ESR of Output Capacitors, RC = 40 mΩ Current Sense Gain, GI = 1 Load Resistance, RO = VOUT / IO Current Sense Resistance, RSN = 167 mΩ Reference Voltage, VFB = 0,8 V Maximum Duty Cycle, DMAX = 0,52 Slope Compensation Ramp, VM = 0,417 V The slope compensation ramp is used to correct for the error between the current at the point in time when it is sensed (at its peak) and the average value of that current. Without this additional slope, PCM switching converters exhibit a sub-harmonic oscillation for duty cycles of 50 % and above. Because the inductor ripple current is higher in a coupled buck than in a normal buck, a higher slope compensation ramp is needed. Very few buck control ICs allow this ramp to be adjusted, hence it is important to evaluate the ideal slope compensation ramp to foresee any possible problems. 2013-09-10, ChrR Page 9 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators For PCM the ideal compensation ramp is equal to the slope of the current during the off-time of the control FET. For simplicity it is assumed that the waveform is a triangle wave, but having the same peak-peak value as the actual trapezoid wave of ∆iP. The ideal compensation slope is then: VSL IDEAL iP RSN GI 0,55 A 0,167 1 0,092V EQ.23 The fixed ramp of 0,417 V used in the TPS54160 is higher than the ideal ramp, eliminating the danger of sub harmonic oscillation in this example. It should be noted that V M cannot be made too high without consequences – if its value is far greater than the actual sensed slope, the converter becomes more of a voltage mode type and requires more careful compensation around the error amplifier to ensure an adequate phase margin. L1EFF is used to correct for the difference between the slope of the triangular waveform of inductor current in a standard buck and the trapezoidal waveform of the inductor current in a coupled buck. L1EFF VIN MIN VOUT 10V 5V DMAX 0,52 9,45H EQ.24 i P f SW 0,55 A 500kHz AFB is the gain reduction that comes from the output divider resistors. KM is the modulator gain, representing the balance between the sensed current slope (the first term in the denominator) and the compensation slope (second term in the denominator.) V AFB 20 log FB VOUT KM 15,9dB EQ.25 1 (0,5 DMAX ) RSN VM 1 L1EFF f SW VIN MIN 24,6 V EQ.26 V The DC gain of the power stage APS can be defined as follows: APS K M RO 14,3dB EQ.27 RO RL RSN K M RSN GI The frequency-dependent terms of the power stage are the load pole ωC, the ESR zero ωZ, and the doublepole derived from the sampling function of the inductor current with corner frequency ωL. C 1 CO 1 1 RO K M RSN GI krad 1,63 EQ.28 s Z 1 krad EQ.29 106 RC CO s RO RC RL RSN K M RSN G I RO RC krad EQ.30 L 534 L1EFF s The control-to-output transfer function can now be written in the LaPlace domain: 1 GPS ( s ) APS 2013-09-10, ChrR s 1 C s Z s s2 1 2 L ( f SW ) EQ.31 Page 10 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators The frequencies in Hertz of the poles and zero are calculated by dividing each in term in radians/sec by (2π): fC C 260 Hz 2 fZ Z 17kHz 2 fL L 85kHz 2 Plotting gain and phase graphically provides an intuitive way to place the gain and phase response needed for the error amplifier. Figure 6 shows the gain of GPS(s): Figure 6: Power Stage Gain and Phase From a graphical perspective, the goal of error amplifier compensation is to create a complete control loop gain that starts at a high level and rolls off at a slope of -20 dB/decade. The bandwidth or “crossover frequency” is defined as the point where the overall loop gain equals 0 dB, and in general bandwidth is made as high as possible while still maintaining a high phase margin. Phase margin turn is defined as the difference between the phase of the loop when gain equals zero dB and -360 º, the point at which the negative feedback becomes positive and implies an unstable control loop. (In practice the result is usually an unstable duty cycle and fluctuations of the output voltage.) A minimum phase margin of 45 º will ensure a transient response with little or no overshoot or oscillation while still allowing the bandwidth to be as much as 1/5th of the switching frequency. In theory the bandwidth could be as much as 1/2 of the switching frequency, but in practice 1/5th is a good upper limit. In this example the compensation will be Type II, consisting of a pole at 0 Hz to ensure high DC gain at low frequencies, one pole whose frequency can be set arbitrarily and a zero whose frequency can also be set as desired. A good starting point is to set the compensation zero frequency f Z1 equal to the load pole fC and to set the compensation pole frequency f P1 equal to the ESR zero frequency fZ. The compensator pole and zero effectively cancel the power stage zero and pole, leaving only the pole at the origin, and in this way the high gain and continuous -20 dB/decade slope is maintained up until the sampling double pole frequency. The remaining variable to be determined is the gain of the compensation between f Z1 and fP1, often called the “mid-band gain”. A good way to start is to set the overall control loop to cross 0 dB and define the bandwidth in this area of flat gain. This can be done by finding the gain of the power stage at the desired crossover frequency and then setting the gain of compensation to be equal in magnitude but opposite in sign, thus forcing the overall bandwidth to be zero at that point. The TPS54160 uses a transconductance error amplifier with a gain of 97 µS along with two external capacitors and one external resistor to set the poles, zero, and mid-band gain. Figure 7 shows the error amplifier and the external components: 2013-09-10, ChrR Page 11 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators VO RTOP RBOT VCOMP + R1 C2 + - VREF C1 Figure 7: Transconductance Error Amplifier The transconductance of the error amplifier can be read from the datasheet of the control IC, and for the TPS54160 this is gM = 97 µS. The mid-band gain, pole and zero frequencies are as follows: f Z1 AMID g M R1 EQ.32 1 EQ.33 2 R1 C1 f P1 1 EQ.34 2 R1 C 2 Referring to Figure 6, the gain of GPS(S) at a conservative 1/10th of the switching frequency, or 50 kHz, is around -14 dB. Here there is one detail that is sometimes overlooked – the gain reduction from the feedback divider resistors, AFB, defined in EQ.25. This term can be summed with the power stage or with the error amplifier – it makes no difference once the loop is closed. This example assumes that A FB is part of the error amplifier, and since it is a DC gain, it affects the mid-band gain: 0,8V AFB 20 log 15,9dB 5V AMID 10 A0 dB ( 1) AFB 20 10 14( 1) 15, 9 20 31,3 V EQ.35 V Now that mid-band gain has been calculated, R1, C1 and C2 can be calculated: R1 31,3V / V 322k 97 S C2 C1 1 1,9nF 2 322k 260 Hz 1 29 pF 2 322k 17kHz The closest 1 % resistor value is 316 kΩ and the closest 10 % capacitor values are 1,8 nF and 27 pF, respectively. These are substituted into the compensation transfer function: GEA ( s ) g M AFB 2013-09-10, ChrR s R1 C1 1 EQ.36 s( s R1 C1 C 2 C1 C 2) Page 12 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators Figure 8 shows the gain and phase of the error amplifier: Figure 8: Error Amp Gain and Phase The final step is to close the loop by multiplying the power stage and error amplifier transfer functions together. The final gain plot shows bandwidth and the final phase plot shows the phase margin: Figure 9: Overall Control Loop Gain and Phase The actual bandwidth is 20 kHz and the phase margin 79 º, indicating a somewhat slow but very stable control loop. The reason bandwidth and phase margin don´t match the predictions perfectly is the relatively low frequency of the compensation pole. This is typical for converters using output capacitors with high ESR such as aluminum electrolytic and tantalum. The frequency of the compensation pole can be increased up to ½ of the switching frequency if desired. In fact, for converters with purely ceramic output capacitors the ESR zero frequency is typically in the MHz range, and in such cases ½ of the switching frequency is the recommended compensation pole frequency. Increasing the mid-band gain and/or increasing the compensation pole frequency will increase the control loop bandwidth, but will also decrease the phase margin. Several iterations of incrementing the bandwidth and monitoring the phase margin can be performed until phase margin has dropped to 45 º, at which point the maximum recommended bandwidth is reached. 2013-09-10, ChrR Page 13 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators 14. L-C Input Filter w ith Damping __________________________________ Buck converters, buck-boost converters and flyback converters are the three principal topologies that draw a discontinuous current from the input source. Even with good quality input capacitors to supply the heavy AC current, the source will supply some current at AC, and the result is conducted electromagnetic interference (EMI) on the input lines. The longer the leads, PCB traces and wiring harnesses that connect these high ripple DC-DC converters to their input sources, the more likely that this conducted EMI also becomes radiated EMI owing to the unwanted antenna behavior of the leads. An input L-C filter placed close to the DC-DC converter is a good way to reduce conducted EMI, and by filtering before the noise can “contaminate” the input leads radiated EMI is reduced as well. Not every laboratory has access to dedicated equipment for measuring and testing conducted EMI, let alone the special antennas and anechoic chambers needed for radiated EMI. The following procedure is based upon correlation of time-domain current waveforms that can be predicted and measured with a common oscilloscope to differential-mode conducted noise in the frequency domain. 15. Estimating Noise Amplitude ___________________________________ The following equation can be used to estimate the amplitude of the first harmonic of the differential mode conducted noise based upon the input current waveform: A1ST I IN AVG sin( DMAX ) 2 C IN f SW EQ.37 20 log 1V Figure 10 shows the input current of the example circuit during maximum load and minimum input voltage – the worst-case for EMI. Figure 10: Primary switch node (yellow), ΔvIN (AC-coupled, blue), IIN (pink) VIN = 10V, IO1 = 500 mA, IO2 = 180 mA 2013-09-10, ChrR Page 14 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators Average input current, or the average height of the pedestals of trapezoid wave input current can be calculated as: I IN AVG I IN DMAX I IN EQ.38 VOUT ( I O1 I O 2 ) 5V (0,5 A 0,2 A) 0,39 A EQ.39 VIN MIN 10V 0,9 Maximum duty cycle was defined in EQ.1 as 0,52, and the average current is then 0,39 / 0,52 = 0,75 A. The total input capacitance CIN is 7 µF. The estimated noise of the first harmonic is: A1ST 0,75 A sin( 0,52) 2 7 F 500kHz 86,6 dBV 20 log 1V The limit for average common mode noise in many standards for conducted EMC, e.g.: IEC55022 in the range of 500 kHz is 46 dBµV, hence the attenuation needed is equal to the noise A1ST minus this limit. For this example the required attenuation is then ATT = 86,6 – 46 = 41 dBµV. 16. Selecting L and C ____________________________________________ Either the inductance or the capacitance of the input filter must be chosen arbitrarily, and for this example it is the inductance that will be chosen first. Values of inductance between 1 and 10 µH provide a good compromise between size, cost, and the resulting resonant frequency of the input L-C filter. The RMS current rating of the inductor must be greater than the input current IIN, and the peak current rating must be higher than the sum of the average current and one-half of the AC ripple: I SAT MIN I IN PK I IN AVG i P 0,55 A 0,75 A 1,02 A EQ.40 2 2 The Würth Elektronik WE-TPC 2828 series 744 025 002 is a shielded 2,2 µH device in a small footprint (2,8 x 2,8 x 2,8 mm) with a DCR of 60 mΩ and current ratings of IRMS = 1,8 A and ISAT = 2,4 A that is well suited for this example. With inductance chosen, two equations exist for selecting the required capacitance. The first is based upon the resonant frequency of the filter, which should be kept to 1/10 th of the switching frequency or less: C F MIN 1 C IN 2f SW C IN LF 1 10 2 7 F 2 500kHz 7 F 2,2H 1 10 2 13,5F EQ.41 The result of EQ.41 may be negative – this would indicate that with the chosen inductor value it is not possible to attain a filter resonance frequency 10x lower than the switching frequency. The inductor value can be increased if desired, though this comes at a penalty of lower efficiency and/or a larger inductor due to the higher DCR and higher core losses that accompany higher inductance. Setting resonant frequency 10x below switching frequency is a guideline, not a hard limit. 2013-09-10, ChrR Page 15 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators The second equation is a hard limit, and it predicts the minimum capacitance needed to ensure that the voltage ripple at the input to the converter is below the limit set by A TT: C F MIN 2 ATT 1 10 40 LF 2f SW 2 1 2,2H 2 41 10 40 5,0F EQ.42 2 500kHz The capacitor chosen should be greater than the larger of the two values given by CF-MIN1 and CF-MIN2, and multi-layer ceramic capacitors are the best choice. When using MLCCs the loss in capacitance due to a DC voltage bias must be taken into account. Depending upon the case size and voltage rating it would likely take two or more 10 µF capacitors in parallel to provide a true 14 µF or more of capacitance. Before selecting the final value for CF, however, the damping of the input filter must be considered. 17. Filter Damping _______________________________________________ Any time an L-C filter feeds into a switching regulator the potential exists for an oscillation (often called “ringing” or also “power supply interaction”) stemming from the output impedance of the filter and the input impedance of the switcher. Properly designed switchers maintain high power efficiency over a range of input voltage, and one effect of this is that as input voltage rises, input current decreases, and vice-versa. The result is effective negative input impedance. If |-ZIN| is less than or equal to ZOUT of the L-C filter, the input line is likely to oscillate, a behavior that is never beneficial. Even in the absence of an input inductor, the input leads have a parasitic inductance, and when switchers use purely MLCC input capacitors with their very low ESR the potential for oscillation is very real. In this example there is a discrete inductor whose inductance and DCR are both known. With these quantities a damping capacitor CD can be selected to go in parallel with CIN, shown in Figure 11. LF + VIN - CF CD CIN Figure 11: L-C filter with damping capacitor Besides stopping any oscillations, CD reduces the ripple voltage at the input, lowering the amplitude of A 1ST and ATT, and in turn reducing the capacitance needed for C F. The following two equations define the minimum capacitance and the minimum ESR needed for CD to critically damp the filter formed by LF and CIN: CD CIN 4 7F 4 28F EQ.43 ESR 2013-09-10, ChrR 1 LF 1 2,2H DCR 0,06 0,22 EQ.44 2 C IN 2 7F Page 16 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators A negative value for the ESR would indicate that the inductor DCR already provides enough damping resistance. The typical choice for CD is an aluminum electrolytic capacitor. Surface mount aluminum capacitors rated to 16 V or 25 V with 33 µF are available at reasonable cost from many different manufacturers. What´s more, for damping the normally “bad” characteristic, high ESR, is actually helpful. A surface mount aluminum electrolytic capacitor rated at 25 V provides 33 µF with an ESR of 0,34 Ω in a compact 6,3 mm radius footprint, or Radial D8 Case. 18. Re-evaluate C F ______________________________________________ The total capacitance of CIN is now equal to (7 µF + 33 µF), so another iteration of the filter capacitor design is needed. For simplicity the results will simply be stated: the final C F value needed is 5,2 µF. Another 25 V rated, nominal 10 µF MLCC identical to CIN will be used, and since the DC bias is the same, this capacitor will provide at least 7 µF of capacitance. 19. Conclusion __________________________________________________ Using a coupled inductor can help derive a second output voltage without the cost of another complete switching power supply, but the tolerance of the secondary output voltage and the DCM threshold in the primary are difficult to predict with pure mathematical expressions. Successful 1:1 coupled buck designs should be lab tested thoroughly, over line, load and temperature. In most cases the secondary output voltage tolerance is too wide, line regulation too high and load regulation too high to be used directly. For all of these reasons a linear regulator is the recommended way to provide a well-regulated secondary output. 20. Graphs and Oscilloscope Captu res _____________________________ Figure 12: Steady State Waveforms. IO1 = 500 mA, IO2 = 100 mA Ch.1 = Primary SW, Ch.2 = VOUT1 AC-coupled, Ch.3 = Secondary SW, Ch.4 = VOUT2 AC-coupled 2013-09-10, ChrR Page 17 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators Figure 13: Load Transient on Primary. IO1 varies from 200 to 500 mA, IO2 = 100 mA Ch.2 = VOUT1 AC-coupled, Ch.3 = IO1, Ch.4 = VOUT2 AC-coupled Figure 14: Load Transient on Secondary. IO1 = 500 mA, IO2 varies from 50 to 100 mA Ch.2 = VOUT1 AC-coupled, Ch.3 = IO1, Ch.4 = VOUT2 AC-coupled 2013-09-10, ChrR Page 18 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators Power Efficiency, IO2 = 25 mA Power Efficiency, IO2 = 50 mA Power Efficiency, IO2 = 100 mA Power Efficiency, IO2 = 200 mA 21. Complete Circuit Schematic _________________________________ 2013-09-10, ChrR Page 19 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators 22. Bill of Materials ____________________________________________ Designat or Quant ity Packag e/Size Part Number Value1 Valu e2 Value3 Manufacture r C10 1 Capacitor SMT Polarized D Case D-Case 10TPB220 M 220u 10V 4mohm SANYO C2 1 Capacitor SMT Polarized D8 Case Radial D8 EEEFK1E330P 33u 25V 0.34ohm Panasonic 3 Terminal Block 3.81mm Wire to Board 7.62x7. 4x 8.5mm 69121431 0002 300Vrm 10A s 3.81mm Würth Elektronik 3 Resistor Thick Film 0603 75V Vishay-Dale R2 1 Resistor Thick Film 0603 75V Vishay-Dale R3 1 Resistor Thick Film 0603 75V Vishay-Dale R4 1 Resistor Thick Film 0603 75V Vishay-Dale R5 1 Resistor Thick Film 0805 150V Vishay-Dale R8 1 Resistor Thick Film 0603 75V Vishay-Dale C1, C3 2 Capacitor MLCC 1210 C4, C5, C6, C11 4 Capacitor MLCC 0603 C7 1 Capacitor MLCC 0603 C8 1 Capacitor MLCC 0603 C9 1 Capacitor MLCC 1210 D1, D2 2 Schottky Diode SMA (DO214AC) SMA L1 1 Inductor Shielded Drum Core L2 1 Inductor U1 1 IC CN1, CN2, CN3 R1, R6, R7 2013-09-10, ChrR Componen Description t Type CRCW060 31002FK CRCW060 31621FK CRCW060 32553FK CRCW060 33163FK CRCW080 51001FK CRCW060 31911FK C3225X5R 1E106M C1608X7R 1E104M C1608CO G1H182J C1608CO G1H270J C3225X5R 1A226M 10k 1.62k 255k 316k 1k 1.91k 0.1 W 0.1 W 0.1 W 0.1 W 0.12 5W 0.1 W 10u 25V X5R TDK 100n 25V X7R TDK 1.8n 25V X5R TDK 27p 25V X5R TDK 22u 10V X5R TDK B120-13-F 20V 1A 0.5V Diodes Inc 2.8x2.8 x 2.8mm 74402500 2 2.2u Shielded 1:1 Coupled 7.3x7.3 x 4mm 74487847 0 47u Non-sync Buck Regulator eMSOP TPS54160 -10 DGQ 1.8A 60mohm 1A 0.6ohm Würth Elektronik Würth Elektronik Texas Instruments Page 20 of 21 APPLICATION NOTE Designing Low-cost, Multiple Output DC-DC Converters Using 1:1 Coupled Inductors with Buck Regulators IMPORTANT NOTICE Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate, where appropriate to investigate and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. 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Moreover WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications, and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applications-related information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETY-CRITICAL APPLICATIONS. USEFUL LINKS Application Notes: http://www.we-online.com/app-notes Component Selector: http://www.we-online.com/component-selector Toolbox: http://www.we-online.com/toolbox Product Catalog: http://katalog.we-online.de/en/ CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web:http://www.we-online.com 2013-09-10, ChrR Page 21 of 21

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