Application Note Evaluation Board 300W SMPS

300 W g en eral purp o se wid e -r a ng e S M PS
PFC + TTF Evaluation Board
Author: Andrea Merello
Application Note
About this document
Scope and purpose
The presented evaluation design is a 300 W Switch Mode Power Supply (SMPS) employing the Infineon
Combi Controller IC ICE1CS02. The system is partitioned in a Power Factor Correction section (PFC stage)
and a Two-Transistor Forward section (TTF stage).
The power supply can be operated from universal input (90 VAC up to 265 VAC at 50~60 Hz) and provides a
regulated output of 42 VDC. Alternatively, suggestions are given to obtain 24 VDC or different outputs.
Note: The power board is operated by a control daughterboard using ICE1CS02G. Infineon also offers an
alternative solution, using ICE3PCS01G as PFC CCM controller and LM5021-2 as TTF controller. Please
see “Addendum to 300 W general purpose wide range SMPS”[7]on Infineon website.
The design originates from general purpose AC line transformer replacement in low voltage motor drive
applications. The design choices are made to allow hard step loads with minimum output drop.
Attention: This board is intended for evaluation purposes only and is not intended to be an end
product
Intended audience
Design engineers approaching a Switch Mode Power Supply (SMPS) intended for non continuous load
conditions like implemented with Infineon’s 300 W SMPS evaluation board.
Table of contents
1
1.1
1.1.1
1.1.2
System and board description ...................................................................................... 3
Topology description .......................................................................................................................... 3
System protection and features ................................................................................................... 6
Schematic ...................................................................................................................................... 7
2
2.1
2.2
AC line input filter and bridge ....................................................................................... 9
Input filter layout ................................................................................................................................ 9
Input rectifier bridge ......................................................................................................................... 10
3
PFC stage ................................................................................................................. 11
1
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
About this document
3.1
3.2
3.3
3.4
3.5
PFC inductor design .......................................................................................................................... 11
PFC MOSFET design .......................................................................................................................... 13
PFC Shunt resistor ............................................................................................................................. 15
PFC diode design ............................................................................................................................... 16
PFC capacitor design......................................................................................................................... 16
4
4.1
4.1.1
4.1.2
4.1.3
4.1.4
4.2
4.2.1
4.2.2
4.2.3
TTF stage ................................................................................................................. 17
TTF primary side................................................................................................................................ 17
TTF transformer .......................................................................................................................... 17
TTF Primary side MOSFETs ......................................................................................................... 19
TTF shunt resistor ....................................................................................................................... 21
MOSFET gate driving ................................................................................................................... 21
TTF secondary side............................................................................................................................ 23
TTF rectifier diodes ..................................................................................................................... 23
TTF output passives .................................................................................................................... 25
TTF feedback ............................................................................................................................... 26
5
Auxiliary power supply .............................................................................................. 28
6
6.1
Control card with ICE1CS02G ...................................................................................... 30
PFC voltage and current loop ........................................................................................................... 31
7
7.1
7.2
7.2.1
7.2.2
7.2.3
7.2.4
Converter test results ................................................................................................ 33
Converter efficiency .......................................................................................................................... 33
Converter waveforms ........................................................................................................................ 34
PFC plots...................................................................................................................................... 34
TTF plots on primary side ........................................................................................................... 38
TTF plots on secondary side ....................................................................................................... 39
Flyback supply............................................................................................................................. 41
8
Final considerations .................................................................................................. 43
9
Bill of Material (BOM)................................................................................................. 44
10
10.1
10.2
10.3
10.4
Appendix A: Average and rms values ........................................................................... 47
Sinusoidal waveform ........................................................................................................................ 47
Rectangular pulse train..................................................................................................................... 48
Sawtooth waveform .......................................................................................................................... 48
Triangular waveforms ....................................................................................................................... 49
11
Appendix B: Chart representation of power MOSFET losses in hard switching.................. 51
12
References and proposed links ................................................................................... 53
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
System and board description
1
System and board description
The presented design consists in a power supply with aim of reducing system volume and weight, while
increasing power factor, standby consumption and efficiency.
The application requirements follow in Table 1:
Table 1
Application requirements
Input
85 VAC to 265 VAC
Input power factor
95% at 150 W @220 VAC(in); 99% full load
99% at 150 W @110 VAC(in); 99% full load
Output voltage
42 V (24 V variant with changes allowed)
Output power
300 W peak
150 W continuous
Efficiency
90% typ. high-line @220 VAC(in) from 150 W
87% typ. low-line @110 VAC(in) from 150 W
Ambient Temperature
Tested at 25°C ambient
Design procedure developed at 70°C ambient
The PFC stage is using Infineon’s high voltage MOSFET 600 V CoolMOS™ C6 and Infineon’s power silicon
diode technology, Rapid 2 Diode. To improve efficiency and reduce commutation noise Infineon’s thinQ!™
Generation 5 SiC Schottky Diode can be used. This stage works at a fixed frequency of 65 kHz, both in
Discontinued Conduction Mode (DCM) and Continued Conduction Mode (CCM).
The TTF stage takes benefit from 500 V CoolMOS™ CE MOSFET family. This stage works at 130 kHz in CCM
mode.
This paper describes a detailed application circuit, the design choices, the PCB drawings, the oscilloscope
waveforms and the components. Furthermore, magnetics, power losses evaluation in active devices and
heatsink design are included.
1.1
Topology description
The following pictures represent the board topology and it’s partitioning with indication of the main
components.
For a better explanation, the power supply can be divided into 5 parts, shown in Figure 1.
1.
2.
3.
4.
5.
The input filter
The PFC stage
The TTF stage
The output stage
The auxiliary supply
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
System and board description
Rapid 2 Diode
2
3
CoolMOS™ C6
1
CoolMOS™ CE
4
ICE1CS02G
2EDL05I06BF
Combi IC
EiceDRIVER™ Compact
ICE QR/FF
5
Figure 1
CoolSET™
300 W SMPS Evaluationboard - simplified schematic
The following Infineon components are used in the presented design:
Table 2
Used Infineon Components
Product family
Part number
Main parameters
Description
Usage
CoolMOS™ P6
IPP60R190P6
190 mΩ, 600 V
High voltage
MOSFET
PFC MOS
thinQ!™ Gen 5
IDH03G65G5
3 A, 650 V
SiC schottky diode
PFC Diode
Rapid2 Diode
IDP08E65D2
8 A, 600 V
Silicon highspeed
diode
PFC Diode
CoolMOS™ CE
IPP50R280CE
280 mΩ, 500 V
High voltage
MOSFET
TTF MOS
CoolSET™
ICE1CS02G
TTF controller
PFC+TTF IC
CoolSET™
ICE3AR10080JZ
Flyback supply
Flyback IC+MOS
Application Note
10 Ω, 800 V
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
System and board description
Figure 2
Top view of 300 W SMPS Evaluation board
1. Input filter
2. Input bridge
3. PFC stage (bridge filter, PFC inductor (from PULSE electronics), 600 V CoolMOS™ C6 and Rapid 2 diode/
thinQ!TM SiC Generation 5 diode
4. TTF stage (bulk and filter capacitor, 500 V CoolMOS™ CE
5. Auxiliary Flyback supply with CoolSET™
6. Control daughter-board with ICE1CS02G Combi PFC-TTF IC
7. Output inductor (PULSE electronics), capacitor and diodes
8. TTF transformer (PULSE electronics)
Attention: HIGH VOLTAGE!
Discharge completely the bulk capacitor C4 before handling!
Application Note
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PFC + TTF Evaluation Board
System and board description
Figure 3
Bottom view of 300 W SMPS Evaluation board (flipped)
1.
2.
3.
4.
5.
6.
Bypass diode
PFC shunts
TTF shunts
TTF gate driver 2EDL05I06BF
TTF free-wheeling diodes
Auxiliary Flyback supply, ICE CoolSET™ design with Fixed Frequency @100 kHz. Quasi Resonant variant
possible. GREEN LED indicates operation. When LED is off, bus voltage falls below typ. 50 V
7. TTF feedback network
8. Output rectifier diode snubber
9. Output rectifier diode snubber
1.1.1
System protection and features
The power supply is featured as follows:
PFC

Cycle-by-cycle overcurrent protection via shunt sensing

Input Brown-out protection and input under-voltage protection at start-up

Output redundant overvoltage protection (double sensing)

Enhanced dynamic response on load jumps and input mains voltage dips

Soft start of DC bus voltage to limit start-up current

Frequency synchronization with PWM TTF stage. PFC=65 kHz; TTF=130 kHz.

Programmable switching frequency by external PWM signal synchronization
Application Note
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PFC + TTF Evaluation Board
System and board description
TTF

Cycle-by-cycle overcurrent protection via shunt sensing

Output short circuit protection

Output overcurrent protection

Output overvoltage protection

Output soft start

DC bus under-voltage protection via PFC

Slope compensation and limited max duty cycle 47% (or programmable 60%)

Gate driving under-voltage lockout (UVLO) via gate driver

Minimum pulse cutout via gate driver
1.1.2
Schematic
The design consists of a Power Board and a Control Card.
Power board and control card schematics are shown respectively in Figure 4 and Figure 5.
For higher resolution, please use the schematic and layout attachment.
Figure 4
Power board schematic of 300 W SMPS Evaluationboard
Application Note
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PFC + TTF Evaluation Board
System and board description
Figure 5
Control card schematic of 300 W SMPS Evaluationboard
Application Note
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PFC + TTF Evaluation Board
AC line input filter and bridge
2
AC line input filter and bridge
The input filter has been designed to cut common mode noise in the first hundreds kHz band. Highest
available values X2 common mode capacitors (C2 and C3) have been equipped, consistently with the
available board space. Differential mode noise filter is not implemented. Fuse protection and surge
protection has been implemented close to the input connector. 5 Ω NTC (RT1) for inrush current limitation is
provided.
Figure 6
2.1
Input filter
Input filter layout
The filter layout should be as symmetrical as possible. A general rule recommends input filter in an
orthogonal placement to other magnetics, in particular the PFC coil. Our proposal follows.
Connection of GND-Earth (Pin 2 of J1 connector) must be connected to metallic enclosure to reduce
common mode noise. In case there is no enclosure, Y-caps (CY1 and CY2) have to be removed.
Figure 7
Input filter placement on board
Input filter is designed pragmatically in order to minimize conducted noise. Space and footprint for a bigger
common mode chocke is available as shown in Figure 8 .
When more than 2-layer board is used, the recommendation is to mount the Y-caps and extend the
GND_EARTH shield plane on the bottom layer in order to couple the switching noise back to the power
GND.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
AC line input filter and bridge
Figure 8
Input filter layout on proposed design
Please note that the fuse (FUSE1) to the surge suppressor (VAR1) net should be shortened in a better layout,
in order to reduce the stray inductance path for very fast/high voltage surges from the net.
2.2
Input rectifier bridge
The proposed input bridge rectifier is a GBU8J, which is sized according to the power losses. The Power
losses are calculated at maximum output power, 300 W, and minimum input voltage, 85 VAC using simple
assumptions.
We assume that efficiency is η=85%, and forward voltage VF=0.85 V, which is the value at 4 ADC.
 () =

 ()
=
 /  300 / 0.85
=
~ 4.15 
 ()
85 
1 = 2 ∙  ∙ () = 2 ∙ 0.85 ∙ 4.15 = 7 
Eq 1
Eq 2
Heatsinking of input bridge is necessary. Average power consumption considering the power supply duty
cycle of 5 min off/ 5 min on = 50% is 4.5 W, but temperature ripples depending on heatsink thermal inertia.
With the provided heatsink, the temperature at bridge case rises up to 115°C.
Provided from GBU8J a Rth(j-c)=4.0 K/W, the junction temperature is close to 150°C at 7 W continuous load.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
3
PFC stage
The controller provides a fixed frequency CCM PFC control algorithm. The PFC stage is designed to operate
at 65 kHz, which is the standard operating frequency generated by the controller ICE1CS02G, mounted on
the daughter board.
This frequency provides a compromise between inductor ripple and switching losses. Moreover keeps the
3rd harmonic of the PFC (190 kHz) in the 1st band of the conducted emissions (EMC).
Frequency can be changed by providing a synchronization signal at SYNC pin of the controller IC (see
ICE1CS02G datasheet, SYNC pin section).
The PFC stage includes a pre-charge silicon diode (D1) which is recommended when D2 is a SiC diode, to
avoid in-rush currents at hot NTC.
Figure 9
3.1
PFC stage schematic
PFC inductor design
PFC sizing is considered at maximum output power of 300 W and low line (VAC,min = 85 VAC), where input
current is maximum.
L1 has then been chosen in order to reach a maximum current ripple of
∆1, = 20% ∙ 1,
Eq 3
Where ΔIL1,max represents the current ripple and IL1,max represents the peak average inductor current in CCM
mode.
Considering the following conditions:

Input mains voltage: 85 VAC

Input power: 300 W / 85% = 350 W
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
1, =
 ∙ √2 350 ∙ √2
=
= 5.82
,
85
Eq 4
Then ΔIL1,max = 1.2 A from Eq 3.
PFC inductance value can be derived from the simplified equation below.
∆1 =

()
=
 − ()
∙ 
1
Eq 5
1

→ 1 =
∙
∙ (1 −  )
1 − 
() ∙ ∆1, 
1 =
380 
∙ 0.5 ∙ (1 − 0.5) = 1.2 
65  ∙ 1.2
The PFC inductor is provided by PULSE Electronics (PH9297NL).
Specification follows in Figure 10.
Figure 10
PFC inductor specification, from Pulse Electronics
Figure 11
PFC inductor mechanical drawing (side)
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
As can be seen in Figure 11, the inductor design has been segmented in 4 sub-windings.
The segmentation is made to reduce inter-winding coupling, by keeping distance between turns which have
high voltage difference. This minimizes stray inter-winding capacitance, which generates high frequency
oscillations. In fact, current oscillations, superimposed on PFC-current, introduce high common mode noise
at high frequency, which is quite difficult to remove with input filters. Current oscillations can be seen by
measuring directly the inductor current. To see comparative measurement see chapter 7.
3.2
PFC MOSFET design
Active components choice relies on many different considerations, which are related in a very short
summary to power dissipation, thermal design and efficiency.
In order to simplify the choice we start from considering the following conditions:
Table 3
Conditions for MOSFET losses calculation in PFC stage
Parameter
Value
Remarks
TA,max
70°C
-
Rth(c-a) (heatsink)
~15°C/W worst case
Heatsink size is chosen mainly
due to space constraints
fsw
65 kHz
Switching frequency
Vbus
380 V
DC bus voltage
Vin
85 VAC
Input AC voltage
Pin
350 W
Considering 85% efficiency
Figure 12
Heatsink choice (space constrained)
PFC main switch Q1 operates in hard switching in CCM, then resulting to high power losses during
operation. MOSFETs with low Qg, Coss, Eon, and Eoff are required to decrease the switching losses. Optimal
RDS(on) for MOSFET selection has been evaluated by total power loss trade-off between conduction and
switching losses, as explained in Chapter 11: Appendix B.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
Attention: The board is provided with heatsink connected
to GND by a screw placed in the middle of the
board on SMD side.
GND is the 400 V negative reference, then,
unless the mains voltage is insulated, the
heatsink is then at high potential!
A 0 Ω resistor (R46) connects the heatsink from
GND and can be removed.
The MOSFET current, which is considered for the plot is

²()1

2 2
2
()()
= ∫  () ∙ 1,
 0
 ∙ sin()
2 2
2
= ∫ (1 −
) ∙ 1,
∙ 2 ()()
 0

1
4 , 2
= −
∙ 1,
2 3 
Eq 6
1 4 ,
1
4 85  ∙ √2
()1 = 1, √ −
= 5.82 √ −
= 3.55 
2 3 
2 3 380 
Eoss is a function of MOSFET drain voltage, which in CCM is output voltage Vbus.
Optimal on-resistance value then should be RDS (on) opt ~75 mΩ, when VIN (rms) = 85 V and PIN = 350 W (worst
case);
However, Q1 RDS (on) has been set to 190 mΩ considering a non-continuous load operation with 50% duty
cycle between full load and low load conditions. In order to reduce the losses and allow full 100% duty load,
a lower RDSon must be considered with the proposed heatsink design.
190 mΩ device provides additionally a good balance between performance and cost.
Figure 13 (described in Chapter 10: Appendix B), represents plots of losses in the power MOSFET as well as
junction temperature. Chart shows total power losses are about 3 W with RDS(on) Q1 = 190 mΩ. Junction
temperature rise will increase the effective RDS(on), by a factor of about 1.5, reaching higher junction
temperature close to 140°C if steady state load is considered.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
MOSFET Power Losses in PFC stage (85VAC in, 300 Wout)
160
3,5
140
3
120
2,5
100
Static Losses
Dynamic Losses
Total Losses
Junction Temperature
2
1,5
80
60
1
40
0,5
20
0
0
0
Figure 13
Junction Temperature (°C)
Power Losses (W)
4
50
100
150
200
250
RDS (on) (mΩ)
300
350
400
450
MOSFET losses plot for the PFC stage
MOSFET proposal is a CoolMOS™ P6, 190 mΩ (max) IPP60R190P6, which main parameters are shown in
Table 4.
Table 4
Main Parameter of CoolMOS™ P6
Parameter
Value
V(BR)DSS
Breakdown voltage @ ambient
600 V
RG
Internal gate resistance
RDS(ON)
Max RDS @ ambient
Co(er)
Energy related output capacitance
56 pF
Coss
Output capacitance
85 pF
Rth(j-c) TO-220
Max junction-case thermal resistance TO-220
6Ω
190 mΩ
0.83 K/W
The above mentioned procedure to make a choice of the PFC MOSFET is simple and fast. It gives more a
qualitative view than a real quantitative solution. Device need to be tested in the environment, in order to
make the correct choice.
3.3
PFC Shunt resistor
Shunt resistor size is determined some constraints.
First is the overcurrent protection threshold determined by the controller.
The power board is designed for a shunt voltage threshold VCSTH=0,65 V giving the following result:
| |
| |
0,65 
=
=
∆
20% ∙ 1 5,82  ∙ (1,1)
1 + 1
1 +
2
2
= 101 Ω
_ ≤
Application Note
15
Eq 7
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
PFC stage
A second point to keep in mind is the power dissipation:
_ = | | ∙ ()1 = 0,65 ∙ 3.55  = 2,3 
Eq 8
In order to fulfill both requirements, 3 shunt resistors of 330 mΩ in parallel have been used. The choice is on
1 W SMD resistors, type “2512” resistors to keep also low the stray inductance, which would cause spikes on
the sensed voltage.
3.4
PFC diode design
PFC diode proposal is also important, and need some considerations. In particular, diode recovery in hard
switching applications is an important parameter.
This design is a CCM PFC design, where diode recovery happens at almost every commutation period when
in high load conditions.
Recovery generates high current spikes that create

High frequency commutation noise that generates conducted common mode noise at mid/hi- frequency

High amplitude spikes in the MOSFET channel during turn-on. Spikes have to be filtered by a leading
edge blanking filter from the controller (usually always present). This current creates additional losses in
the MOSFET, flowing through the channel at turn-on
The output power of the proposed converter is such that a small SiC diode can be used. Here we have
chosen a 3 A device in 5th Generation thinQ!™ technology: IDH03G65C5.
Table 5
IDH03G65C5 summary of parameters. Please refer to component datasheet for details
Parameter
Value
VRRM
Repetitive peak reverse voltage
IF
Continuous forward current
IF(max)
Non-repetitive peak forward current
VF
Diode forward voltage @ ambient
QC
Total capacitive charge
Rth(j-c) TO-220
Max junction-case thermal resistance TO-220
3.5
650 V
3A
178 A
1.5 VpF
5 nC
3.6 K/W
PFC capacitor design
PFC bulk capacitor is designed according to required hold-up time. The requirement is usually to keep a
minimum voltage of 340 V for 20 ms line drop at maximum output power.
Assuming a constant discharge of the bulk capacitor, Eq 9 can be applied to calculate the minimum
Capacitance required.
1
300  ∙ 20 
2
2
 (
− ,min
) ≥ , ∙ ∆ →  ≥
= 208 
2
3802  − 3402 
Eq 9
Bus capacitor is a 220 µF, 450 V electrolytic, with additional 100 nF high frequency film cap bypass.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
4
TTF Stage
The controller provides a fixed frequency forward current mode control algorithm. The gating output PWM
OUT can be used to drive directly a single MOSFET for a Single Transistor Forward design, or, alternatively,
can drive a Gate Driver IC in order to drive a Two Transistor Forward topology.
The switching frequency is 130 kHz, high enough to reduce the size of magnetic components (TTF
transformer as well as output inductors) and keep reduced switching losses.
Frequency can be changed by providing a synchronization signal at SYNC pin of the controller IC (see
ICE1CS02G datasheet, SYNC pin section).
The TTF stage can be split into primary side (left hand-side of Figure 14) and secondary (insulated) side
(right-hand side of Figure 14).
Figure 14
4.1
TTF stage. Primary and secondary side
TTF primary side
On primary side, the bulk voltage Vbus is controlled by the former PFC stage to 380 V typ. In order to reduce
solution cost, but still with a good safety margin on MOSFETs breakdown voltage, the decision is to use
500 V devices.
500 V CoolMOS™ CE is the perfect fit, with best switching performance / system cost ratio.
The two sides of the TTF stage are analyzed in the following chapters.
4.1.1
TTF transformer
In order to size the transformer for a 42 V output starting from 380 V bulk voltage, we must consider the
maximum duty cycle limit of 50%. Allowing margin for step load response, typical working duty cycle is set
to 40%. Moreover, the higher the duty cycle is, the lower will be the secondary side output of the
transformer.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
Ideally Vout and Vbus are linked by the following Eq 10.
Eq 10


=
∙
  


42 
1
=
=
=
  ∙  380 ∙ 0.4 3.62
Eq 11
Transformer primary inductance LTRA1 value has been set in order to ensure enough magnetizing current
to properly set / reset the core. During the primary-side turn-on period, the magnetizing inductance is
charged by a current IMAG which must be reset during the turn-off time. IMAG peak is set to about 400 mA in
order to keep small the recirculation diodes and limit the recirculation current which creates EMI current
loops. 1 A fast-diodes D3 and D4 are used.
1 ≥
 
380 
0,4
∙
=
∙
= 2,9 
 () 400  130 
Eq 12
Specification from magnetics manufacturer follows on Figure 15 where primary side inductance is higher
and allows lower values of IMAG.
The choice is on Pulse Electronics (PH9326NL), which is a transformer without shield. Shield design for the
transformer was not studied in details, but it might be necessary in certain cases.
Figure 15
TTF transformer specification 42 V, from Pulse Electronics
Primary side current derives from choice of secondary side current ripple.
The choice is done in such a way that at 15% load the output stage is in Boundary Condition Mode, which
means the ripple current touches zero at the end of each cycle. Then, in order to have an output current
ripple of about 30% of the maximum load current, ripple current is calculated from Eq 14:
() =
 300 
=
= 7.15 

42 
 = 30% ∙ () = 30% ∙ 7.15  = 2.2 
Application Note
18
Eq 13
Eq 14
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
4.1.2
TTF Primary side MOSFETs
To size the primary side components, the primary side currents have to be calculated. The following
equations show how to calculate the most important currents of the primary side used to size the switches.
 =


380 
0,4
∙
=
∙
= 334 
1 () 3,5  130 
Magnetizing
current

1
= 6.05  ∙
= 1.68 

3.59

1
= o() ∙
= 8.25  ∙
= 2.3 

3.59
Load current
seen by
primary side
(sec _) = o() ∙
(sec _)
() = MOS(sec _) +  = 2.63 
Primary MOS
peak current
Eq 15
Eq 16
Eq 17
() =
2
2
√((max) + ((sec _) +  ) + (max) ∙ ((sec _) +  )) MOS rms
current
=
3
=2
Eq 18
The following Figure 16 shows the naming conventions used.
IMOSmax
IMOS(sec_max)
IMOS(sec_min)
IMAG
Io(max)
Io(ripple)
Io(avg)
Io(min)
Ton=DPWMT
T
Figure 16
Current shapes on primary MOSFET (IMOS) and output inductor (Io)
As well as for the PFC, for the TTF stage, MOSFETs Q2 and Q3 work in hard switching, and then selection
criteria are the same as in the PFC stage: low Qg, Coss, Eon, and Eoff are required to decrease switching losses.
We are going to use the same graphical procedure as or the PFC.
Application Note
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PFC + TTF Evaluation Board
TTF Stage
The chart shown in Figure 17 points out the power losses of the TTF MOSFETs, showing that a minimum can
be found with RDS(on)= 125 mΩ. For a better cost/performance compromise the decision falls on a 280 mΩ
(max) device, like the IPP50R280CE, which still shows a good thermal behavior and acceptable losses.
Estimated losses for each MOSFET are about 1.5 W, which may reach some above 2 W in hot conditions. Still
Junction temperature is far below 150°C.
As well as in the case of the PFC MOSFET, the chart provides qualitative rather than quantitative information
of the power losses. Nevertheless, the provided indications demonstrated a good matching with the
hardware.
Please refer to chapter 11: APPENDIX B for notes on calculation of the mentioned charts.
MOSFET Power Losses in TTF stage (380 VDC, 300 Wout)
4
160
Static Losses
Total Losses
3
Power Losses (W)
140
Dynamic Losses
120
Junction Temperature
2,5
100
2
80
1,5
60
1
40
0,5
20
0
0
0
Figure 17
Junction Temperature (°C)
3,5
50
100
150
200
250
RDS (on) (mΩ)
300
350
400
450
MOSFET power losses for the TTF stage
Here below are listed the environment conditions used to plot the chart.
Table 6
Conditions for MOSFET losses calculation in TTF stage
Parameter
Value
Remarks
TA,max
70°C
-
Rth(c-a) (heatsink)
~15°C/W worst case
Heatsink choice from space constraints
Value considers that the heatsink is shared and
a multiplication factor 1.5 for non-ideal
mounting, vs datasheet value in Figure 12.
fsw
130 kHz
Switching frequency
Vbus
380 V
DC bus voltage, at switch-on half Vbus is applied
Pin
350 W
Considering 85% efficiency
Application Note
20
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
MOSFET proposal is a CoolMOS™ CE, 280 mΩ (max) IPP50R280CE, which main parameters are shown in
Table 7.
Table 7
IPP50R280CE summary of parameters. Please refer to component datasheet for details
Parameter
Value
V(BR)DSS
Breakdown voltage @ ambient
500 V
RG
Internal gate resistance
RDS(ON)
Max RDS @ ambient
Co(er)
Energy related output capacitance
40 pF
Coss
Output capacitance
49 pF
Rth(j-c) TO-220
Max junction-case thermal resistance TO-220
3Ω
280 mΩ
1.36 K/W
A note shall be pointed out for the choice of the freewheeling diodes D3 and D4.
Note: Even though the magnetization current IMAG they bring may appear small, and the switch off smoothly
(ZCS) it is important that the choice falls on fast reverse recovery diodes. D3 and D4 are MURS160.
Note: Layout design of TTF is not easy, and demagnetization path is usually longer than main path. This path
is partially responsible for emissions and must be carefully studied, starting for the device choice.
Layout of the board described in this application note, is unfortunately not perfectly optimized in this
path.
4.1.3
TTF shunt resistor
Same procedure as for the PFC stage has been taken for the TTF shunt resistor choice.
The power board is designed for a shunt voltage threshold Vcsth=1 V giving the following result:
_ ≤
| |
()
=
1
= 380 ℎ
2,63 
Eq 19
A second point to keep in mind is the power dissipation:
_ = | | ∙ () = 1  ∙ 2 = 2 
Eq 20
In order to fulfill both requirements, and to keep some flexibility, 3 shunt resistors of 1 Ω in parallel have
been used. The choice is on 1 W SMD resistors, type “2512” to keep also low the stray inductance, which
would cause spikes on the sensed voltage.
4.1.4
MOSFET gate driving
The gate driving of the TTF power MOSFETs is done using an Infineon EiceDRIVER™ gate driver IC
2EDL05I06BF. The gate driver IC is a non-interlocking driver, so that high side and low side can be turnedon simultaneously, which is necessary in a TTF topology.
Application Note
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PFC + TTF Evaluation Board
TTF Stage
The gate driving section is shown in Figure 18, where input of the IC is “PWMOUT” (which comes from the
control board) and G3, G2, S2 represent respectively the low-side MOSFET gate and the high-side MOSFET
gate and source.
D11 and D12 are optional turn-off paths and can be removed. There is no Miller induced turn-off effect in
TTF, since both MOSFETs are turned on simultaneously. Turn-on can be soft, since the topology allows zero
load current switching and only capacitive spike take place at turn-on.
Trimming on gate driving resistance are mainly related to a compromise between switching behavior and
high frequency oscillations that may be induced by layout problems.
Figure 18
Gate driver IC schematic
The use of a gate driver versus a pulse transformer (widely used in this topology), is the capability to control
and keep well above the threshold voltage the gate of the MOSFETs, allowing a safe driving in all supply
conditions.
Especially in case of failures or difficult layout situations, the gate driver IC solves grounding problems and
helps placing the drive buffers closer to the MOSFETs to be controlled. In this power supply the choice of a
gate driver IC solution pairs with the use of a controller daughterboard, where there may be different kind of
controllers.
Moreover it avoids the typical effect of pulse transformers, where there can be an effect of offset shifting of
the gate driving signals, which is dependent on duty cycle as shown in the following picture.
Pulse transformer
Pulse transformer
Figure 19
Gate driver
Typical offset shifting when using pulse transformer
Please note that the bootstrap supply cannot be applied to TTF topology to supply the hi-side buffer.
Auxiliary power supply is necessary.
Application Note
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PFC + TTF Evaluation Board
TTF Stage
4.2
TTF secondary side
Secondary side schematic is shown in Figure 20.
Figure 20
4.2.1
TTF secondary side schematic
TTF rectifier diodes
Secondary output voltage VSEC is then rectified by Schottky Barrier Rectifier D5 and D6 (30CTH02PBF): diodes
in common cathode configuration, having then a rectifier (in series with the transformer) and a freewheeling
path. Synchronous rectification (SR) could also be available with two MOSFETs for power loss reduction
with an external driver.
Two snubber protections are also present (R2 - C5 and R5 - C9), in order to reduce inductor L3 overvoltage
and dump the oscillations. R2 – C5 in particular are redundant and can be used for testing.
RC pole should be located in the 10 ... 50 MHz range, and have to be trimmed as a function of snubbering
effect, by watching at the waveforms across the diodes.
Diodes theoretical reverse voltage VSRD is derived from the following Eq 21:
 =

42 
=
= 105 

0.4
Purely theoretical value
Eq 21
But the equation doesn’t take into account the spikes and overshoots due to stray components.
The board is equipped with 200 V diodes, and the measured peak voltage at maximum load gets repetitively
voltages up to 160 V for few nanoseconds, with the proposed snubbers.
Output diodes are one of the main source of losses and shottky diodes are necessary. A second and equally
important characteristic of the diode is the switching behavior, which has to be ultrafast and smooth.
Several kind of diodes has to be proven before a final choice, because the switching behavior strongly
influences the primary side, through the transformer and the conducted and emitted noise.
Rectifier diode losses are due to secondary current ISEC (avg). For rough estimation of power losses for the
secondary side, it might be considered the following chart and calculation at peak output power:
Application Note
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PFC + TTF Evaluation Board
TTF Stage
IDR
Io(ripple)
Io(MAX)
IDF
Io(MIN)
VSRD(ideal)
DR
VSRD
VOUT=42V
VOUT
Ton=DPWMT
T
DF
Figure 21
Output ideal waveforms and circuit
0.4 ∙ (6.052 + 6.05 ∙ 8.25 + 8.252 )
 =  ∙ () = 0.65  ∙ √
 = 0.65  ∙ 4.54 
3
= 2.95 

0.6 ∙ (6.052 + 6.05 ∙ 8.25 + 8.252 )
=  ∙ () = 0.65  ∙ √
 = 0.65  ∙ 5.56 
3
= 3.61 
Eq 22
Eq 23
Calculation of diodes average currents in Eq 22 and Eq 23 follow examples in Chapter 9: APPENDIX A.
The calculation takes into account only the conduction losses.
A very rough estimation of diode switching losses can be done considering the datasheet value of Qrr=120 nC
@125°C vs reverse commutated voltage VSRD=105 V at fsw=130 kHz, which gives about PDSW=1,64 W additional
per diode.
The power board hosts space for mounting two pairs of common cathode diodes.
In order to allow power dissipation for the diodes in continuous load conditions can be done the following
estimation for the required heatsink Rthja =Rthjc+Rthca:
 <
 − 
 − 
150° − 70°
=
=

 +  + 2 ∙  2.95  + 3.61  + 2 ∙ 1.64 
= 8°/
Application Note
24
Eq 24
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
4.2.2
TTF output passives
Design for LC output inductor and capacitor tank starts from the assumption that the entire ripple
component in inductor current ΔIL3 flows through C7, while its average component flows through the load.
During off-time toff (PWM) = (1 – DPWM) ∙ Tsw (PWM), inductor ripple current is:
∆3 =

 (1 −  )
∙ (1 −  ) ∙  =
∙
3
3
()
Eq 25
and the output inductor is calculated as follows:
3 =
(1 −  ) 42  (1 − 0.4)

∙
=
∙
= 88 
∆3 ()
2.2  130 
Eq 26
The output inductor was manufactured by Pulse Electronics (PH9327NL), as well. The core is toroidal and
specification from manufacturer follows in Figure 22:
Figure 22
Output inductor specification, from Pulse Electronics
To find an indication about sizing the output capacitor, the output voltage ripple must be fixed. Let’s start
fixing ΔVout = 0.01%·Vout = 40 mV. Then let’s consider that the inductor current charges and discharges by ±ΔQ
the output capacitor, so that the output ripple can be calculated as follows:
∆ =
∆ 1 ∆3 ∙ ()
= ∙(
)


8
Eq 27
Replacing ΔIL3 from Eq 25 the relative ripple can be obtained as:
∆ 1
1 − 
= ∙
≤ 0.01%
2

8  ∙ 
()
Eq 28
And the output capacitor:
≥
1 1 − 
1
1
1 − 0.4
1
∙
∙
=
∙
∙
= 550 
2
8  ∙ 
8 80  ∙ (130 )2 0.01%
() 0.01%
Eq 29
This value may be able to keep the ripple, but the ESR of the capacitor plays and important role. During load
jumps, the major drops and jumps are due to stray inductances and resistive behavior.
Best choice may is a parallel of smaller electrolytic capacitors and a good low ESR cap.
Our board as a general purpose demonstrator is equipped with a single 1000 µF capacitor and a small high
frequency ceramic capacitor.
Application Note
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PFC + TTF Evaluation Board
TTF Stage
An additional LC output filter is necessary for EMI/EMC compliance, when the output is connected to a load
with cables.
Purpose of this work is to provide a supply for different application, but in particular in conjunction with an
on-board motor drive system. For this reason, the LC filter has not been provided.
4.2.3
TTF feedback
The loop feedback is using a standard approach with TL431 and opto-coupler circuit, as shown in.
Figure 23
TTF voltage feedback
In particular output voltage can be scaled as follows:
 = 2.5  ∙ [1 +
(21 + 22)
] + 3 = 42 
33
Eq 30
Note on DZ3:

TL431A absolute maximum voltage is 37 V and current is 150 mA. DZ3 is used to avoid voltage/current stress
on the regulator.

If different output voltage is desired it can be either chosen a different zener voltage for DZ3 according to
Table 8, or DZ3 can be replaced with a 0 Ohm resistor and it can be changed the resistor divider.
Attention:
The power board output stage is designed for 42 V, 300 W peak power.
The board can work for lower output voltages without changing the
main components, but:

Changing the output voltage requires reducing the output current. Approximate output power is indicated
in Table 7.

When keeping the same output power, output diodes, output inductor and transformer MUST be redesigned
due to higher currents.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
TTF Stage
Table 8
Output voltage and power scaling
Desired output voltage
Action
Approximate Max
Critical points and
output power
optimization
(according to 50% duty)
24 V
Replace DZ3 with 0Ohm
resistor
150 W
Output diodes cooling
Un-efficient usage of
trafo
36 V
DZ3=12 Vz
220 W
Output diodes cooling
42 V
DZ3=18 Vz
300 W
48 V
DZ3=24 Vz
300 W
Following the above mentioned table, primary side relaxes to lower power consumption. The main limit is
the power dissipation of the output diodes, the current of which is inverse-proportionally increasing as
output voltage decrease.
The same transformer can be used for the indicated output voltages, since the controller will act on duty
cycle to provide the desired output, BUT it will work with higher output voltages and peak currents than
needed. A correct transformer design would generate lower output voltage at rectification diodes, which
allows the choice of better rectifiers with lower voltage and current peaks.
A TTF transformer design for 24 V application is also provided by Pulse Electronics (PH9298NL) which
specification is shown in Figure 24.
Figure 24
TTF transformer specification 24 V, from Pulse Electronics
Application Note
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PFC + TTF Evaluation Board
Auxiliary power supply
5
Auxiliary power supply
A flyback circuit has been implemented in order to provide forward converter start-up and supply. Its input
is the bus voltage VBUS, and provides a ground referenced VCC = 15 VDC for the controller board, as well as an
insulated VCC = 15 VDC for the high side gate driving for the TTF.
The Auxiliary power supply is designed to provide a max of about 3 W, and the schematic is shown below.
Figure 25
ICE CoolSET™ Auxiliary power supply schematic
The schematic provides the possibility to use both a quasi-resonant flyback as well as a fixed frequency
flyback. The choice for this design is the ICE3AR10080JZ, which is a 10 Ω, 100 kHz CoolSET™ controller with
integrated 800 V MOSFET, in DIP7 package.
The switching frequency of 100 kHz allows the use of a small transformer, with DIP8 footprint. The
transformer was designed to be with functional insulation, since it doesn’t supply secondary side circuits.
The transformer was provided by Pulse Electronics (PH9323NL) with the attached specifications:
Figure 26
Flyback transformer specification, from Pulse Electronics
Application Note
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PFC + TTF Evaluation Board
Auxiliary power supply
Design guidelines for the flyback design can be found in Infineon website, at CoolSET™ fixed frequency
controller page, searching for ICE3AR10080JZ.
The auxiliary supply is taken from the bus voltage and start-up is guaranteed from 80 VAC input. As soon as
the PFC starts working, the Bus voltage goes to 380 V.The auxiliary power supply works mainly in burst
mode, which allows low power consumption and less emission.
The advantage of a flyback supply is the possibility to add windings and provide insulated supply voltage to
the secondary side. This requires a different insulation for the transformer, but helps especially when a
complete turn-off of the main power stage (PFC+TTF) is required by the secondary side controller, while the
auxiliary works as a low consumption standby supply.
This Application Note does not implement the above mentioned proposal, which is anyway recommended
to achieve very low standby consumption and full control of the power stage from the secondary side.
The auxiliary flyback can be configured in a quasi-resonant configuration allowing to use the ICE2QR4765Z.
The list below shows the modifications on the power board, which are necessary to use one or the other
solution.
Note: While the Fixed Frequency configuration has been tested and optimized, the Quasi-Resonant
configuration is provided by theory and has not been tested.
Table 9
Fixed Frequency vs Quasi-Resonant Flyback component configuration
Component
Fixed frequency flyback
Quasi resonant flyback
Device
ICE3AR10080JZ
ICE2QR4780Z
R35
Mount
Do Not Mount
R17
Do Not Mount
Mount
C16
Do Not Mount
Mount
R22
Mount
Mount
Application Note
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PFC + TTF Evaluation Board
Control card with ICE1CS02G
6
Control card with ICE1CS02G
The control card schematic is shown in Figure 5. The design of the pin-out is such that it would be easy to
use another controller to manage the power board.
Control card is equipped with ICE1CS02G combi (PFC+Forward) controller which features can be found in
the product datasheet available at www.infineon.com.
The control card is shown in Figure 27 and can come soldered to the power-board.
Figure 27
Control card
Connector is described in the following table:
Table 10
1
Control card connector
2
3
VBUS
4
VCC
5
6
PWMout PFCout
7
8
VCC
9
GND PWMcs
10
FB
11
12
PFCcs GND
13
14
RECac+
Pin 1 and Pin 14 are connected to high voltage, please be careful when handling the control cards while
connected to the power board.
A alternative daughter-board using PFC CCM controller ICE3PCS01G + separate TTF controller is also
available. Please ask Infineon representative, in case of need.
In the following chapter it is describer how to size the main control inputs to manage the power board.
Application Note
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PFC + TTF Evaluation Board
Control card with ICE1CS02G
6.1
PFC voltage and current loop
Pin 14: RECac+
Pins VINS (Input Voltage Sensing) and VINS HYS (Input Voltage Sensing Hysteresis) of the combi combi
controller derive from RECac+ thorough a resistor divider. These controller pins sense, in fact, the rectified
AC main line signal RECAC+.
VINS senses the input mains voltage protecting the system from brown-out (BOP): brown-out happens when
the input voltage VINRMS falls below the minimum designed input voltage (i.e. VINRMS < 85.0 VAC). Brown-out
protection avoids large currents flowing in the PFC stage in low-line voltage conditions. When input voltage
is under minimum designed input voltage, i.e. VINRMS < 85.0 V, PFC gate signal is stopped and then resumed
with standard start-up procedure when proper AC input voltage is restored.
Maximum value of RECAC+ (~ √2 ∙ VINRMS) is hence detected by this pin and maintained by a capacitor: when
proper input voltage is applied (85 VAC ≤ VINRMS ≤ 265 VAC), stand-by mode is switched-off. Hysteresis pin (VIN
HYS) prevents the system to oscillate between normal and stand-by mode.
Pin 1: VBUS and Pin 6: PFC out
Here are connected the pins of PFC VSENSE (Bus Voltage Sense / Feedback) and PFC OVP (Overvoltage
Sense Input). Output bus voltage VBUS is sensed via a resistive divider. PFC OUT is then enabled and PFC
duty cycle DPFC is gradually increased by soft-start until VBUS reaches the desired voltage, thus also
enabling the PWM TTF controller section.
On the other hand, PFC OVP provides overvoltage protection by reducing or stopping the PFC duty cycle.
Pin 11: PFC CS
Associated to PFC CS is the controller pin PFC ISENSE (Current Sense Input). This input senses the PFC
MOSFET current by resistance RSHUNT_PFC. RSHUNT_PFC has been designed on the ICE1CS02 PFC peak
current limitation (PCL, VPCL) protection: when the PFC ISENSE signal reaches the PCL threshold, PFC gate
switching will shut down.
Leading Edge Blanking time (220 ns, LEB) is integrated in order to prevent current limit protection from
distortions caused by edge spikes; moreover; an additional filtering can be set by adding a capacitor
between PFC ISENSE and ground. According to IC datasheet, PCL threshold for PFC stage is about –0.60 V.
Check for shunt resistor sizing in chapter 3.3 (“PFC Shunt resistor”) on page 15.
Pin 10: FB and Pin 5: PWM OUT
Pin FB is the TTF stage voltage feedback, associated with Pin 12 – PWM FB (PWM Feedback): output voltage
VOUT is sensed by PWM FB via opto- opto-coupler. PWM OUT (controller pin 8) regulates duty cycle of the
TTF converter in order to achieve a regulated output voltage VOUT. Feedback loop and controller are
electrically isolated: error voltage signal is provided to the controller by a 1:1 current from opto-coupler and
compared to the internal reference voltage (VREF=5 V). PWM OUT is provided to Q2 and Q3 by a gate driver
IC 2EDL05I06BF which is described in chapter 4 (“TTF gate driver 2EDL05I06BF”) on page 6.
Pin 9: PWM CS:
PWM CS provides the TTF current sense information to the controller IC PWM CS pin (PWM Current Sense).
RSHUNT_PWM value has been designed to provide peak current limitation (Vcsth) via ICE1CS02 PWM
protection. According to IC datasheet, PWM threshold for PFC stage Vcsth = 1.00 V. Shunt sizing is explained in
chapter 4.1.3 (“TTF shunt resistor”) on page 21.
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Control card with ICE1CS02G
In case noise filtering is necessary, it can be set by adding a RC-network (R18 and C12); in this board,
τ = R18∙C12 ~ 300 ns.
When the PWM ISENSE signal at pin 9 reaches the limit threshold for a period set externally through pin 11
(PWM Pre-Short), the TTF control is switched off. Timing of Preshort is set to about ~10 ms with C9 = 100 nF.
There is no restart of the system, unless the input line is cycled. Please refer to ICE1CS02G datasheet for
details.
Application Note
32
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
7
Converter test results
Power supply test results will be briefly shown in this chapter. Tests were performed at ambient
temperature with various loads. Collection of data is done on a single prototype (no statistical data was
performed).
7.1
Converter efficiency
Converter efficiency is measured at ambient temperature with the parts proposed in this application note
and it is published here below for 220 VAC input as well as 110 VAC input.
Efficiency
Efficiency
BOM: SiC 3 A G5, 190 mΩ E6 + 280 mΩ CE + Shielded TTF
95
90
85
80
75
70
65
60
55
50
45
40
35
Vin = 230Vac / 60Hz
Vin = 110Vac / 60Hz
0
50
100
150
200
250
300
POUT
Figure 28
Efficiency chart at 42 V output
Efficiency
Power Factor
BOM: SiC 3 A G5, 190 mΩ E6 + 280 mΩ CE + Shielded TTF
1
0,98
0,96
0,94
0,92
0,9
0,88
0,86
0,84
0,82
0,8
Vin = 230Vac / 60Hz
Vin = 110Vac / 60Hz
0
50
100
150
200
250
300
POUT
Figure 29
Power Factor chart at 42 V output
Application Note
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PFC + TTF Evaluation Board
Converter test results
7.2
Converter waveforms
7.2.1
PFC plots
PFC inductor plays a great role in the PFC design. Since this is not an Application Note for magnetics, we are
not going to spend much time on the inductor design, but some important things must be noticed.
The following PFC waveforms are taken with different winding styles, bringing to quite different results.
Yellow: PFC inductor current
Yellow: PFC inductor current
Blue:
Blue:
VDC BUS voltage
VDC BUS voltage
Green: PFC MOSFET Drain Voltage
Green: PFC MOSFET Drain Voltage
Figure 30
Figure 31
Original design
Re-winded manually in 4 sectors
Yellow: PFC inductor current
Yellow: PFC inductor current
Blue:
Blue:
VDC BUS voltage
VDC BUS voltage
Green: PFC MOSFET Drain Voltage
Green: PFC MOSFET Drain Voltage
Figure 32
Re-winded from manufacturer in
4 sectors (final)
Figure 33
Application Note
34
Other manufacturer design
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
Waveform explanation follows:
Figure 30: Original design
Strong inter-winding coupling generate oscillations in the 10 MHz range (superimposed to yellow trace),
clearly visible in EMC test. Beginning and end of winding are superimposed in large part of the toroidal core.
Figure 31: Re-winded manually in 4 sectors
Manual winding didn’t allow same number of turns to test the absence of inter-winding coupling.
Much less coupling reduces drastically oscillations, which eventually shifted to very high frequency and
lower amplitude. Space is left intentionally between beginning and end of winding to reduce the capacitive
coupling between the physical leads.
Figure 32: Re-winded from manufacturer in 4 sectors (final)
Automatic winding with required inductance value in 4 sectors reduced greatly the inter-winding coupling,
with a visible effect of oscillation reduction. Still some high frequency oscillations are present.
Figure 33: Other manufacturer design
Another manufacturer design has been tested, which is shown here to provide different scenarios.
Capacitive coupling between beginning and end of winding is clearly visible at current peak in the big dip
circled in red.
Very high frequency oscillations at MOSFET turn-on are produced instead by the cross-winding coupling,
which in this design is extended along the toroidal shape (blue square).
(Purple trace is PFC diode current, superimposed to inductor current)
Following plots are taken at 85 VAC input. Please note that the horizontal scales may change!
Inductor current
Q1 VDS voltage
Figure 34
Vrec_AC +
PFC 85 VAC, 5 Wout
Application Note
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PFC + TTF Evaluation Board
Converter test results
Inductor current
Q1 VDS voltage
Figure 35
Vrec_AC+
PFC 85 VAC, 100 Wout
Inductor current
Q1 VDS voltage
Figure 36
Vrec_AC+
PFC 85 VAC, 300 Wout, PFC current limiting
At 300 W output with 85 VAC input, PFC stage is at current limit. Cycle-by-cycle limitation get triggered.
Peak voltage at MOSFET drain is less than 500 V.
Application Note
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PFC + TTF Evaluation Board
Converter test results
Inductor current
Vrec_AC+
Q1 VDS voltage
Figure 37
PFC 220 VAC , 5 Wout
Inductor current
Vrec_AC+
Q1 VDS voltage
Figure 38
PFC 220 VAC, 100 Wout
Application Note
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PFC + TTF Evaluation Board
Converter test results
Inductor current
Vrec_AC +
Q1 VDS voltage
Figure 39
7.2.2
PFC 220 VAC, 300 Wout
TTF plots on primary side
Primary side plots are shown below. Gate driver signal for TTF is also plotted in green.
Q1 VDS voltage (PFC)
Q2 source voltage
(TTF HS)
Q3 VDS voltage
(TTF LS)
TTF gating signal
Figure 40
TTF 5 W output load
Application Note
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300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
Q1 VDS voltage (PFC)
Q2 source voltage
(TTF HS)
Q3 VDS voltage
(TTF LS)
TTF gating signal
Figure 41
TTF 5 W output load, pulse skip
Figure 40 shows that PFC drain pulses (in yellow) are synchronized at half TTF pulses as expected.
In Figure 41 effect of pulse skipping is visible in light load conditions. Pulse skipping is not a due to the
control algorithm: it is implemented by the gate driver IC to avoid short pulses on MOSFET gates. The
minimum input pulse-width of the 2EDL05I06BF gate driver IC is specified in the datasheet as follows in
tFILIN=192 ns typ. parameter. A shorter pulse causes a skipped commutation of the outputs. It is to be noticed
that a very small change in the input pulse-width while close to the tFILIN time may create either the high-side
or the low-side to skip the pulse. In the example shown in Figure 41, the high-side pulse is skipped.
From the control point of view, at low loads the system will be kept by the feedback loop around the tFILIN
point, with some more ripple at the output, but a safe operation of the MOSFETs.
7.2.3
TTF plots on secondary side
On TTF output side measurement is done with respect to output ground line.
Output is shown in load-step condition under the following transitions:
Table 11
Load step test
Time [s] – load step duration
Output load current [A]
Output power [W]
0.1
0.1
4
4
7.3
300
1
3
130
4
7.3
300
0.2
0.1
4
Application Note
39
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
(2)
(1)
42 Vout
Output load current
Figure 42
Load step test, overview
Numbered squares (1,2) are detailed in the Figures 43 and 44.
42 Vout, AC coupling, 2.5 V step drop
Load current (forced by electronic
load)
Secondary winding current
42 Vout, AC coupling, initial transient
Load current (forced by electronic load)
Secondary winding current
Figure 43
Step response detail, 4 W to 300 W (1)
Application Note
40
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
42 Vout, AC coupling, 2 V step jump
Load current (forced by
electronic load)
Secondary winding current
42Vout, AC coupling
Load current (forced by
electronic load)
Figure 44
Secondary winding current
Step response detail, 300 W to 4 W (2)
Free-wheeling diode voltage
(SRD voltage), 160 V peak
42 Vout
Transformes secondary side voltage
(ref. to output GND)
Secondary side winding current
Figure 45
7.2.4
300 W load, output waveforms
Flyback supply
Flyback circuit generates the 15 VDC voltage to supply the control board and the low side driver IC supply as
well as the floating 15 VDC supply for the high side TTF MOSFET.
The flyback supply works in active burst mode in order to reduce power consumption to minimum, while
the consequent 15 VDC ripple is well ignored by the control stage.
Application Note
41
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Converter test results
15 VDC output of flyback supply for controller board
CoolSET™ DRAIN pin
voltage
Figure 46
Flyback supply waveforms
Burst frequency changes from output no-load condition to load condition, due to continuous PWM activity
at PFC and TTF stages in the second case.
Flyback waveform is taken as an example in Figure 46. There is margin to load the flyback supply up to some
Watt power.
The flyback supply is designed to start at 85 VAC input, and it is connected to 400 VDC bus.
Application Note
42
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Final considerations
8
Final considerations
The presented demonstrator is designed to provide a regulated output voltage under step-load change
conditions, such as supplying motor drive stages or switching un-constant loads. It has been therefore
indicated as “general purpose” design.
The Combi Control IC card, equipped with ICE1CS02G, can be replaced with different control cards in order
to evaluate different strategies. Some possible alternatives include XMC ARM microcontroller family from
Infineon, which may allow maximum flexibility of the control algorithm.
Points of improvement for the demonstrator are listed:

Provide better cooling for the output diodes, especially for continuous output load

Output filter and EMI tests are not provided: the demonstrator is not designed to be a stand-alone power
supply, but to be included in a wider system. In case of stand-alone usage, the output filter must be
included to reduce Conducted EMI.

The TTF transformer is not shielded. Better EMI figures can be obtained with a copper-band shield in
between primary and secondary side.

In a board layout redesign, it is suggested to reduce TTF freewheeling paths to reduce thermal
dissipation and current loop.

The input GND_EARTH shield can be effective if connected to a metallic enclosure surrounding the
demonstrator, alternatively keep GND_EAERTH disconnected. The demonstrator may come not
equipped with Y caps.

A harder connection of the PFC and TTF heatsink H1 to GND (VBUS C4-minus) should be done in case of a
layout redesign in order to provide better natural shielding to the commutation noise of the input
stages.
In order to improve standby consumption, there can be some alternatives like:

Shut-off the PFC-TTF stages completely (it is needed a redesign of the control board) and keep only the
aux-flyback supply active.

Alternatively, in case a secondary side controller is used, provide an insulated signal to shut-off the
PFC-TTF stages from the secondary side controller. It is needed a redesign the aux-flyback transformer
to provide an insulated supply to the secondary side, which keeps the standby power for the secondary
side controller.
Application Note
43
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Bill of Material (BOM)
9
Bill of Material (BOM)
Table 12
Bill of Material
Designator
Mount
Footprint
Description
Comment
Manufacturer
BR1
No
D_BRIDGE - slim
BRIDGE RECTIFIER
GBU8J 8A 600V
GBU8J
Vishay
C1
No
C_MKT7X18X12.5
MKP cap 100 nF 630 V
DC 250 VAC
100 nF 630 V
Epcos
C2, C3
No
C_MKT8.5X26X15
MKP cap 0.47 uF
0.47 µF
Epcos
C4
No
C_POL30_SNAP10
Aluminum Electrolyte 220 µF 450 V
220 uF 450 V 3000 h
105°
Panasonic
C5, C9
Yes
C_E1 size - Panasonic
1 nF 250 V, 1 nF 250 V
1 nF
Panasonic
C6
No
C_MKT7X18X12.5
MKP cap 100 nF
450 VDC
100 nF 450 V
Epcos
C7
No
C_POL_D10/12.5
Elco, 1000 µF, 63 V
1000 µF, 63 V
Nichion
C8
Yes
1206C
MLCC 4.7 µF X7R
1206 >=63 V
4.7 µF, 63 V
AVX
C10, C11
Yes
1206C
MLCC 4.7 µF X7R
1206 35 V
4.7 µF
TDK
C12
Yes
1206C
MLCC 1 nF X7R 1206
35 V
1 nF
C13
No
C_FKP2_4.5x6x7.2
polypro 220 pF,
>500 V
220 pF, 630 V
Kemet
C14
Yes
1206C
2.2 µF 35 V
2.2 µF
Murata
C15
Yes
1206C
MLCC 2.7 nF X7R
1206 50 V
2.7 nF
C16
No
C_FKP2_4.5x6x7.2
33 pF, 1000 V, FKP2
33 pF, 1000 V NM
C17
Yes
1206C
MLCC 68 nF X7R 1206
50 V
68 nF
C18
Yes
1206C
100 nF 35 V
100 nF
C19
Yes
C_POL_6032-28_Csize
Tantalum 4.7 µF 35 V
4.7 µF
Vishay
C20
Yes
C_POL_D5
Elco, 10 µF, 35 V
10 µF
Panasonic
C21
Yes
1206C
MLCC 2.2 nF X7R
1206 35 V
2.2 nF NM
C22
Yes
1206C
47 pF 35 V
47 pF
C25
Yes
1206C
MLCC 1 nF X7R 1206
35 V
NM
CY1, CY2
No
C_MKT12..5X4X10
Y2 2.2 nF 250 V PET
Y2 2.2 nF
CY3, CY4
No
C_MKT12..5X4X10
B81123 2.2 nF 250 V
PET
2.2 nF 250 V
Epcos
D1
Yes
D0214AB-W
DIODE VISHAY 3 A
600 V trr=2.5 µs
S3J
Vishay
D2
No
TO220-2
DIODE 600 V 6 A
TO220R2L
IDP06E60
Infineon
Application Note
44
Wima
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Bill of Material (BOM)
Designator
Mount
Footprint
Description
Comment
Manufacturer
D3, D4
Yes
D0214AC
DIODE SMA 600 V 1 A
trr=1.8 µs
S1J
Vishay
D5, D6
No
TO220_ST
DIODE 2X 15 A 200 V
30CTH02
Vishay
D7
Yes
SOD80
Diode 800 V, 0.5 A
RGL34K
Vishay
D8, D10
Yes
D0214AA
200 V SMB
ES2D
Fairchild
D9, D11, D12
Yes
SOD80
100 V signal, 100 V
signal diode, 100 V
signal
LL4148
Fairchild
DZ1
Yes
SOD80
12 V 05 W 5% 200 mA
SOD80
12 Vz
Vishay
DZ3
Yes
SOD80
18 V 05 W 5% 200 mA
18 Vz
Vishay
EMI1
No
L_EMI
12.5X17.5variante
COMMON MODE
CHOKE
B82724J2602N041
Epcos
F1
No
FUSE5X20
FUSE 5 A 232 5X20
5A
RS
H1
No
PAD
H1b
No
SK48084 – 84 mm
Heatsink
Heatsink
Fisher
Elektronik
I1
Yes
DSO8
INFINEON 600 V gate
driver IC, 15 VCC 0.5 A,
NO interlock
2EDL05I06BF
Infineon
I2
No
DIP7
INFINEON Flyback
ICE2QR4780Z
IC1
No
DIP4/10
SFH617A-2
SFH617A-2
Vishay
IC3
Yes
SOT23R
VOLTREG. TL431 1%
TL431A
Texas
Instruments
J1
No
CONN7.5mm
CON3
J2
No
CON2
CON2
J3
No
CONN SIN 14 ICE1CS02G CB
ICE1CS02G CB
ICE1CS02G Control
Board
L1
No
L_TOR
OUTPUT CHOKE
0.7 mH Pulse
electronics
Pulse
L3
No
L_TOR
Pulse electronics
47 µH Pulse
electronics
Pulse
LED1
Yes
LED - TOPLed
Green Led, TOPLED
Green
OSRAM
Q1
No
TO220_ST
POWER MOSFET
IPP60R190E6
Infineon
Q2, Q3
No
TO220_ST
POWER MOSFET
IPP60R280E6
Infineon
Q4
Yes
SOT323
BJT NPN 50 V hfe=330 BC847CW
R1, R3, R4
Yes
1206R
RESISTOR 10K 1206
5%
10K
R2, R5
Yes
2010
RESISTOR 10R 5%
2010
10R
R6, R7, R11
Yes
2512R
RESISTOR 0R33 2512
1%
0R33
Panasonic
R8, R9, R10
Yes
2512R
RESISTOR 1R 2512
1%
1R
Panasonic
R12, R13
Yes
1206R
RESISTOR 68K 1206
68K
Application Note
Heatsink
45
Infineon
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Bill of Material (BOM)
Designator
Footprint
Description
5%
Comment
R14, R15, R28, R29, Yes
R46
1206R
RESISTOR 0R0 1206
5%
0R0
R16, R26
Yes
1206R
RESISTOR 1K0 1206
5%
1K0
R17, R21, R37
Yes
1206R
RESISTOR 15K 1206
5%
15k
R18
Yes
1206R
RESISTOR 22K 1206
5%
22K
R19
Yes
1206R
RESISTOR 22R 1206
5%
22R
R20
Yes
1206R
RESISTOR 680R 1206
5%
680R
R22
Yes
1206R
RESISTOR 3K9 1206
5%
3K9
R23
Yes
1206R
RESISTOR 2.2K 1206
5%
2.2K
R24
Yes
1206R
RESISTOR 7K5 1206
5%
7K5
R25
Yes
2512R
RESISTOR 6R8 2512
5%
6R8
R27, R42, R45
Yes
1206R
RESISTOR 3R3 1206
5%
3R3
R33
Yes
1206R
RESISTOR 4K3 1206
5%
4K3
R35
Yes
1206R
RESISTOR 15K 1206
5%
NM
RT1
No
NTC7.5
NTC THERMISTOR 5
passo 7.5 mm 21 mm
diam 8.5 A
B57364S509M
Epcos
TRA1
No
T_ETD34
ETD34 TTF
transformer
Pulse electronics
Pulse
TRA2
No
EE10 - 1pri 2sec
Flyback transformer
Pulse electronics
Pulse
VAR1
No
VARISTOR
S10K300
S10K300
Epcos
Application Note
Mount
46
Manufacturer
Panasonic
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix A: Average and rms values
10
Appendix A: Average and rms values
In typical electric / electronic circuits, voltage v(t) and current i(t) vary as a function of time. If a generic
function x(t) repeats with a time period T in steady state (x (t + T) = x (t)), then average function value Xavg
and root-mean square (rms) value Xrms can be calculated as:
 =


1
∙ ∫  () 

0
Eq
31
Eq
32

1
= √ ∙ ∫  2 () 

0
Typically, Xavg and Xrms are different; only if x (t) is constant, then Xav = Xrms. Standard waveforms (sinusoidal,
rectangular and saw-tooth and triangular waves) will be further analyzed. Time origin is arbitrary, as only
periodic functions will be analyzed; eventually, offset amplitude values can be easily added by slightly
modifying formulas, as shown below.
10.1
Sinusoidal waveform
T=Period
Peak value=XP
RMS value=XRMS
Figure 47
Nomenclature in sinusoidal function
As shown in this Figure 47, it is easy to evaluate average and rms values: average value Xavg = 0 because there
is no offset and sinusoidal waveform is positive for half period (T / 2) and negative in a simmetric way.
However, rms value of a sinusoidal waveform x (t) = sin (t) is not 0, as shown below:
 =

=√


1
1
∙ ∫  ()  =
∙ ∫  ∙  ()  = 0


0
0
Eq 33



1
1
1 1
2
2
√
∙ ∫  ()  =
∙ ∫ ( ∙ sin())  = √
∙ ∙ 2 ∙  =


 2
√2
0
0
Application Note
47
Eq 34
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix A: Average and rms values
10.2
Rectangular pulse train
T=Period
t=Pulse
Figure 48
Peak
value=XP
Nomenclature in pulse train function
In this case, duty cycle D = t / T makes the difference between average and rms value in a rectangular pulse
train: D = [0, 1] and it is easy to understand that if D = 1, then x (t) is constant and x (t) = Xavg = Xrms = A. If D < 1,
then integrals are evaluated between 0 and t = D ∙ T.




1
1
1
= ∙ ∫  ()  = ∙ ∫   = ∙  ∙  =  ∙ 

 0

0
Eq 35

=∙
1
1
1
= √ ∙ ∫  2 ()  = √ ∙ ∫
 2  = √  2 ∙  ∙  =  ∙ √



0
0
10.3
Eq 36
Sawtooth waveform
Saw-tooth waves can be expressed analytically as:
 () =  ∙


0< <  = ∙
 () = 0 ℎ
Eq 37
T=Period
t=Pulse
Figure 49
Peak
value=XP
Nomenclature in saw-tooth train function
Hence, Xavg and Xrms can be calculated as follows:
 =


1
1

1

( ∙ )2

∙ ∫  ()  = ∙ ∫  ∙  = ∙
∙
=  ∙

 0

  ∙ 
2
2
0



1
1
 2

=√
∙ ∫  2 ()  = √ ∙ ∫  2 ∙ ( )  =  ∙ √



3
0
0
Eq 38
Eq 39
If an offset B exists, then:
Application Note
48
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix A: Average and rms values
 () =  ∙

+

0< <  = ∙
 () = 0 ℎ
Eq 40
T=Period
t=Pulse
Peak
value=XP
Offset=B
Figure 50
Nomenclature saw-tooth train function with offset
Hence, Xavg and Xrms can be calculated as follows:
 =



1
1


∙ ∫  ()  = ∙ ∫ (( − ) ∙ + )  = ∙ ( + )

 0

2
0
Eq 41
Eq 42


2
1
1

= √ ∙ ∫  2 ()  = √ ∙ ∫ (( − ) ∙ ( ) + ) 



0
0

= √ ∙ √ 2 +  ∙  +  2
3
10.4
Triangular waveforms
Triangular waves can be expressed as:
 () =  ∙


 () = −  ∙
0< <  = ∙
−
−
Eq 43
<< 
T=Period
t
Figure 51
Peak
value=XP
Nomenclature in triangular train function
Xavg and Xrms can then be calculated as follows:




1
1

−

()
= ∙ ∫ 
 = ∙ (∫  ∙  + ∫ − ∙
) =



−
2
0
0

Application Note
49
Eq 44
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix A: Average and rms values




1
1
 2
− 2
2
2
= √ ∙ ∫  ()  = √
∙ [∫  ∙ ( )  + ∫ (− ∙
) ]



−
0
0


=
√3
Eq 45
If an offset B exists, then:
 () = ( − ) ∙

+ 

 () = −( − ) ∙
0< <  = ∙
−
+ 
−
Eq 46
<< 
T=Period
t
Peak
value=XP
Offset=B
Figure 52
Nomenclature in triangular train function with offset
Xavg and Xrms can then be calculated as follows:
 =

1
∙ ∫  () 

0


1

−
(
= ∙ (∫  − ) ∙ +   + ∫ −( − ) ∙
+  )


−
0

 − 
 + 
=
+ =
2
2
Eq 47
Eq 48

2
=√
2


1

−
∙ [∫ (( − ) ∙ ( ) + )  + ∫ (−( − ) ∙
+ ) ]


−
0

1
= √ ∙ √ 2 +  ∙  +  2
3
Application Note
50
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix B: Chart representation of power MOSFET losses in hard switching
11
Appendix B: Chart representation of power MOSFET
losses in hard switching
MOSFETs selection in DC/DC applications strongly impacts on converter efficiency: transistors choice is
mainly determined by power losses, which can be divided in conduction/static and switching/dynamic
losses. MOSFETs conduction losses (Pstat) have been estimated by multiplying their on-resistance RDS (on) and
their squared rms drain current value ID(rms) (see Eq 49). Switching losses in MOSFETs (Pdyn) have been
calculated as described in Eq 51, where VDS is the drain-to-source voltage, Eoss (VDS) represents the energy
stored in the output capacitance (Coss) and it is function of VDS, and fsw is the switching frequency. Total
power losses (Ptot) are then the sum of conduction losses Pstat and switching losses Pdyn: optimal RDS (on) for
MOSFETselection has then been evaluated by total power loss trade-off between Pstat and Pdyn.
2
 = () ∙ ()
Eq 49
 =  ( ) ∙ 
Eq 50
2
 = () ∙ ()
+  ( ) ∙ 
Eq 51
For a same technology it can be assumed that the product FOM=RDS(on)·Eoss is nearly constant while fixing the
desired VDS voltage and measurement characteristics.
This exercise is done for CoolMOS™ C6 and CoolMOS™ CE in the following tables by taking the datasheet
values.
Then Eq 51 can be rewritten as follows:
 = () ∙  ( )
2
2
 = () ∙ ()
+  ( ) ∙  = () ∙ ()
+
Figure 53
Eq 52

∙
() 
Eq 53
Eoss chart for 190 mΩ CooMOS™ P6 600V IPP60R190P6
Table 13 and Table 14 shows the calculation of indicative FOM from datasheet values.
Application Note
51
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
Appendix B: Chart representation of power MOSFET losses in hard switching
Table 13
RDS(on), Eoss (@400 V) and evaluated FOM for 600 V CoolMOS™ P6 transistors
600 V CoolMOSTM P6
power transistor
RDS (on)
[mΩ]
Eoss (@ 400 V)
[µJ]
FOM (400 V) = RDS (on) ∙ Eoss
[Ω ∙ µJ]
IPP60R125P6
113
7.2
0.81
IPP60R160P6
144
5.7
0.82
IPP60R190P6
171
4.9
0.84
IPP60R230P6
207
4.2
0.87
IPP60R280P6
252
3.5
0.88
IPP60R330P6
297
3
0.89
Average
0.86
Table 14
RDS (on), Eoss (@400 V) and evaluated FOM for 500 V CoolMOS™ CE transistors
500 V CoolMOSTM CE
power transistor
RDS (on)
[Ω]
Eoss
(@ 400 V/200 V)
[µJ]
FOM (@400 V/200 V)
= RDS (on) ∙ Eoss
[µΩ ∙ J]
IPP50R190CE
0.17
4.4 / 2.3
0.75 / 0.39
IPP50R280CE
0.25
3.2 / 1.6
0.80 / 0.4
IPP50R380CE
0.35
2.5 / 1.2
0.87 / 0.42
IPP50R500CE
0.45
2.0 / 0.95
0.9 / 4.3
Plotting equation Eq 53 with the corresponding data, produces graphs in Figure 13 on page 15 and Figure 17
on page 20.
Note: It must be clear that the Eoss measurement is done in certain conditions, which are of course different
from the conditions found in a different design. The purpose of the above mentioned calculation is then
to get a draft indication on the expected power losses in order to have a good starting point for the
choice of the power mosfet.
Gate resistance and driving, as well as the stray capacitances connected to the mosfet drain will
contribute to increase the calculated losses.
Additionally the counterpart freewheeling/rectifier diode, when a path is activated, have a big impact
on losses, which are here neglected.
Application Note
52
Revision 1.1, 2015-10-20
300 W general purpose wide-range SMPS
PFC + TTF Evaluation Board
References and proposed links
12
References and proposed links
[1] CoolMOS™ high voltage MOSFETs product main page
www.infineon.com/coolmos
[2] thinQ™! Silicon Carbide Schottky Diodes main page
www.infineon.com/thinq
[3] Rapid Silicon Diodes main page
http://www.infineon.com/cms/en/product/transistor-and-diode/diode/silicon-powerdiode/channel.html?channel=ff80808112ab681d0112ab6a527f04a6
[4] CoolSET™ auxiliary and SMPS controller ICs main page
www.infineon.com/coolset
[5] ICE1CS02G application note: “300W Evaluation board using ICE1CS02”:
http://www.infineon.com/dgdl?folderId=5546d4694909da4801490a2652e6286a&fileId=db3a3043
1c69a49d011c8e8e3df1048f
[6] ICE1CS02G datasheet from Infineon product page
http://www.infineon.com/dgdl?folderId=5546d4694909da4801490a07012f053b&fileId=db3a3043
1c69a49d011c8e917a2d0494
[7] Addendum to 300 W general purpose wide range SMPS
http://www.infineon.com/dgdl/InfineonApplicationNote_Addendum_to_300W_general_purpose_wide-range_SMPS-AN-v01_00EN.pdf?fileId=5546d46250cc1fdf01513392e96c3537
Revision History
Major changes since the last revision
Page or Reference
-Revision 1.1
Application Note
Description of change
First Release
Alternative solution for daughterboard added; See Note on page 1
53
Revision 1.1, 2015-10-20
Trademarks of Infineon Technologies AG
AURIX™, C166™, CanPAK™, CIPOS™, CIPURSE™, CoolGaN™, CoolMOS™, CoolSET™, CoolSiC™, CORECONTROL™, CROSSAVE™, DAVE™, DI-POL™, DrBLADE™,
EasyPIM™, EconoBRIDGE™, EconoDUAL™, EconoPACK™, EconoPIM™, EiceDRIVER™, eupec™, FCOS™, HITFET™, HybridPACK™, ISOFACE™, IsoPACK™, iWafer™, MIPAQ™, ModSTACK™, my-d™, NovalithIC™, OmniTune™, OPTIGA™, OptiMOS™, ORIGA™, POWERCODE™, PRIMARION™, PrimePACK™,
PrimeSTACK™, PROFET™, PRO-SIL™, RASIC™, REAL3™, ReverSave™, SatRIC™, SIEGET™, SIPMOS™, SmartLEWIS™, SOLID FLASH™, SPOC™, TEMPFET™,
thinQ!™, TRENCHSTOP™, TriCore™.
Other Trademarks
Advance Design System™ (ADS) of Agilent Technologies, AMBA™, ARM™, MULTI-ICE™, KEIL™, PRIMECELL™, REALVIEW™, THUMB™, µVision™ of ARM
Limited, UK. ANSI™ of American National Standards Institute. AUTOSAR™ of AUTOSAR development partnership. Bluetooth™ of Bluetooth SIG Inc. CATiq™ of DECT Forum. COLOSSUS™, FirstGPS™ of Trimble Navigation Ltd. EMV™ of EMVCo, LLC (Visa Holdings Inc.). EPCOS™ of Epcos AG. FLEXGO™ of
Microsoft Corporation. HYPERTERMINAL™ of Hilgraeve Incorporated. MCS™ of Intel Corp. IEC™ of Commission Electrotechnique Internationale. IrDA™ of
Infrared Data Association Corporation. ISO™ of INTERNATIONAL ORGANIZATION FOR STANDARDIZATION. MATLAB™ of MathWorks, Inc. MAXIM™ of
Maxim Integrated Products, Inc. MICROTEC™, NUCLEUS™ of Mentor Graphics Corporation. MIPI™ of MIPI Alliance, Inc. MIPS™ of MIPS Technologies, Inc.,
USA. muRata™ of MURATA MANUFACTURING CO., MICROWAVE OFFICE™ (MWO) of Applied Wave Research Inc., OmniVision™ of OmniVision Technologies,
Inc. Openwave™ of Openwave Systems Inc. RED HAT™ of Red Hat, Inc. RFMD™ of RF Micro Devices, Inc. SIRIUS™ of Sirius Satellite Radio Inc. SOLARIS™ of
Sun Microsystems, Inc. SPANSION™ of Spansion LLC Ltd. Symbian™ of Symbian Software Limited. TAIYO YUDEN™ of Taiyo Yuden Co. TEAKLITE™ of CEVA,
Inc. TEKTRONIX™ of Tektronix Inc. TOKO™ of TOKO KABUSHIKI KAISHA TA. UNIX™ of X/Open Company Limited. VERILOG™, PALLADIUM™ of Cadence
Design Systems, Inc. VLYNQ™ of Texas Instruments Incorporated. VXWORKS™, WIND RIVER™ of WIND RIVER SYSTEMS, INC. ZETEX™ of Diodes Zetex
Limited.
Last Trademarks Update 2014-07-17
www.infineon.com
Edition 2015-10-20
Published by
Infineon Technologies AG
81726 Munich, Germany
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Document reference
ANEVAL_201410_PL52_002
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