1 W to 5 W LED Driver for MR16 LED

TND373/D
Rev. 0, Oct -- 2009
1 W to 5 W LED Driver for MR16 LED
Reference Design Documentation Package
Disclaimer: ON Semiconductor is providing this reference design documentation package “AS IS” and the recipient
assumes all risk associated with the use and/or commercialization of this design package. No licenses to ON Semiconductor’s
or any third party’s Intellectual Property is conveyed by the transfer of this documentation. This reference design
documentation package is provided only to assist the customers in evaluation and feasibility assessment of the reference design.
It is expected that users may make further refinements to meet specific performance goals.
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1 W to 5 W LED Driver for
MR16 LED
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TECHNICAL NOTE
Introduction
boost topologies. Once understood, the buck--boost
topology can offer many advantages for cost effective HB
LED lighting when Vin overlaps Vf.
The market for High--Brightness LED’s (HB--LED)
continues to rise rapidly. Over the last few years LED
efficacy (lm/W) has more than doubled while flux per
package continues to rise rapidly making them more useful
for many new applications. We have seen revolutionary new
products in handheld flashlights, architectural lighting and
street lighting. The challenge continues to be making a more
cost effective system versus incandescent and compact
fluorescent bulbs.
In many applications this challenge involves powering the
HB--LED’s from a wide input--voltage range source. This is
especially true in general illumination applications like
track lighting where the power source is a 12 Vac or +12 Vdc
source that can be very loosely regulated. The LED’s need
to be driven by a current source rather than a voltage source
since the forward voltage (3.4 V nominal) can vary more
than ±20% over process tolerance and temperature.
Moreover given the flux of current 1 W warm white power
LEDs, it is common to need 3--4 LEDs to replace the light
output of a 20 W incandescent. To obtain predictable and
matched luminosity and chromaticity it is also desirable to
drive the LED’s with a constant current. The buck--boost
topology meets this requirement from an architectural
perspective but it not as common as the standard buck or
Overview
This reference document describes a built and tested,
GreenPoint® solution for a 1 to 5 W LED driver for MR16
LED replacement. The circuit is proposed for driving
HB--LED (high--brightness LED) in a variety of lighting
applications but is configured in size and features for an
MR16 LED replacement. Configurations like this are found
in 12 Vac / 12 Vdc track lighting applications, automotive
applications, and low voltage AC landscaping applications
as well as task lighting such as under--cabinet lights and desk
lamps that might be powered from standard off--the--shelf
Vac wall adapters.
The circuit is based around the ON Semiconductor
NCP3065 operating at ~150 kHz in a non--isolated
configuration. A key consideration in this design was
achieving flat current regulation across input line variation
and output voltage variation with a 12 Vac input. It also
features an auto--detect circuit in combination with the
NCP3065 which allows input from a 12 Vdc or 12 Vac
supply and still maintain targeted output current regulation.
12 VACIN
NCP3065
Compensation
Network
Figure 1. Simplified Block Diagram
Actual Size
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Basic Power Topology
energy in L. In this stage, the capacitor C supplies energy to
the output load;
While in the Off--state, the inductor is connected to the
output load and capacitor through the Output Diode, so
energy is transferred to the load.
The principle of the Buck--Boost converter is fairly simple
(see Figure 2):
While in the On--state, the input voltage source is directly
connected to the inductor (L). This results in accumulating
Vin
ID
IQ
Vsw
Vgate
Vout
C
IL
RLoad
Ton
Vgate
From Inductor Volt Second V = L di
Balance and:
dt
Toff
Vin
Vi(Ton)
Vsw
Vo--Vf
L
=
Vo(Toff)
L
VinD = Vo(1 − D)
IQ
ID
Vo
Vin
IL
Figure 2. Buck--Boost Operation
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=
D
(1 − D)
TND373/D
Figure 3. Vswitch Node Waveform
TSD
NC
ILimit
Comp
Set
Switch
dominant
Collector
R Q
S
Ipk
Sense
S
Switch
Q
Emitter
Set
R
dominant
0.2V
Oscillator
Ct
Vcc
Vref
Ct
GND
Comp
Inv
Figure 4. NCP3065 Burst Mode Controller
Burst Mode Control
R8 is used to sense the inductor current and is fed to the
FB pin of the NCP3065.
This application produces OFF time instantaneous
(Ivalley) inductor current control (see Figure 5). A cycle of
switch ON time is only allowed to start once the OFF time
Inductor current crosses the Vref threshold.
The basic control loop consists of a 235 mV internal
Reference, a Feedback Comparator, and two Set--Dominant
RS Latches. Basically the NCP3065 allows the Power FET
for the Buck--Boost stage to switch ON as the Feedback
Voltage falls below the reference voltage. The Power FET
will be then be forced OFF unconditionally during Ct Ramp
down.
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Ipeak
Ivalley
Ton
Ton
Toff
Toff
Average Load Current = Area During Toff
Figure 5. Buck--Boost Inductor Current
0.4
Since the controller does not provide integral PWM
control and utilizes only a comparator trip point for
feedback, the peak to average load current is not in direct
proportion as in a Buck Converter, but rather follows the
following formula:
⎧V 1 −
⎪L
⎩
1
valley +
2
o
Vo
V o+V
F
⎫⎫
⎪⎪1 − V
⎭⎭
in
Iave, (Vin)
I ave
⎧
=⎪I
⎩
0.3

Vo
o + V in
0.2
0.1
Where, Ivalley is the lowest inductor current point. Plotting
Iave vs Vin shows a dramatic curve which would cause a
significant change in light output of the LED (see Figure 6).
0
10
12
14
16
18
Figure 7. Average LED Current vs Vin DC
(With Vin Compensation)
1.2
Iave, (Vin)
8
Vin, N
1.4
A resistive divider network consisting of R3, R5 and
summing resistor R4 are used to add Vin proportional
voltage to the FB pin in order to reduce the load current as
Vin is increased. This has the effect of flattening the curve of
Figure 6 and reduces the overall current error (see Figure 7).
This average line can be DC shifted with R8 and the ends can
be aligned by adjusting R5, R3 and R4.
R9 and C6 are used to limit the gate to source voltage on
the external switch at high input voltage. The resistor divider
network of R9 and R2 are used to program and gate to source
maximum.
1
0.8
0.6
0.4
6
0
5
10
Vin, N
15
20
Figure 6. Average LED Current vs Vin DC
(Without Vin Compensation)
V gs = V in −
Therefore an input voltage feed--forward compensation
network is used to reduce the error due to the nonlinear
response of the Iout vs Vin curve.
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
V in ⋅ R9

R9 + R2
(eq. 1)
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Pulsed Feedback Resistor
R7 and D5 add current to the Ct timing capacitor C2. This
effectively limits the maximum achievable duty cycle of the
NCP3065. When conditions warrant low duty cycle, R7 and
D5 make higher than desired duty cycles unavailable. D7 is
necessary to block voltage during the OFF time, since this
is Buck--Boost Topology. More information on Pulsed
Feedback compensation is available in the NCP3065 data
sheet.
R7 and D5 are used to reduce the possibility of pulse
skipping (see Figure 8). Since burst mode control involves
only one feedback voltage, cross--detection per cycle and
does not involve the use of a window comparator, it is
possible to have skipped pulses which do not effect the DC
regulation but could be visible as flicker in an LED
application if the pulsing had a low frequency component.
Figure 8. Pulsed Feedback Resistor
AC Operation vs DC
for some finite portion ~80% of the 120 Hz line cycle, and
then no output for ~20%. This has the effect of reducing the
average current by ~20% when operating with AC input.
Thermal consideration should be taken when running with
> 12 Vac. In most applications the module is potted to
increase thermal dissipation.
An additional AC compensation network is added to the
Vin Compensation to account for the different operating
point (see Figure 9).
Since there is a half sine wave input to the Buck--Boost
stage, there is a different operating point as compared with
pure DC input. Since small size is a goal for this design very
little input capacitance is used past the full bridge rectifier.
As a result the line voltage can drop to as little as 3 V
depending on the input capacitance selection. Therefore, the
input to the converter is a full wave rectified sine wave.
Since the regulator is non--functional below ~4 V there are
dead spots in the regulation. So we end up with regulation
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Figure 9. Pulsed Feedback Resistor
Protection
the power FET with reasonable voltage margin. This may
require some trial and error to select since the clamp voltage
will stretch depending on how much energy needs to be
absorbed.
Z1 and R1, along with the Current limit function of the
NCP3065, are used for open circuit protection. In the event
of an open circuit at the load, the loop will try to increase the
output voltage in order to satisfy the current demand which
feeds back zero current. When (Vin + Vout) exceeds the
voltage of Z1, current will flow in R1 which triggers the
current limit function of the NCP3065.
Short circuit protection is handled with a fuse, F1, on the
input. Surge protection from inductive loads is an important
consideration specifically in transformer fed systems that
carry significant source inductance such as found with
magnetic transformers used in landscape lighting
applications. The surge device needs to be selected to a
voltage that will never exceed the gate to source voltage of
Increasing Output Current
The reference design is configured for 350 mA average
LED current. Increasing the current regulation point on the
reference board is as simple as cutting the current sense
resistor R8 in half from 250 mΩ to 125 mΩ. Also, the input
fuse must be increased to accommodate the increased input
current draw. Heat sinking may be required depending on
the implementation of the housing and the environmental
characteristics when moving to the higher power design.
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TND373/D
PC BOARD
Figure 10. Component Placement (Top)
Figure 11. Traces (Top View)
Figure 12. Component Placement (Bottom)
Figure 13. Traces (Bottom View)
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Figure 14. Reference Design
0.457” x 1.148” (11 mm x 29 mm)
Remember this is an inverting output. So the negative
output will connect to the anode of the LED, and the positive
output will connect to the cathode of the LED.
Also note, when trying to make measurements with a
scope probe, that ground is NOT ground. The scope will
need to be floating (ground connection removed from the
AC wall source) or there will be a ground loop / short circuit
that will cause the device to turn off.
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Figure 15.
SCHEMATIC
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Table 1. BILL OF MATERIALS
Qty
Ref
Value
Part Number
Description
Manufacturer
1
F1
4A
0457004.NR
Fuse
LittleFuse
1
C3
10 mF
GRM31MF51A106ZA01L
Ceramic Chip Capacitor
Murata
1
C6
1 nF
GRM188R71H102KA01D
Ceramic Chip Capacitor
Murata
1
C1
1 mF
GRM188R61E105KA12D
Ceramic Chip Capacitor
Murata
1
C2
5.6 nF
GRM188R71H562KA01D
Ceramic Chip Capacitor
Murata
1
C4
10 mF
GRM32NF51E106ZA01L
Ceramic Chip Capacitor
Murata
1
C5
10 mF
GRM32NF51E106ZA01L
Ceramic Chip Capacitor
Murata
1
D1
1 A, 30 V
MBR130T1G
DIODE, SCHOTTKY
ON Semiconductor
1
D2
1 A, 30 V
MBR130T1G
DIODE, SCHOTTKY
ON Semiconductor
1
D3
1 A, 30 V
MBR130T1G
DIODE, SCHOTTKY
ON Semiconductor
1
D4
1 A, 30 V
MBR130T1G
DIODE, SCHOTTKY
ON Semiconductor
1
D6
2 A, 60 V
MBRS260T3
DIODE, SCHOTTKY
ON Semiconductor
1
Q1
PNP
MBT3946DW1T1
General Purpose NPN Transistor
ON Semiconductor
1
D5
0.2 A,
100 V
MMSD4148T1
Diode, Small Signal
ON Semiconductor
1
D8
0.2 A,
100 V
MMSD4148T1
Diode, Small Signal
ON Semiconductor
1
Z1
36 V
MM5Z36VT1
DIODE, ZENER
ON Semiconductor
1
L1
68 mH
MSS1278-683MLD
INDUCTOR, SM
Coilcraft
1
U1
40 V 1.5 A
NCP3065DR2G
Switching Regulator
ON Semiconductor
1
M1
P--FET
NTGS4111PT1G
MOSFET, P
ON Semiconductor
1
R4
1.2k
CRCW04021K20FKED
Resistor
Vishay / Dale
1
R1
100
CRCW0402100RFKED
Resistor
Vishay / Dale
1
R3
162k
CRCW0402162KFKED
Resistor
Vishay / Dale
1
R6
196
CRCW0402196RFKED
Resistor
Vishay / Dale
1
R7
22k
CRCW040222K0FKED
Resistor
Vishay / Dale
1
R5
22k
CRCW040222K0FKED
Resistor
Vishay / Dale
1
R2
1k
CRCW04021K00FKED
Resistor
Vishay / Dale
1
R9
200
CRCW0402200RFKED
Resistor
Vishay / Dale
1
R8
0.25
CSR1/20.25FICT-ND
Resistor
Vishay / Dale
1
D7
P6SMB22CAT3
ZENER, BACK TO BACK
ON Semiconductor
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MEASUREMENTS
(12 Vac Data)
0.4
0.8
0.75
0.38
0.7
0.36
0.65
0.34
0.6
0.32
0.55
Iout
0.3
8
9
10
11
12
13
14
Efficiency
15
0.5
7
Figure 16. Iout vs Vac
9
11
13
15
17
19
Figure 17. Efficiency vs Vdc
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