DATASHEET

SMBus Level 2 Battery Charger
ISL88731
Features
The ISL88731 is a highly integrated Lithium-ion battery charger
controller, programmable over the SMBus system management
bus (SMBus). The ISL88731 is intended to be used in a smart
battery charger (SBC) within a smart battery system (SBS) that
throttles the charge power such that the current from the
AC-adapter is automatically limited. High efficiency is achieved
with a DC/DC synchronous-rectifier buck converter, equipped
with diode emulation for enhanced light load efficiency and
system bus boosting prevention. The ISL88731 charges one to
four Lithium-ion series cells, and delivers up to 8A charge
current. Integrated MOSFET drivers and bootstrap diode result
in fewer components and smaller implementation area. Low
offset current-sense amplifiers provide high accuracy with
10mΩ sense resistors. The ISL88731 provides 0.5%
end-of-charge battery voltage accuracy.
• 0.5% Battery Voltage Accuracy
The ISL88731 provides a digital output that indicates the
presence of the AC adapter as well as an analog output which
indicates the adapter current within 4% accuracy.
The ISL88731 is available in a small 5mmx5mm 28 Ld Thin
(0.8mm) QFN package. An evaluation kit is available to reduce
design time. The ISL88731 is available in Pb-Free packages.
Pin Configuration
• SMBus 2-Wire Serial Interface
• Battery Short Circuit Protection
• Fast Response for Pulse-Charging
• Fast System-Load Transient Response
• Monitor Outputs
- Adapter Current (3% Accuracy)
- AC-Adapter Detection
• 11-Bit Battery Voltage Setting
• 6 Bit Charge Current/Adapter Current Setting
• 8A Maximum Battery Charger Current
• 11A Maximum Adapter Current
• +8V to +28V Adapter Voltage Range
• Pb-Free (RoHS Compliant)
Applications
• Tablet PCs
CSSP
CSSN
VCC
BOOT
UGATE
PHASE
DCIN
• Portable Equipment with Rechargeable Batteries
28
27
26
25
24
23
22
Ordering Information
LGATE
VREF
3
19
PGND
ICOMP
4
18
CSOP
NC
5
17
CSON
VCOMP
6
16
NC
NC
7
15
VFB
8
9
10
11
12
13
14
NC
20
ACOK
2
GND
ACIN
VDDSMB
VDDP
SCL
21
SDA
1
ICM
NC
1
• 3% Charge Current Accuracy
• Notebook Computers
ISL88731
(28 LD TQFN)
TOP VIEW
June 8, 2011
FN9258.3
• 3% Adapter Current Limit Accuracy
PART
NUMBER
(Notes 1, 2, 3)
PART
MARKING
TEMP
RANGE (°C)
PACKAGE
(Pb-Free)
ISL88731HRZ ISL887 31HRZ -10 to +100 28 Ld 5x5 TQFN
PKG.
DWG. #
L28.5x5B
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel
specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free
material sets, molding compounds/die attach materials, and 100%
matte tin plate plus anneal (e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations). Intersil Pb-free products are MSL classified at Pb-free peak
reflow temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page
for ISL88731. For more information on MSL please see techbrief TB363.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 |Copyright Intersil Americas Inc. 2006, 2009, 2011. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL88731
VCC
DCIN
VDDSMB
SMBUS
SDA
11
DACV
DACV
6
DACS
DACS
DACI
DACI
6
SCL
LINEAR
REGULATOR
REFERENCE
VDDP
REF
VREF
ACOK
+
-
EN
ACIN
ICM
BUFF
CSSP
LEVEL
SHIFTER
20x
CSSN
DACS
EN
GMS
+
BOOT
ICOMP
CSO
CSOP
LEVEL
SHIFTER
20x
CSON
DACI
UGATE
GMI
PHASE
DC-DC
CONVERTER
+
LVB
+
DACV
VDDP
LVB
GMV
LGATE
VFB
500k
100k
PGND
EN
CSSP
GND
VCOMP
FIGURE 1. FUNCTIONAL BLOCK DIAGRAM
AC ADAPTER
TO SYSTEM
RS1
CSSP
CSSN
UGATE
PHASE
ACIN
DCIN
RS2
TO BATTERY
ISL88731
BOOT
LGATE
CSOP
CSON
VFB
AGND
ICOMP
VCOMP
VDDP
PGND
VCC
GND
ACOK
ICM
SDA
SCL
VDDSMB
PGND
HOST
VREF
AGND
FIGURE 2. TYPICAL APPLICATION CIRCUIT
2
FN9258.3
June 8, 2011
ISL88731
Absolute Maximum Ratings
Thermal Information
DCIN, CSSP, CSSN, CSOP, CSON, VFB . . . . . . . . . . . . . . . . . . . -0.3V to +28V
CSSP-CSSN, CSOP-CSON, PGND-GND . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
PHASE to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -6V to +30V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
UGATE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . PHASE - 0.3V to BOOT + 0.3V
LGATE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . PGND - 0.3V to VDDP + 0.3V
ICOMP, VCOMP, VREF, to GND . . . . . . . . . . . . . . . . . . . . . -0.3V to VCC + 0.3V
VDDSMB, SCL, SDA, ACIN, ACOK . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
VDDP, ICM, VCC to GND, VDDP to PGND . . . . . . . . . . . . . . . . . . -0.3V to +6V
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
28 Ld TQFN Package (Notes 4, 5) . . . . . . .
36
6
Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . . -55°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . . . . . . . . -10°C to +100°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V,
CVDDP = 1µF, IVDDP = 0mA, TA = -10°C to +100°C. Boldface limits apply over the operating temperature range, -10°C to +100°C.
PARAMETER
CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6) UNITS
16.716
16.8
16.884
V
0.5
%
CHARGE VOLTAGE REGULATION
Battery Full Charge Voltage and Accuracy
ChargeVoltage = 0x41A0
-0.5
ChargeVoltage = 0x3130
12.529 12.592 12.655
-0.5
ChargeVoltage = 0x20D0
8.350
ChargeVoltage = 0x1060
4.163
0.5
%
8.4
8.450
V
0.6
%
4.192
4.221
V
0.7
%
-0.6
-0.7
Battery Undervoltage Lockout Trip Point for Trickle
Charge
VFB rising
V
2.55
2.7
2.85
V
100
250
400
mV
78.22
80.64
83.06
mV
RS2 = 10mΩ (see Figure 2)
ChargingCurrent = 0x1f80
7.822
8.064
8.306
A
3
%
RS2 = 10mΩ (see Figure 2)
ChargingCurrent = 0x0f80
3.809
4.126
A
4
%
RS2 = 10mΩ (see Figure 2)
ChargingCurrent = 0x0080
64
220
mA
Based on charge current = 128mA and 8.064A
-1.6
1.4
%
0
19
V
135
200
µA
0.2
2
µA
Battery Undervoltage Lockout Trip Point Hysteresis
CHARGE CURRENT REGULATION
CSOP to CSON Full-Scale Current-Sense Voltage
Charge Current and Accuracy
Charge Current Gain Error
CSOP/CSON Input Voltage Range
Battery Quiescent Current
-3
-4
Adapter present, not charging,
ICSOP + ICSON + IPHASE + ICSSP + ICSSN + IFB
VPHASE = VCSON = VCSOP = VDCIN = 19V, VACIN = 5V
Adapter Absent
ICSOP + ICSON + IPHASE + ICSSP + ICSSN + IFB
VPHASE = VCSON = VCSOP = 19V, VDCIN = 0V
3
3.968
-1
128
FN9258.3
June 8, 2011
ISL88731
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V,
CVDDP = 1µF, IVDDP = 0mA, TA = -10°C to +100°C. Boldface limits apply over the operating temperature range, -10°C to +100°C. (Continued)
PARAMETER
CONDITIONS
Adapter Quiescent Current
MIN
(Note 6)
IDCIN + ICSSP + ICSSN
Vadapter = 8V to 26V, Vbattery 4V to 16.8V
TYP
MAX
(Note 6) UNITS
3
5
mA
110
113.3
mV
INPUT CURRENT REGULATION
CSSP to CSSN Full-Scale Current-Sense Voltage
CSSP = 19V
Input Current Accuracy
RS1 = 10mΩ (see Figure 2)
Adapter Current = 11004mA or 3584mA
-3
3
%
RS1 = 10mΩ (see Figure 2)
Adapter Current = 2048mA
-5
5
%
-1.5
1.5
%
Input Current Limit Offset
-1
1
mV
CSSP/CSSN Input Voltage Range
8
26
V
Input Current Limit Gain Error
Based on InputCurrent = 1024mA and 11004mA
ICM Gain
VCSSP-CSSN = 110mV
ICM Accuracy
VCSSP-CSSN = 110mV
ICM Max Output Current
106.7
20
V/V
-2.5
2.5
%
VCSSP-CSSN = 55mV or 35mV
-4
4
%
VCSSP-CSSN = 20mV
-8
8
%
500
µA
26
V
5.1
5.23
V
35
100
mV
5.5
V
VCSSP-CSSN = 0.1V
SUPPLY AND LINEAR REGULATOR
DCIN, Input Voltage Range
8
VDDP Output Voltage
8.0V < VDCIN < 28V, no load
VDDP Load Regulation
0 < IVDDP < 30mA
5.0
VDDSMB Range
2.7
VDDSMB UVLO Rising
2.4
2.5
2.6
V
VDDSMB UVLO Hysteresis
40
100
150
mV
20
27
µA
3.168
3.2
3.232
V
2
8
VDDSMB Quiescent Current
VDDP = SCL = SDA = 5.5V
V REFERENCE
VREF Output Voltage
0 < IVREF < 300µA
ACOK
ACOK Sink Current
VACOK = 0.4V, ACIN = 1.5V
ACOK Leakage Current
VACOK = 5.5V, ACIN = 3.7V
mA
1
µA
ACIN
ACIN rising Threshold
3.15
3.2
3.25
V
ACIN Threshold Hysteresis
40
60
90
mV
ACIN Input Bias Current
-1
1
µA
SWITCHING REGULATOR
Frequency
BOOT Supply Current
UGATE High
330
400
440
kHz
170
290
400
µA
PHASE Input Bias Current
VDCON = 28V, VCSON = VPHASE = 20V
0
2
µA
UGATE On-Resistance Low
IUGATE = -100mA
0.9
1.6
Ω
UGATE On-Resistance High
IUGATE = 10mA
1.4
2.5
Ω
LGATE On-Resistance High
ILGATE = +10mA
1.4
2.5
Ω
LGATE On-Resistance Low
ILGATE = -100mA
0.9
1.6
Ω
4
FN9258.3
June 8, 2011
ISL88731
Electrical Specifications
DCIN = CSSP = CSSN = 18V, CSOP = CSON = 12V, VDDP = 5V, BOOT-PHASE = 5.0V, GND = PGND = 0V,
CVDDP = 1µF, IVDDP = 0mA, TA = -10°C to +100°C. Boldface limits apply over the operating temperature range, -10°C to +100°C. (Continued)
PARAMETER
CONDITIONS
Dead Time
MIN
(Note 6)
TYP
35
50
80
ns
200
250
300
µA/V
Falling UGATE to rising LGATE or
falling LGATE to rising UGATE
MAX
(Note 6) UNITS
ERROR AMPLIFIERS
GMV Amplifier Transconductance
GMI Amplifier Transconductance
40
50
60
µA/V
GMS Amplifier Transconductance
40
50
60
µA/V
GMI/GMS Saturation Current
15
21
25
µA
GMV Saturation Current
10
17
30
µA
200
300
400
mV
0.8
V
0.25V < VICOMP, VCOMP < 3.5V
ICOMP, VCOMP Clamp Voltage
LOGIC LEVELS
SDA/SCL Input Low Voltage
VDDSMB = 2.7V to 5.5V
SDA/SCL Input High Voltage
VDDSMB = 2.7V to 5.5V
2
SDA/SCL Input Bias Current
VDDSMB = 2.7V to 5.5V
-1
SDA, Output Sink Current
VSDA = 0.4V
7
V
1
15
µA
mA
NOTE:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
SMBus Timing Specification
VDDSMB = 2.7V to 5.5V
PARAMETERS
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
100
kHz
SMBus Frequency
FSMB
10
Bus Free Time
TBUF
4.7
µs
Start Condition Hold Time from SCL
THD:STA
4
µs
Start Condition Setup Time from SCL
TSU:STA
4.7
µs
Stop Condition Setup Time from SCL
TSU:STO
4
µs
SDA Hold Time from SCL
THD:DAT
300
ns
SDA Setup Time from SCL
TSU:DAT
250
ns
SCL Low Timeout (Note 7)
TTIMEOUT
22
SCL Low Period
TLOW
4.7
µs
SCL High Period
THIGH
4
µs
Maximum Charging Period Without a SMBus Write to
ChargeVoltage or ChargeCurrent Register
140
25
180
30
220
ms
s
NOTE:
7. If SCL is low for longer than the specified time, the charger is disabled.
5
FN9258.3
June 8, 2011
ISL88731
Typical Operating Performance
DCIN = 20V, 3S2P Li-Battery, TA = +25°C, unless otherwise noted.
5.15
3.23
5.10
3.22
5.05
3.21
1.0%
5.00
VREF (V)
VDDP (V)
0.5%
4.95
0.0%
3.20
3.19
-0.5%
3.18
4.90
4.85
20
40
60
80
0
100
50
100
150
I VREF (µA)
VDDP LOAD CURRENT (mA)
FIGURE 4. VREF LOAD REGULATION
FIGURE 3. VDD LOAD REGULATION
15
13.0
10
12.5
3.5
3.0
2.5
BATTERY VOLTAGE
5
0
-5
-10
12.0
VCHG (V)
1.5
11.0
1.0
10.5
0.5
ICHG (A)
-15
1
2
3
4
5
6
7
8
2.0
11.5
10.0
0
20
40
AC-ADAPTER CURRENT (A)
BATTERY CURRENT
ICM ACCURACY (%)
-1.0%
200
3.17
0
60
80
100
120
140
0.0
160
TIME (MINUTES)
FIGURE 5. ICM ACCURACY vs AC-ADAPTER CURRENT
VCOMP
ICOMP
FIGURE 6. TYPICAL CHARGING VOLTAGE AND CURRENT
ICOMP
VCOMP
CHARGE
CURRENT
CHARGE
CURRENT
INDUCTOR
CURRENT
FIGURE 7. CHARGE ENABLE
6
INDUCTOR
CURRENT
FIGURE 8. CHARGE DISABLE
FN9258.3
June 8, 2011
ISL88731
Typical Operating Performance
UGATE
DCIN = 20V, 3S2P Li-Battery, TA = +25°C, unless otherwise noted.
UGATE
LGATE
INDUCTOR
CURRENT
PHASE
LGATE
INDUCTOR
CURRENT
PHASE
FIGURE 10. SWITCHING WAVEFORMS IN CC MODE
FIGURE 9. SWITCHING WAVEFORMS AT DIODE EMULATION
CSON/
V BATTERY
CSON/
V BATTERY
BATTERY
CURRENT
BATTERY
CURRENT
FIGURE 12. BATTERY INSERTION
FIGURE 11. BATTERY REMOVAL
100
SYSTEM
LOAD
CHARGE
CURRENT
ADAPTER
CURRENT
90
EFFICIENCY (%)
BATTERY
VOLTAGE
95
16.8V BATTERY
85
12.6V BATTERY
80
8.4V BATTERY
75
4.2V BATTERY
70
0
FIGURE 13. LOAD TRANSIENT RESPONSE
7
2
4
CHARGE CURRENT (A)
8
6
FIGURE 14. EFFICIENCY vs CHARGE CURRENT AND BATTERY
VOLTAGE (EFFICIENCY DCIN = 20V)
FN9258.3
June 8, 2011
ISL88731
Functional Pin Descriptions
PGND
BOOT
Power Ground. Connect PGND to the source of the low side
MOSFET.
High-Side Power MOSFET Driver Power-Supply Connection.
Connect a 0.1µF capacitor from BOOT-to -PHASE.
VCC
UGATE
High-Side Power MOSFET Driver Output. Connect to the high-side
N-channel MOSFET gate.
LGATE
Low-Side Power MOSFET Driver Output. Connect to low-side N
channel MOSFET. LGATE drives between VDDP and PGND.
PHASE
High-Side Power MOSFET Driver Source Connection. Connect to
the source of the high-side N-Channel MOSFET.
Power input for internal analog circuits. Connect a 4.7Ω resistor
from VCC to VDDP and a 1µF ceramic capacitor from VCC to
ground.
VDDP
Linear Regulator Output. VDDP is the output of the 5.2V linear
regulator supplied from DCIN. VDDP also directly supplies the
LGATE driver and the BOOT strap diode. Bypass with a 1µF
ceramic capacitor from VDDP to PGND.
ICOMP
Charge Current-Sense Positive Input.
Compensation Point for the charging current and adapter current
regulation Loop. Connect 0.01µF to GND. See “Charge Current
Control Loop” on page 18. for details of selecting the ICOMP
capacitor.
CSON
VCOMP
Charge Current-Sense Negative Input.
Compensation Point for the voltage regulation loop. Connect
4.7kΩ in series with 0.01µF to GND. See “Voltage Control Loop”
on page 19 for details on selecting VCOMP components.
CSOP
CSSP
Input Current-Sense Positive Input.
VFB
CSSN
Feedback for the Battery Voltage.
Input Current-Sense Negative Input.
VDDSMB
DCIN
SMBus interface Supply Voltage Input. Bypass with a 0.1µF
capacitor to GND.
Charger Bias Supply Input. Bypass DCIN with a 0.1µF capacitor to
GND.
ACIN
AC Adapter Detection Input. Connect to a resistor divider from the
AC adapter output.
ACOK
AC Detect Output. This open drain output is high impedance
when ACIN is greater than 3.2V. The ACOK output remains low
when the ISL88731 is powered down. Connect a 10k pull-up
resistor from ACOK to VDDSMB.
ICM
Input Current Monitor Output. ICM voltage equals 20 x
(VCSSP - VCSSN).
SDA
SMBus Data I/O. Open-drain Output. Connect an external pull-up
resistor according to SMBus specifications.
SCL
SMBus Clock Input. Connect an external pull-up resistor
according to SMBus specifications.
GND
Analog Ground. Connect directly to the backside paddle. Connect
to PGND close to the output capacitor.
Back Side Paddle
Connect the backside paddle to GND.
VREF
NC
VREF is a reference output pin. It is internally compensated. Do
not connect a decoupling capacitor.
No Connect. Pins 1, 5, 7 and 14 are not connected.
8
FN9258.3
June 8, 2011
ISL88731
Theory of Operation
Introduction
The ISL88731 includes all of the functions necessary to charge 1
to 4 cell Li-Ion and Li-polymer batteries. A high efficiency
synchronous buck converter is used to control the charging
voltage up to 19.2V and charging current up to 8A. The ISL88731
also has input current limiting up to 11A. The Input current limit,
charge current limit and charge voltage limit are set by internal
registers written with SMBus. The ISL88731 “Typical Application
Circuit” is shown in Figure 2.
The ISL88731 charges the battery with constant charge current,
set by the ChargeCurrent register, until the battery voltage rises to
a voltage set by the ChargeVoltage register. The charger will then
operate at a constant voltage. The adapter current is monitored
and if the adapter current rises to the limit set by the InputCurrent
register, battery charge current is reduced so the charger does not
reduce the adapter current available to the system.
The ISL88731 features a voltage regulation loop (VCOMP) and 2
current regulation loops (ICOMP). The VCOMP voltage regulation
loop monitors VFB to limit the battery charge voltage. The ICOMP
current regulation loop limits the battery charging current
delivered to the battery to ensure that it never exceeds the
current set by the ChargeCurrent register. The ICOMP current
regulation loop also limits the input current drawn from the
AC-adapter to ensure that it never exceeds the limit set by the
InputCurrent register, and to prevent a system crash and
AC-adapter overload.
PWM Control
The ISL88731 employs a fixed frequency PWM control
architecture with a feed-forward function. The feed-forward
function maintains a constant modulator gain of 11 to achieve fast
line regulation as the input voltage changes.
The duty cycle of the buck regulator is controlled by the lower of
the voltages on ICOMP and VCOMP. The voltage on ICOMP and
VCOMP are inputs to a Lower Voltage Buffer (LVB) who’s output is
the lower of the 2 inputs. The output of the LVB is compared to an
internal 400kHz ramp to produce the Pulse Width Modulated
signal that controls the UGATE and LGATE drivers. An internal
clamp holds the higher of the 2 voltages (0.3V) above the lower
voltage. This speeds the transition from voltage loop control to
current loop control or vice versa.
The ISL88731 can operate up to 99.6% duty cycle if the input
voltage drops close to or below the battery charge voltage (drop
out mode). The DC/DC converter has a timer to prevent the
frequency from dropping into the audible frequency range.
To prevent boosting of the system bus voltage, the battery
charger drives the lower FET in a way that prevents negative
inductor current.
An adaptive gate drive scheme is used to control the dead time
between two switches. The dead time control circuit monitors the
LGATE output and prevents the upper side MOSFET from turning
on until 20ns after LGATE falls below 1V VGS, preventing
cross-conduction and shoot-through. The same occurs for LGATE
turn on. In order for the dead time circuit to work properly, there
must be a low resistance, low inductance path from the LGATE
9
driver to MOSFET gate, and from the source of MOSFET to PGND.
An internal Schottky diode between the VDDP pin and BOOT pin
keeps the bootstrap capacitor charged.
AC-Adapter Detection
Connect the AC-adapter voltage through a resistor divider to ACIN
to detect when AC power is available, as shown in Figure 2. ACOK
is an open-drain output and is active low when ACIN is less than
Vth,fall, and high when ACIN is above Vth,rise. The ACIN rising
threshold is 3.2V (typ) with 57mV hysteresis.
Current Measurement
Use ICM to monitor the adapter current being sensed across
CSSP and CSSN. The output voltage range is 0 to 2.5V. The
voltage of ICM is proportional to the voltage drop across CSSP
and CSSN, and is given by Equation 1:
(EQ. 1)
ICM = 20 ⋅ I INPUT ⋅ R S1
where Iadapter is the DC current drawn from the AC adapter. It is
recommended to have an RC filter at the ICM output for
minimizing the switching noise.
VDDP Regulator
VDDP provides a 5.2V supply voltage from the internal LDO
regulator from DCIN and can deliver up to 30mA of continuous
current. The MOSFET drivers are powered by VDDP. VDDP also
supplies power to VCC through a low pass filter as shown in the
”Typical Application Circuit” section on page 2. Bypass VDDP and
VCC with a 1µF capacitor.
VDDSMB Supply
The VDDSMB input provides power to the SMBus interface.
Connect VDDSMB to VCC, or apply an external supply to VDDSMB.
Bypass VDDSMB to GND with a 0.1µF or greater ceramic
capacitor.
The typical application connects VDDSMB to the same power
source as the SMBus master. This supply should be active and
greater than 2.5V when either the adapter or the battery is
present.
ISL88731 does not function when VDDSMB is below its specified
Under Voltage Lockout (UVLO) voltage. All of the SMBus registers
in ISL88731 are powered by VDDSMB and are set to zero when it
is below the UVLO threshold. Other functions are unpredictable
when VDDSMB is below the UVLO threshold.
Short Circuit Protection and 0V Battery
Charging
Since the battery charger will regulate the charge current to the
limit set by the ChargeCurrent register, it automatically has short
circuit protection and is able to provide the charge current to
wake up an extremely discharged battery. Undervoltage trickle
charge folds back current if there is a short circuit on the output.
FN9258.3
June 8, 2011
ISL88731
Undervoltage Detect and Battery Trickle
Charging
If the voltage at CSON falls below 2.5V ISL88731 reduces the
charge current limit to 128mA to trickle charge the battery.
When the voltage rises above 2.7V the charge current reverts to
the programmed value in the ChargeCurrent register.
START and STOP Conditions
As shown in Figure 16, START condition is a HIGH-to-LOW transition
of the SDA line while SCL is HIGH.
The STOP condition is a LOW-to-HIGH transition on the SDA line
while SCL is HIGH. A STOP condition must be sent before each
START condition.
Over Temperature Protection
If the die temp exceeds +150°C, it stops charging. Once the die
temp drops below +125°C, charging will start up again.
The System Management Bus
SDA
SCL
The System Management Bus (SMBus) is a 2-wire bus that
supports bidirectional communications. The protocol is described
briefly here. More detail is available from www.smbus.org.
S
P
START
CONDITION
STOP
CONDITION
FIGURE 16. START AND STOP WAVEFORMS
General SMBus Architecture
Acknowledge
VDDSMB
SMBUS SLAVE
INPUT
SCL
OUTPUTCONTROL
SMBUS MASTER
INPUT
SDA
OUTPUTCONTROL
INPUT
SCL
CONTROL OUTPUT
CPU
INPUT
SDA
CONTROL OUTPUT
STATE
MACHINE,
REGISTERS,
MEMORY,
ETC
SMBUS SLAVE
INPUT
SCL
OUTPUT CONTROL
SDA
S CL
INPUT
SDA
OUTPUT CONTROL
STATE
MACHINE,
REGISTERS,
MEMORY,
ETC
Each address and data transmission uses 9-clock pulses. The ninth
pulse is the acknowledge bit (ACK). After the start condition, the
master sends 7-slave address bits and a R/W bit during the next 8clock pulses. During the ninth clock pulse, the device that recognizes
its own address holds the data line low to acknowledge. The
acknowledge bit is also used by both the master and the slave to
acknowledge receipt of register addresses and data (see Figure 17).
SCL
1
2
8
9
SDA
MSB
START
ACKNOWLEDGE
FROM SLAVE
TO OTHER
SLAVE DEVICES
FIGURE 17. ACKNOWLEDGE ON THE I2C BUS
Data Validity
The data on the SDA line must be stable during the HIGH period
of the SCL, unless generating a START or STOP condition. The
HIGH or LOW state of the data line can only change when the
clock signal on the SCL line is LOW. Refer to Figure 15.
SDA
SCL
DATA LINE CHANGE
STABLE
OF DATA
DATA VALID ALLOWED
FIGURE 15. DATA VALIDITY
10
SMBus Transactions
All transactions start with a control byte sent from the SMBus
master device. The control byte begins with a Start condition,
followed by 7-bits of slave address (0001001 for the ISL88731)
followed by the R/W bit. The R/W bit is 0 for a write or 1 for a read. If
any slave devices on the SMBus bus recognize their address, they
will Acknowledge by pulling the serial data (SDA) line low for the last
clock cycle in the control byte. If no slaves exist at that address or
are not ready to communicate, the data line will be 1, indicating a
Not Acknowledge condition.
Once the control byte is sent, and the ISL88731 acknowledges it,
the 2nd byte sent by the master must be a register address byte
such as 0x14 for the ChargeCurrent register. The register address
byte tells the ISL88731 which register the master will write or
read. See Table 1 for details of the registers. Once the ISL88731
receives a register address byte it responds with an acknowledge.
FN9258.3
June 8, 2011
ISL88731
Write To A Register
S
SLAVE
ADDR + W
A
REGISTER
ADDR
LO BYTE
DATA
A
A
HI BYTE
DATA
A
A
LO BYTE
DATA
P
Read From A Register
S
SLAVE
ADDR + W
A
REGISTER
ADDR
A
P
S
SLAVE
ADDR + R
A
HI BYTE
DATA
N
S
START
A
ACKNOWLEDGE
DRIVEN BY THE MASTER
P
STOP
N
NO ACKNOWLEDGE
DRIVEN BY ISL88731
P
FIGURE 18. SMBus/ISL88731 READ AND WRITE PROTOCOL
Byte Format
Every byte put on the SDA line must be eight bits long and must
be followed by an acknowledge bit. Data is transferred with the
most significant bit first (MSB) and the least significant bit last
(LSB).
ISL88731 and SMBus
The ISL88731 receives control inputs from the SMBus interface.
The serial interface complies with the SMBus protocols as
documented in the System Management Bus Specification V1.1,
which can be downloaded from www.smbus.org. The ISL88731
uses the SMBus Read-Word and Write-Word protocols (Figure 18)
to communicate with the smart battery. The ISL88731 is an
SMBus slave device and does not initiate communication on the
bus. It responds to the 7-bit address 0b0001001_ (0x12).
Read address = 0b00010011 and
Write address = 0b00010010.
In addition, the ISL88731 has two identification (ID) registers: a
16-bit device ID register and a 16-bit manufacturer ID register.
The data (SDA) and clock (SCL) pins have Schmitt-trigger inputs
that can accommodate slow edges. Choose pull-up resistors for
SDA and SCL to achieve rise times according to the SMBus
specifications. The ISL88731 is controlled by the data written to
the registers described in Table 1.
Battery Charger Registers
The ISL88731 supports five battery-charger registers that use
either Write-Word or Read-Word protocols, as summarized in
Table 1. ManufacturerID and DeviceID are “read only” registers
and can be used to identify the ISL88731. On the ISL88731,
ManufacturerID always returns 0x0049 (ASCII code for “I” for
Intersil) and DeviceID always returns 0x0001.
Enabling and Disabling Charging
After applying power to ISL88731, the internal registers contain
their POR values (see Table 1). The POR values for charge current
and charge voltage are 0x0000. These values disable charging.
To enable charging, the ChargeCurrent register must be written
with a number >0x007F and the ChargeVoltage register must be
written with a number >0x000F. Charging can be disabled by
writing 0x0000 to either of these registers.
TABLE 1. BATTERY CHARGER REGISTER SUMMARY
REGISTER
ADDRESS
REGISTER NAME
READ/WRITE
DESCRIPTION
POR STATE
0x14
ChargeCurrent
Read or Write
6-bit Charge Current Setting
0x0000
0x15
ChargeVoltage
Read or Write
11-bit Charge Voltage Setting
0x0000
0x3F
InputCurrent
Read or Write
6-bit Charge Current Setting
0x0080
0xFE
ManufacturerID
Read Only
Manufacturer ID
0x0049
0xFF
DeviceID
Read Only
Device ID
0x0001
11
FN9258.3
June 8, 2011
ISL88731
Setting Charge Voltage
Charge voltage is set by writing a valid 16-bit number to the
ChargeVoltage register. This 16-bit number translates to a
65.535V full-scale voltage. The ISL88731 ignores the first 4 LSBs
and uses the next 11 bits to set the voltage DAC. The charge
voltage range of the ISL88731 is 1.024V to 19.200V. Numbers
requesting charge voltage greater than 19.200V result in a
ChargeVoltage of 19.200V. All numbers requesting charge
voltage below 1.024V result in a voltage set point of zero, which
terminates charging. Upon initial power-up or reset, the
ChargeVoltage and ChargeCurrent registers are reset to 0 and
the charger remains shut down until valid numbers are sent to
the ChargeVoltage and ChargeCurrent registers. Use the WriteWord protocol (Figure 18) to write to the ChargeVoltage register.
The register address for ChargeVoltage is 0x15. The 16-bit binary
number formed by D15–D0 represents the charge voltage set
point in mV. However, the resolution of the ISL88731 is 16mV
because the D0–D3 bits are ignored as shown in Table 2. The
D15 bit is also ignored because it is not needed to span the
1.024V to 19.2V range. Table 2 shows the mapping between the
charge-voltage set point and the 16-bit number written to the
ChargeVoltage register. The ChargeVoltage register can be read
back to verify its contents.
TABLE 2. CHARGEVOLTAGE (REGISTER 0x15)
BIT
BIT NAME
DESCRIPTION
0
Not used.
1
Not used.
2
Not used.
3
Not used.
4
Charge Voltage, DACV 0
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 16mV of charger voltage.
5
Charge Voltage, DACV 1
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 32mV of charger voltage.
6
Charge Voltage, DACV 2
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 64mV of charger voltage.
7
Charge Voltage, DACV 3
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 128mV of charger voltage.
8
Charge Voltage, DACV 4
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 256mV of charger voltage.
9
Charge Voltage, DACV 5
0 = Adds 0mV of charger voltage, 1024mV min.
1 = Adds 512mV of charger voltage.
10
Charge Voltage, DACV 6
0 = Adds 0mA of charger voltage.
1 = Adds 1024mV of charger voltage.
11
Charge Voltage, DACV 7
0 = Adds 0mV of charger voltage.
1 = Adds 2048mV of charger voltage.
12
Charge Voltage, DACV 8
0 = Adds 0mV of charger voltage.
1 = Adds 4096mV of charger voltage.
13
Charge Voltage, DACV 9
0 = Adds 0mV of charger voltage.
1 = Adds 8192mV of charger voltage.
14
Charge Voltage, DACV 10
0 = Adds 0mV of charger voltage.
1 = Adds 16384mV of charger voltage, 19200mV max.
15
Not used. Normally a 32768mV weight.
12
FN9258.3
June 8, 2011
ISL88731
Setting Charge Current
ISL88731 has a 16-bit ChargeCurrent register that sets the
battery charging current. ISL88731 controls the charge current
by controlling the CSOP-CSON voltage. The register’s LSB
translates to 10µV at CSON-CSOP. With a 10mΩ charge current
Rsense resistor (RS2 in ”Typical Application Circuit” on page 2),
the LSB translates to 1mA charge current. The ISL88731 ignores
the first 7 LSBs and uses the next 6 bits to control the current
DAC. The charge-current range of the ISL88731 is 0 to 8.064A
(using a 10mΩ current-sense resistor). All numbers requesting
charge current above 8.064A result in a current setting of
8.064A. All numbers requesting charge current between 0mA to
128mA result in a current setting of 0mA. The default charge
current setting at Power-On Reset (POR) is 0mA. To stop
charging, set ChargeCurrent to 0. Upon initial power up, the
ChargeVoltage and ChargeCurrent registers are reset to 0 and
the charger is disabled. To start the charger, write valid numbers
to the ChargeVoltage and ChargeCurrent registers. The
ChargeCurrent register uses the Write-Word protocol (Figure 18).
The register code for ChargeCurrent is 0x14 (0b00010100).
Table 3 shows the mapping between the charge current set point
and the ChargeCurrent number. The ChargeCurrent register can
be read back to verify its contents.
The ISL88731 includes a fault limiter for low battery conditions.
If the battery voltage is less than 2.5V, the charge current is
temporarily set to 128mA. The ChargeCurrent register is
preserved and becomes active again when the battery voltage is
higher than 2.7V. This function effectively provides a foldback
current limit, which protects the charger during short circuit and
overload.
TABLE 3. CHARGE CURRENT (REGISTER 0x14) (10mΩ SENSE RESISTOR, RS2)
BIT
BIT NAME
DESCRIPTION
0
Not used.
1
Not used.
2
Not used.
3
Not used.
4
Not used.
5
Not used.
6
Not used.
7
Charge Current, DACI 0
0 = Adds 0mA of charger current.
1 = Adds 128mA of charger current.
8
Charge Current, DACI 1
0 = Adds 0mA of charger current.
1 = Adds 256mA of charger current.
9
Charge Current, DACI 2
0 = Adds 0mA of charger current.
1 = Adds 512mA of charger current.
10
Charge Current, DACI 3
0 = Adds 0mA of charger current.
1 = Adds 1024mA of charger current.
11
Charge Current, DACI 4
0 = Adds 0mA of charger current.
1 = Adds 2048mA of charger current.
12
Charge Current, DACI 5
0 = Adds 0mA of charger current.
1 = Adds 4096mA of charger current, 8064mA max.
13
Not used.
14
Not used.
15
Not used.
13
FN9258.3
June 8, 2011
ISL88731
Setting Input-Current Limit
The total power from an AC adapter is the sum of the power
supplied to the system and the power into the charger and battery.
When the input current exceeds the set input current limit, the
ISL88731 decreases the charge current to provide priority to
system load current. As the system load rises, the available charge
current drops linearly to zero. Thereafter, the total input current
can increase to the limit of the AC adapter.
The internal amplifier compares the differential voltage between
CSSP and CSSN to a scaled voltage set by the InputCurrent
register. The total input current is the sum of the device supply
current, the charger input current, and the system load current.
The total input current can be estimated as shown in Equation 2.
I INPUT = I SYSTEM + [ ( I CHARGE × V BATTERY ) ⁄ ( V IN × η ) ]
(EQ. 2)
Where η is the efficiency of the DC/DC converter (typically 85%
to 95%).
The ISL88731 has a 16-bit InputCurrent register that translates
to a 2mA LSB and a 131.071A full scale current using a 10mΩ
current-sense resistor (RS1 in Figure 2). Equivalently, the 16-bit
InputCurrent number sets the voltage across CSSP and CSSN
inputs in 20µV per LSB increments. To set the input current limit
use the SMBus to write a 16-bit InputCurrent register using the
data format listed in Table 4. The InputCurrent register uses the
Write-Word protocol (see Figure 18). The register code for
InputCurrent is 0x3F (0b00111111). The InputCurrent register
can be read back to verify its contents.
The ISL88731 ignores the first 7 LSBs and uses the next
6 bits to control the input-current DAC. The input-current range of
the ISL88731 is from 256mA to 11.004A. All 16-bit numbers
requesting input current above 11.004A result in an inputcurrent setting of 11.004A. All 16-bit numbers requesting input
current between 0mA to 256mA result in an input-current setting
of 0mA. The default input-current-limit setting at POR is 256mA.
When choosing the current-sense resistor RS1, carefully
calculate its power rating. Take into account variations in the
system’s load current and the overall accuracy of the sense
amplifier. Note that the voltage drop across RS1 contributes
additional power loss, which reduces efficiency. System currents
normally fluctuate as portions of the system are powered up or
put to sleep. Without input current regulation, the input source
must be able to deliver the maximum system current and the
maximum charger-input current. By using the input-current-limit
circuit, the output-current capability of the AC wall adapter can
be lowered, reducing system cost.
TABLE 4. INPUT CURRENT (REGISTER 0x3F) (10mΩ SENSE RESISTOR, RS1)
BIT
BIT NAME
DESCRIPTION
0
Not used.
1
Not used.
2
Not used.
3
Not used.
4
Not used.
5
Not used.
6
Not used.
7
Input Current, DACS 0
0 = Adds 0mA of input current.
1 = Adds 256mA of input current.
8
Input Current, DACS 1
0 = Adds 0mA of input current.
1 = Adds 512mA of input current.
9
Input Current, DACS 2
0 = Adds 0mA of input current.
1 = Adds 1024mA of input current.
10
Input Current, DACS 3
0 = Adds 0mA of input current.
1 = Adds 2048mA of input current.
11
Input Current, DACS 4
0 = Adds 0mA of input current.
1 = Adds 4096mA of input current.
12
Input Current, DACS 5
0 = Adds 0mA of input current.
1 = Adds 8192mA of input current, 11004mA max.
13
Not used.
14
Not used.
15
Not used.
14
FN9258.3
June 8, 2011
ISL88731
Charger Timeout
The ISL88731 includes 2 timers to insure the SMBus master is
active and to prevent overcharging the battery. ISL88731 will
terminate charging if the charger has not received a write to the
ChargeVoltage or ChargeCurrent register within 175s or if the
SCL line is low for more than 25ms. If a time-out occurs, either
ChargeVoltage or ChargeCurrent registers must be written to reenable charging.
ISL88731 Data Byte Order
Each register in ISL88731 contains 16-bits or 2, 8 bit bytes. All
data sent on the SMBus is in 8 bit bytes and 2 bytes must be
written or read from each register in ISL88731. The order in
which these bytes are transmitted appears reversed from the
way they are normally written. The LOW byte is sent first and the
HI byte is sent second. For example, When writing 0x41A0, 0xA0
is written first and 0x41 is sent second.
Writing to the Internal Registers
In order to set the charge current, charge voltage or input current,
valid 16-bit numbers must be written to ISL88731’s internal
registers via the SMBus.
To write to a register in the ISL88731, the master sends a control
byte with the R/W bit set to 0, indicating a write. If it receives an
Acknowledge from the ISL88731 it sends a register address byte
setting the register to be written (i.e., 0x14 for the ChargeCurrent
register). The ISL88731 will respond with an Acknowledge. The
master then sends the lower data byte to be written into the
desired register. The ISL88731 will respond with an
Acknowledge. The master then sends the higher data byte to be
written into the desired register. The ISL88731 will respond with
an Acknowledge. The master then issues a Stop condition,
indicating to the ISL88731 that the current transaction is
complete. Once this transaction completes the ISL88731 will
begin operating at the new current or voltage.
ISL88731 does not support writing more than one register per
transaction
Reading from the Internal
Registers
The ISL88731 has the ability to read from 5 internal registers.
Prior to reading from an internal register, the master must first
select the desired register by writing to it and sending the registers
address byte. This process begins by the master sending a control
byte with the R/W bit set to 0, indicating a write. Once it receives
an Acknowledge from the ISL88731 it sends a register address
byte representing the internal register it wants to read. The
ISL88731 will respond with an Acknowledge. The master must
then respond with a Stop condition. After the Stop condition the
master follows with a new Start condition, then sends a new
control byte with the ISL88731 slave address and the R/W bit set
to 1, indicating a read. The ISL88731 will Acknowledge then send
the lower byte stored in that register. After receiving the byte, the
master Acknowledges by holding SDA low during the 9th clock
pulse. ISL88731 then sends the higher byte stored in the register.
After the second byte neither device holds SDA low (No
15
Acknowledge). The master will then produce a Stop condition to
end the read transaction.
ISL88731 does not support reading more than 1 register per
transaction.
Application Information
The following battery charger design refers to the “Typical
Application Circuit” (see Figure 2), where typical battery
configuration of 3S2P is used. This section describes how to select
the external components including the inductor, input and output
capacitors, switching MOSFETs and current sensing resistors.
Inductor Selection
The inductor selection has trade-offs between cost, size, cross
over frequency and efficiency. For example, the lower the
inductance, the smaller the size, but ripple current is higher. This
also results in higher AC losses in the magnetic core and the
windings, which decreases the system efficiency. On the other
hand, the higher inductance results in lower ripple current and
smaller output filter capacitors, but it has higher DCR (DC
resistance of the inductor) loss, lower saturation current and has
slower transient response. So, the practical inductor design is
based on the inductor ripple current being ±15% to ±20% of the
maximum operating DC current at maximum input voltage.
Maximum ripple is at 50% duty cycle or VBAT = VIN,MAX/2. The
required inductance for ±15% ripple current can be calculated
from Equation 3:
V IN, MAX
L = ------------------------------------------------------4 ⋅ F SW ⋅ 0.3 ⋅ I L, MAX
(EQ. 3)
Where VIN,MAX is the maximum input voltage, FSW is the
switching frequency and IL,MAX is the max DC current in the
inductor.
For VIN,MAX = 20V, VBAT = 12.6V, IBAT,MAX = 4.5A, and
fs = 400kHz, the calculated inductance is 9.3µH. Choosing the
closest standard value gives L = 10µH. Ferrite cores are often the
best choice since they are optimized at 400kHz to 600kHz
operation with low core loss. The core must be large enough not
to saturate at the peak inductor current IPeak in Equation 4:
1
I PEAK = I L, MAX + --- ⋅ I RIPPLE
2
(EQ. 4)
Inductor saturation can lead to cascade failures due to very high
currents. Conservative design limits the peak and RMS current in
the inductor to less than 90% of the rated saturation current.
Crossover frequency is heavily dependent on the inductor value.
FCO should be less than 20% of the switching frequency and a
conservative design has FCO less than 10% of the switching
frequency. The highest FCO is in voltage control mode with the
battery removed and may be calculated (approximately) from
Equation 5:
5 ⋅ 11 ⋅ R SENSE
F CO = ---------------------------------------2π ⋅ L
(EQ. 5)
Output Capacitor Selection
The output capacitor in parallel with the battery is used to absorb
the high frequency switching ripple current and smooth the
FN9258.3
June 8, 2011
ISL88731
output voltage. The RMS value of the output ripple current IRMS
is given by Equation 6:
V IN, MAX
I RMS = ---------------------------------- ⋅ D ⋅ ( 1 – D )
12 ⋅ L ⋅ F SW
(EQ. 6)
Where the duty cycle D is the ratio of the output voltage (battery
voltage) over the input voltage for continuous conduction mode
which is typical operation for the battery charger. During the
battery charge period, the output voltage varies from its initial
battery voltage to the rated battery voltage. So, the duty cycle
varies from 0.53 for the minimum battery voltage of 7.5V
(2.5V/Cell) to 0.88 for the maximum battery voltage of 12.6V.
The maximum RMS value of the output ripple current occurs at
the duty cycle of 0.5 and is expressed as Equation 7:
V IN, MAX
I RMS = -----------------------------------------4 ⋅ 12 ⋅ L ⋅ F SW
(EQ. 7)
For VIN,MAX = 19V, VBAT = 16.8V, L = 10µH, and fs = 400kHz, the
maximum RMS current is 0.19A. A typical 20µF ceramic
capacitor is a good choice to absorb this current and also has
very small size. Organic polymer capacitors have high
capacitance with small size and have a significant equivalent
series resistance (ESR). Although ESR adds to ripple voltage, it
also creates a high frequency zero that helps the closed loop
operation of the buck regulator.
EMI considerations usually make it desirable to minimize ripple
current in the battery leads. Beads may be added in series with
the battery pack to increase the battery impedance at 400kHz
switching frequency. Switching ripple current splits between the
battery and the output capacitor depending on the ESR of the
output capacitor and battery impedance. If the ESR of the output
capacitor is 10mΩ and battery impedance is raised to 2Ω with a
bead, then only 0.5% of the ripple current will flow in the battery.
MOSFET Selection
The Notebook battery charger synchronous buck converter has
the input voltage from the AC-adapter output. The maximum ACadapter output voltage does not exceed 25V. Therefore, 30V logic
MOSFET should be used.
The high side MOSFET must be able to dissipate the conduction
losses plus the switching losses. For the battery charger
application, the input voltage of the synchronous buck converter
is equal to the AC-adapter output voltage, which is relatively
constant. The maximum efficiency is achieved by selecting a
high side MOSFET that has the conduction losses equal to the
switching losses. Switching losses in the low-side FET are very
small. The choice of low-side FET is a trade-off between
conduction losses (rDS(ON)) and cost. A good rule of thumb for
the rDS(ON) of the low-side FET is 2x the rDS(ON) of the high-side
FET.
The LGATE gate driver can drive sufficient gate current to switch
most MOSFETs efficiently. However, some FETs may exhibit cross
conduction (or shoot-through) due to current injected into the
drain-to-source parasitic capacitor (Cgd) by the high dV/dt rising
edge at the phase node when the high side MOSFET turns on.
Although LGATE sink current (1.8A typical) is more than enough
to switch the FET off quickly, voltage drops across parasitic
impedances between LGATE and the MOSFET can allow the gate
16
to rise during the fast rising edge of voltage on the drain.
MOSFETs with low threshold voltage (<1.5V) and low ratio of
Cgs/Cgd (<5) and high gate resistance (>4Ω) may be turned on
for a few ns by the high dV/dt (rising edge) on their drain. This
can be avoided with higher threshold voltage and Cgs/Cgd ratio.
Another way to avoid cross conduction is slowing the turn-on
speed of the high-side MOSFET by connecting a resistor between
the BOOT pin and the bootstrap capacitor.
For the high-side MOSFET, the worst-case conduction losses
occur at the minimum input voltage, as shown in Equation 8:
V OUT
2
P Q1, conduction = ------------- ⋅ I BAT ⋅ r DS ( ON )
V IN
(EQ. 8)
The optimum efficiency occurs when the switching losses equal
the conduction losses. However, it is difficult to calculate the
switching losses in the high-side MOSFET since it must allow for
difficult-to-quantify factors that influence the turn-on and turn-off
times. These factors include the MOSFET internal gate
resistance, gate charge, threshold voltage, stray inductance and
the pull-up and pull-down resistance of the gate driver.
The following switching loss calculation (Equation 9) provides a
rough estimate.
P Q1, Switching =
⎛ Q gd ⎞ 1
⎛ Q gd ⎞
1
-⎟ + --- V IN I LP f sw ⎜ ----------------⎟ + Q rr V IN f sw
--- V IN I LV f sw ⎜ ----------------------2
⎝ I g, source⎠ 2
⎝ I g, sin k⎠
(EQ. 9)
where the following are the peak gate-drive source/sink current
of Q1, respectively:
• Qgd: drain-to-gate charge,
• Qrr: total reverse recovery charge of the body-diode in low-side
MOSFET,
• ILV: inductor valley current,
• ILP: Inductor peak current,
• Ig,sink
• Ig,source
Low switching loss requires low drain-to-gate charge Qgd.
Generally, the lower the drain-to-gate charge, the higher the
ON-resistance. Therefore, there is a trade-off between the
ON-resistance and drain-to-gate charge. Good MOSFET selection
is based on the Figure of Merit (FOM), which is a product of the
total gate charge and ON-resistance. Usually, the smaller the
value of FOM, the higher the efficiency for the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum battery voltage and maximum input voltage
as shown in Equation 10.
V OUT⎞
⎛
2
P Q2 = ⎜ 1 – -------------⎟ ⋅ I BAT ⋅ r DS ( ON )
V IN ⎠
⎝
(EQ. 10)
Choose a low-side MOSFET that has the lowest possible onresistance with a moderate-sized package like the 8 Ld SOIC and
is reasonably priced. The switching losses are not an issue for the
low-side MOSFET because it operates at zero-voltage-switching.
Ensure that the required total gate drive current for the selected
MOSFETs should be less than 24mA. So, the total gate charge for
the high-side and low-side MOSFETs is limited by Equation 11:
FN9258.3
June 8, 2011
ISL88731
(EQ. 11)
Where IGATE is the total gate drive current and should be less
than 24mA. Substituting IGATE = 24mA and fs = 400kHz into the
Equation 11 yields that the total gate charge should be less than
80nC. Therefore, the ISL88731 easily drives the battery charge
current up to 8A.
Snubber Design
ISL88731's buck regulator operates in discontinuous current
mode (DCM) when the load current is less than half the
peak-to-peak current in the inductor. After the low-side FET turns
off, the phase voltage rings due to the high impedance with both
FETs off. This can be seen in Figure 9. Adding a snubber (resistor
in series with a capacitor) from the phase node to ground can
greatly reduce the ringing. In some situations a snubber can
improve output ripple and regulation.
The snubber capacitor should be approximately twice the
parasitic capacitance on the phase node. This can be estimated
by operating at very low load current (100mA) and measuring the
ringing frequency.
CSNUB and RSNUB can be calculated from Equations 12 and 13:
2
C SNUB = -----------------------------------2
( 2πF ring ) ⋅ L
R SNUB =
(EQ. 12)
2⋅L
----------------C SNUB
(EQ. 13)
Transconductance Amplifiers GMV, GMI and
GMS
ISL88731 uses several transconductance amplifiers (also known
as gm amps). Most commercially available op amps are voltage
controlled voltage sources with gain expressed as A = VOUT/VIN.
gm amps are voltage controlled current sources with gain
expressed as gm = IOUT/VIN. gm will appear in some of the
equations for poles and zeros in the compensation.
PWM Gain Fm
The Pulse Width Modulator in the ISL88731 converts voltage at
VCOMP to a duty cycle by comparing VCOMP to a triangle wave
(duty = VCOMP/VP-P RAMP). The low-pass filter formed by L and
CO convert the duty cycle to a DC output voltage
(Vo = VDCIN*duty). In ISL88731, the triangle wave amplitude is
proportional to VDCIN. Making the ramp amplitude proportional
to DCIN makes the gain from VCOMP to the PHASE output a
constant 11 and is independent of DCIN. For small signal AC
analysis, the battery is modeled by its internal resistance. The
total output resistance is the sum of the sense resistor and the
internal resistance of the MOSFETs, inductor and capacitor.
Figure 19 shows the small signal model of the pulse width
modulator (PWM), power stage, output filter and battery.
VDD
RAMP GEN
VRAMP = VDD/11
-
Input Capacitor Selection
The input capacitor absorbs the ripple current from the
synchronous buck converter, which is given by Equation 14:
V OUT ( V IN – V OUT )
I rms = I BAT -------------------------------------------------V IN
(EQ. 14)
This RMS ripple current must be smaller than the rated RMS
current in the capacitor datasheet. Non-tantalum chemistries
(ceramic, aluminum, or OSCON) are preferred due to their
resistance to power-up surge currents when the AC-adapter is
plugged into the battery charger. For Notebook battery charger
applications, it is recommended that ceramic capacitors or
polymer capacitors from Sanyo be used due to their small size
and reasonable cost.
Loop Compensation Design
ISL88731 has three closed loop control modes. One controls the
output voltage when the battery is fully charged or absent. A
second controls the current into the battery when charging and
the third limits current drawn from the adapter. The charge
current and input current control loops are compensated by a
single capacitor on the ICOMP pin. The voltage control loop is
compensated by a network on the VCOMP pin. Descriptions of
these control loops and guidelines for selecting compensation
components will be given in the following sections. Which loop
controls the output is determined by the minimum current buffer
and the minimum voltage buffer shown in the Block Diagram.
These three loops will be described separately.
17
+
L
DRIVERS
I GATE
Q GATE ≤ -------------F SW
CO
PWM
INPUT
PWM
GAIN=11
L
RSENSE
11
RFET_RDSON
RL_DCR
PWM
INPUT
CO
RBAT
RESR
FIGURE 19. SMALL SIGNAL AC MODEL
In most cases the Battery resistance is very small (<200mΩ)
resulting in a very low Q in the output filter. This results in a
frequency response from the input of the PWM to the inductor
current with a single pole at the frequency calculated in
Equation 15:
( R SENSE + r DS ( ON ) + R DCR + R BAT )
F POLE1 = ---------------------------------------------------------------------------------------------2π ⋅ L
(EQ. 15)
The output capacitor creates a pole at a very high frequency due
to the small resistance in parallel with it. The frequency of this
pole is calculated in Equation 16:
FN9258.3
June 8, 2011
ISL88731
1
F POLE2 = -----------------------------------2π ⋅ C o ⋅ R BAT
(EQ. 16)
Charge Current Control Loop
When the battery is less than the fully charged, the voltage error
amplifier goes to it’s maximum output (limited to 0.3V above
ICOMP) and the ICOMP voltage controls the loop through the
minimum voltage buffer. Figure 21 shows the charge current
control loop.
The compensation capacitor (CICOMP) gives the error amplifier
(GMI) a pole at a very low frequency (<<1Hz) and a zero at FZ1.
FZ1 is created by the 0.25*CA2 output added to ICOMP. The
frequency can be calculated from Equation 17:
4 ⋅ gm2
F ZERO = ------------------------------------( 2π ⋅ C ICOMP )
gm2 = 50μA ⁄ V
(EQ. 17)
L
PHASE
11 ⋅ R SENSE
A DC = ---------------------------------------------------------------------------------------------( R SENSE + r DS ( ON ) + R DCR + R BAT )
(EQ. 20)
11 ⋅ R SENSE
F CO = A DC ⋅ F POLE = -------------------------------2π ⋅ L
(EQ. 21)
The Bode plot of the loop gain, the compensator gain and the
power stage gain is shown in Figure 21.
Adapter Current Limit Control Loop
If the combined battery charge current and system load current
draws current that equals the adapter current limit set by the
InputCurrent register, ISL88731 will reduce the current to the
battery and/or reduce the output voltage to hold the adapter
current at the limit. Above the adapter current limit the minimum
current buffer equals the output of GMS and ICOMP controls the
charger output. Figure 22 shows the adapter current limit control
loop.
60
Compensator
Modulator
11
RL_DCR
RFET_RDSON
40
+
S
Σ
CA2
+
0.25
-
-
GMI
+
CF2
DACI
CICOMP
CO
Loop
ZERO
20
RS2
CSON
-
ICOMP
RF2
CSOP
20X
RBAT
GAIN (dB)
Σ
F
0
-20
RESR
F
-40
F
POLE1
F
FIGURE 20. CHARGE CURRENT LIMIT LOOP
Placing this zero at a frequency equal to the pole calculated in
Equation 16 will result in maximum gain at low frequencies and
phase margin near 90°. If the zero is at a higher frequency
(smaller CICOMP), the DC gain will be higher but the phase
margin will be lower. Use a capacitor on ICOMP that is equal to or
greater than the value calculated in Equation 18. The factor of
1.5 is to ensure the zero is at a frequency lower than the pole
including tolerance variations.
1.5 ⋅ 4 ⋅ ( 50μA ⁄ V ) ⋅ L
C ICOMP = ---------------------------------------------------------------------------------------------( R SENSE + r DS ( ON ) + R DCR + R BAT )
(EQ. 18)
0.1
1
(EQ. 19)
10
1000
DCIN
L
PHASE
RS1
11
RFET_RDSON
CF1
+
0.25
Σ
S
CSSN
CSSP
-
CA2
+
20X
-
- 20
+
ICOMP
RL_DCR
RF2
CSOP
CF2
CO
+
RS2
CSON
CA1
RBAT
RESR
-
18
100
FIGURE 21. CHARGE CURRENT LOOP BODE PLOTS
GMS
The crossover frequency is determined by the DC gain of the
modulator and output filter and the pole in Equation 16. The DC
gain is calculated in Equation 20 and the cross over frequency is
calculated with Equation 21:
POLE2
FREQUENCY (kHz)
RF1
A filter should be added between RS2 and CSOP and CSON to
reduce switching noise. The filter roll-off frequency should be
between the crossover frequency and the switching frequency
(~100kHz). RF2 should be small (<10Ω) to minimize offsets due
to leakage current into CSOP. The filter cutoff frequency is
calculated using Equation 19:
1
F FILTER = ----------------------------------------( 2π ⋅ C F2 ⋅ R F2 )
-60
0.01
FILTER
DACS
CICOMP
FIGURE 22. ADAPTER CURRENT LIMIT LOOP
FN9258.3
June 8, 2011
ISL88731
The loop response equations, bode plots and the selection of
CICOMP are the same as the charge current control loop with loop
gain reduced by the duty cycle and the ratio of RS1/RS2. In other
words, if RS1 = RS2 and the duty cycle D = 50%, the loop gain will
be 6dB lower than the loop gain in Figure 22. This gives lower
crossover frequency and higher phase margin in this mode. If
RS1/RS2 = 2 and the duty cycle is 50% then the adapter current
loop gain will be identical to the gain in Figure 22.
A filter should be added between RS1 and CSIP and CSIN to
reduce switching noise. The filter roll off frequency should be
between the cross over frequency and the switching frequency
(~100kHz).
Voltage Control Loop
When the battery is charged to the voltage set by ChargeVoltage
register the voltage error amplifier (GMV) takes control of the
output (assuming that the adapter current is below the limit set
by ACLIM). The voltage error amplifier (GMV) discharges the cap
on VCOMP to limit the output voltage. The current to the battery
decreases as the cells charge to the fixed voltage and the voltage
across the internal battery resistance decreases. As battery
current decreases the 2 current error amplifiers (GMI and GMS)
output their maximum current and charge the capacitor on
ICOMP to its maximum voltage (limited to 0.3V above VCOMP).
With high voltage on ICOMP, the minimum voltage buffer output
equals the voltage on VCOMP.
The voltage control loop is shown in Figure 23.
L
PHASE
s ⎞
⎛ 1 – ------------⎝
ω ESR⎠
A LC = --------------------------------------------------------⎛ s2
⎞
s
⎜ ----------- + ----------------------- + 1⎟
ω
ω
(
⋅
Q
)
⎝ DP
⎠
LC
1
ω ESR = ----------------------------( R ESR ⋅ C o )
1
ω LC = ----------------------( L ⋅ Co )
L
Q = R o ⋅ -----Co
(EQ. 22)
The resistance RO is a combination of MOSFET rDS(ON), inductor
DCR, RSENSE and the internal resistance of the battery (normally
between 50mΩ and 200mΩ) The worst case for voltage mode
control is when the battery is absent. This results in the highest Q
of the LC filter and the lowest phase margin.
The compensation network consists of the voltage error amplifier
GMV and the compensation network RVCOMP, CVCOMP which give
the loop very high DC gain, a very low frequency pole and a zero
at FZERO1. Inductor current information is added to the feedback
to create a second zero FZERO2. The low pass filter RF2, CF2
between RS2 and ISL88731 add a pole at FFILTER. R3 and R4 are
internal divider resistors that set the DC output voltage. For a 3cell battery, R3 = 500kΩ and R4 = 100kΩ. The equations
following relate the compensation network’s poles, zeros and
gain to the components in Figure 23. Figure 25 shows an
asymptotic Bode plot of the DC/DC converter’s gain vs.
frequency. It is strongly recommended that FZERO1 is
approximately 30% of FLC and FZERO2 is approximately 70% of
FLC.
11
S
Σ
+
RF2
CSOP
20x
-
GAIN (dB)
+
0.25
-
CA2
NO BATTERY
RL_DCR
RFET_RDSON
CF2
RBATTERY
= 200mΩ
RBATTERY
= 50mΩ
RS2
CSON
VCOMP
R3
-
CO
GMV
+
R4
RESR
PHASE (°)
CVCOMP
RBAT
DACV
RVCOMP
FIGURE 23. VOLTAGE CONTROL LOOP
FREQUENCY
Output LC Filter Transfer Functions
FIGURE 24. FREQUENCY RESPONSE OF THE LC OUTPUT FILTER
The gain from the phase node to the system output and battery
depend entirely on external components. Typical output LC filter
response is shown in Figure 24. Transfer function ALC(s) is shown
in Equation 22:
19
FN9258.3
June 8, 2011
ISL88731
Compensator
1. Top Layer: signal lines, or half board for signal lines and the
other half board for power lines
Modulator
2. Signal Ground
60
40
Loop
4. Bottom Layer: Power MOSFET, Inductors and other Power
traces
LC
20
GAIN (dB)
3. Power Layers: Power Ground
FPOLE1
F
Separate the power voltage and current flowing path from the
control and logic level signal path. The controller IC will stay on
the signal layer, which is isolated by the signal ground to the
power signal traces.
0
FFILTER
-20
F
-40
ZERO1
F
Component Placement
ZERO2
F
The power MOSFET should be close to the IC so that the gate
drive signal, the LGATE, UGATE, PHASE, and BOOT, traces can be
short.
ESR
-60
0.1
1
10
FREQUENCY (kHz)
100
1000
FIGURE 25. ASYMPTOTIC BODE PLOT OF THE VOLTAGE
CONTROL LOOP GAIN
Compensation Break Frequency Equations
1
F ZERO1 = ----------------------------------------------------------------( 2π ⋅ C VCOMP ⋅ R 1COMP )
(EQ. 23)
R VCOMP
⎛
⎞ ⎛ R 4 ⎞ gm1
F ZERO2 = ⎜ -------------------------------------------⎟ ⋅ ⎜ --------------------⎟ ⋅ ⎛ ------------⎞
2π
⋅
R
⋅
C
R + R 3⎠ ⎝ 5 ⎠
⎝
SENSE
o⎠ ⎝ 4
(EQ. 24)
Place the components in such a way that the area under the IC
has less noise traces with high dv/dt and di/dt, such as gate
signals and phase node signals.
Signal Ground and Power Ground Connection
At minimum, a reasonably large area of copper, which will shield
other noise couplings through the IC, should be used as signal
ground beneath the IC. The best tie-point between the signal
ground and the power ground is at the negative side of the output
capacitor on each side, where there is little noise; a noisy trace
beneath the IC is not recommended.
GND and VCC Pin
1
F LC = -----------------------------( 2π L ⋅ C o )
(EQ. 25)
1
F FILTER = ----------------------------------------( 2π ⋅ R F2 ⋅ C F2 )
At least one high quality ceramic decoupling capacitor should be
used to cross these two pins. The decoupling capacitor can be
put close to the IC.
(EQ. 26)
LGATE Pin
1
F POLE1 = -----------------------------------------------( 2π ⋅ R SENSE ⋅ C o )
(EQ. 27)
1
F ESR = ----------------------------------------( 2π ⋅ C o ⋅ R ESR )
(EQ. 28)
Choose RVCOMP equal or lower than the value calculated from
Equation 29.
⎛ R 3 + R 4⎞
5
R VCOMP = ( 0.7 ⋅ F LC ) ⋅ ( 2π ⋅ C o ⋅ R SENSE ) ⋅ ⎛ ------------⎞ ⋅ ⎜ --------------------⎟
⎝ gm1⎠ ⎝ R
4 ⎠
(EQ. 29)
Next, choose CVCOMP equal or higher than the value calculated
from Equation 30.
1
C VCOMP = ---------------------------------------------------------------------( 0.3 ⋅ F LC ) ⋅ ( 2π ⋅ R VCOMP )
PGND Pin
PGND pin should be laid out to the negative side of the relevant
output capacitor with separate traces. The negative side of the
output capacitor must be close to the source node of the bottom
MOSFET. This trace is the return path of LGATE.
PHASE Pin
(EQ. 30)
PCB Layout Considerations
Power and Signal Layers Placement on the
PCB
As a general rule, power layers should be close together, either
on the top or bottom of the board, with signal layers on the
opposite side of the board. As an example, layer arrangement on
a 4-layer board is shown below:
20
This is the gate drive signal for the bottom MOSFET of the buck
converter. The signal going through this trace has both high dv/dt
and high di/dt, and the peak charging and discharging current is
very high. These two traces should be short, wide, and away from
other traces. There should be no other traces in parallel with
these traces on any layer.
This trace should be short, and positioned away from other weak
signal traces. This node has a very high dv/dt with a voltage
swing from the input voltage to ground. No trace should be in
parallel with it. This trace is also the return path for UGATE.
Connect this pin to the high-side MOSFET source.
UGATE Pin
This pin has a square shape waveform with high dv/dt. It
provides the gate drive current to charge and discharge the top
MOSFET with high di/dt. This trace should be wide, short, and
FN9258.3
June 8, 2011
ISL88731
away from other traces, similar to the LGATE.
DCIN Pin
BOOT Pin
This pin connects to AC-adapter output voltage, and should be
less noise sensitive.
This pin’s di/dt is as high as the UGATE; therefore, this trace
should be as short as possible.
H IG H
CU RRENT
TR AC E
SENSE
R E S IS T O R
H IG H
CU RR ENT
TRAC E
K E L V IN C O N N E C T IO N T R A C E S
T O T H E L O W P A S S F IL T E R A N D
C SO P AN D CSO N
FIGURE 26. CURRENT SENSE RESISTOR LAYOUT
CSOP, CSON, CSSP and CSSN Pins
Accurate charge current and adapter current sensing is critical
for good performance. The current sense resistor connects to the
CSON and the CSOP pins through a low pass filter with the filter
capacitor very near the IC (see Figure 2). Traces from the sense
resister should start at the pads of the sense resister and should
be routed close together, through the low pass filter and to the
CSOP and CSON pins (see Figure 26). The CSON pin is also used
as the battery voltage feedback. The traces should be routed
away from the high dv/dt and di/dt pins like PHASE, BOOT pins.
In general, the current sense resistor should be close to the IC.
These guidelines should also be followed for the adapter current
sense resister and CSSP and CSSN. Other layout arrangements
should be adjusted accordingly.
Copper Size for the Phase Node
The capacitance of PHASE should be kept very low to minimize
ringing. It would be best to limit the size of the PHASE node
copper in strict accordance with the current and thermal
management of the application.
Identify the Power and Signal Ground
The input and output capacitors of the converters, the source
terminal of the bottom switching MOSFET PGND should connect
to the power ground. The other components should connect to
signal ground. Signal and power ground are tied together at one
point.
Clamping Capacitor for Switching MOSFET
It is recommended that ceramic capacitors be used closely
connected to the drain of the high-side MOSFET, and the source
of the low-side MOSFET. This capacitor reduces the noise and the
power loss of the MOSFET.
For additional products, see www.intersil.com/product_tree
Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted
in the quality certifications found at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be
accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
21
FN9258.3
June 8, 2011
ISL88731
Package Outline Drawing
L28.5x5B
28 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 1, 10/07
4X 3.0
5.00
24X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
28
22
1
5.00
21
3 .25 ± 0 . 10
15
(4X)
7
0.15
8
14
TOP VIEW
0.10 M C A B
28X 0.55 ± 0.05
4 28X 0.25 ± 0.05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 75 ± 0.05
C
BASE PLANE
SEATING PLANE
0.08 C
( 4. 65 TYP )
( 24X 0 . 50)
(
SIDE VIEW
3. 25)
(28X 0 . 25 )
C
0 . 2 REF
5
0 . 00 MIN.
0 . 05 MAX.
( 28X 0 . 75)
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
22
FN9258.3
June 8, 2011
Similar pages