DATASHEET

ISL6597
®
Data Sheet
May 4, 2007
Dual Synchronous Rectified MOSFET
Drivers
The ISL6597 integrates two ISL6596 drivers and is
optimized to drive two independent power channels in a
synchronous-rectified buck converter topology. These
drivers, combined with an Intersil multiphase PWM
controller, form a complete high efficiency voltage regulator
solution.
The IC is biased by a single low voltage supply (5V),
minimizing driver switching losses in high MOSFET gate
capacitance and high switching frequency applications.
Each driver is capable of driving a 3nF load with less than
10ns rise/fall time. Bootstrapping of the upper gate driver is
implemented via an internal low forward drop diode,
reducing implementation cost, complexity, and allowing the
use of higher performance, cost effective N-Channel
MOSFETs. Adaptive shoot-through protection is integrated
to prevent both MOSFETs from conducting simultaneously.
The ISL6597 features 4A typical sink current for the lower
gate driver, enhancing the lower MOSFET gate hold-down
capability during PHASE node rising edge, preventing power
loss caused by the self turn-on of the lower MOSFET due to
the high dV/dt of the switching node.
The ISL6597 also features an input that recognizes a highimpedance state, working together with Intersil multi-phase
3.3V or 5V PWM controllers to prevent negative transients
on the controlled output voltage when operation is
suspended. This feature eliminates the need for the schottky
diode that may be utilized in a power system to protect the
load from negative output voltage damage.
ISL6597CRZ
Features
• 5V Quad N-Channel MOSFET Drives for Two
Synchronous Rectified Bridges
• Adaptive Shoot-Through Protection
• Programmable Deadtime for Efficiency Optimization
• 0.4Ω On-Resistance and 4A Sink Current Capability
• Supports High Switching Frequency
- Fast Output Rise and Fall
- Ultra Low Tri-State Hold-Off Time (20ns)
• Low VF Internal Bootstrap Diode
• Low Bias Supply Current
• Support 3.3V and 5V PWM Input
• Enable Input and Power-On Reset
• QFN Package
- Compliant to JEDEC PUB95 MO-220 QFN-Quad Flat
No Leads-Product Outline
- Near Chip-Scale Package Footprint; Improves PCB
Utilization and Thinner in Profile
• Pb-Free Plus Anneal Available (RoHS Compliant)
Applications
• Core Voltage Supplies for Intel® and AMD®
Microprocessors
• High Frequency Low Profile High Efficiency DC/DC
Converters
• High Current Low Voltage DC/DC Converters
• Synchronous Rectification for Isolated Power Supplies
Ordering Information
PART
NUMBER
(Note)
FN9165.1
Related Literature
PART
MARKING
TEMP.
RANGE
(°C)
PKG.
DWG. #
• Technical Brief TB389 “PCB Land Pattern Design and
Surface Mount Guidelines for QFN (MLFP) Packages”
65 97CRZ
0 to +70 16 Ld 4x4 QFN L16.4x4
• Technical Brief TB363 “Guidelines for Handling and
Processing Moisture Sensitive Surface Mount Devices
(SMDs)
PACKAGE
(Pb-Free)
Add “-T” suffix for tape and reel.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100%
matte tin plate termination finish, which are RoHS compliant and
compatible with both SnPb and Pb-free soldering operations. Intersil
Pb-free products are MSL classified at Pb-free peak reflow
temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006, 2007. All Rights Reserved
AMD® is a registered trademark of Advanced Micro Devices, Inc. All other trademarks mentioned are the property of their respective owners.
ISL6597
Pinout
PWM2
PWM1
VCC
PHASE1
ISL6597
(16 LD QFN)
TOP VIEW
16
15
14
13
GND 1
12 UGATE1
17
LGATE1 2
11 BOOT1
PGND
10 BOOT2
PVCC 3
5
6
7
8
LGATE2
VCTRL
PHASE2
9
PGND
EN 4
UGATE2
Block Diagram
ISL6597
VCC PVCC
VCTRL
BOOT1
UGATE1
SHOOTTHROUGH
PROTECTION
3.5K
PHASE1
CHANNEL 1
PVCC1
PWM1
LGATE1
3.5K
PGND
EN
CONTROL
LOGIC
VCTRL
PGND
PVCC
BOOT2
UGATE2
3.5K
PWM2
SHOOTTHROUGH
PROTECTION
3.5K
GND
PHASE2
CHANNEL 2
PVCC
LGATE2
PGND
PAD
2
FN9165.1
May 4, 2007
ISL6597
Typical Application - Multiphase Converter Using ISL6597 Gate Drivers
BOOT1
+3.3V
+12V
UGATE1
VCTRL
PHASE1
+5V
LGATE1
+3.3V
PVCC
VCC
DUAL
DRIVER
ISL6597
BOOT2
COMP
FB
VCC
VSEN
+12V
EN
UGATE2
ISEN1
PGOOD
PWM1
EN
PWM2
VID
PWM1
PHASE2
PWM2
LGATE2
MAIN ISEN2
CONTROL
ISL65xx
GND
PGND
+VCORE
ISEN3
FS/DIS
PWM3
PWM4
GND
+3.3V
BOOT1
+12V
ISEN4
UGATE1
VCTRL
PHASE1
+5V
LGATE1
PVCC
VCC
DUAL
DRIVER
ISL6597
BOOT2
+12V
EN
UGATE2
PWM1
PHASE2
PWM2
LGATE2
GND
3
PGND
FN9165.1
May 4, 2007
ISL6597
Absolute Maximum Ratings
Thermal Information
Supply Voltage (PVCC, VCC) . . . . . . . . . . . . . . . . . . . . -0.3V to 7V
Input Voltage (VEN, VPWM) . . . . . . . . . . . . . . . -0.3V to VCC + 0.3V
BOOT Voltage (VBOOT-GND). . . -0.3V to 25V (DC) or 36V (<200ns)
BOOT To PHASE Voltage (VBOOT-PHASE) . . . . . . -0.3V to 7V (DC)
-0.3V to 9V (<10ns)
PHASE Voltage . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 15V (DC)
GND -8V (<20ns Pulse Width, 10μJ) to 30V (<100ns)
UGATE Voltage . . . . . . . . . . . . . . . . VPHASE - 0.3V (DC) to VBOOT
VPHASE - 5V (<20ns Pulse Width, 10μJ) to VBOOT
LGATE Voltage . . . . . . . . . . . . . . . GND - 0.3V (DC) to VCC + 0.3V
GND - 2.5V (<20ns Pulse Width, 5μJ) to VCC + 0.3V
Ambient Temperature Range . . . . . . . . . . . . . . . . . .-40°C to +125°C
HBM ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2kV
Thermal Resistance (Notes 1 and 2)
θJA(°C/W)
θJC(°C/W)
QFN Package . . . . . . . . . . . . . . . . . .
46
8.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C
Recommended Operating Conditions
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . 0°C to +70°C
Maximum Operating Junction Temperature. . . . . . . . . . . . . +125°C
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5V ±10%
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
+150°C max junction temperature is intended for short periods of time to prevent shortening the lifetime. Constantly operated at +150°C may shorten the life of the part.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features.
2. θJC, “case temperature” location is at the center of the package underside exposed pad. See Tech Brief TB379 for details.
Electrical Specifications
These specifications apply for TA = 0°C to +70°C, unless otherwise noted
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
PWM pin floating, VVCC = VPVCC = 5V
-
350
-
μA
FPWM = 300kHz, VVCC = VPVCC = 5V
-
1.7
-
mA
POR Rising
-
3.4
4.2
V
POR Falling
2.6
3.0
-
V
-
400
-
mV
0.3
0.6
0.7
V
2.5
2.8
-
V
-
100
-
mV
EN LOW Threshold
1.00
1.34
-
V
EN HIGH Threshold
1.40
1.60
-
V
EN Hysteresis
100
260
-
mV
VCC SUPPLY CURRENT
Bias Supply Current
IVCC+PVCC
Hysteresis
BOOTSTRAP DIODE
Forward Voltage
VF
Forward bias current = 2mA
VCTRL INPUT
Turn-On Threshold
Hysteresis
ENABLE INPUT
PWM INPUT
Sinking Impedance
RPWM_SNK
-
3.5
-
kΩ
Source Impedance
RPWM_SRC
-
3.5
-
kΩ
VVCC = 3.3V (120mV Hysteresis)
-
1.15
1.4
V
VVCC = 5V (300mV Hysteresis)
-
1.55
1.75
V
Tri-State Lower Threshold
Tri-State Upper Threshold
Tri-State Shutdown Holdoff Time
VVCC = 3.3V (110mV Hysteresis)
1.65
1.85
-
V
VVCC = 5V (300mV Hysteresis)
3.00
3.18
-
V
-
80
-
ns
tTSSHD
SWITCHING TIME (Note 3, See Figure 1)
UGATE Rise Time
tRU
VVCC = 5V, 3nF Load
-
8.0
-
ns
LGATE Rise Time
tRL
VVCC = 5V, 3nF Load
-
8.0
-
ns
UGATE Fall Time
tFU
VVCC = 5V, 3nF Load
-
8.0
-
ns
4
FN9165.1
May 4, 2007
ISL6597
Electrical Specifications
These specifications apply for TA = 0°C to +70°C, unless otherwise noted (Continued)
PARAMETER
SYMBOL
MIN
TYP
MAX
UNITS
tFL
VVCC = 5V, 3nF Load
-
4.0
-
ns
UGATE Turn-Off Propagation Delay
tPDLU
VVCC = 5V, Unloaded,
-
18
-
ns
LGATE Turn-Off Propagation Delay
tPDLL
VVCC = 5V, Unloaded,
-
25
-
ns
UGATE Turn-On Propagation Delay
tPDHU
VVCC = 5V, Unloaded,
-
18
-
ns
LGATE Turn-On Propagation Delay
tPDHL
VVCC = 5V, Unloaded,
-
23
-
ns
tPTS
VVCC = 5V, Unloaded
-
30
-
ns
Upper Drive Source Resistance
RUG_SRC
250mA Source Current
-
1.0
2.5
Ω
Upper Drive Sink Resistance
RUG_SNK
250mA Sink Current
-
1.0
2.5
Ω
Lower Drive Source Resistance
RLG_SRC
250mA Source Current
-
1.0
2.5
Ω
Lower Drive Sink Resistance
RLG_SNK
250mA Sink Current
-
0.4
1.0
Ω
LGATE Fall Time
Tri-state to UG/LG Rising Propagation Delay
TEST CONDITIONS
OUTPUT (Note 3)
NOTE:
3. Guaranteed by Characterization. Not 100% tested in production.
Functional Pin Description
PACKAGE
PIN #
PIN
SYMBOL
1
GND
2
LGATE1
3
PVCC
4
EN
5
PGND
6
LGATE2
Lower gate drive output of Channel 2. Connect to gate of the low-side power N-Channel MOSFET.
7
VCTRL
This pin sets the PWM logic threshold. Connect this pin to 3.3V source for 3.3V PWM input and pull it to 5V source for
5V PWM input.
8
PHASE2
Connect this pin to the SOURCE of the upper MOSFET and the DRAIN of the lower MOSFET in Channel 2. This pin
provides a return path for the upper gate drive.
9
UGATE2
Upper gate drive output of Channel 2. Connect to gate of high-side power N-Channel MOSFET.
10
BOOT2
Floating bootstrap supply pin for the upper gate drive of Channel 2. Connect the bootstrap capacitor between this pin
and the PHASE2 pin. The bootstrap capacitor provides the charge to turn on the upper MOSFET. See “Bootstrap
Considerations” on page 7 for guidance in choosing the capacitor value.
11
BOOT1
Floating bootstrap supply pin for the upper gate drive of Channel 1. Connect the bootstrap capacitor between this pin
and the PHASE1 pin. The bootstrap capacitor provides the charge to turn on the upper MOSFET. See “Bootstrap
Considerations” on page 7 for guidance in choosing the capacitor value.
12
UGATE1
Upper gate drive output of Channel 1. Connect to gate of high-side power N-Channel MOSFET.
13
PHASE1
Connect this pin to the SOURCE of the upper MOSFET and the DRAIN of the lower MOSFET in Channel 1. This pin
provides a return path for the upper gate drive.
14
VCC
Connect this pin to a +5V bias supply. It supplies power to internal analog circuits. Place a high quality low ESR ceramic
capacitor from this pin to GND.
15
PWM1
The PWM signal is the control input for the Channel 1 driver. The PWM signal can enter three distinct states during operation,
see “Tri-State PWM Input” on page 6 for further details. Connect this pin to the PWM output of the controller.
16
PWM2
The PWM signal is the control input for the Channel 2 driver. The PWM signal can enter three distinct states during operation,
see “Tri-State PWM Input” on page 6 for further details. Connect this pin to the PWM output of the controller.
17
PAD
FUNCTION
Bias and reference ground. All signals are referenced to this node.
Lower gate drive output of Channel 1. Connect to gate of the low-side power N-Channel MOSFET.
This pin supplies power to both the lower and higher gate drives. Place a high quality low ESR ceramic capacitor from
this pin to PGND.
Enable input pin. Connect this pin high to enable and low to disable the driver.
It is the power ground return of both low gate drivers.
Connect this pad to the power ground plane (PGND) via thermally enhanced connection.
5
FN9165.1
May 4, 2007
ISL6597
Timing Diagram
2.5V
PWM
tPDHU
tPDLU
tTSSHD
tRU
tRU
tFU
tPTS
1V
UGATE
LGATE
tPTS
1V
tRL
tTSSHD
tPDHL
tPDLL
tFL
FIGURE 1. TIMING DIAGRAM
Operation and Adaptive Shoot-Through Protection
Designed for high speed switching, the ISL6597 MOSFET
driver controls both high-side and low-side N-Channel FETs
from one externally provided PWM signal.
absorb the current injected into the lower gate through the
drain-to-gate (CGD) capacitor of the lower MOSFET and
help prevent shoot through caused by the self turn-on of the
lower MOSFET due to high dV/dt of the switching node.
A rising transition on PWM initiates the turn-off of the lower
MOSFET (see Figure 1). After a short propagation delay
[tPDLL], the lower gate begins to fall. Typical fall times [tFL]
are provided in the Electrical Specifications. Adaptive shootthrough circuitry monitors the LGATE voltage and turns on
the upper gate following a short delay time [tPDHU] after the
LGATE voltage drops below ~1V. The upper gate drive then
begins to rise [tRU] and the upper MOSFET turns on.
Tri-State PWM Input
A falling transition on PWM indicates the turn-off of the upper
MOSFET and the turn-on of the lower MOSFET. A short
propagation delay [tPDLU] is encountered before the upper
gate begins to fall [tFU]. The adaptive shoot-through circuitry
monitors the UGATE-PHASE voltage and turns on the lower
MOSFET a short delay time, tPDHL, after the upper
MOSFET’s gate voltage drops below 1V. The lower gate
then rises [tRL], turning on the lower MOSFET. These
methods prevent both the lower and upper MOSFETs from
conducting simultaneously (shoot-through), while adapting
the dead time to the gate charge characteristics of the
MOSFETs being used.
The ISL6597 also features the adaptable tri-state PWM
input. Once the PWM signal enters the shutdown window,
either MOSFET previously conducting is turned off. If the
PWM signal remains within the shutdown window for longer
than the gate turn-off propagation delay of the previously
conducting MOSFET, the output drivers are disabled and
both MOSFET gates are pulled and held low. The shutdown
state is removed when the PWM signal moves outside the
shutdown window. The PWM rising and falling thresholds
outlined in the Electrical Specifications determine when the
lower and upper gates are enabled. During normal operation
in a typical application, the PWM rise and fall times through
the shutdown window should not exceed either output’s turnoff propagation delay plus the MOSFET gate discharge time
to ~1V. Abnormally long PWM signal transition times through
the shutdown window will simply introduce additional dead
time between turn off and turn on of the synchronous
bridge’s MOSFETs. For optimal performance, no more than
50pF parasitic capacitive load should be present on the
This driver is optimized for voltage regulators with large step
down ratio. The lower MOSFET is usually sized larger
compared to the upper MOSFET because the lower
MOSFET conducts for a longer time during a switching
period. The lower gate driver is therefore sized much larger
to meet this application requirement. The 0.4Ω on-resistance
and 4A sink current capability enable the lower gate driver to
6
A unique feature of the ISL6597 is the programmable PWM
logic threshold set by the control pin (VCTRL) voltage. The
VCTRL pin should connect to the controller’s VCC so that
the PWM logic thresholds follow with the VCC voltage level.
For applications using single rail 5V to power up both
controller and driver, this pin can be tied to the driver VCC,
simplifying the trace routing.
FN9165.1
May 4, 2007
ISL6597
PWM line of ISL6597 (assuming an Intersil PWM controller
is used).
Bootstrap Considerations
This driver features an internal bootstrap diode. Simply
adding an external capacitor across the BOOT and PHASE
pins completes the bootstrap circuit.
The following equation helps select a proper bootstrap
capacitor size:
Q GATE
C BOOT_CAP ≥ -------------------------------------ΔV BOOT_CAP
(EQ. 1)
Q G1 • PVCC
Q GATE = ------------------------------------ • N Q1
V GS1
P Qg_TOT = 2 • ( P Qg_Q1 + P Qg_Q2 ) + I Q • VCC
where QG1 is the amount of gate charge per upper MOSFET
at VGS1 gate-source voltage and NQ1 is the number of
control MOSFETs. The ΔVBOOT_CAP term is defined as the
allowable droop in the rail of the upper gate drive.
As an example, suppose two HAT2168 FETs are chosen as
the upper MOSFETs. The gate charge (QG) from the data
sheet is 12nC at 5V (VGS) gate-source voltage. Then the
QGATE is calculated to be 26.4nC at 5.5V PVCC level. We
will assume a 100mV droop in drive voltage over the PWM
cycle. We find that a bootstrap capacitance of at least
0.264μF is required. The next larger standard value
capacitance is 0.33µF. A good quality ceramic capacitor is
recommended.
2.0
1.8
1.6
CBOOT_CAP (µF)
1.2
1.0
0.8
0.6
QGATE = 100nC
0.4
50nC
20nC
0.0
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
Q G1 • PVCC 2
P Qg_Q1 = --------------------------------------- • F SW • N Q1
V GS1
Q G2 • PVCC 2
P Qg_Q2 = --------------------------------------- • F SW • N Q2
V GS2
⎛ Q G1 • N Q1 Q G2 • N Q2⎞
I DR = 2 • ⎜ ------------------------------ + ------------------------------⎟ • F SW + I Q
V GS2 ⎠
⎝ V GS1
(EQ. 3)
where the gate charge (QG1 and QG2) is defined at a
particular gate to source voltage (VGS1and VGS2) in the
corresponding MOSFET datasheet; IQ is the driver’s total
quiescent current with no load at both drive outputs; NQ1
and NQ2 are number of upper and lower MOSFETs,
respectively. The factor 2 is the number of active channels.
The IQ VCC product is the quiescent power of the driver
without capacitive load and is typically negligible.
1.0
ΔVBOOT (V)
FIGURE 2. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE
VOLTAGE
Power Dissipation
Package power dissipation is mainly a function of the
switching frequency (FSW), the output drive impedance, the
external gate resistance, and the selected MOSFET’s
internal gate resistance and total gate charge. Calculating
the power dissipation in the driver for a desired application is
critical to ensure safe operation. Exceeding the maximum
7
(EQ. 2)
The total gate drive power losses are dissipated among the
resistive components along the transition path. The drive
resistance dissipates a portion of the total gate drive power
losses, the rest will be dissipated by the external gate
resistors (RG1 and RG2, should be a short to avoid
interfering with the operation shoot-through protection
circuitry) and the internal gate resistors (RGI1 and RGI2) of
MOSFETs. Figures 3 and 4 show the typical upper and lower
gate drives turn-on transition path. The power dissipation on
the driver can be roughly estimated as:
1.4
0.2
allowable power dissipation level will push the IC beyond the
maximum recommended operating junction temperature of
+125°C. The maximum allowable IC power dissipation for
the 16 lead 4x4 QFN packages, with an exposed heat
escape pad, is around 2W. See Layout Considerations
paragraph for thermal transfer improvement suggestions.
When designing the driver into an application, it is
recommended that the following calculation is used to
ensure safe operation at the desired frequency for the
selected MOSFETs. The total gate drive power losses due to
the gate charge of MOSFETs and the driver’s internal
circuitry and their corresponding average driver current can
be estimated with Equations 2 and 3, respectively,
P DR = 2 • ( P DR_UP + P DR_LOW ) + I Q • VCC
(EQ. 4)
R HI1
R LO1
⎛
⎞ P Qg_Q1
P DR_UP = ⎜ -------------------------------------- + ----------------------------------------⎟ • --------------------R
+
R
R
+
R
2
⎝ HI1
EXT1
LO1
EXT1⎠
R HI2
R LO2
⎛
⎞ P Qg_Q2
P DR_LOW = ⎜ -------------------------------------- + ----------------------------------------⎟ • --------------------2
⎝ R HI2 + R EXT2 R LO2 + R EXT2⎠
R GI1
R EXT2 = R G1 + ------------N
Q1
R GI2
R EXT2 = R G2 + ------------N
Q2
FN9165.1
May 4, 2007
ISL6597
PVCC
overcharging, exceeding the device rating. Low-profile
MOSFETs, such as Direct FETs and multi-SOURCE leads
devices (SO-8, LFPAK, PowerPAK), have low parasitic lead
inductances and are preferred.
BOOT
D
CGD
RHI1
RLO1
G
UGATE
RG1
CDS
RGI1
CGS
Q1
S
PHASE
FIGURE 3. TYPICAL UPPER-GATE DRIVE TURN-ON PATH
PVCC
D
CGD
RHI2
LGATE
RLO2
G
CDS
RGI2
RG2
CGS
Q2
S
GND
FIGURE 4. TYPICAL LOWER-GATE DRIVE TURN-ON PATH
MOSFET Selection
The parasitic inductances of the PCB and of the power
devices’ packaging (both upper and lower MOSFETs) can
cause serious ringing, exceeding absolute maximum rating
of the devices. The negative ringing at the edges of the
PHASE node could increase the bootstrap capacitor voltage
through the internal bootstrap diode, and in some cases, it
may overstress the upper MOSFET driver. Careful layout,
proper selection of MOSFETs and packaging can go a long
way toward minimizing such unwanted stress.
BOOT
D
RHI1
RLO1
G
Q1
UGATE
S
PHASE
RPH = 1Ω to 2Ω
FIGURE 5. PHASE RESISTOR TO MINIMIZE SERIOUS
NEGATIVE PHASE SPIKE
The D2-PAK, or D-PAK packaged MOSFETs, have large
parasitic lead inductances and are not recommended unless
a phase resistor (RPH), as shown in Figure 5, is
implemented to prevent the bootstrap capacitor from
8
A good layout helps reduce the ringing on the switching
node (PHASE) and significantly lower the stress applied to
the output drives. The following advice is meant to lead to an
optimized layout and performance:
• Keep decoupling loops (VCC-GND, PVCC-PGND and
BOOT-PHASE) short and wide, at least 25 mils. Avoid
using vias on decoupling components other than their
ground terminals, which should be on a copper plane with
at least two vias.
• Minimize trace inductance, especially on low-impedance
lines. All power traces (UGATE, PHASE, LGATE, PGND,
PVCC, VCC, GND) should be short and wide, at least 25
mils. Try to place power traces on a single layer,
otherwise, two vias on interconnection are preferred
where possible. For no connection (NC) pins on the QFN
part, connect it to the adjacent net (LGATE2/PHASE2) can
reduce trace inductance.
• Shorten all gate drive loops (UGATE-PHASE and LGATEPGND) and route them closely spaced.
• Minimize the inductance of the PHASE node. Ideally, the
source of the upper and the drain of the lower MOSFET
should be as close as thermally allowable.
Application Information
PVCC
Layout Considerations
• Minimize the current loop of the output and input power
trains. Short the source connection of the lower MOSFET
to ground as close to the transistor pin as feasible. Input
capacitors (especially ceramic decoupling) should be
placed as close to the drain of upper and source of lower
MOSFETs as possible.
• Avoid routing relatively high impedance nodes (such as
PWM and ENABLE lines) close to high dV/dt UGATE and
PHASE nodes.
In addition, connecting the thermal pad of the QFN package
to the power ground through multiple vias is recommended.
This is to improve heat dissipation and allow the part to
achieve its full thermal potential.
Upper MOSFET Self Turn-On Effects At Startup
Should the driver have insufficient bias voltage applied, its
outputs are floating. If the input bus is energized at a high
dV/dt rate while the driver outputs are floating, due to the
self-coupling via the internal CGD of the MOSFET, the
UGATE could momentarily rise up to a level greater than the
threshold voltage of the MOSFET. This could potentially turn
on the upper switch and result in damaging inrush energy.
Therefore, if such a situation (when input bus powered up
before the bias of the controller and driver is ready) could
conceivably be encountered, it is a common practice to
place a resistor (RUGPH) across the gate and source of the
FN9165.1
May 4, 2007
ISL6597
R = R UGPH + R GI
VCC
(EQ. 5)
C iss = C GD + C GS
C rss = C GD
VIN
BOOT
D
CBOOT
CGD
DU
DL
UGATE
RUGPH
The coupling effect can be roughly estimated with the
following equations, which assume a fixed linear input ramp
and neglect the clamping effect of the body diode of the
upper drive and the bootstrap capacitor. Other parasitic
components such as lead inductances and PCB
capacitances are also not taken into account. These
equations are provided for guidance purpose only.
Therefore, the actual coupling effect should be examined
using a very high impedance (10MΩ or greater) probe to
ensure a safe design margin.
–V
DS
⎛
----------------------------------⎞
dV
⎜
-----⋅
R
⋅ C iss⎟
dV
⎟
V GS_MILLER = ------- ⋅ R ⋅ C rss ⎜ 1 – e dt
⎜
⎟
dt
⎜
⎟
⎝
⎠
ISL6597
upper MOSFET to suppress the Miller coupling effect. The
value of the resistor depends mainly on the input voltage’s
rate of rise, the CGD/CGS ratio, as well as the gate-source
threshold of the upper MOSFET. A higher dV/dt, a lower
CDS/CGS ratio, and a lower gate-source threshold upper
FET will require a smaller resistor to diminish the effect of
the internal capacitive coupling. For most applications, the
integrated 20kΩ typically sufficient, not affecting normal
performance and efficiency.
G
CDS
RGI
CGS
QUPPER
S
PHASE
FIGURE 6. GATE TO SOURCE RESISTOR TO REDUCE
UPPER MOSFET MILLER COUPLING
9
FN9165.1
May 4, 2007
ISL6597
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L16.4x4
16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220-VGGC ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
A3
b
0.23
D
0.28
9
0.35
5, 8
4.00 BSC
D1
D2
9
0.20 REF
-
3.75 BSC
1.95
2.10
9
2.25
7, 8
E
4.00 BSC
-
E1
3.75 BSC
9
E2
1.95
e
2.10
2.25
7, 8
0.65 BSC
-
k
0.25
-
-
-
L
0.50
0.60
0.75
8
L1
-
-
0.15
10
N
16
2
Nd
4
3
Ne
4
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 5 5/04
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
10
FN9165.1
May 4, 2007
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