DATASHEET

ISL6520B
®
Data Sheet
July 23, 2007
Single Synchronous Buck Pulse-Width
Modulation (PWM) Controller
The ISL6520B makes simple work out of implementing a
complete control scheme for a DC/DC stepdown converter.
Designed to drive N-channel MOSFETs in a synchronous
buck topology, the ISL6520B integrates the control, output
adjustment and monitoring functions into a single 8 Lead
package.
The ISL6520B provides simple, single feedback loop,
voltage-mode control with fast transient response. The
output voltage can be precisely regulated to as low as 0.8V,
with a maximum tolerance of ±1.5% over-temperature and
line voltage variations. A fixed frequency oscillator reduces
design complexity, while balancing typical application cost
and efficiency.
The error amplifier features a 15MHz gain-bandwidth
product and 8V/μs slew rate which enables high converter
bandwidth for fast transient performance. The resulting
PWM duty cycles range from 0% to 100%.
Ordering Information
PART
MARKING
TEMP.
RANGE
(°C)
6520 BCB
0 to +70 8 Ld SOIC
M8.15
ISL6520BCBZ* 6520 BCBZ 0 to +70 8 Ld SOIC
(Note)
(Pb-free)
M8.15
PART
NUMBER
ISL6520BCB*
ISL6520BCR*
65 20BCR
PACKAGE
PKG.
DWG. #
0 to +70 16 Ld 4x4 QFN L16.4x4
ISL6520BCRZ* 65 20BCRZ 0 to +70 16 Ld 4x4 QFN L16.4x4
(Note)
(Pb-free)
ISL6520BIR*
65 20BIR
-40 to +85 16 Ld 4x4 QFN L16.4x4
ISL6520BIRZ* 65 20BIRZ -40 to +85 16 Ld 4x4 QFN L16.4x4
(Note)
(Pb-free)
ISL6520EVAL1 Evaluation Board
*Add “-T” suffix for tape and reel. Please refer to TB347 for details
on reel specifications.
NOTE: Intersil Pb-free products employ special Pb-free material
sets; molding compounds/die attach materials and 100% matte tin
plate termination finish, which are RoHS compliant and compatible
with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
FN9083.3
Features
• Operates from +5V Input
• 0.8V to VIN Output Range
- 0.8V Internal Reference
- ±1.5% Over Line Voltage and Temperature
• Drives N-Channel MOSFETs
• Simple Single-Loop Control Design
- Voltage-Mode PWM Control
• Fast Transient Response
- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Cycle
• Small Converter Size
- 300kHz Fixed Frequency Oscillator
- Internal Soft Start
- 8 Ld SOIC or 16Ld 4mmx4mm QFN
• QFN Package:
- Compliant to JEDEC PUB95 MO-220 QFN - Quad Flat
No Leads - Package Outline
- Near Chip Scale Package footprint, which improves
PCB efficiency and has a thinner profile
• Pb-free Plus Anneal Available (RoHS compliant)
Applications
• Power Supplies for Microprocessors
- PCs
- Embedded Controllers
• Subsystem Power Supplies
- PCI/AGP/GTL+ Buses
- ACPI Power Control
- SSTL-2 and DDR SDRAM Bus Termination Supply
• Cable Modems, Set-Top Boxes, and DSL Modems
• DSP and Core Communications Processor Supplies
• Memory Supplies
• Personal Computer Peripherals
• Industrial Power Supplies
• 5V-Input DC/DC Regulators
• Low-Voltage Distributed Power Supplies
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2003, 2005, 2007. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6520B
Pinouts
ISL6520B
(16 LD QFN)
TOP VIEW
NC
16
15
14
13
2
11 COMP/SD
GND
3
10 NC
NC
4
9
5
6
7
8
NC
UGATE
VCC
12 NC
NC
1
LGATE
BOOT
5 VCC
LGATE 4
PHASE
7 COMP/SD
6 FB
GND 3
NC
8 PHASE
BOOT 1
UGATE 2
NC
ISL6520B
(8 LD SOIC)
TOP VIEW
FB
Block Diagram
VCC
POR AND
SOFTSTART
BOOT
UGATE
+
PWM
COMPARATOR
ERROR
AMP
+
0.8V
-
INHIBIT
PHASE
GATE
CONTROL
LOGIC
PWM
+
-
-
VCC
FB
LGATE
COMP/SD
20μA
OSCILLATOR
FIXED 300kHz
GND
Typical Application
VIN
5V
CDCPL
DBOOT
RPULLUP
VCC
5
COMP/SD
SHUTDOWN
7
1
2
ISL6520B
8
RF
CI
6
CF
FB
3
4
CBULK
CHF
BOOT
CBOOT
QU
UGATE
LOUT
PHASE
QL
LGATE
VOUT
COUT
GND
ROFFSET
RS
2
FN9083.3
July 23, 2007
ISL6520B
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6.0V
Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . +15.0V
Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . 7.0V (DC)
8.0V (<10ns Pulse Width, 10μJ)
Input, Output or I/O Voltage . . . . . . . . . . . GND -0.3V to VCC +0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance
θJA (°C/W)
θJC (°C/W)
SOIC Package (Note 1) . . . . . . . . . . . .
95
N/A
QFN Package (Notes 2, 3). . . . . . . . . .
45
7
Maximum Junction Temperature
(Plastic Package) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . -65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
Ambient Temperature Range - ISL6520BC . . . . . . . . . 0°C to +70°C
Junction Temperature Range. . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
2. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
3. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
IVCC
2.6
3.2
3.8
mA
POR
4.19
4.30
4.50
V
-
0.25
-
V
ISL6520BC, VCC = 5V
250
300
340
kHz
ISL6520BI, VCC = 5V
230
300
340
kHz
-
1.5
-
VP-P
ISL6520BC
-1.5
-
+1.5
%
ISL6520BI
-2.5
+2.5
%
VCC SUPPLY CURRENT
Nominal Supply
POWER-ON RESET
Rising VCC POR Threshold
VCC POR Threshold Hysteresis
OSCILLATOR
Frequency
fOSC
ΔVOSC
Ramp Amplitude
REFERENCE
Reference Voltage Tolerance
Nominal Reference Voltage
VREF
-
0.800
-
V
(Note 4)
-
88
-
dB
GBWP
(Note 4)
-
15
-
MHz
SR
(Note 4)
-
8
-
V/μs
ERROR AMPLIFIER
DC Gain
Gain-Bandwidth Product
Slew Rate
GATE DRIVERS
Upper Gate Source Current
IUGATE-SRC VBOOT - VPHASE = 5V, VUGATE = 4V
-
-1
-
A
Upper Gate Sink Current
IUGATE-SNK
-
1
-
A
Lower Gate Source Current
ILGATE-SRC VVCC = 5V, VLGATE = 4V
-
-1
-
A
Lower Gate Sink Current
ILGATE-SNK
-
2
-
A
VDISABLE
-
0.8
-
V
DISABLE
Disable Threshold
NOTE:
4. Limits should be considered typical and are not production tested
3
FN9083.3
July 23, 2007
ISL6520B
Functional Pin Description
Functional Description
VCC
Initialization
This pin provides the bias supply for the ISL6520B, as well
as the lower MOSFET’s gate. Connect a well-decoupled 5V
supply to this pin.
The ISL6520B automatically initializes upon receipt of power.
The Power-On Reset (POR) function continually monitors the
bias voltage at the VCC pin. The POR function initiates the soft
start operation.
FB
This pin is the inverting input of the internal error amplifier.
Use this pin, in combination with the COMP/SD pin, to
compensate the voltage-control feedback loop of the
converter.
GND
This pin represents the signal and power ground for the IC.
Tie this pin to the ground island/plane through the lowest
impedance connection available.
PHASE
Connect this pin to the upper MOSFET’s source.
UGATE
Connect this pin to the upper MOSFET’s gate. This pin
provides the PWM-controlled gate drive for the upper
MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the upper
MOSFET has turned off.
BOOT
This pin provides ground referenced bias voltage to the
upper MOSFET driver. A bootstrap circuit is used to create a
voltage suitable to drive a logic-level N-channel MOSFET.
Soft Start
The ISL6520B is held in reset with both UGATE and LGATE
driven to ground until the POR threshold on VCC has been
reached and the COMP/SD pin has been pulled above 0.8V. If
COMP is not actively pulled high following POR the internal
20μA current sink will hold COMP/SD low and the device will
remain in reset. COMP/SD can either be statically tied to VCC
through a pullup resistor or driven high through a resistor to
terminate reset. The recommended range of resistor values to
use as the pullup resistor is between 50kΩ and 100kΩ.
Following reset the ISL6520B provides a 1024 clock cycle
settling period (~3.4ms) prior to initiating softstart. At the
conclusion of the settling period the COMP/SD pin is driven
to 0.8V for 24 clock cycles (~75μs) to discharge the
compensation network. Soft start of the regulated output is
generated by imposing an internal offset on the FB pin which
ramps down from 0.8V to 0V over the next 2048 clock cycles
(~6.8ms). Total time from end of reset to completion of soft-start
is 10.2ms.
Pulling COMP/SD below 0.8V or VCC dropping below
minimum POR initiates another reset.
COMP/SD
This pin is the output of the error amplifier. Use this pin, in
combination with the FB pin, to compensate the voltagecontrol feedback loop of the converter.
Pulling COMP/SD to a level below 0.8V disables the
controller. Disabling the ISL6520B causes the oscillator to
stop, the LGATE and UGATE outputs to be held low, and the
softstart circuitry to re-arm. The COMP/SD pin must be
pulled above 0.8V to terminate shutdown. This may be done
through a pullup resistor tied between VCC and COMP/SD.
The recommended range of resistor values to use as the pullup
resistor is between 50kΩ and 100kΩ.
LGATE
VOUT
500mV/DIV.
VCOMP/SD
1V/DIV.
TIME (2ms/DIV.)
FIGURE 1. SOFT START INTERVAL
Connect this pin to the lower MOSFET’s gate. This pin
provides the PWM-controlled gate drive for the lower
MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the lower
MOSFET has turned off.
Current Sinking
The ISL6520B incorporates a MOSFET shoot-through
protection method which allows a converter to sink current
as well as source current. Care should be exercised when
designing a converter with the ISL6520B when it is known
that the converter may sink current.
When the converter is sinking current, it is behaving as a
boost converter that is regulating it’s input voltage. This
means that the converter is boosting current into the VCC
4
FN9083.3
July 23, 2007
ISL6520B
as close as practical to the BOOT and PHASE pins. All
components used for feedback compensation should be
located as close to the IC a practical.
BOOT
+VIN
D1
Q1
CBOOT
ISL6520B
LO
VOUT
PHASE
VCC
+5V
Q2
LOAD
rail, which supplies the bias voltage to the ISL6520B. If there
is nowhere for this current to go, such as to other distributed
loads on the VCC rail, through a voltage limiting protection
device, or other methods, the capacitance on the VCC bus
will absorb the current. This situation will allow voltage level
of the VCC rail to increase. If the voltage level of the rail is
boosted to a level that exceeds the maximum voltage rating
of the ISL6520B, then the IC will experience an irreversible
failure and the converter will no longer be operational.
Ensuring that there is a path for the current to follow other
than the capacitance on the rail will prevent this failure
mode.
CO
CVCC
GND
Application Guidelines
Layout Considerations
As in any high frequency switching converter, layout is very
important. Switching current from one power device to another
can generate voltage transients across the impedances of the
interconnecting bond wires and circuit traces. These
interconnecting impedances should be minimized by using
wide, short printed circuit traces. The critical components
should be located as close together as possible, using ground
plane construction or single point grounding.
VIN
ISL6520B
LO
PHASE
LGATE
Feedback Compensation
Figure 4 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The
error amplifier (Error Amp) output (VE/A) is compared with
the oscillator (OSC) triangular wave to provide a
pulse-width modulated (PWM) wave with an amplitude of
VIN at the PHASE node. The PWM wave is smoothed by the
output filter (LO and CO).
PWM
COMPARATOR
VOUT
CIN
Q2
CO
VIN
DRIVER
OSC
Q1
LOAD
UGATE
FIGURE 3. PRINTED CIRCUIT BOARD SMALL SIGNAL
LAYOUT GUIDELINES
LO
-
ΔVOSC
DRIVER
+
PHASE
RETURN
Figure 2 shows the critical power components of the converter.
To minimize the voltage overshoot, the interconnecting wires
indicated by heavy lines should be part of a ground or power
plane in a printed circuit board. The components shown in
Figure 2 should be located as close together as possible.
Please note that the capacitors CIN and CO may each
represent numerous physical capacitors. Locate the ISL6520B
within 3 inches of the MOSFETs, Q1 and Q2 . The circuit traces
for the MOSFETs’ gate and source connections from the
ISL6520B must be sized to handle up to 1A peak current.
Figure 3 shows the circuit traces that require additional
layout consideration. Use single point and ground plane
construction for the circuits shown. Minimize any leakage
current paths on the COMP/SD pin and locate the resistor,
ROSCET close to the COMP/SD pin because the internal
current source is only 20μA. Provide local VCC decoupling
between VCC and GND pins. Locate the capacitor, CBOOT
5
CO
ESR
(PARASITIC)
ZFB
FIGURE 2. PRINTED CIRCUIT BOARD POWER AND
GROUND PLANES OR ISLANDS
VOUT
VE/A
ZIN
-
+
ERROR
AMP
REFERENCE
DETAILED COMPENSATION COMPONENTS
ZFB
C2
C1
VOUT
ZIN
C3
R2
R1
COMP/SD
-
R3
FB
+
ISL6520B
REFERENCE
FIGURE 4. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
FN9083.3
July 23, 2007
ISL6520B
Modulator Break Frequency Equations
1
f LC = ------------------------------------------2π x L O x C O
1
f ESR = -------------------------------------------2π x ESR x C O
100
40
20
The compensation network consists of the error amplifier
(internal to the ISL6520B) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 4. Use these guidelines for
locating the poles and zeros of the compensation network:
-40
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
5. Place 2ND Pole at Half the Switching Frequency.
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
1
F Z1 = -----------------------------------2π x R 2 x C 1
1
F P1 = --------------------------------------------------------⎛ C 1 x C 2⎞
2π x R 2 x ⎜ ----------------------⎟
⎝ C1 + C2 ⎠
1
F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3
1
F P2 = -----------------------------------2π x R 3 x C 3
(EQ. 2)
Figure 5 shows an asymptotic plot of the DC/DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak due to the high Q factor of the output filter and is not
shown in Figure 5. Using the above guidelines should give a
Compensation Gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
Check the compensation gain at FP2 with the capabilities of
the error amplifier. The Closed Loop Gain is constructed on
the graph of Figure 5 by adding the Modulator Gain (in dB) to
the Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
compensation transfer function and plotting the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
6
FP2
OPEN LOOP
ERROR AMP GAIN
20LOG
(R2/R1)
20LOG
(VIN/ΔVOSC)
0
-20
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
FP1
60
(EQ. 1)
1. Pick Gain (R2/R1) for desired converter bandwidth.
FZ1 FZ2
80
GAIN (dB)
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ΔVOSC .
COMPENSATION
GAIN
MODULATOR
GAIN
CLOSED LOOP
GAIN
FLC
-60
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 5. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
Modern components and loads are capable of producing
transient load rates above 1A/ns. High frequency capacitors
initially supply the transient and slow the current load rate
seen by the bulk capacitors. The bulk filter capacitor values
are generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
FN9083.3
July 23, 2007
ISL6520B
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
ΔI =
VIN - VOUT
Fs x L
x
VOUT
ΔVOUT = ΔI x ESR
VIN
(EQ. 3)
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6520B will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
(EQ. 4)
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q1 turns on. Place the
small ceramic capacitors physically close to the MOSFETs
and between the drain of Q1 and the source of Q2 .
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
7
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.
For a through hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge currentrating. These capacitors
must be capable of handling the surge-current at power-up.
Some capacitor series available from reputable manufacturers
are surge current tested.
MOSFET Selection/Considerations
The ISL6520B requires 2 N-Channel power MOSFETs. These
should be selected based upon rDS(ON) , gate supply
requirements, and thermal management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
switching losses seen when sinking current. When sourcing
current, the upper MOSFET realizes most of the switching
losses. The lower switch realizes most of the switching
losses when the converter is sinking current (see the
equations below). These equations assume linear voltagecurrent transitions and do not adequately model power loss
due the reverse-recovery of the upper and lower MOSFET’s
body diode. The gate-charge losses are dissipated by the
ISL6520B and don't heat the MOSFETs. However, large gatecharge increases the switching interval, tSW which increases
the MOSFET switching losses. Ensure that both MOSFETs
are within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications. A separate heatsink
may be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
Losses while Sourcing Current
2
1
P UPPER = Io × r DS ( ON ) × D + --- ⋅ Io × V IN × t SW × F S
2
PLOWER = Io2 x rDS(ON) x (1 - D)
Losses while Sinking Current
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io × r DS ( ON ) × ( 1 – D ) + --- ⋅ Io × V IN × t SW × F S
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
FS is the switching frequency.
(EQ. 5)
Given the reduced available gate bias voltage (5V),
logic-level or sub-logic-level transistors should be used for
both N-MOSFETs. Caution should be exercised with devices
exhibiting very low VGS(ON) characteristics. The shoot-
FN9083.3
July 23, 2007
ISL6520B
Figure 6 shows the upper gate drive (BOOT pin) supplied by a
bootstrap circuit from VCC . The boot capacitor, CBOOT,
develops a floating supply voltage referenced to the PHASE
pin. The supply is refreshed to a voltage of VCC less the boot
diode drop (VD) each time the lower MOSFET, Q2 , turns on.
through protection present aboard the ISL6520B may be
circumvented by these MOSFETs if they have large parasitic
impedences and/or capacitances that would inhibit the gate
of the MOSFET from being discharged below it’s threshold
level before the complementary MOSFET is turned on.
+5V
DBOOT
VCC
ISL6520B DC/DC Converter Application
Circuit
+5V
+ VD -
Figure 7 shows an application circuit of a DC/DC Converter.
Detailed information on the circuit, including a complete Billof-Materials and circuit board description, can be found in
Application Note AN9932.
BOOT
CBOOT
ISL6520B
Q1
UGATE
PHASE
-
NOTE:
VG-S ≈ VCC -VD
Q2
LGATE
+
NOTE:
VG-S ≈ VCC
GND
FIGURE 6. UPPER GATE DRIVE BOOTSTRAP
+5V
+
CIN
2 x 330μF
0.1μF
2 x 1μF
VCC
ISL6520B
50kΩ
5
D1
POR
AND
SOFT START
1
2 UGATE
COMP/SD 7
REF
8 PHASE
10.0kΩ
0.1μF
Q1
L1
+
470pF
-
8200pF
4
+
-
FB 6
OSC
1.00kΩ
BOOT
U1
VOUT
LGATE
Q2
3
+
COUT
3 x 330μF
0.1μF
GND
3.16kΩ
60.4Ω
18000pF
Component Selection Notes:
CIN - Each 330mF 6.3WVDC, Sanyo 6TPB330M or Equivalent.
COUT - Each 330mF 6.3WVDC, Sanyo 6TPB330M or Equivalent.
D1 - 30mA Schottky Diode, MA732 or Equivalent
L1 - 3.1μH Inductor, Panasonic P/N ETQ-P6F2ROLFA or Equivalent.
Q1 , Q2 - Fairchild MOSFET; HUF76143.
FIGURE 7. 5V TO 3.3V 15A DC/DC CONVERTER
8
FN9083.3
July 23, 2007
ISL6520B
Small Outline Plastic Packages (SOIC)
M8.15 (JEDEC MS-012-AA ISSUE C)
N
8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE
INDEX
AREA
0.25(0.010) M
H
B M
INCHES
E
SYMBOL
-B-
1
2
3
L
SEATING PLANE
-A-
A
D
h x 45°
-C-
e
A1
B
0.25(0.010) M
C
0.10(0.004)
C A M
MIN
MAX
MIN
MAX
NOTES
A
0.0532
0.0688
1.35
1.75
-
A1
0.0040
0.0098
0.10
0.25
-
B
0.013
0.020
0.33
0.51
9
C
0.0075
0.0098
0.19
0.25
-
D
0.1890
0.1968
4.80
5.00
3
E
0.1497
0.1574
3.80
4.00
4
e
α
B S
0.050 BSC
-
0.2284
0.2440
5.80
6.20
-
h
0.0099
0.0196
0.25
0.50
5
L
0.016
0.050
0.40
1.27
6
α
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
1.27 BSC
H
N
NOTES:
MILLIMETERS
8
0°
8
8°
0°
7
8°
Rev. 1 6/05
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact.
9
FN9083.3
July 23, 2007
ISL6520B
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L16.4x4
16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220-VGGC ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
A3
b
0.23
D
0.28
9
0.35
5, 8
4.00 BSC
D1
D2
9
0.20 REF
-
3.75 BSC
1.95
2.10
9
2.25
7, 8
E
4.00 BSC
-
E1
3.75 BSC
9
E2
1.95
e
2.10
2.25
7, 8
0.65 BSC
-
k
0.25
-
-
-
L
0.50
0.60
0.75
8
L1
-
-
0.15
10
N
16
2
Nd
4
3
Ne
4
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 5 5/04
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
10
FN9083.3
July 23, 2007