dm00207043

AN4720
Application note
Half bridge resonant LLC converters and
primary side MOSFET selection
Alfio Scuto
Introduction
These days, vast amounts of data are stored and shared by innumerable users around the world,
leading to the vertiginous increase of data centers that house clusters of servers, storage devices,
networks and telecommunications systems servicing users around the world.
The conflicting requirement of limiting or reducing total carbon emissions deriving from the enormous
corresponding energy demand imposes the need to constantly improve the efficiency of the associated
electronic systems.
The power supplies that feed all of these systems are pivotal in this respect and are required to satisfy
the following requirements:
•
•
•
higher efficiency
higher power density
higher component density
Among several types of switched-mode power supplies, resonant power converters with LLC half-bridge
configurations are receiving a lot of interest because of their intrinsic capacity to lower switching losses
while increasing switching frequencies.
LLC resonant converters convert power with frequency modulation and Zero Voltage Switching (ZVS)
for power MOSFETs.
They require switching frequencies as close as possible to the resonant frequency, as any deviation
increases the current circulating in the resonant tank which translates into increased electrical losses,
compromising the advantages of this topology.
In addition, current and/or voltage peaks can further stress the MOSFET devices when the circuit works
in resonance.
The challenge, therefore, is to select the right power MOSFETs to simultaneously satisfy low switching
loss and high reliability requirements.
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www.st.com
Contents
AN4720
Contents
1
LLC resonant half-bridge converters: topology and
characteristics.......................................................................................... 4
2
Circuit analysis and operation ....................................................... 5
3
The role of MOSFETs in achieving the ZVS condition .................. 8
4
MOSFET output capacitance (COSS) ............................................... 9
4.1
Q for ZVS at full load and min Vin ................................................... 10
4.2
Q for ZVS at no load and max Vin................................................... 10
5
MOSFET body diodes ................................................................... 11
6
Power MOSFET failure mechanisms............................................ 12
6.1
Failure mode 1: static dv/dt associated with the parasitic BJT ........ 12
6.2
Failure mode 2: reverse recovery dv/dt ........................................... 12
6.2.1
7
Body diode reverse recovery ............................................................ 15
Non-standard operations in LLC resonance HB ......................... 17
7.1
Capacitive region operations ........................................................... 17
7.2
Hard switching at start-up ............................................................... 19
7.3
Hard switching caused by supply disconnection ............................. 20
8
Hard switching caused by fast load transition ............................ 22
9
Conclusions ................................................................................... 23
10
References ..................................................................................... 24
11
Revision history ............................................................................ 25
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List of figures
List of figures
Figure 1: Basic LLC resonant half-bridge converter ................................................................................... 4
Figure 2: Fundamental Harmonic Analysis (FHA) ...................................................................................... 5
Figure 3: AC equivalent circuit for the LLC resonant converter .................................................................. 6
Figure 4: Operating regions for an LLC resonant converter ....................................................................... 7
Figure 5: Role of capacitor COSS and dead time TD .................................................................................... 9
Figure 6: Typical waveforms for ZVS........................................................................................................ 11
Figure 7: MOSFET device with its parasitic components ......................................................................... 12
Figure 8: HB switching .............................................................................................................................. 13
Figure 9: Current spikes in HB .................................................................................................................. 14
Figure 10: Current through body diode ..................................................................................................... 15
Figure 11: Body diode reverse recovery charge ....................................................................................... 15
Figure 12: MOSFETs with standard (left) and fast intrinsic body diodes (right) ....................................... 16
Figure 13: capacitive and inductive areas ................................................................................................ 18
Figure 14: Hard capacitive mode .............................................................................................................. 19
Figure 15: Dangerous operating conditions exceeding datasheet maximum ratings ............................... 20
Figure 16: Hard switching caused by supply disconnection ..................................................................... 20
Figure 17: Hard switching caused by fast load transition ......................................................................... 22
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LLC resonant half-bridge converters: topology and
characteristics
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LLC resonant half-bridge converters: topology and
characteristics
A basic LLC resonant half-bridge converter is shown below.
Figure 1: Basic LLC resonant half-bridge converter
The circuit consists of:
•
•
•
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A square-wave generator: two power MOSFETs, Q1 (High Side) and Q2 (Low Side),
are configured to produce a unipolar square-wave voltage. To prevent any crossconduction through ground and allow sufficient time to realize the ZVC, the two
devices are driven with a small dead time Td.
Resonant Tank: the resonant network is formed by capacitor Cr and inductors Lr and
Lm. In particular, Lm represents the transformer’s magnetizing inductance. The LLC
resonant converter looks very similar to an LC series resonant converter (SRC), apart
from the addition of the inductor Lm.
Rectifier and filter: on the secondary side of a converter, the rectifier consists of two
diodes for full-wave rectification and an output capacitor Co to smooth the rectified
voltage to the load RL. In some “synchronous rectification” configurations, the two
diodes are replaced with MOSFETs to help reduce conduction losses, especially in
low-voltage and high current applications.
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Circuit analysis and operation
Circuit analysis and operation
LLC converters have been the subject of extensive research, with various developments in
analysis and modeling techniques.
Fundamental harmonic analysis (FHA) represents a consolidated method to derive the
transfer function of the resonant tank and the output rectifier stage [1]. The FHA frequency
domain method sacrifices model accuracy to simplify the topology through sinusoidal
waveform approximations and AC equivalent circuits.
Assuminga 50% duty-cycle (180° saturation and 180° cut-off for each MOSFET) for Q1 e
Q2, the voltage applied to the resonant tank is a periodical square-wave voltage between 0
and Vbus.
Using the Fourier series, we can break up the square-wave voltage in input and output
through their harmonic components (see Figure 2: "Fundamental Harmonic Analysis
(FHA)")
When the resonance tank operates at its series resonance frequency, we can assume that
the bulk of the energy is transferred to the output through the fundamental component of
this square wave.
Moreover, if we consider a Q-factor of the resonator tank high enough to neglect the high
order harmonics, the current in the cell can be considered as purely sinusoidal.
At resonance, the amount of energy provided to the resonator tank is mainly dependent on
the value of the resonant circuit impedance at that frequency for a given load impedance
RL.
In power amplifiers, the configuration of the two MOSFETs and the electrical characteristics
of Figure 1 determine what is known as a ‘quasi-complementary Class D’ circuit because it
uses two identical transistors (N-channel MOSFETs). A true-complementary configuration
would require N- and P-channel MOSFETs.
The second side of the transformer is center tapped and the square-wave voltage is
bipolar.
Figure 2: "Fundamental Harmonic Analysis (FHA)" shows the input and output voltages
when implementing the FHA hypothesis at resonance frequency.
Figure 2: Fundamental Harmonic Analysis (FHA)
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Circuit analysis and operation
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To include the load resistance in the AC circuit, it needs to be converted to an equivalent
AC impedance on the primary side of the transformer (see Figure 3).
As the frequency of the square-wave generator is varied, the resonant circuit impedance
varies. In the LLC, the circuit peak resonance frequency is a function of the load. When
there is no load, the resonance frequency of the circuit is fp. As the load increases, the
circuit resonance frequency approaches the f0 limit representing a short-circuit load.
Hence, LLC impedance adjustment follows a family of curves with fp ≤ fres ≤ f0 during the
normal operation of the circuit.
Figure 3: AC equivalent circuit for the LLC resonant converter
From the equivalent AC circuit in Figure 3: "AC equivalent circuit for the LLC resonant
converter", the DC voltage gain can be derived using the FHA.
In Figure 4: "Operating regions for an LLC resonant converter", the operating regions of the
LLC resonant converter are divided into two primary switching type regions: the ZCS region
and the ZVS region. When the converter switches at higher than resonant frequencies fr1, it
always runs in ZVS mode. When the converter switches at lower than resonant frequencies
fr2, it always runs in ZCS mode.
When the converter is switching between the resonant frequencies fr1 and fr2, the load
condition determines whether the converter operates in ZVS or ZCS mode.
Under normal operating conditions, the LLC resonant converter operates at slightly higher
than the resonant frequency fr1, which is optimal for high efficiency.
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Circuit analysis and operation
Figure 4: Operating regions for an LLC resonant converter
One interesting feature of the gain curves is that they all converge on the point where the
switching frequency is equal to the resonant frequency; in other words, this unity gain point
is load independent and the converter operating at this point does not need to change its
switching frequency for any level of output power as long as the input voltage is the same.
The principal features of the FHA model are:
•
•
The resonant tank responds primarily to fundamental component of the applied
square-wave voltage, then tank waveforms are approximated into their fundamental
components.
A secondary rectifier + low-pass filter effect is incorporated into load.
The disadvantages of the FHA model are:
•
•
It does not accurately predict operation frequency and tank current amplitude,
especially in Discontinuous Conduction Mode (DCM).
The FHA model is not sufficiently accurate for overcurrent or overpower protection
design, so appropriate design margins need to be considered.
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The role of MOSFETs in achieving the ZVS
condition
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The role of MOSFETs in achieving the ZVS condition
Having briefly analyzed the resonant tank circuit, we can now focus on how to drive the
MOSFETs to create the ZVS condition for soft-switching at turn-on.
In Figure 1: "Basic LLC resonant half-bridge converter", the body diodes of Q1 and Q2 are
highlighted because they play an important role in the functionality of the circuit.
Moreover, the drain-to-source capacitances Coss1 and Coss2 provide an additional
contribution to the voltage transient of the node HB (midpoint between Q1 and Q2).
It is important to bear in mind that Coss1 and Coss2 are nonlinear capacitors; their value is a
function of the drain-to-source voltage [2].
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4
MOSFET output capacitance (COSS)
MOSFET output capacitance (COSS)
At the HB midpoint node, the total capacitance Czvs is the sum of the output MOSFET
capacitors COSS and the parasitic capacitance Cstray of the power MOSFET cases, the heat
sink, the intra-winding capacitance of the resonant inductor, etc.
Equation 1
Let’s see how the capacitance Czvs can influence the ZVS condition through the Q quality
factor [3].
Figure 5: Role of capacitor COSS and dead time TD
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MOSFET output capacitance (COSS)
4.1
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Q for ZVS at full load and min Vin
In order to guarantee ZVS, the tank current at the end of the first half cycle (see Figure 5:
"Role of capacitor Coss and dead time TD") must exceed the minimum value necessary to
deplete the equivalent capacitance within the dead time interval TD.
Equation 2
This current equals the peak value of the reactive current flowing through the resonant tank
and determines the reactive power level.
Moreover, the RMS component of the tank current associated with the active power is:
Equation 3
From Equation 2, Equation 3 and the resonant tank input impedance Zn(fn, K, Q) we obtain:
Equation 4
Solving Equation 4, we determine the quality factor Qzvs1 that ensures ZVS behavior at full
load and minimum input voltage.
4.2
Q for ZVS at no load and max Vin
The ZVS condition also needs to be satisfied under no load and maximum input voltage.
For this operating condition, it is possible to find an additional constraint for the maximum
quality factor to guarantee ZVS.
If Zin OL is the impedance of the resonance tank under no load, the condition to operate in
ZVS is:
Equation 5
The above equation forms the constraint for obtaining the quality factor Qzvs2 for the ZVS
under no load and maximum input voltage, thus:
Equation 6
Therefore, in order to guarantee ZVS over the whole operating range of the resonant
converter, we have to choose a maximum quality factor value lower than the smaller of Q=
min [Qzvs1; Qzvs2]
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MOSFET body diodes
MOSFET body diodes
In Figure 6: "Typical waveforms for ZVS", we see the two gate signals with the added dead
time to avoid Q1 and Q2 being ON simultaneously.
During this delay time, the current flows through the body diode of the each MOSFET to
guarantee ZVS operation, while during the ON state the body diodes are OFF.
Figure 6: Typical waveforms for ZVS
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Power MOSFET failure mechanisms
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Power MOSFET failure mechanisms
Figure 7: "MOSFET device with its parasitic components" shows the MOSFET device and
its parasitic components.
6.1
Failure mode 1: static dv/dt associated with the parasitic
BJT
Figure 7: MOSFET device with its parasitic components
6.2
Failure mode 2: reverse recovery dv/dt
The load current is negative while the MOSFET is ON and thus flows through the sourcedrain diode in the MOSFET structure (see Figure 7: "MOSFET device with its parasitic
components").
The base-emitter junction is reverse biased and the parasitic BJT is OFF.
When the MOSFET is turned OFF, a voltage step with a certain dv/dt is applied across DS. The resulting displacement current ID flows through the drain-base capacitance CDB and
the P-base finite resistance RB.
At the same time, the diode reverse recovery IRR flows through RB itself in order to remove
the charge stored in the drain region; IRR is not generated by dV/dt, but accompanies it.
The parasitic bipolar could be turned ON. This would reduce the clamping voltage and
potentially cause the device to enter an avalanche state.
As these dV/dt issues are common for bridge topologies, we need to carefully control the
switching speed of the MOSFET. In the HB, the switching MOSFET generates the VDS
variation across the device in the OFF state (see Figure 8: "HB switching ").
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Power MOSFET failure mechanisms
Q1 is ON and the current is flowing through the parasitic diode of Q1 (D1) due to its
direction; when Q1 is turned OFF, the load current still flows through D1 for the whole deadtime duration.
Q2 is turned ON with a high |dv/dt| (the dv/dt is negative on Q2 and depends on RG-ON(Q2))
so the current switches from D1 to Q2. The same |dv/dt | with a positive value is applied to
Q1.
The base-drain diode reverse recovery increases the risk of parasitic BJT turn-on, so the
dv/dt could be very dangerous and cause Q1 failure. The same problem could affect Q2
during Q1 turn-on.
Figure 8: HB switching
Generally, MOSFETs with fast recovery diodes are not necessary for normal ZVS
operation, but MOSFET failure in LLC resonant converters can sometimes be caused by
high current spikes (shoot through) due to the poor reverse recovery characteristics of the
body diode at particular instants (see Figure 9: "Current spikes in HB").
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Power MOSFET failure mechanisms
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Figure 9: Current spikes in HB
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6.2.1
Power MOSFET failure mechanisms
Body diode reverse recovery
Figure 10: "Current through body diode" shows the cross section of a MOSFET device with
the intrinsic diode between the body and drain.
Figure 10: Current through body diode
MOSFET parasitic body diode reverse recovery occurs during diode switching from the ON
to the OFF state because its stored minority charges must be removed either actively via
negative current or passively via recombination inside the device. The typical dynamic
parameters listed in the datasheet for diode reverse recovery are depicted in Figure 11:
"Figure 11: Body diode reverse recovery charge".
Figure 11: Body diode reverse recovery charge
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Power MOSFET failure mechanisms
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The highlighted area in the picture represents the body diode reverse recovery charge
(Qrr). Figure 12: "MOSFETs with standard (left) and fast intrinsic body diodes (right)" shows
a comparison between MOSFETs with standard intrinsic diodes and MOSFETs with fast
intrinsic diodes (low Qrr).
Figure 12: MOSFETs with standard (left) and fast intrinsic body diodes (right)
As already mentioned, MOSFETs with fast recovery diodes are not necessary in normal
ZVS operation, which is why STMicroelectronics has developed the MDmesh™ M2 and
MDmesh™ M2-EP series.
Both series are well suited for LLC applications and the MDmesh™ M2-EP series in
particular has advanced features to maximize power system efficiency with special
emphasis on high frequency operation and low load conditions.
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Non-standard operations in LLC resonance HB
Non-standard operations in LLC resonance HB
For specific situations where it is not possible to ensure safe device operation,
STMicroelectronics offers the special MDmesh™ DM2 series with improved body diode
performance, offering both very low reverse recovery charge (Qrr) and short recovery time
(trr).
7.1
Capacitive region operations
An example case is when the system operates in steady state under a low load. In this
condition, the system frequency is near the lower resonant frequency and the ZVS is
obtained.
If the load changes from a low to a high value, the switching frequency should follow the
new resonant frequency; if this doesn’t occur we might fall into region 3 (see Figure 4:
"Operating regions for an LLC resonant converter")
The MOSFET is turned OFF while the current is circulating through the body diode and,
since the antagonist MOSFET is turning ON, a body diode recovery can occur [4].
In this case, there is additional power dissipation due to the current and voltage of the
conducting body diode.
This gives rise to a potentially lethal shoot-through condition for the half-bridge led, due to
the simultaneous parasitic turn-on of both MOSFETs; moreover, the recovery of the body
diodes generates large and energetic positive current spikes.
In order to reduce this kind of risk, several solutions are implemented. Dedicated gate
driver controllers able to manage the dead time or appropriate network circuits able to
increase the dead time or higher Rgate values can generally resolve this problem.
Chip manufacturers can also help solve this problem with dedicated MOSFET devices
featuring short recovery times. The STMicroelectronics MDmesh™ DM2 MOSFET
technology re[resents a robust solution with enhanced body diode recovery time
performance, rated at less than 200ns.
As previously mentioned, a resonant converter operates in capacitive and inductive
regions, as depicted below in Figure 13: "capacitive and inductive areas".
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Non-standard operations in LLC resonance HB
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Figure 13: capacitive and inductive areas
When the system operates in the inductive region, the switching is in ZVS and during the
transition where the main switch passes from the ON to the OFF state, its current Ip has a
positive value (violet area) and flows from drain to source. When the system functions in
the capacitive region, the operations occur at ZCS. In this case (yellow area), the current
on the main switch goes from source to drain, also involving the physical diode on the
MOSFET structure.
The LLC system can experience capacitive mode operation in the following circumstances:
•
•
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Soft capacitive mode: occurs when the tank current phase progressively approaches
zero, for example at power down, with max load when the input voltage goes low.
Generally, resonant controllers like the L6699A [5] by STMicroelectronics have an
advanced protection feature (anti capacitive mode) that increase the switching
frequency as if there were an overload condition, thus raising the tank current phase.
Hard capacitive mode: this can occur when the tank current phase becomes zero or
negative from one cycle to another as in the case of a short at the output (see Figure
14: "Hard capacitive mode"). In this situation, the MOSFET is turned OFF and the
converter is stopped and no hard switching takes place.
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Non-standard operations in LLC resonance HB
Figure 14: Hard capacitive mode
7.2
Hard switching at start-up
During start-up, the ZVS condition can be lost and the MOSFETs will undergo hardswitching, potentially causing a very high diode reverse recovery current.
At start-up, the voltage across resonant capacitor is initially discharged and needs a
number of switching cycles before being charged to the steady-state value Vin/2. During the
initial transient, abnormally high tank current peak values may appear. The tank current
does not reverse during the first one or two switching cycles. In this condition, we may
experience the potentially hazardous capacitive-mode and hard-switching operations, even
if for a very short time interval. The MOSFETs could exceed their maximum dv/dt and di/dt
ratings (Figure 15: "Dangerous operating conditions exceeding datasheet maximum
ratings"), leading to potential failures.
This issue is also addressed in the L6699A resonant controller thanks to an enhanced softstart procedure to smooth the start-up.
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Non-standard operations in LLC resonance HB
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Figure 15: Dangerous operating conditions exceeding datasheet maximum ratings
Another possible solution to these potential failures is to slow down the dynamics of the
circuit using a combination of a diode and resistors in series with the two gates.
7.3
Hard switching caused by supply disconnection
Hard switching can also occur during normal SMPS operation. In fact, if the main supply is
disconnected, the system could be forced to operate in capacitive mode, as shown in
Figure 16: "Hard switching caused by supply disconnection".
Figure 16: Hard switching caused by supply disconnection
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Non-standard operations in LLC resonance HB
The figure shows that when the main supply is removed (point A), the dead-time fixed by
the driver is not enough to maintain inductive operation, so the current continues to rise
due to the shoot-through between the two devices.
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Hard switching caused by fast load transition
8
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Hard switching caused by fast load transition
In this case, the system might not change the switching frequency quickly enough and so
allow capacitive mode operation for the time required by the control system to restore
normal inductive operation of the SMPS.
Figure 17: Hard switching caused by fast load transition
In the above figure, the purple Vgssignal shows how the working frequency changes when
there is a fast load transition. On the left side of the figure, the current (cyan line) has a
typical shape for a resonant LLC, while on the right side, the voltage Vds (green line) lags
the current, which is typical of a capacitive network.
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Conclusions
Conclusions
This application note reviews the underlying theories for the characteristics of a resonance
LLC topology and, in particular, the key role of MOSFET devices in circuit operation.
Particular attention was paid to the operating conditions that could determine potential
failures of the MOSFETs. For these special situations, STMicroelectronics suggests its
dedicated DM2 series with improved body diode performance.
From this report, it is clear that special consideration should be given to MOSFET selection
for new LLC designs requiring body diodes with fast recovery times, low voltage drops and
rugged, dynamic dv/dt parameters.
In the context of the above requirements for soft and resonant applications, important
benefits are offered by the MDmesh™ M2-EP series technology, which optimizes both the
shape and the absolute value of the output capacitance, proven to be key requirements in
high efficiency SMPS applications.
For additional information regarding the MDmesh™ M2-EP series technology, please refer
to AN4742 [9]
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References
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References
1.
2.
3.
4.
5.
6.
7.
8.
9.
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Steigerwald, R. L. A comparison of half-bridge resonant converter topologies. Vols.
vol. 3, pp. 174-182, 1988.
Giuseppe Consentino, Antonino Gaito. Which Power MOSFET technologies in LLC
HB converters ? Nuremberg : s.n., 8 - 10 May; 2012 .
www.st.com LLC resonant half-bridge converter design guideline. AN2450.
Antonino Gaito Why use a Fast Diode MOSFET in a LLC topology.
STMicroelectronics.
www.st.com L6699A - Datasheet.
Mohan, N., Undeland, T. M. and Robbins, W. P.Power Electronics, converters,
applications, and design. s.l. : John Willey & Son, Inc. New York 1995, 2nd Edition.
www.st.com. Design LLC resonant converters for ottimum efficiency. paper #530, s.l.
: EPE 2009, EPE, p. 530.
www.st.com An introduction to LLC resonant half-bridge converter AN2644.
www.st.com MDmesh™ M2-EP: an additional improvement to MDmesh™ M2 ST
super-junction technology AN4742.
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Revision history
Revision history
Table 1: Document revision history
Date
Revision
05-Aug-2015
1
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Changes
Initial release.
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