AN39

AN39
Current measurement applications handbook
by Peter Abiodun Bode, Snr. Applications Engineer
Contents
1 Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.1
Application types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.2
Closed loop systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.3
Open loop systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.4
Measuring methods . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.5
Summary of methods and performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.6
Factors that determine the type of methods used . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
2 Detailed discussion of methods . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.1
Optically isolated resistive method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2
Magnetic method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.3
Resistive method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.3.1 Low-side resistive measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.3.2 High-side resistive measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
3 Properties of translation circuits for current monitors . . . . . . . . . . . . . . . . . . . . . . . 10
3.1
Accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.2
Frequency or transient response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.3
Power consumption . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.4
Maximum allowed sense voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
3.5
Considerations for sense resistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.5.2 Type of resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
4 Zetex ZXCT series product description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.1
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.2
Current Output Current Monitor (COCM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.3
Voltage Output Current Monitor (VOCM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
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5 Design examples and procedures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
5.1
Basic calculations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
5.2
Transient protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
5.3
Extended operating range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.3.1 Simplest circuit for VSUPPLY > VMAX . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.3.2 Improved circuit for VSUPPLY > VMAX . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5.4
Bi-directional current sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
5.5
Driving an H bridge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.5.1 With two independent current monitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.5.2 With anti-parallel current monitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
5.6
Short circuit protection/over-current protection applications . . . . . . . . . . . . . . . . . . . . 23
5.7
Latching over-current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
5.8
ZXCT1030 application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.9
Short circuit detection and continued operation (ZXCT1050, 1051) . . . . . . . . . . . . . . . 27
5.9.1 Extending the common mode input range of the ZXCT1050 beyond VCC using at a higher voltage than VMAX . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
5.9.2 Cost of precision resistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
5.9.3 Using ZXCT1050 at high voltages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
5.10 High speed and reverse transient considerations of ZXCT series . . . . . . . . . . . . . . . . . 32
5.11 Reverse transient . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
6 Appendix - Current monitor summary chart . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
7 Recommended further reading . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
8 Glossary of terms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
9 List of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
10 List of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
11 List of Equations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
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Introduction
Current measurement or monitoring within electronic circuitry is a common requirement
spanning many types of applications. These may include anything from portable, handheld
equipment through to automotive applications. This application handbook explores factors that
are relevant for AC and DC current measurement and the implications on cost and performance
for different approaches, and how to best use ZETEX current monitors in your applications. Basic
application topologies are explored including typical example calculations.
1.1 Application types
There are two basic application types,
•
Closed loop
•
Open loop
1.2 Closed loop systems
In closed loop systems, current is measured compared with a reference value and then modified
as necessary by some control element. Response time can be critical here especially where
immediate actions need to be taken based on instantaneous current value.
Examples of closed loop current monitoring include:
•
Switching power supplies with current limiting functions or switch mode battery charging
circuits.
•
PWM control of solenoids (automotive valve applications).
•
RF transmit control loops in portable cellular equipment, where transmitted power is
adapted with distance.
•
Control of bias currents in (RF) power amplifiers.
•
Electronic fuses for internal fault limiting in equipment on distributed power systems.
•
Auto shutdown functions for DC motor control (replacing slipping mechanical clutches).
1.3 Open loop systems
Open loop current monitoring systems are characterised by the fact that the measured value is
not acted upon immediately. It may, for example, be made available for some other system,
usually less time critical. Examples include,
•
Current measurement in instrumentation (e.g. bench power supplies, ammeters,
current probes).
•
Power consumption indication, especially portable battery powered consumer items.
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1.4 Measuring methods
There are basically three methods of monitoring current. Which of these three is used will depend
on a number of factors both intrinsic and extrinsic to the application.
These requirements may sometimes also be conflicting. Therefore a careful balancing of
requirements to select an optimum method is required.
These three basic methods, Resistive, Optical and Magnetic are tabulated in Table 1 to give a
quick overview comparison in terms of their benefits and weaknesses.
1.5 Summary of methods and performance
Table 1 below shows the methods described in this handbook and a quick comparison between
the most common factors found in current measuring application.
Table 1
Type
Common current measuring methods
Current
range
Isolated
Accuracy
AC
response
NonIntrusive
Cost
Resistive
Very Low/
High
No
High
Medium/
High
No
Low
Optically
isolated
resistive
Medium/
High
Yes
Low/
Medium
Low/
Medium
No
Low/
Medium
Magnetic
Medium/
Very High
Yes
Medium
Medium/
High
Yes
High
1.6 Factors that determine the type of methods used
Deciding which of the methods is most appropriate for any given application would involve
weighting these factors namely,
•
Magnitude of current
•
The need for galvanic isolation
•
Accuracy
•
Response time
•
Cost
Regardless of which choice is made, all will require some signal pre- or post processing.
The applications addressed in this document include steady state DC and those requiring
recovery of high frequency components from DC currents as well as bi-directional ones. We will
very briefly consider each of the methods in turn.
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Detailed discussion of methods
This Application Note is primarily concerned with resistive method because this is by far the most
frequently used and also since it is supported by Zetex' wide range of current monitors. However,
for completeness, the other two methods will be very briefly discussed before we look extensively
at the resistive method compared to Zetex' range of current monitors.
2.1 Optically isolated resistive method
In the strictest sense, this can not really be considered as a current sensing method in its own
right. This is because the opto-isolating device (usually an optically isolated transistor) does not
directly measure the current but merely transfers the already sensed current information across
a galvanically isolated barrier. It is discussed here to illustrate an often used method in isolated
current monitoring.
Various configurations may be used but the simplest method is illustrated in Figure 1. Resistor R
is the current sensing resistor. It is chosen such that, at the current that we want to limit the output
to, it develops a voltage equal in magnitude to VF, the forward voltage drop of the opto-isolator,
IC1. VF is typically approximately 2V.
This circuit is however so simple and basic that it is of rather limited use. The main reason being
the 2V or so VF that is needed to drive the opto-coupler. For example, if this were to be a power
supply required to supply a very modest 5A, R would have to dissipate at least 10W of power.
This is just not acceptable for many practical reasons. If the output voltage were 5V, it would
mean 40% of the available power is lost in the current sensing resistor!
Therefore, in practice, a much more lower voltage of the order of 50mV or less would be
developed across R which will then be amplified and used to drive the optocoupler.
Where isolation is required, the use of an optical method provides the most cost effective
solution. Its main limitation is that it is relatively low speed.
The example in Figure 1 (and Figure 2) is also a very good illustration of a closed loop current
monitoring.
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R
Controller
GND
Isolated
PSU
Elements
I
Load
VF
Limit
0V
IC1
Figure 1
Principle of optical and closed loop current monitoring
2.2 Magnetic method
Like the optical method, the magnetic method also offers isolation but, unlike it, also directly
senses its own current without the need for a current sensing resistor.
The magnetic method commonly uses a current transformer which produces an output voltage
that is proportional to the current. A change in topology is immediately obvious when we
compare Figure 2 with Figure 1. This type of magnetic method can only be used with AC
measurements unlike resistive and optical methods which can be either. Even so, the use of the
magnetic method is only practical at high frequencies rather than low frequencies. This is
because the current transformer that would be required at low frequency would be so bulky and
expensive as to make it a non-practical solution. To put things into perspective, you could imagine
a scenario where the current monitoring transformer could be nearly as big or bigger than the
circuit to be monitored.
For high frequencies however (e.g. switch mode power supplies) magnetic current monitoring
becomes feasible and is often used although, even here, it is being replaced by other more cost
effective methods such as the use of intelligent FET's to implement cycle-by-cycle current limiting
and indeed the now ubiquitous current monitoring solutions from Zetex.
The main advantage of the magnetic method is that it is a relatively loss-less technique. Hence it
is very useful where it is required to monitor very high currents in the hundreds or even
thousands of amperes.
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I
Controller
GND
Isolated
PSU
Elements
CT
Load
DC
Recovery
Circuit
Limit
0V
VCT
PostProcessing
Circuitry
Figure 2
Principle of magnetic current monitoring
2.3 Resistive method
This is the simplest, cheapest and the most basic method of current sensing. It is also by
definition the most accurate and linear method of all. Inserting a resistance into the current path
has the advantage of converting that current into voltage in a linear way that inherently follows
Ohm's law of V = I x R.
It is however not without its own faults although these can be minimised for many given
applications. The first and obvious one of these drawbacks is that it introduces additional
resistance into an electrical circuit. This can result in unacceptable power loss manifested as heat
and loss of efficiency.
Since power dissipation is a square function of resistance (P=I2R), this power loss increases as an
exponential function of current which is why the resistive method is rarely used beyond the low/
medium current application. Figure 3 illustrates just how very quickly power dissipation builds up
in a circuit using resistive current monitoring.
Another drawback is that the method inherently increases the source output resistance. The effect
of this may range from the mildly undesirable (such as slightly reduced terminal voltage) to
catastrophic, especially where the introduction of the resistor would interrupt the circuit from the
ground plane (e.g. a very noisy design which fails to meet statutory EMC requirements).
Either of these problems could be alleviated by using a resistance as close to zero as possible. In
accordance with this, the lowest value of resistance will produce the lowest power dissipation.
However, with an extremely low resistance, the voltage developed across it is also very low and
becomes comparable to circuit offset voltages, compromising accuracy. Therefore a balance
between required accuracy and power dissipated for a given current must be found. Shunt
resistors with values as low as 0.5 ⍀ (500µ⍀) are available commercially.
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Figure 3
Power dissipated vs current measured
2.3.1 Low-side resistive measurement
Low-side (negative or ground potential) measuring circuits generally offer the simplest solutions
because the resulting signal is already ground referenced. One such method is to insert a small
resistance into the ground plane between the supply's ground and the load to be measured as
illustrated in Figure 4. The resulting proportional voltage developed across that resistance can be
used directly or amplified.
Care must be taken to add further circuitry on the correct side of the sense resistor. Circuit C will
contribute additional current to that of Circuit B. This may or may not be desirable. To avoid this
happening, other circuits should be placed in the position of circuit A so that their currents do not
pass through the sense resistor.
A more subtle consequence of placing resistance in the ground plane is that any signal current
that passes through RSENSE will generate an offset voltage relative to true ground, 'VE rail' is now
a virtual ground at the potential of VSENSE, which will change with the load current. This may be
unacceptable with many analogue signals or high frequency circuits from an EMC standpoint.
An important property of the current measurement circuitry is that it should be able to accept
VSENSE, a signal of typically tens of millivolts from ground. Input offset voltage can be a
consideration here.
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ILOAD
Supply
Load/
Circuit
A
Load/
Circuit
B
Current
measurement
circuitry
Load/
Circuit
C
VSENSE = ISENSE x RSENSE
RSENSE
‘TE’ (True earth)
Figure 4
‘VE’ (Virtual earth)
Low side resistive method
2.3.2 High-side resistive measurement
In some applications, the loss of true ground cannot be tolerated because of the inter-circuit
offsets incurred, as mentioned earlier. Sometimes, regardless of whether the effects of disrupting
the ground plane could be ignored or tolerated, it is not possible to adopt this method for physical
reasons. For example, where a system uses a common ground in the form of a metal chassis (as
in an automotive environment) components that are connected mechanically to it may not have
the first method as an option. It is then necessary to measure the current in the supply (or high)
side. In many applications, noisy sources, for EMC reasons, need to be firmly tied to true ground
with as low impedance as is possible.
RSENSE
ILOAD
Supply
VSENSE
S+
S-
Zetex Current
Monitor
OUT
Current
measurement
circuitry
VOUT ∝ vSENSE
Load/
Circuit
RG
Figure 5
High side current measurement
The challenge now is to translate this small signal with a high common mode element on the
supply rail to a ground referenced signal, as further circuitry in most cases will require this.
Zetex has created a wide range of very compact integrated circuit Current Monitors, the
ZXCT series, for this requirement.
This means the complex task of processing the small current-related voltage signal is taken care
of and the designer can concentrate instead on making use of this processed signal.
With Zetex' range of current monitors, high- or low-side current monitoring is now a very simple
task. The devices are very small too coming in either SOT23 or SOT23-5 package, with very few
exceptions in bigger packages because of higher pin count requirements.
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3
Properties of translation circuits for current monitors
3.1 Accuracy
Accuracy is conditional on the size of the sense voltage for a given circuit design. Consider a fixed
input offset of say 2mV in a translation circuit. A 100mV sense voltage would then have a
theoretical best accuracy of 2% while a 10mV sense voltage has 20%.
Discrete transistor circuits are not going to give more than a modest performance in this
application. For any tolerance better than 5%, operational amplifiers and dedicated ICs are
needed.
It is worth, at this point, calling attention to dynamic range limitations which will have to be taken
into consideration always. For example a current monitor required to monitor current varying
from, say, 100mA to 1A has a dynamic range of 10:1. What this means is that the offset error
when measuring at 100mA would be at least ten times worse than when measuring at 1A. This
relationship obviously gets worse the wider the dynamic range.
3.2 Frequency or transient response
This may not be at all important in some circuits. For example, automatic shutdown of DC motors
probably won't require less than a few milliseconds response time. Whereas cycle by cycle
current limiting in a switching power supply might well need small signal responses in the order
of a microsecond or even down to the tens of nanoseconds. Closed loop power supply systems
which are monitoring average current will require response times anything in the range of 10s to
1000s of microseconds.
3.3 Power consumption
Current is required to drive parasitic and load capacitances so the speed requirement will always
be a balance with supply current usually only important in battery portable applications. For
example to drive a signal of 1 volt into 5pF of load capacitance at 10kHz you need only hundreds
of nanoamps of signal current and so bias currents will dominate. If you want to drive a signal
with 10ns edges at a volt into the same load you'll need hundreds of microamps, changing the
biasing of the circuit in a way which makes it less suitable for battery applications.
One circuit will not optimally provide both microamp operating currents and excellent AC
performance, so a choice has to be made.
3.4 Maximum allowed sense voltage
Even if the magnitude of the sense voltage is not critical from a power dissipation perspective
(because of the small currents being measured) there may be a maximum voltage drop for supply
headroom in the measured circuit. For example, in a 3.3V circuit, it would not be desirable for the
current monitor to drop 500mV which would represent a loss of 15% in both voltage and power
terms.
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3.5 Considerations for sense resistors
Power dissipation in resistors causes heating effects, the question is how much can be tolerated
and what further consequences are experienced. For currents under 1A, in general this is unlikely
to pose any problems as a typical sense voltage would yield power dissipations of less than
100mW which is easily handled by most surface mount and through-hole resistors. As Figure 3
shows, the power dissipation is directly linked to the accuracy required. It would be more critical
for a large current to have a low sense voltage in order to minimise power dissipation and so, for
a given accuracy, the translation circuit tolerances must be lower.
Example:
20A current must be sensed with maximum 500mW of dissipated power and with a best 3%
accuracy. We must determine the allowable input offset.
P = I 2 ⋅ Rs
Equation 1
P 0.5
=
I 2 400
Equation 2
Rs = 1.25mΩ
Equation 3
VSENSE = I ⋅ Rs = 25mV
Equation 4
Rs =
Therefore, for a best accuracy of 3% the input offset voltage can not be more than 3% of 25mV,
that is, 0.75mV.
Many of Zetex' range of current monitors have much better offset voltages than this and thus
would either give a better accuracy at sense voltage of 25mV or could be used at lower sense
voltages. For example the ZXCT1021 and ZXCT1022 are rated at 3% accuracy at a 10mV sense
voltage.
Another important reason to keep the self-heating of the resistors low is to minimise any further
loss of accuracy caused by the temperature coefficient of the sense resistor. Component to ambient
thermal resistances of surface mount resistors on a PCB could be in the order of 200°C/W - i.e. a
250mW dissipation would cause a temperature rise of 50 degrees above ambient.
Consider a resistor with a temperature coefficient of 100ppm/°C, this would give a further 0.5%
error from the self-heating effects which must be factored into the assessment of circuit
tolerances, this does not take into account the ambient operating temperature range.
3.5.2 Type of resistor
In most cases surface mount resistors are preferred and, for larger currents and their associated
power levels, small arrays of series or parallel resistors can be used to share the dissipation. Wirewound resistors provide higher operating temperatures, but usually have higher temperature
coefficients and higher inductance, which might cause problems for high frequency signals.
Beware of carbon film (or, much less common these days, carbon composition) resistors. Their
temperature coefficient can be thousands of parts per million!
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Zetex ZXCT series product description
4.1 Introduction
The Zetex ZXCT series current monitors have been designed specifically for resistive DC current
measurement discussed in chapter 1. Most of these are line powered, meaning that there is no
need for a separate power source when using them to monitor current. However some, due to
the need to offer enhanced features such as short-circuit current and/or low-side current
monitoring, have provisions for a separate VCC supply.
All of the devices in the series provide a basic function by taking a very small differential voltage
with a relatively high common mode content and converting it to an amplified ground referenced
voltage. All devices are high-side sensing capable whilst some are also capable of either low-side
sensing or are able to continue to provide an output in case of a high-side short circuit.
Table 2 in the Appendix (page 37) shows the range of current monitoring solutions that are
available from Zetex. The range includes, broadly, two types: Current Output Current Monitors
(COCM) and Voltage Output Current Monitors (VOCM). There are also the special function current
monitors (SFCM) which are basically current monitors with added functions for specialised
applications such as the ZXCT1030 (with built-in voltage reference and comparator) and the
ZXCT1032 (for inrush current control and use as an electronic fuse).
4.2 Current Output Current Monitor (COCM)
The current output device converts the sense voltage into an output current. This arrangement
offers unique advantages in that the overall gain of the circuit can now be set by a single resistor
connected to the output pin. Furthermore, although this resistor will in most cases be grounded
for a ground referenced output, in reality it does not have to be, making it possible for the output
to be referenced to some other arbitrary level within the circuit.
The simplest devices in the series are high-side current sense monitors with only 3 pins (SOT23
- see Table 2 on page 37). They are intended for cost effective applications. The current output
versions produce a constant current output which is proportional to the voltage across a sense
resistor. This allows a wide current range to be measured accurately when the appropriate sense
resistor is chosen.
For example, consider a 50mV sense voltage. A 1mA current could be sensed with a 50⍀ sense
resistor or a 50A current could be measured with a 1m⍀ sense resistor. This output current is then
converted into a voltage with a single user defined output resistor allowing flexible scaling.
In general, the COCM has a transfer function given by:
I OUT = GT ⋅ VSENSE = GT ⋅ RS ⋅ I LOAD
Equation 5
Where GT is the transconductance value in A/V or Siemens (S).
Since VSENSE is proportional to the current being measured, it is obvious that the monitor's
output current is also proportional to it. Hence, IOUT is a scaled down image of the load current.
For some devices (e.g. ZXCT1008, ZXCT1009), GT is internally set, typically at 0.01S. For other
devices (e.g. ZXCT1011, ZXCT1020), GT is set by a single external resistor whilst overall gain is
set by another resistor.
Consider Figure 6 below. The output current is proportional to the sense voltage (Equation 5
above) which itself can be seen to be proportional to the load current, ILOAD. Combining this
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information and noting that the output voltage is a product of the output current and RG, we have
an overall transfer function given by
VOUT = RG ⋅ GT ⋅ RS ⋅ I LOAD
Equation 6
(Transfer function for COCM
This shows that the output voltage, VOUT, is directly proportional to the current being measured.
RS
VSUPPLY
ILOAD
VSENSE
2
3
S+
S-
ZXCT1008 / 9
OUT
1
IOUT
VOUT
RG
Figure 6
Simplest current output current monitor (COCM)
From this basic and simple concept, some other devices with some additions and/or refinements
are offered.
For example, the ZXCT1010 adds a separate ground connection to the ZXCT1009 to remove the
IC's quiescent supply current from the output, reducing the output current offset component.
Some devices (e.g. ZXCT1050) provide for a VCC pin separate from the supply voltage whose
current is being monitored. The advantage of this arrangement is that the common mode range
of the sensed current can now include ground thus allowing low-side current monitoring as well
as high-side. Table 2 (page 37) gives a complete overview of all current monitors.
4.3 Voltage Output Current Monitor (VOCM)
The voltage output device is straightforward in its operation, it simply translates the small high
side (or low side in some cases) sense voltage into an amplified ground referenced
representation. The gain is fixed, typically 10 or 100.
For a VOCM device the gain setting resistor (and transconductance resistor) has been integrated.
Therefore the transfer function reduces to
Vout = k ⋅ VSENSE
Equation 7
Where k is a gain constant, typically 10 (e.g. ZXCT1021) or 100 (e.g. ZXCT1022). Compared with
the transfer function of a COCM (VOUT = RG • GT • VSENSE) it can be seen that k = RG • GT.
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RS
VSUPPLY
ILOAD
VSENSE
S+
GND
SOUT
VOUT
Figure 7
Simplest voltage-output current monitor (VOCM)
Again combining the information above with Equation 7, we see that the transfer function of the
VOCM is given by
Vout = k ⋅ RS ⋅ I LOAD
Equation 8
(Transfer function for COCM)
Unlike its COCM counterpart, it is clear that it is not possible to have a 3-terminal VOCM.
However, other than the sense resistor RS, the VOCM does not require any external component
at all whereas the COCM always requires at least one. Therefore, where space and component
count are critical, the VOCM will have a definite advantage.
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5
Design examples and procedures
5.1 Basic calculations
COCM
Example 1
Consider a requirement to measure a current of 5A to be represented by a 1V output.
Using a purely resistive method on its own would require a 200m⍀ resistor. The supply voltage
would be dropped by 1V and the power loss would be 5W. Not an ideal set of circumstances for
most applications.
However using one of Zetex' current monitors changes all of this. Let us consider the ZXCT1008
with reference to Figure 6 on page 13. For best accuracy, we require a VSENSE of 200mV. Therefore
the required value of RS is given by
RS =
0.2V
= 40mΩ
5A
Equation 9
This will give us our required VSENSE of 200mV. From Equation 5 (page 12) and noting that the
ZXCT1008 has a GT of 0.01S, we calculate the output current as
I out = 0.01⋅ 0.2 = 2mA
Equation 10
This only now requires choosing an appropriate value of RG to obtain the required output, thus
RG =
1V
= 500Ω
2mA
Equation 11
With this design the supply voltage would only now lose 0.2V and the power loss in RS would
only be 1W. It should be pointed out that this major improvement is, by comparison, modest
when compared with what can now be achieved with Zetex' latest range of current monitors
which can give reasonable accuracy at sense voltages as low as 10mV.
VOCM
The gain for a VOCM is given on the datasheet and we therefore only need to determine RS as per
Equation 7 and Equation 8, decide what gain is required and pick a device with that gain.
Example 2
For example, consider a requirement for a 20A current to be sensed with a maximum of 500mW
dissipated power within an accuracy of 3%.
Power = I2 x R
Therefore,
RS = 0.5/400 = 1.25m⍀
(4 x 5m⍀ resistors in parallel could be used for instance).
VSENSE at 20A = 25mV
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In order to meet our requirement, the VOCM must have a typical input offset of 3% at 25mV of
less than 750µV. The ZXCT1021/22 were created to meet design requirements such as these. They
are rated at ±3% at 10mV sense voltage. So, requirement is met.
The ZXCT1021 would give an output of VSENSE x 10 or 250mV, whilst the ZXCT1022 would give
an output of VSENSE x 100 or 2.5V.
Example 3
10A is to yield a resulting output voltage of approximately 2.56V for an ADC input. RS must be
smaller than 10m⍀ to minimize power loss.
We have the choice of gain of 10 or 100. Which would give the best value of VSENSE?
VSENSE =VOUT / Gain
VSENSE = 2.56/10 = 256mV or
VSENSE = 2.56/100 = 25.6mV
To obtain these two VSENSE values at 10A, RS must be:
RS = 256mV / 10A = 25.6m⍀ or
RS = 25.6mV / 10A = 2.56m⍀
Since RS is to be less than 10m⍀, we choose a gain of 100 i.e. the ZXCT1022 with a 2.5m⍀ sense
resistor, which could be two 5m⍀ resistors in parallel. This yields an output of 2.50V at 10A.
5.2 Transient protection
In some applications, especially those containing inductive elements, high voltage transients can
be present. Because of this, it may be necessary to provide some protection for the device, its
connected load or both. The following circuits are suggested ways of accomplishing this.
In Figure 8, the zener diode, Z1, is chosen such that any transient voltage that is greater than (VZ +
VOUT) is commuted to the output, therefore protecting the ZXCT device from any harm. This scheme
assumes that the load is tolerant of this transient or over-voltage event.
An alternative scheme that does not transfer the strain to the output as much is shown in Figure 9.
Here, an additional resistor, RLIM, is added in series with RG to limit the transient output current. Any
circuit connected to VOUT will therefore be protected from input voltage transients. A zener diode can
also be placed between VOUT and ground as optional protection for the connected circuitry. In this
case, ensure that the zener diode voltage is sufficiently higher than VOUT(max) so as not to interfere
with normal operation.
RS
VSUPPLY
ILOAD
2
SOUT
1
Z1
IOUT
Figure 8
ZXCT1008 / 9
1
IOUT
RLIM
IOUT
RLIM
VOUT
RG
SOUT
OUT
1
3
S+
S-
ZXCT1008 / 9
ZXCT1008 / 9
ILOAD
2
3
S+
RS
VSUPPLY
VSENSE
2
3
S+
ILOAD
VSENSE
VSENSE
Z1
RS
VSUPPLY
VOUT
RG
Z1
VOUT
RG
Z2
Transient protection Figure 9 Transient protection
Figure 10 Better transient
for device
for device and load
protection for device and load
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The value of RLIM will have to be carefully determined. At the lower end of its value, it is limited
by the maximum current that can be supplied by the device, whilst at the higher end the limiting
factor is the available circuit compliance. The two limits are determined as follows.
RLIM (min) =
RLIM (max) =
VPK − VMAX
I OUT (max)
Equation 12
RG (VSUPPLY (min) − (VDO + VOUT (max) ) ) Equation 13
VOUT (max)
Where,
VPK = Peak transient voltage to be withstood
VMAX = Maximum operating voltage (20V in most cases)
IOUT(max) = Max continuous output current (25mA)
VSUPPLY(min) = Minimum supply operating voltage,
VDO = Drop-out voltage
VOUT(max) = Maximum required output voltage
For practical determination of RLIM, since the value has to lie somewhere between these two
limits, it is just as well to pick a value nearest mid-point between them. Therefore:
RLIM =
RLIM (min) + RLIM (max)
Equation 14
2
If the two values are very close together, then use either of the values. Note that the two values
can converge and this is acceptable. If, as can sometimes happen, RLIM(min) is larger than
RLIM(max), this would indicate that the maximum output voltage, VOUT(max), required from the
circuit is too high for the set of circumstances. This can be corrected by doing any or all of the
following.
•
Reducing VOUT(max) which means lowering the value of RG.
•
Increasing minimum supply voltage VSUPPLY(min).
•
Reducing IOUT. Note that this can only be done by reducing RS which will result in an increase
in error which may not be desirable.
The solutions in Figure 8 and Figure 9 can be combined as in Figure 10 to get the best of both.
Figure 8 and Figure 10 in particular and, to a lesser degree Figure 9, have the advantage of
protecting the current monitor against any negative excursions of the supply voltage.
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5.3 Extended operating range
The ZXCT1009, for example, has a maximum operating voltage of 20V1. Where higher voltages
are required, the following circuits can be used to extend the operating voltage range of the
current output current monitor.
5.3.1 Simplest circuit for VSUPPLY > VMAX
Figure 11 shows the simplest way to extend the working voltage range of a current monitor above
its design value. The zener diode, Z1, is simply inserted in series with RG.
The zener diode's value is chosen taking into consideration the range of the supply voltage and
ensuring that the following conditions are met.
VSUPPLY (min) ≥ VDO + VOUT (max) + VZ
Equation 15
VSUPPLY (max) ≤ VMAX + VZ
Equation 16
where, VDO = device drop-out voltage, VZ = zener voltage, VOUT(max) is the maximum output
voltage, VMAX = maximum working voltage.
In reality this scheme is not so much an "extension" of the working voltage range as a "lifting" of
it by an amount equal in magnitude to the zener voltage. It nonetheless does allow the device to
be used beyond its designed maximum working voltage.
It may be noticed that the circuit in Figure 11 is quite similar to Figure 9 with RLIM being replaced
with a zener diode. Indeed Figure 9 can be used as a simple supply voltage range extender where
the supply range is lifted by an amount given by IOUT x RLIM instead of VZ.
1 20V is used as an example in most cases as many of the ZXCT range work at this maximum supply voltage, although the range has now
expanded to include 40V and 60V devices. The same principle will still apply to these other devices.
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5.3.2 Improved circuit for VSUPPLY > VMAX
A true extension of the supply voltage range is provided by Figure 12. This circuit has the
advantage of also offering very good transient protection.
ILOAD
RS
VSUPPLY
3
S-
R1
ZXCT1008 / 9
S-
Z1
ZXCT10xx
GND
OUT
1
S+
RS
VSUPPLY
VSENSE
VSENSE
2
S+
ILOAD
OUT
ILOAD
VSENSE
15V - 20V
RS
VSUPPLY
S+
SZXCT10xx
GND
OUT
IOUT
IOUT
Z1
IOUT
TR1
TR1
VOUT
VOUT
R2
RG
Figure 11 Simplest supply
range extension
VOUT
R2
RG
Figure 12 Improved supply
range extension
RG
Figure 13
Best supply range
extension
Suitable devices: All COCM's.
Note that the ZXCT1050 has a VCC pin which needs to be at least 2V above VSUPPLY. Whilst this is
not impossible it might present a challenge in some situations. In that case the circuit shown in
Figure 28 (page 28) and Figure 32 (page 31) is recommended instead especially for short circuit
measurement. Also, the common mode range of these circuits does not include ground, the more
reason why Figure 32 should be used for the ZXCT1050 if low side sensing is required.
TR1 is used in the common base configuration and is used to drop most of the supply voltage
between collector and emitter. When the current gain is reasonably high (>100), IC ≈IE and IOUT
still flows through RG and hence VOUT can still be calculated in the normal way.
Ideally, R1 must be chosen to preserve the ZXCT's normal supply range, large enough in value to
provide the minimum operating voltage to the device at the lowest supply voltage but not too
large that the maximum device operating voltage is exceeded at the highest input voltage.
Procedure 1 - Design steps for Figure 12
1.
2.
3.
Determine or estimate IOUT (it doesn’t need to be precise at this stage)
Determine the required minimum supply voltage, VSUPPLY(min).
Determine device’s maximum working voltage, VMAX.
4.
Calculate transistor bias current IB from I B = I OUT
hFE (min)
5.
Calculate bias resistor RB from
RB =
6.
7.
(VSUPPLY (min) − VDO − Veb )
IB
⎛
Calculate R1 from R1 = ⎜
⎜
=
(VSUPPLY (min) − VDO − Veb ) ⋅ hFE (min)
I OUT
VSUPPLY (max)
⎝ VSUPPLY (max) − VMAX
Calculate R2 from R 2 = ⎛⎜ VSUPPLY (max) ⎞⎟ ⋅ RB
⎜ V
⎟
⎝
⎠
MAX
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=
R1 ⋅ R 2
R1 + R 2
⎞
⎟ ⋅ RB
⎟
⎠
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In a situation where a higher supply voltage is required or where the supply voltage varies over
a wide range, the scheme in Figure 13 could be used where resistor R1 in Figure 12 is replaced
with a zener diode rated within the maximum working voltage of the COCM. The design steps are
similar to those in Procedure 1 but slightly simpler.
Procedure 2 - Design steps for Figure 13
4.
Determine or estimate IOUT (it doesn’t need to be precise at this stage)
Determine device’s maximum working voltage, VMAX.
Chose the value of Z1 to be within VMAX e.g. VZ=15V for a 20VMAX
device. In general, make sure (VDO + Vbe ) < VZ ≤ VMAX .
Determine the required minimum supply voltage, VSUPPLY(min).
5.
Calculate transistor bias current IB from I B = I OUT
1.
2.
3.
hFE (min)
6.
Calculate resistor R2 from
R2 =
(VSUPPLY (min) − VZ )
=
IB
(VSUPPLY (min) − VZ ) ⋅ hFE (min)
.
I OUT
5.4 Bi-directional current sensing
Two COCM's can be connected in anti-parallel fashion to provide bi-directional current
monitoring as shown in Figure 14 below.
It will be noticed from Figure 14 below that only amplitude information is available, which is
acceptable if all that is needed is amplitude data as is often the case in many applications.
However, sometimes it may also be required to preserve polarity information after rectification.
The arrangement in Figure 14 does not have this feature. Zetex has produced the ZXCT1041 to
fulfil this role. The ZXCT1041 is a VOCM and not only converts bi-directional current into a
unipolar output voltage but also has an output flag which indicates the current polarity.
RS
3.5
ILOAD
-Ve
+VSENSE
S+
3
+Ve
-VSENSE
S-
ZXCT10xx
GND OUT
+IOUT
S+
2.5
IOUT (mA)
UPPLY
SZXCT10xx
GND OUT
2
1.5
1
-IOUT
0.5
VOUT
0
-300 -250 -200 -150 -100 -50
RG
Figure 14
0
50
100 150 200 250 300
V SENSE (mV)
Bi-directional current monitoring
Figure 15 Output response for bi-directional COCM
Suitable devices: ZXCT1008, 1009, 1010, 1012, 1020
Note that some VOCM's (e.g. 1021, 1022) can also be used but the outputs need to be kept
separate. This in fact means that both amplitude and direction information can now be captured.
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Some points are worth noting regarding Figure 14.
•
If the ZXCT1008 or ZXCT1009 are used, the quiescent current offset error will be doubled. For
this reason, the ZXCT1010 and 1012 may be preferable as they have separate ground pin
which diverts the quiescent current away from the output.
•
All current monitors have an internal reverse diode across the S+ and S- pins (shown in-laid
on diagram. Thus, in the anti-parallel mode example given above, it is important to make
sure that VSENSE is kept below ±500mV so that these diodes do not become forward biased.
This is not a problem as it is in any case advisable to keep VSENSE well below this level for
other reasons.
•
While one half of the devices is active, the other device' internal output stages are driven into
a saturation mode from which it will take a finite time for them to recover. It will therefore be
found that this application will not be suitable for very fast (sub-milliseconds) circuits. Again
the ZXCT1041 has kept this delay to a minimum, less than 5µs.
5.5 Driving an H bridge
5.5.1 With two independent current monitors
A very common application of bi-directional current monitoring is in a full or H bridge circuit. If
there is access to the supply rails to the bridge, the optimum way to measure the load current is
to use either low or high side sensing as depicted by RX and RY in Figure 16. However there are
situations when this may not be possible (if for example there is no access to the supply rails) or
the right thing to do. For example, it may be required to know which direction the load current is
flowing. Measuring with RX and RY will not necessarily indicate this as the current through them
is always unidirectional regardless of the direction of the load current. However this is only true
if the load is resistive. The situation gets even more complicated if the load is inductive as is the
case with driving a motor. In this case, the current in RX or RY will change direction when
commutating the FET switches whilst the current through the inductive load is still flowing in the
same direction! For these three different reasons, the most reliable way, therefore, to monitor the
load current is at the load.
Since the current is bidirectional, the designer may be tempted to use the simple two device
arrangement illustrated in Figure 14 above. This however would not work due to the fact that both
devices would need to have a common mode input range which includes ground. Therefore the only
effective way to measure the bidirectional current in this bridge would be as shown in Figure 16.
Two separate COCM's (VOCM's are also possible) are used to alternately monitor the load current
depending on the direction. Whilst one device is active, the other is inactive due to it being biased
off, hence the two devices together provide both amplitude and polarity information.
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VSUPPLY
RY
+
S+
-
S-
ZXCT10xx
GND
OUT
RX
Figure 16
M
S-
VOUT+
S+
ZXCT10xx
GND
OUT
VSUPPLY
VOUT-
Measuring bidirectional motor current in a full bridge driver
5.5.2 With anti-parallel current monitors
It has been argued above that the circuit of Figure 14 would not be suitable for the H-bridge
application in Figure 16. Whilst this is true, some of the latest devices, the ZXCT1050, ZXCT1051,
which can also work in the low-side mode can indeed be configured for anti-parallel operation in
this application. However, doing this only results in saving one sense resistor and a separate VCC
supply would have to be provided. The user needs to make a judgement whether this is
acceptable or not.
VCC > VSUPPLY + 2V
VSUPPLY
RY
-
+ M
RS
RGT
RX
S+
SZXCT105x VCC
GND
OUT
VOUT-
Figure 17
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RGT
VCC
S-
S+
ZXCT105x
GND
OUT
VOUT+
Measuring bidirectional motor current with anti-parallel current monitors
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5.6 Short circuit protection/over-current protection applications
R1
VSUPPLY
ILOAD
R2
1k
C4
100n
VSENSE
2
3
S+
U1
SZXCT1009
OUT
1
C1
2200µF
IOUT
C2
100µF
VOUT
Q1
FMMT618
Open collector overcurrent output
U2
ZR431L
R3
1k
C3
10n
Figure 18
Over-current protection circuit
Transistor Q1 serves to reduce the voltage seen by U1 and operates as a zener diode in its reverse
Vbe breakdown mode. Q1 is not needed when operating with supplies below the device's VMAX.
The voltage developed across R3 is then fed to U2 (ZR431L) which is a programmable voltage
reference used as a comparator. When the voltage on the Vref pin exceeds 1.24 volts, the device
conducts, pulling the open collector output low. The required pull-up resistor can be connected
to any supply rail of choice up to 20V. The advantage of using the ZR431L as a level detector in
preference to a transistor Vbe or a FET VGth is that the voltage sense level is virtually independent
of temperature and varies less from device to device.
The sensitivity of the current limit can be increased or reduced by adjusting the value of R3 to
develop the required 1.24 volts at different currents. C3 provides a time delay to prevent false
triggering.
The value of current that causes the output to switch can be calculated by rearranging the
datasheet formula:
VOUT =Vref = 0.01 x VSENSE x RG (Vref = 1.24V, RG =R3 and VSENSE =ILOAD x R1)
Hence, the trip current will be:
I trip =
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0.01 ⋅ R1 ⋅ R3
23
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For the example in Figure 18 Itrip = 5.6A
When monitoring an inductive load where voltage transients may be present, R2 and C4 provide
filtering to ensure correct operation. If the load is purely resistive these components can be
omitted.
For a more integrated solution for over-current applications see the ZXCT1030 with internal
comparator and reference.
5.7 Latching over-current protection
Figure 19 below shows how any standard COCM may be used to implement a latching over-current
protection circuit. This particular circuit is aimed at low power applications but can easily be adapted
for a higher power one by suitably substituting Q5 with a high power p-channel MOSFET. With a little
modification, it can also be adapted for use with a VOCM. The circuit consists of three distinct stages.
Stage A is easily recognised as a compliance-boosted current monitor as previously discussed in
Figure 13 with a slight modification of including a grounded second zener diode, Z2, to generate a
reference supply.
Stage B is a comparator consisting of Q2 and Q3. It compares the current monitor's output at R3 with
a reference, VREF, derived from Z2. Once this is reached, the comparator turns on Q4 which turns off
the series pass transistor, Q5, thus removing power from the load. Q4 and Q5 form Stage C.
Q5 at the same time provides a positive feedback via R10 to the current monitor forcing it into an
over-drive mode that will eventually saturate it. The circuit will remain locked in this state until both
power and the overload are removed.
A
B
S+
R1
1k
S-
ILOAD
R10
220k
Q2
5k
5.1V
R3
R4
R7
100k
Q3
R9
Q4
10k
R5
100k
Figure 19
ZXTN2038F
R6
10k
10k
2 x ZXTP2039F
Q1
LOAD
ZXCT10xx
GND OUT
ZXTP2039F
5.1V
0.51R
Z2
ZXTN2038F
Q5
R11
10k
R2
48V
Z1
C
R8
100k
Latching over-current protection
The transfer function of this circuit (i.e. the current at which the circuit trips) is deduced and is
given by:
I trip
⎞
⎛
⎞
V ⎛
R8
R8
⎟⎟ = 100.VZ 2 ⎜⎜
⎟⎟
= Z 2 ⎜⎜
GT ⎝ R 2 ⋅ R3( R7 + R8) ⎠
⎝ R 2 ⋅ R3( R7 + R8) ⎠
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Equation 17
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With the values shown in Figure 19, the circuit would trip out at around 100mA. This trip current
value can be extended up to 2A by substituting transistor Q5 with the two transistor network as
shown below in Figure 20. Also, this simplified circuit (when compared with an op-amp
implementation instead of the ZXCT COCM) can be further much simplified when implemented
using one of Zetex' later addition to the range, the ZXCT1030 (see "ZXCT1030 application" below).
Q5
ZXTP25060BFH
1
3
1
2
Q6
ZXTN2038F
2
Q5
3
R12
3.3k
Figure 20
Replace Q5 in Figure 19 above with this network to extend the trip
current range up to 2A
5.8 ZXCT1030 application
The ZXCT1030, with its integrated bandgap reference and comparator, expands the functionality
of the Zetex current monitors range. It is essentially a VOCM with a fixed gain of 10 and housed
in an SO8 package. It is also classed as a special function current monitor (SFCM) because of the
additional in-built features.
RS
VSUPPLY
3
VCC
1
S+
VCC
GND
ILOAD
2
SOUT
7
Vref
6
+
VOUT
8
Cout (o/c)
Vcomp
GND
Figure 21
5
-
4
k (typically) = 10
The ZXCT1030 VOCM with comparator and bandgap reference
The ZXCT1030 has all of the basic functions of a VOCM and can be used to implement the latching
current monitor of Figure 19.
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AN39
D
C
ZXMP3A17E6
Q5
ILOAD
R2
20V
0.51R
R1
1k
VCC
OUT
ZXCT1030
VCOMP
COUT
VREF
GND
68k
R10
Q4
LOAD
S-
ZXMN3B14F
S+
R11
4.7k
R7
10k
R8
51k
Figure 22
A much reduced version of Figure 19 using the ZXCT1030
It can be seen from above that the entire A and B section in Figure 19 (13 components) has been
replaced by just section D in Figure 22 (5 components).
The transfer function, i.e. the trip current, of this simplified circuit can be shown to be
I trip =
1.24 ⋅ R8
10 ⋅ R 2(R7 + R8)
Equation 18
Both Figure 23 and Figure 24 show the ZXCT1030 being used in motor drive applications.
RS
5V
R1
S+
S-
VCC
OUT
FB
COUT
SD
ZXCT1030
VCOMP
VREF
V
Motor
Controller
DR
GND
GND
R2
M
R3
Figure 23
ZXCT1030 in a motor control application
In Figure 23 the ZXCT1030 monitors the motor current and feeds it output into a generic dedicated
motor controller which is able to modulate the motor based on the current information received
from the ZXCT1030. The comparator's output in this circuit is used to determine the maximum
current at which to send a shut-down signal to the SD (shut down) input of the controller.
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RS
5V
RS
5V
R1
R1
M
S-
VCC
S+
OUT
OUT
ZXCT1030
VCOMP
VREF
S-
VCC
ZXCT1030
VCOMP
COUT
GND
VREF
R2
R2
R3
R3
Figure 24
Load
S+
IC1
Motor current limit
COUT
GND
To isolated
part of circuit
Figure 25 Driving optocoupler for
isolated application
Figure 24 is a bit simpler and relies entirely on the ZXCT1030 and an external MOSFET to control
the motor. The motor is simply powered on by the device. If the motor current, due to loading,
ever rises to the reference level set by R2 and R3, it shuts down the MOSFET which then removes
power from the motor. When the overload is removed, the motor automatically restarts.
5.9 Short circuit detection and continued operation (ZXCT1050, 1051)
Most Zetex current monitors are line powered. The ZXCT1050 and ZXCT1051 have a separate
supply rail (VCC) enabling them to continue operating even if the rail being sensed is shorted to
ground. Both devices have a common mode range that includes ground which means they can
be used for either low or high side current monitoring.
The ZXCT1050 is a COCM with an external transconductance setting resistor (offering greater
versatility) whilst the ZXCT1051 is a VOCM with a fixed gain of 10 (offering economy of
component count and board space).
RS
VSUPPLY
VSUPPLY
RS
ILOAD
RGT
VCC
S+
SVCC ZXCT1050
GND
OUT
VCC
Vin
Load
VCC ZXCT1051
GND
OUT
IOUT
VOUT
VOUT
RG
Figure 26
Basic ZXCT1050 configuration
Figure 27
VOUT = RG ⋅ GT ⋅ RS ⋅ I LOAD
Basic ZXCT1051 configuration
VOUT = 10 ⋅ RS ⋅ I LOAD
Equation 19
where GT = 20/RGT.
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Although RGT can be freely chosen, it is recommended that its value be kept at 7.5k or similar
since the internal design of the ZXCT1050 has been optimised for this value. Any other value may
impact on linearity and accuracy.
VSUPPLY which is effectively the common mode input voltage can vary between 0 and VCC-2V.
A very useful circuit to extend this common mode input range to VCC and beyond is given in
Figure 28 below.
5.9.1 Extending the common mode input range of the ZXCT1050 beyond VCC - using at a higher
voltage than VMAX
In some situations, it may be advantageous to operate the ZXCT1050 with a supply range in
excess of VCC. Happily this is something that can be easily realised with just three additional
resistors as shown in Figure 28 below.
A resistor, R3, is connected from the S- pin to ground so as to form a potential divider with the
transconductance resistor, RGT. The S+ pin is similarly connected to another potential divider
formed by R1, R2. It must be strictly ensured that the ratios (not necessarily absolute values) of
the two potential dividers are exact. In other words, R1/R2 must be equal to RGT/R3. Failure to
observe this rule will result in massive common mode error that would render the scheme
practically useless. In addition, the resistors themselves need to be very closely matched to better
than 1%. In a practical implementation of this circuit (Figure 29), 0.1% tolerance resistors were
used and the performance shown in Figure 30 was obtained. It will be noticed that there is
virtually no common mode error in this example.
RS
VSUPPLY
R1
RGT
S+
SVCC ZXCT1050
GND
OUT
VCC
IOUT
VOUT
RG
R2
Figure 28
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R3
Extending the CM range of the ZXCT1050
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Extending CM range for ZXCT1050
RS
ILOAD
7.00
VSUPPLY
6.00
RGT
R1
Vcm = 0
Vcm =10
Vcm = 20
Vcm = 30
R2
VCC
RS = 1R
R1 = RGT = 7k5
R2 = R3 = 2k49
RG = 7k5
VOUT (V)
5.00
R3
S+
SVCC ZXCT1050
GND OUT
4.00
3.00
2.00
VOUT
1.00
0.00
RG
0
Figure 29
20
40
60
Test circuit
80
100 120 140 160 180 200 220 240 260 280 300
Vsense (mV)
Figure 30
Output response
With the values shown, it can be seen that the circuit has a gain of 20.
An analysis of Figure 28 will readily show that the transfer function of the ZXCT is not affected by
the addition of R1, R2 and R3 and remains that given by Equation 19 in Figure 26.
It will be noticed from Figure 30 above that there is virtually no common mode error in this circuit
for a CM input ranging from zero to 30V
5.9.2 Cost of precision resistors
Although readily available, precision resistors cost a lot more than standard tolerance ones. That
fact may make this proposed solution less desirable for some low-volume low-cost applications.
However there is a way round the problem. Remember that what is important is the relative ratios
of the two potential dividers and not their absolute values.
Hence, one resistor could be replaced by a trimmable resistor to balance both legs. This way, less
than precise values could be used to start with as shown in Figure 31. Here, R2 has been replaced
by the combination of a fixed and a variable resistor1. Now, the resistors do not have to be low
tolerance ones and standard 1% or even 2% resistors can be used. What is more important is
stability. So, in any case, always make sure that high stability resistors are used. Metal film
resistors are generally very good in this regard.
RS
VSUPPLY
R1
VCC
RGT
S+
SVCC ZXCT1050
GND
OUT
R2V
IOUT
R2F
RG
VOUT
R2
Figure 31
R3
Using non-precision resistors to extend CM range
1
Note that it is not recommended to make all of R2 variable as this would result in very low resolution, increased potential
for long term drift and make the circuit more susceptible to thermal and mechanical shock effects.
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It needs to be kept in mind that there is a very small error associated with the scheme in
Figure 31 (and Figure 28) due to the small quiescent current that now flows in RGT and R3. This
error is quantified by the following expression.
I Error
⎛
VSUPPLY
= ⎜⎜
⎝ RGT + R 2 + RS
⎞ RS
⎟⎟ ⋅
⎠ RGT
Equation 20
In most cases, this error current will be quite small and negligible but it should be verified that
this is so in all cases.
Procedure 3 - Design steps for extending CM range Figure 28 and Figure 31
1. Determine the maximum required supply voltage, VS(max).
2. Calculate R3 from
R3 =
RGT
⎛ VS (max) ⎞
⎜⎜
⎟⎟ − 1
⎝ VCC − 2 ⎠
3. Make R3 the nearest lower preferred value. E.g. if the result of 2 above were 69.35k, choose
68k as the nearest lower preferred value.
R1 RGT
=
4. Next, determine R1 and R2 from
.
R 2 R3
The easiest thing to do is to simply make R1=RGT and R2=R3 but you may also make
R1=nRGT and R2=nR3 where n is any number preferably not less than 1. The advantage of
making n greater than 1 is that the current down the potential divider network formed by
R1,R2 can be kept to a minimum. Be careful however not to make n too high as it then begins
to introduce offset errors into the circuit. A value of n between 1 and 10 is quite reasonable.
This is all that is required as far as using high precision resistors is concerned (Figure 28). In order
to use standard resistors however (Figure 31) the following steps are required as well.
5. Determine the tolerance, Tol, of resistors being used, e.g. 1%.
6. Calculate R2V from R 2V ≥ 8 ⋅ Tol ⋅ R 2 and select the nearest higher preferred value.
100
7. Calculate R2F from R 2 F ≤ ⎛⎜1 − 4 ⋅ Tol ⎞⎟ ⋅ R 2 and select the nearest lower preferred value.
⎝
100 ⎠
Make sure that R2V is a good quality variable resistor (e.g. cermet type). If the circuit is going to
be subjected to a wide temperature range, it would also be advisable to make sure that the
temperature coefficient of R2V is comparable to that of the fixed resistors.
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Worked example:
It is required to measure the current of a load which is powered from a maximum supply of 30V.
Design the monitoring circuit around the ZXCT1050 using a VCC supply of 12V. Assume RGT to
be 7.5k.
Solution;
R3 =
RGT
⎛ VS (max) ⎞
⎜⎜
⎟⎟ − 1
V
−
2
⎝ CC
⎠
=
7.5
= 3.75kΩ
⎛ 30 ⎞
⎜
⎟ −1
⎝ 12 − 2 ⎠
Therefore,
R3 = 3.6k - picking the lower nearest preferred value in the E24 series.
Hence,
R1 = RGT = 7.5k, and
R2 = R3 = 3.6k
5.9.3 Using ZXCT1050 at high voltages
The technique discussed above provides a very convenient way to use the ZXCT1050 at a higher
operating voltage than its normal VMAX of 20V would allow.
An example to extend the operating voltage to 60V is show in Figure 32 below. Note that the 12V
VCC is derived from VSUPPLY which has a maximum value of 60V. At this voltage, the voltage on
the S+ and S- pins is 10V which is 2V less than VCC. Thus operational requirement will be met.
Because VCC is dependent on VSUPPLY, this circuit's common mode range will not include ground.
This can be changed by simply supplying VCC from a different source.
There is no theoretical limit to how high VSUPPLY (hence common mode input changes) can be as
long as the resistor values are scaled accordingly. However the higher the common mode input
voltage, the more accurate the resistors will have to be and, if the circuit in Figure 31 is used, the
more difficult it will be to trim out common mode errors.
There is also the limit imposed by power dissipation requirement of the resistors especially the
zener bias resistor which needs to drop most of the excess voltage and still supply the operating
current of the current monitor.
60V(max)
7.5k
7.5k (RGT)
5.6k
S+
SVCC ZXCT1050
GND
OUT
IOUT
12V
VOUT
RG
1.5k
Figure 32
1.5k
Using the ZXCT1050 at a higher voltage (alternative to using the schemes
in Figure 12 and Figure 13)
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5.10 High speed and reverse transient considerations of ZXCT series
The ZXCT series have very good bandwidth and good high speed performance even though they
are aimed primarily at low speed applications. For example, the bandwidth of the ZXCT1009
operating with a VSENSE of 100mV is 2MHz.
However in cases where the current is bi-directional or where there is a lot of high frequency
noise resulting in the device being exposed (even momentarily) to negative VSENSE, the ZXCT
device can be dramatically slowed down. This is because it ceases to operate in the linear part of
its range and takes considerable time (several microseconds) to recover. This can sometimes
result in a big part of the output being suppressed or even "disappearing" altogether. Therefore
if quick response is required, it is recommended to ensure that the device is never exposed to a
negative VSENSE. The magnitude of reverse voltage to cause this need not be large at all (2mV or
less) due to the very high internal gain stages in the device.
An example of where this can happen is where an inductive load is being driven and the load
current is monitored on the high (or low) side as shown in Figure 35 later. During the dead-band
between one pair of transistors switching off and the other switching on, the stored energy in the
inductive load causes current to continue to flow. This current reverses direction in RS and is
dumped into the power supply. Thus a negative sense voltage is developed across the inputs of
the current monitor. Since this is real current that is flowing in the reverse direction, merely
protecting the input of the current monitor will not have much effect except to slow down the
device generally. This will also happen when current flows in the forward direction.
One solution that has been found to work fine is to bias the device so that it never sees a negative
voltage across its input even when current flows in the reverse direction. Figure 36 to Figure 38
show the effects that negative VSENSE has on the speed of a current monitor. In contrast to these
Figure 40 shows what a dramatic improvement is obtained when biasing is applied to the S- pin
as shown in Figure 39.
This method of applying a compensating offset will only work provided the negative current flow
is consistent, which it usually is. In this case we only need to determine the current and from it
decide the required bias voltage as follows:
VBIAS = I REVERSE ⋅ RS
Equation 21
(Determining bias voltage)
We need a reference voltage, VREF, from which to obtain the required Vbias. With reference to
Figure 39 for example, where VREF is 1.24V (ZTLV341)1, we can see that
VBIAS =
Equation 22
VREF ⋅ R1
(Obtaining bias voltage from VREF)
R1 + R 2
Combining Equation 21 and Equation 22 and transposing results in
R2 ⎛
VREF
= ⎜⎜
R1 ⎝ I REVERSE ⋅ RS
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32
⎞
⎟⎟ − 1
⎠
Equation 23
(Determining R1 and R2)
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Since all in the RHS are known quantities, we can fix R1 and calculate R2 or vice versa. The
simplest thing to do is perhaps determine R1 as follows.
1. Determine bias current IREF for the reference and compute R3 from R3 =
VSUPPLY − VREF
.
I REF
2. Calculate R1 from R1 = k ⋅ VBIAS where k should be chosen between 2 and 10 to ensure that
I REF
the bias current is only a fraction (1/k) of IREF. The value of k you choose will be dependent
on how stable VSUPPLY is.
⎛⎛
VREF
3. Finally calculate R2 from R 2 = R1⎜ ⎜
⎜⎜ I
⎝ ⎝ REVERSE ⋅ RS
⎞ ⎞
⎟⎟ − 1⎟
⎟
⎠ ⎠
In the final analysis, as in most designs, a one-time tweak of VBIAS may be required for the design.
5.11 Reverse transient
If the reverse current is of a relatively short duration compared to the signal of interest, it can be
considered to be a transient. In that case it may be possible instead to use an RC network to filter
out this transient and thus prevent it from interacting with the current monitor's operation as
shown in Figure 33 and Figure 34 below. This method could be used for any of the current
monitors and these two examples demonstrate that specifically for the ZXCT1050 and generically
for all others.
ILOAD
RS
R1
RG
GT
C1
C1
VCC
S+
SVCC ZXCT1050
GND
OUT
S+
SZXCT10xx
GND
OUT
IOUT
IOUT
VOUT
VOUT
RG
Figure 33
ILOAD
RS
VSUPPLY
VSUPPLY
RG
RC noise suppression for ZXCT1050
Figure 34 RC noise suppression for a
generic current monitor
Notice in Figure 33 that the transconductance resistor is also used as the snubber resistor whilst
in most other cases this would be an additional component.
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VSUPPLY
RS
3
2
S-
S+
ZXCT1008 / 9
OUT
1
Inductive
Load
IOUT
VOUT
RG
Figure 35
Example of reverse VSENSE in a current monitor
5.0V
OUTPUT
0V
SEL>>
-5.0V
V(IOUT)
500mV
INPUT (Vsense)
0V
-500mV
0s
20us
40us
60us
80us
100us
120us
140us
160us
V(IN+,IN-)
Time
Figure 36 Response with only a 2.5mV of reverse sense voltage
Bottom trace = VSENSE; Top trace = Output; freq = 20kHz
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5.0V
OUTPUT
0V
SEL>>
-5.0V
V(IOUT)
500mV
INPUT (Vsense)
0V
-500mV
0s
20us
40us
60us
80us
100us
120us
140us
160us
V(IN+,IN-)
Time
Figure 37 Response with a 5mV reverse sense voltage
Bottom trace = VSENSE; Top trace = Output; freq = 20kHz
5.0V
OUTPUT
0V
SEL>>
-5.0V
V(IOUT)
500mV
INPUT (Vsense)
0V
-500mV
0s
20us
40us
60us
80us
100us
120us
140us
160us
V(IN+,IN-)
Time
Figure 38 Response with a 50mV reverse sense voltage
Bottom trace = VSENSE; Top trace = Output; freq = 20kHz
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AN39
VSUPPLY
10V
RS 50mΩ
R1
2.6k
3
2
S-
ZTLV431
1.24V
REF
S+
ZXCT1008 / 9
OUT
R2
10k
1
Inductive
Load
IOUT
R3
10k
VOUT
RG
1k
Figure 39
Bias applied to S- pin to prevent negative VSENSE
8.0V
4.0V
OUTPUT
0V
V(IOUT)
500mV
INPUT (Vsense)
0V
SEL>>
-500mV
0s
V(IN+,IN-)
20us
V(R2:2,IN-)
40us
60us
80us
100us
120us
140us
160us
Time
Figure 40
A much improved response by introducing an offset voltage onto the S- pin.
Bottom trace = VSENSE; Top trace = Output; freq = 20kHz
Notice the additional superimposed straight line trace on the bottom plot representing the offset
voltage on pin S- with respect to S+.
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6
Appendix - Current monitor summary chart
Table 2
Monitor
Output type
Current
Current monitor summary/selection chart
Voltage gain2
Sensing
rail1
x10
LS
x100
External
Resistors
Special feature
Target
application
Package
HS
ZXCT1008
•
•
1
Simplest COCM3 +
Extended temperature range
General
SOT23
ZXCT1009
•
•
1
Simplest COCM
General
SOT23 or
SM8
ZXCT1010
•
•
1
Ground pin4
Battery
discharge/
General
SOT23-5
ZXCT1011
•
•
2
Ground pin4 + External
transconductance resistor + Extended
temperature range
General
SOT23-5
ZXCT1012
•
•
1
Ground pin4
Battery
discharge/
General
TSOT23-5
TDFN3-5
ZXCT1020
•
•
2
Ground pin4 + External
transconductance resistor + Extended
temperature range
General
SOT23-5
•
0
Simplest VOCM5
General
SOT23-5
•
0
Simplest VOCM
General
SOT23-5
•
0
High speed + Bandgap reference + OC
Comparator
General
SO8
•
2
Series transistor drive + Programmable
threshold + Output flag
In-rush or
over-current
protection
SO8
ZXCT1021
•
•
ZXCT1022
ZXCT1030
ZXCT1032
www.zetex.com
•
37
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Monitor
Output type
Current
Voltage gain2
x10
External
Resistors
Special feature
Target
application
Package
•
0
Bidirectional + Output flag
General
SOT23-5
•
•
2
Separate VCC pin + current gain (x20) +
Short circuit operation
General
SOT23-5
•
•
0
Separate VCC pin + Extended
temperature + Short circuit operation
General
SOT23-5
0
Separate VCC pin + 60V CM + Extended
temperature
Automotive
SOT23-5
0
Separate VCC pin + 40V CM + Extended
temperature
LS
•
ZXCT1041
ZXCT1050
x100
Sensing
rail1
•
ZXCT1051
•
ZXCT1080
•
ZXCT1081
•
HS
•
SOT23-5
NOTES;
1
LS = Low Side Sensing, HS = High Side Sensing.
2
Output/VSENSE.
3
Current Output Current Monitor.
4
Eliminates quiescent current error.
5
Voltage Output Current Monitor
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7
Recommended further reading
1. AN45 - High voltage current monitoring using the ZXCT series in power supplies
2. DN77 - Transient and noise protection for current monitors
8
Glossary of terms
COCM
Current-Output Current Monitor
SFCM
Special Function Current Monitor
VOCM
Voltage-Output Current Monitor
RHS
Right Hand Side
9
List of Figures
Figure 1 - Principle of optical and closed loop current monitoring . . . . . . . . . . . . . . . . . . . . . . . . 6
Figure 2 - Principle of magnetic current monitoring . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Figure 3 - Power dissipated vs current measured . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Figure 4 - Low side resistive method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Figure 5 - High side current measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Figure 6 - Simplest current output current monitor (COCM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 7 - Simplest voltage-output current monitor (VOCM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Figure 8 - Transient protection for device . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 9 - Transient protection for device and load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 10 - Better transient protection for device and load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 11 - Simplest supply range extension . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Figure 12 - Improved supply range extension . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Figure 13 - Best supply range extension . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Figure 14 - Bi-directional current monitoring . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Figure 15 - Output response for bi-directional COCM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Figure 16 - Measuring bidirectional motor current in a full bridge driver . . . . . . . . . . . . . . . . . . 22
Figure 17 - Measuring bidirectional motor current with anti-parallel current monitors . . . . . . . 22
Figure 18 - Over-current protection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Figure 19 - Latching over-current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Figure 20 - Replace Q5 in Figure 19 above with this network to extend the trip
current range up to 2A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Figure 21 - The ZXCT1030 VOCM with comparator and bandgap reference . . . . . . . . . . . . . . . . 25
Figure 22 - A much reduced version of Figure 19 using the ZXCT1030 . . . . . . . . . . . . . . . . . . . . 26
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Figure 23 - ZXCT1030 in a motor control application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Figure 24 - Motor current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 25 - Driving optocoupler for isolated application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 26 - Basic ZXCT1050 configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 27 - Basic ZXCT1051 configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 28 - Extending the CM range of the ZXCT1050 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Figure 29 - Test circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Figure 30 - Output response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Figure 31 - Using non-precision resistors to extend CM range . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Figure 32 - Using the ZXCT1050 at a higher voltage (alternative to using the schemes
in Figure 12 and Figure 13) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Figure 33 - RC noise suppression for ZXCT1050 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Figure 34 - RC noise suppression for a generic current monitor . . . . . . . . . . . . . . . . . . . . . . . . . 33
Figure 35 - Example of reverse VSENSE in a current monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Figure 36 - Response with only a 2.5mV of reverse sense voltage . . . . . . . . . . . . . . . . . . . . . . . 34
Figure 37 - Response with a 5mV reverse sense voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Figure 38 - Response with a 50mV reverse sense voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Figure 39 - Bias applied to S- pin to prevent negative VSENSE . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Figure 40 - A much improved response by introducing an offset voltage onto the S- pin. . . . . 36
10 List of Tables
Table 1 - Common current measuring methods . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Table 2 - Current monitor summary/selection chart . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
11 List of Equations
Equation 1. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Equation 2. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Equation 3. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Equation 4. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Equation 5. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Equation 6. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Equation 7. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Equation 8. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Equation 9. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
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Equation 10. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Equation 11. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Equation 12. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Equation 13. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Equation 14. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Equation 15. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Equation 16. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Equation 17. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Equation 18. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Equation 19. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Equation 20. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Equation 21. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Equation 22. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Equation 23. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
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Definitions
Product change
Zetex Semiconductors reserves the right to alter, without notice, specifications, design, price or conditions of supply of any product or
service. Customers are solely responsible for obtaining the latest relevant information before placing orders.
Applications disclaimer
The circuits in this design/application note are offered as design ideas. It is the responsibility of the user to ensure that the circuit is fit for
the user’s application and meets with the user’s requirements. No representation or warranty is given and no liability whatsoever is
assumed by Zetex with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights
arising from such use or otherwise. Zetex does not assume any legal responsibility or will not be held legally liable (whether in contract,
tort (including negligence), breach of statutory duty, restriction or otherwise) for any damages, loss of profit, business, contract,
opportunity or consequential loss in the use of these circuit applications, under any circumstances.
Life support
Zetex products are specifically not authorized for use as critical components in life support devices or systems without the express written
approval of the Chief Executive Officer of Zetex Semiconductors plc. As used herein:
A. Life support devices or systems are devices or systems which:
1. are intended to implant into the body
or
2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the
labelling can be reasonably expected to result in significant injury to the user.
B. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to
cause the failure of the life support device or to affect its safety or effectiveness.
Reproduction
The product specifications contained in this publication are issued to provide outline information only which (unless agreed by the
company in writing) may not be used, applied or reproduced for any purpose or form part of any order or contract or be regarded as a
representation relating to the products or services concerned.
Terms and Conditions
All products are sold subjects to Zetex’ terms and conditions of sale, and this disclaimer (save in the event of a conflict between the two
when the terms of the contract shall prevail) according to region, supplied at the time of order acknowledgement.
For the latest information on technology, delivery terms and conditions and prices, please contact your nearest Zetex sales office .
Quality of product
Zetex is an ISO 9001 and TS16949 certified semiconductor manufacturer.
To ensure quality of service and products we strongly advise the purchase of parts directly from Zetex Semiconductors or one of our
regionally authorized distributors. For a complete listing of authorized distributors please visit: www.zetex.com/salesnetwork
Zetex Semiconductors does not warrant or accept any liability whatsoever in respect of any parts purchased through unauthorized sales channels.
ESD (Electrostatic discharge)
Semiconductor devices are susceptible to damage by ESD. Suitable precautions should be taken when handling and transporting devices.
The possible damage to devices depends on the circumstances of the handling and transporting, and the nature of the device. The extent
of damage can vary from immediate functional or parametric malfunction to degradation of function or performance in use over time.
Devices suspected of being affected should be replaced.
Green compliance
Zetex Semiconductors is committed to environmental excellence in all aspects of its operations which includes meeting or exceeding
regulatory requirements with respect to the use of hazardous substances. Numerous successful programs have been implemented to
reduce the use of hazardous substances and/or emissions.
All Zetex components are compliant with the RoHS directive, and through this it is supporting its customers in their compliance with
WEEE and ELV directives.
Product status key:
“Preview”
Future device intended for production at some point. Samples may be available
“Active”
Product status recommended for new designs
“Last time buy (LTB)”
Device will be discontinued and last time buy period and delivery is in effect
“Not recommended for new designs” Device is still in production to support existing designs and production
“Obsolete”
Production has been discontinued
Datasheet status key:
“Draft version”
This term denotes a very early datasheet version and contains highly provisional information, which
may change in any manner without notice.
“Provisional version”
This term denotes a pre-release datasheet. It provides a clear indication of anticipated performance.
However, changes to the test conditions and specifications may occur, at any time and without notice.
“Issue”
This term denotes an issued datasheet containing finalized specifications. However, changes to
specifications may occur, at any time and without notice.
Zetex sales offices
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Corporate Headquarters
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Telefon: (49) 89 45 49 49 0
Fax: (49) 89 45 49 49 49
[email protected]
Zetex Inc
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USA
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Zetex Semiconductors plc
Zetex Technology Park, Chadderton
Oldham, OL9 9LL
United Kingdom
Telephone: (1) 631 360 2222
Fax: (1) 631 360 8222
[email protected]
Telephone: (852) 26100 611
Fax: (852) 24250 494
[email protected]
Telephone: (44) 161 622 4444
Fax: (44) 161 622 4446
[email protected]
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