cd00004271

AN1453
APPLICATION NOTE
®
NEW FAMILY OF 150V POWER SCHOTTKY
By F. GAUTIER
INTRODUCTION
Nowadays, the Switch Mode Power Supply
(SMPS) is becoming more widespread as a result
of computer, telecom and consumer applications.
The constant increase in services (more
peripherals) and performance, which offers us
these applications, tends to move conversion
systems towards higher output power.
In addition to these developments dictated by the
market, SMPS manufacturers are in competition,
their battlefield being the criteria of power density,
efficiency, reliability and cost, this last being factor
very critical.
Today, SMPS designers of 12V-24V output have
practically the choice between a 100V Schottky or
a 200V bipolar diode.
The availability of an intermediate voltage has
become necessary to gain in design optimization.
In the following examples, the conduction losses
between a 150V Schottky and a 200V bipolar
diode in a Flyback and a Forward converter will be
compared.
The conduction losses in the diode are calculated
from the classical formula:
2
Pcond = VT0 ⋅ IF(AV) + Rd ⋅ IIF(RMS)
Vt0 :threshold voltage with VF(@ IF) = VT0 + Rd .IF
Rd : dynamic resistance with Rd = ∆VF / ∆IF
where VT0 and Rd are calculated from the current
range of current view by the diode (Fig. 1), for
better accuracy.
Figure 1 shows also, the typical current through
the rectification diode and the corresponding IF(AV)
and I2IF(RMS) :
Fig. 1: Typical current through a rectification diode
This is why STMicroelectronics is introducing a
new family of 150V POWER SCHOTTKY diodes,
intended for 12V and more secondary rectification,
in applications such as desktops, file servers or
adaptors for notebook.
Consequently, this application note will underline
the advantages of a 150V Schottky technology
compared to a 200V ultra fast diode.
In order to do this, the example of a Flyback
converter will be used, and the static and dynamic
parameters of the 150V Schottky will be detailed,
as well as their influence in this converter.
1. CONDUCTION LOSSES & EFFICIENCY GAIN
Schottky diodes are mainly used for output
rectification. In a typical SMPS working with a
switching frequency lower than 100kHz,
conduction losses are generally the main losses in
the diode. They are directly linked to the curve of
forward voltage (VF) versus forward current (IF),
and obviously the best gain in efficiency will be
obtained with the lowest VF .
July 2001
ID
Ima x
Imin
t
0
αID.T
T
α ID
(Imax + Imin )
2
α
2
I2F(RMS) = ID (I2max + Imin
+ Imax ⋅ Imin )
3
V
− VF(@ Imin)
Rd = F(@ Imax)
VT0 = VF(@ imax) − Rd ⋅ Imax
Imax − Imin
IF(AV) =
NB:
-In the datasheet, the VT0 and Rd are maximum
values given for IF and 2 IF at 125°C.
-In discontinuous mode Imin=0.
1/9
APPLICATION NOTE
1.1. Example 1: FLYBACK
1.2. Example 2: FORWARD
The first example is a 24V/48W Flyback converter
working in continuous mode (Vmains=90V) with
the following conditions:
α ID = 0.4, Imax ID = 6.66A, Imin ID = 3.33A, Iout = 2A
In the following example, the conduction losses in
a 12V/96W Forward converter are simulated:
Fig. 4: Rectification diode in a Forward converter
ID
Io u t
IL Io u
D1
Fig. 2: Rectification diode in a Flyback converter
Vin
Vin
D2
α D1 = 0.3, ILmax = 9A, ILmin = 7A, Iout = 8A
Calculations per diode give:
IF(AV)per diode = 1A and IF(RMS) per diode = 1.6A
Calculations per diode give:
IF(AV)D1 = 2.4A, IF(RMS)D1 = 4.39A
We can now calculate the efficiency gain (∆η(%)=
ηref - η) for this Flyback converter which has a
reference (ref) efficiency of 85% with
STPR1020CT:
Fig. 3: Example of efficiency gain in Flyback
converter
Pout=48W
Vout=24V
VT0 Rd
typ(V) mΩ
1.5A, 3A,
125°C
Pcond
(W)
∆P
(W)
η=85
%
∆η%
STPR102CT
2x5A / 200V
PN diode
0.58 46.5
1.4
0 (ref) 0 (ref)
STPR162CT
2x8A / 200V
PN diode
0.54 46.5
1.32
-0.08 +0.12
STPS10150CT
2x5A / 150V
0.50
Schottky diode
43
1.22
-0.18 +0.27
STPS16150CT
2x8A / 150V
0.47
Schottky diode
40
1.14
-0.26 +0.39
2/9
IF(AV)D2 = 5.6A, IF(RMS)D2 = 6.71A
The difference of efficiency between a
STPR1620CT (2x8A, 200V Ultrafast) and a
STPS16150CT (2x8A, 150V Schottky) for a 12V
output, are given in table Fig. 5:
Fig. 5: Example of efficiency gain in Flyback
converter
Pout=96W
Vout=12V
STPR1620CT
VT0 Rd
typ(V) mΩ
7A, 9A,
125°C
Pcond
(W)
∆P
(W)
Ref
0.8
20
6.48
STPS16150CT 0.68
20
5.60
η=85
%
∆η%
Ref
-0.95 +0.72
These two examples show that whatever the type
of converter, a significant efficiency gain can be
achieved only by replacing a 200V bipolar diode by
a 150V Schottky.
APPLICATION NOTE
2. REVERSE LOSSES AND TJMAX
2.1. Reverse losses: Prev
The reverse losses can be determined by:
Prev = VR ⋅ IR ⋅ (1 − α )
with:
(1- ): duty cycle when the reverse voltage (VR) is
applied
IR: leakage current versus VR and operating
junction temperature (Tj)
VR: reapplied voltage accross the diode
Fig. 6 shows an example of reverse losses in a
Flyback converter with the following conditions:
(1− α ) = 0.4, VR = 80V, Tj = 125 ° C
Fig. 6: Example of reverse losses in a Flyback
converter
STPS10150CT
IRtyp per diode
100V, 125°C
Prev
per diode
130µA
4.2mW
Thus, the reverse losses are very low due to the
low value of the leakage current.
The following paragraph will show that due to
these low values of reverse current, the thermal
runaway limit is only reached for high junction
temperature.
2.2. Tjmax before thermal instability is reached
Remembering that the stability criterion is given
by:
dPrev
1
<
dTj
Rth(j − a)
Example:
Flyback converter with 2 diodes in parallel
(1− α ) = 0.4, c = 0.069, VR = 80V
Rth(j − c)total = 2.4 ° C / W, Rth(c − a) = 7.6 ° C / W
Fig. 7: Example of Tjmax with STPS10150CT
For a dual
diode
IR(VR,Tjmax)
IRmax
(80V, 125°C)
STPS10150CT
1.3mA
Tjmax
45.28mA 176.5°C
This example shows that in a typical application, a
150V Schottky can be used up to 175°C.
STMicroelectronics specifies in the datasheet
Tjmax at 175°C.
3. SWITCHING BEHAVIOUR
3.1. Turn-on behaviour
The behaviour at turn-on is characterized by a low
value of peak forward voltage (VFP) and forward
reverse recovery time (tfr) (Fig. 8).
Fig. 8: VFP and tfr for STPS16150CT
IF=16A
dIF/dt=100A/µs
Tj=25°C
Per diode
STPS16150CT
tfr
(ns)
VFP
(V)
100
2.2
These values depends mainly on the dIF/dt. The
switching losses at turn-on are always negligible.
with:
Prev = VR .IR(VR,Tjmax) .(1 − α )
The above formulae give the critical value of the
leakage current before the thermal runaway limit is
reached:
1
IR(VR,Tjmax) =
VR ⋅ c.Rth(j − a) ⋅ (1 − α )
The evolution of the leakage current versus Tj and
VR is given by:
IR(VR ,Tj) = IR(VR ,125) exp c(Tj − 125)
From these physical laws, it can be deduced that:
Tjmax = 125 +
3.2. Turn-off behaviour
The turn-off behaviour is a transitory phenomenon
(ns), but repetitive depending on the switching
frequency. It is a source of spike voltage, noise
and for high switching frequency, of non-negligible
switching losses.
In order to illustrate this phenomenon, the example
of a Flyback converter will be used once again.
The difference in behaviour between a 150V
Schottky and 200V bipolar diode will be compared
for the three following points: spike voltage, EMC
and switching losses.
IR(VR ,Tjmax)
1
⋅ In
c
IR max (VR ,125 ° C)datasheet
3/9
APPLICATION NOTE
3.2.1. Difference of spike voltage between a
150V Schottky and 200V PN diode
In a Flyback converter, the reverse voltage (VR
used in §2) across the diode will be maximum, for
the maximum mains voltage (VINmax):
n
VR = VINmax ⋅ s + Vout
np
3.2.1.1 Turn-off behaviour for a PN diode
In the datasheet are specified the main turn-off
parameters (Qrr, IRM, trr…). These parameters are
represented in Fig. 10:
Fig. 10: Key parameters at turn-off for a bipolar
diode without snubber
(cf Fig. 9)
I
In addition to this nominal reverse voltage (VR),
generally an overvoltage spike at the turn-off of the
diode is observed (Fig. 9). It can be shown that
with a conventional bipolar diode, this spike is
more important for a Flyback converter working in
continuous mode than in a discontinuous mode.
In the case of a high spike voltage, the Maximum
Repetive Reverse Voltage (VRRM) of the diode has
to be oversized, compared with the real need (VR)
defined in Fig. 9.
To limit this peak and to preserve a "guard band"
with the VRRM (in order to avoid reaching the
breakdown voltage), the designer places a
snubber circuit (RS, CS) in parallel with the diode.
Generally, the "guard band" is such that the
maximum voltage reapplied to the diode does not
exceed 80% of the VRRM.
Fig. 9: Spike voltage across the rectification diode
Rs Cs
np
Vin
ID
VD
ns
ID
Vs
VP
Vo u t
VD
VR = VIN
Turn-off diode
dI F/dt
dI R/dt
ta
V
tb
t
Qrrb
Qrra
VR
IRM
VRma x
The following oscillogram shows the turn-off
behaviour for a bipolar diode (STPR1620CT) with
snubber and without snubber, in a 24V/45W
Flyback working in continuous mode.
To observe the phenomenon correctly, it is
necessary to compensate the delay time between
the voltage and the current, (by temporal shift) due
to the measuring equipment Fig. 11.
Fig. 11: Switching behaviour of a 200V bipolar
diode
dIF/dt=130A/µs
IRM
Tj=100°C
t0
ns
+ Vout
np
VRmax
VRRM
I
dIR/dt=600A/µs
IRM=4A
V
Compensative curve
VRmax=250V
This spike voltage is due to the leakage inductance
of the transformer (Lf) and to the nature of the
recovery charge of the diode, which itself depends
on the diode technology: bipolar diode or Schottky
diode.
V
I
delay time
t0
Rs=22ohms
Cs=2.2nF
I
dV/dt
V
I
VR=42V
VRmax=90V
4/9
Qrr = Qrra + Qrrb
trr = ta + tb
t
S= b
ta
V
20V/div
2A/div
50ns/div
APPLICATION NOTE
Without a snubber, in this example the diode is
repeatedly in conduction because the oscillation is
very strong. Furthermore, the voltage is close to
the breakdown voltage. This means that the
system is no longer reliable and a snubber circuit is
necessary.
On these 2 oscillograms, we can see that the value
of the maximum reverse current (IRM) is defined
when the reverse voltage rises (typical behaviour
of a bipolar diode). At this time the voltage is not
fixed by the diode.
The curve Qrr, IRR versus dIF/dt and Tj is given in
the datasheet. For example in Fig. 12, the
evolution of IRM versus dIF/dt for a STPR1620CT
can be observed.
Fig. 12: Peak reverse recovery current versus
dIF/dt (per diode)
STPR1620CG/CT
IRM(A)
20
IF=IF(av) 90% confidence
Tj=125°C
10
1
10
20
50
100
dIF/dt(A/µs)
200
500
It can be also noticed, that the parameter IRM
significantly increases with the temperature.
In continuous mode the dIF/dt (few hundred A/µs)
is fixed by the leakage inductance and the reverse
voltage (VR):
dIF VR
n
with VR = s ⋅ VIN + Vout
=
dt
Lf
np
It is many time higher than in discontinuous mode
(lower than 1A/µs):
dIF
Vout
=
withL S 〉〉 L f
dt
LS + Lf
(LS: Secondary inductance)
Thus, with this curve we can see that, in
continuous mode (high dIF/dt), the bipolar diode
must evacuate a non-negligible charge, which
means a higher IRM. This is verified on oscillogram
Fig. 11.
With this value of IRM, an equivalent model at t0
with a snubber circuit can be established:
Fig. 13: Equivalent model at t0 for a bipolar diode
Rs Cs
np
LP
ns
Ls
Lf
VD
IL =I
f RM
Vo u t
Cs
Vs
CQrrb
Cj
Rs
VD
VR = V S + V o u t
Where:
Vs: secondary voltage
ns
⋅ VIN
np
Lf: leakage inductance of the transformer
Cj: junction capacitance
CQrrb: equivalent capacitance modeling the
reverse charge, necessary for the establishment of
the potential barrier, which supports the reverse
voltage.
Vout: output voltage
VS =
With the following initial conditions at t=t0:
IL f = IRM bipolar and VD ≈ 0
The equivalent schematic can be used to define
VD = VR max
NB:
1) Without snubber, there is a L f , C circuit
( C = C j + C Qrrb ) which lead to a second order
differential equation:
d 2 VC
+ ω 20 ⋅ VC + ω 20 ⋅ VR = 0 and ω 20 = 1/ L f ⋅ C
dt 2
with initial conditions at t=t0:
IL f = IRM and VC 0 = VD = 0
In this equation, an approximation is made with C
constant, because in reality Cj and CQrrb vary with
the voltage applied.
The solution of the differential equation gives us:
VR max = VD = VR +
VR
2

Lf 
+  IRM ⋅

C 

2
Therefore we can see that the VRmax depends the
leakage inductance (Lf) and on recovery charge
(IRM). Thus, VR max is very dependent on the
temperature.
5/9
APPLICATION NOTE
If C is a low value of capacitance, the expression
leads to a first order differential equation:
VR max = VD = VR + L f ⋅ dIR / dt
with:
Fig. 14: Turn-off for a perfect Schottky diode
without snubber
ID
dIR / dt = IRM / t b and t b = Q rrb / (IRM / 2 )
dIR / dt = IRM / (2 ⋅ Q rrb )
2
If the diode is of a snap-off type, we have
t b → 0, Q rrb → 0 and dIR/dt is very high,
consequently VR max is considerable. There is a risk
to reach the breakdown voltage of the diode. (It is
not guaranteed by the manufacturer)
dt 2
+ 2mω 0
dVC S
dt
+ ω 20 ⋅ VR = ω 02 .VR
where:
m: the absorption coefficient
0:
natural frequency ω 0 =
m=
RS
2
CS
Lf
1
Lf ⋅CS
There are 3 possible cases:
- m>1 behaviour without oscillation (over damping)
- m<1 behaviour with oscillation damped (under
damping), where the frequency oscillation can be
determined by:
ω r = ω 0 1 − m2
- m=1 limit of behaviour without oscillation (critical
damping)
t
VR
2) With a snubber, if it is supposed that
C S >> C j + C Qrrb (more true for an ultrafast PN
diode), we have a simple Rs, Lf, CS circuit to define
by the second order differential equation:
d 2 VC S
t0
VD
VRma x
In this perfect case at t=t0, we have:
IL f = IRM = 0 and VD ≈ 0
Without a snubber circuit, the new solution of the
differential equation gives:
VD = VR max = 2 ⋅ VR
Unfortunately, it is difficult to realize a perfect high
voltage Schottky. The reason is the presence of a
parasitic bipolar diode in parallel with the Schottky.
If it is polarized, the recovery charge is added at
the turn-off. The phenomenon begins to appear for
a 100V technology.
In the same conditions as before, the
STPS16150CT is used:
Fig. 15: Switching behaviour of a STPS16150CT
dIF/dt=130A/µs
Tcase=100°C
t0
dIR/dt=250A/µs
I
I
V
IRM=1.6A
Like this, in this case, it is possible to suppress the
oscillation across the diode with m>1.
V
VRmax=120V
3.2.1.2 Turn-off behaviour for a Schottky diode
For an ideal Schottky diode, there is no recovery
charge (Qrr=0). Therefore, it is said that the diode
has a capacitive type recovery Fig. 14.
Rs=22ohms Cs=2.2nF
I
t0
I
VR=42V
VRmax=58V
dV/dt
6/4
V
V
20V/div
2A/div
50ns/div
APPLICATION NOTE
It can be observed that this is not an ideal
Schottky. In fact, when the voltage rises at , we
have a value of IRM. The charge QrrB is not easily
identifiable because it is embedded in the
capacitive current.
However, the slope dIR/dt can be observed.
Unlike a PN diode, we can see that with the 150V
Schottky the maximum reverse voltage (VRmax)
and the maximum reverse current (IRM) are
distinctly lower.
The equivalent model at t0 for a STPR1620CT and
a STPS16150CT is the same, with the lower initial
conditions at t0:
IL f = IRM schottky and VD ≈ 0
In Fig. 16, we can see the curve Cj versus VR for
the 200V bipolar and 150V Schottky diode.
Whatever the reverse voltage, the junction
capacitance of the 150V Schottky is always higher
than for a PN diode. This justifies, the lower
observed with the Schottky diode.
Fig. 16: Cj versus VR
F=1MHz,Tj=125°C
STPR1620CT
STPS16150CT
300.0
Thus, a different efficiency of the snubber circuit
with a bipolar diode and a Schottky diode is
observed. In most cases, we can say that the 150V
Schottky behaves better at turn-off, due to its
larger capacity and its softness recovery.
Model showed in Fig. 13 can be used to define the
snubber circuit.
NB:
In the case of a Forward converter with multiple
outputs (12V, 5V, 3.3V…) and cross regulation
with coupled inductor, the poor behaviour at the
turn-off with a bipolar diode on 12V output, will be
reflected on the other coupled outputs (that means
an overvoltage on rectification diode of 5V output).
A 150V Schottky will decrease the coupled effects.
3.2.2. EMC Comparison between a 150V
Schottky and a 200V bipolar diode
The better the behaviour at turn-off of the 150V
Schottky in comparison with a 200V bipolar diode,
the better performance in the EMC.
Fig. 16 shows the comparison of electromagnetic
disturbance conducted in a 45W/24V Flyback
converter (in continuous mode) between a
STPR1620CT and a STPS16150CT with a
snubber circuit.
Fig. 17: Electromagnetic disturbances conducted
between a STPR1620CT and a STPS16150CT
250.0
C( pF)
200.0
150.0
STPR1620CT
100.0
50.0
0.0
1
10
100
1000
VR(V)
In summary, when we compare the different
parameters with those of the bipolar diode, we
have:
IRM bipolar > IRM schottky
C j bipolar < C j schottky
(dI / dt )
(dV / dt )
R
bipolar
bipolar
> (dIR / dt )
> (dV / dt )
VRmax bipolar > VRmax schottky
schottky
schottky
STPS16150CT
We can see near 30MHz, that there is a difference
of -10dB. This difference is partially explained by
the higher dV/dt with a bipolar diode at turn-off
than with a 150V Schottky. In fact, the lower
capacitance junction of the PN diode favors the
high dV/dt at t0, and therefore the common current
mode. ( i CM = C ⋅ dV / dt, C equivalent capacitance
junction-heatsink)
The other high dV/dt, which could take place due
to the strong oscillation, are suppressed by a good
choice of the snubber circuit.
7/9
APPLICATION NOTE
In the case of an EMC problem, the first solution is
to reduce the current slope (dIF/dt) by adding a
gate resistance (RG about 10 ohms). In this way,
IRM and dIR/dt decrease as well as the VR max .
3.2.3. Switching losses
We have evaluated the consequences of poor
behaviour at the turn-off: spike reverse voltage,
possible oscillations and EMC problem. For these
reasons, the designer may wish to use a soft
recovery PN diode, but which, in return, will
increase the switching time and particularly the
switching losses at turn-off.
Switching losses at turn-off due to the diode are
the sum of losses inside the diode and the energy
dissipated in the other elements of the circuit. In
fact, during this time the IRM current, due to the
recovery charge, flows through the transformer,
the power MOS transistor and the primary bulk
capacitor. Thus, there are additional losses. The
distribution of power losses at turn-off can be
detailed:
- Losses inside the diode:
1
Pturn − off diode = ⋅ t b ⋅ IRM ⋅ VR ⋅ F
2
- Losses due to the energy store in the leakage
inductance:
1
WLf = ⋅ L f ⋅ I2 RM
2
which is mainly dissipated in the snubber resistor
(RS).
- Losses due to the eddy current in the transformer
(view AN1262)
- Losses due to the all additional resistor of circuit,
defined by:
Pturn − off R = I2 RMS (IRM) ⋅ ∑ R
As described before (in §1), in a typical converter
working with a switch frequency lower than
100kHz, these different losses can be considered
negligible compared to the conduction losses.
However in applications such as the DC/DC
converter (12V-48V) working with a switching
frequency around 300kHz, these losses can be
predominant, and a 150V Schottky can be very
interesting to reduce the switching losses.
8/9
4. RESULT OF EXPERIMENTS
Experimental measurements (Fig. 18) were
carried out in a 45W/24V Flyback converter
working in the following conditions:
VIN = 90V Pout = 45W Vout = 24V
Fs = 100kHz Tc = 100°C
These experiments confirm the interest in a 150V
Schottky in comparison with a 200V bipolar diode.
Fig. 18: Experiments of efficiency in a Flyback
converter
η%
∆P(W)
STPR1620CT (2x8A)
84.04
0.41
STPS16150CT (2x8A)
84.4
0.18
STPS20150CT (2x10A)
84.69
Ref
CONCLUSION
We have been able to highlight that when we have
the choice between 150V Schottky and 200V PN
diode, the 150V Schottky is the best choice for the
safety of the component and the environment, the
limitation of parasitic effects and for the efficiency
of the converter.
In fact, in addition to the low VF, the 150V Schottky
has a better switching behaviour, due to its
essentially capacitive recovery (less sensibility to
the temperature). We have the advantage of a soft
recovery diode in terms of EMC and the Schottky
is preferable to a fast recovery diode in terms of
losses. The 150V Schottky diode is the better
choice versus the 200V bipolar as for EMC and
losses at turn-off are concerned. Experimental
measurements confirm this.
Moreover, future advancements will mean that this
product will be developed.
In fact, with the arrival of the EN6100-3-2 standard
and the introduction of the PFC, whatever the input
voltage is, there will be a continuous voltage on the
primary. This will lead to a reduction of the
transformation ratio, and in the same time, the
reverse voltage of the diode.
Consequently, a lower breakdown voltage diode
will be needed in the future to replace a 200V PN
diode used today.
Also, the tendency is for the output power of
adaptors to increase. This involves an increase in
the output voltages. The voltage requirements of
the diode in this case will be higher than 100V and
a 150V diode is likely to be the appropriate
component.
APPLICATION NOTE
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of
use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by
implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to
change without notice. This publication supersedes and replaces all information previously supplied.
STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
The ST logo is a registered trademark of STMicroelectronics
© 2001 STMicroelectronics - Printed in Italy - All rights reserved.
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9/9
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