AN1335 Phase-Shifted Full-Bridge (PSFB) Quarter Brick DC/DC Converter Reference Design Using a dsPIC® DSC Author: Ramesh Kankanala Microchip Technology Inc. ABSTRACT This application note provides the digital implementation of a telecom input 36 VDC-76 VDC to output 12 VDC, 200W Quarter Brick DC/DC Brick Converter using the Phase-Shifted Full-Bridge (PSFB) topology. This topology combines the advantages of Pulse-Width Modulation (PWM) control and resonant conversion. The dsPIC33F “GS” family series of Digital Signal Controllers (DSCs) was introduced by Microchip Technology Inc., to digitally control Switched Mode Power Converters. The dsPIC33F “GS” family of devices consists of an architecture that combines the dedicated Digital Signal Processor (DSP) and a microcontroller. These devices support all of the prominent power conversion technologies that are used today in the power supply industry. In addition, the dsPIC33F “GS” family of devices controls the closed loop feedback, circuit protection, fault management and reporting, soft start, and output voltage sequencing. A DSC-based Switched Mode Power Supply (SMPS) design offers reduced component count, high reliability and flexibility to have modular construction to reuse the designs. Selection of peripherals such as the PWM module, Analog-to-Digital Converter (ADC), Analog Comparator, Oscillator and communication ports are critical to design a good power supply. MATLAB® based simulation results are compared to the actual test results and are discussed in subsequent sections. INTRODUCTION Recently, Intermediate Bus Converters (IBCs) have become popular in the telecom power supply industry. Most telecom and data communication systems contain ASIC, FPGAs and integrated high-end processors. These systems require higher currents at multiple lowlevel voltages with tight load regulations. Traditionally, bulk power supplies deliver different load voltages. In the conventional Distributed Power Architecture (DPA), the front-end AC/DC power supply generates 24V/48V and an individual isolated Brick Converter supports the required low system voltages. These systems become inefficient and costly where very low voltages are © 2010 Microchip Technology Inc. required. In the Intermediate Bus Architecture (IBA), the IBC generates 12V/5V. Further, these voltages are stepped down to the required load voltages by Point of Loads (PoLs). In IBA, the high-density power converters, IBC and PoLs are near to the load points, which bring considerable financial gains with the improved performance. Because these converters are at the load points, PCB design will be simpler with reduction in losses. Electromagnetic Interference (EMI) is also considerably reduced due to minimum routing length of high current tracks. Due to the position of these converters, the transient response is good and the system performance is improved. Modern systems require voltage sequencing, load sharing between the converters, external communication and data logging. Conventional Switched Mode Power Supplies are designed with Analog PWM control to achieve the required regulated outputs, and an additional microcontroller performs the data communication and load sequencing. To maximize the advantages of IBC, the converter must be designed with reduced component count, higher efficiency, and density with lower cost. These requirements can be achieved by integrating the PWM controller, communication and load sharing with the single intelligent controller. The dsPIC33F “GS” family series of DSCs have combined these design features in a single chip that is suitable for the bus converters. Some of the topics covered in this application note include: • DC/DC power module basics • Topology selection for the Quarter Brick DC/DC Converter • DSC placement choices and mode of control • Hardware design for the isolated PSFB Quarter Brick DC/DC Converter • Planar magnetics design • Digital PSFB Quarter Brick DC/DC Converter design • Digital control system design • Digitally controlled load sharing • MATLAB modeling • Digital nonlinear control techniques • Circuit schematics and laboratory test results • Test demonstration DS01335A-page 1 AN1335 DISTRIBUTED POWER ARCHITECTURE (DPA) Isolation Barrier FIGURE 1: Load DC/DC Brick Converter 3.3 VDC Load Load AC/DC Power Supply 24V/48V Bus DC/DC Brick Converter 2.5 VDC Load Load DC/DC Brick Converter 1.8 VDC Load INTERMEDIATE BUS ARCHITECTURE (IBA) Isolation Barrier FIGURE 2: AC/DC Power Supply 24V/48V Bus Intermediate Bus Converter (IBC) 1.3 VDC PoL Load 1.8 VDC PoL 12V/5V Bus Load 1.5 VDC Load PoL 1.2 VDC PoL Load 1.0 VDC PoL Load 0.8 VDC PoL Load PoL = Point of Load DS01335A-page 2 © 2010 Microchip Technology Inc. AN1335 QUARTER BRICK CONVERTER Remote ON/OFF Control The Distributed-Power Open Standards Alliance (DOSA) defines the specifications for the single output pin Quarter Brick DC/DC Converter. These specifications are applicable to all Quarter Bricks (unregulated, semi-regulated and fully regulated) for an output current range up to 50A. Remote ON/OFF control is used to enable or disable the DC/DC converter through an external control signal. The most common method to enable or disable the converter is from the primary side (input side). Because the controller exists in the secondary side of the isolated barrier, an isolation circuit must be used to transfer the signal from the primary side to the secondary side. This can be achieved using the opto-isolator, which is illustrated in Figure 3. The AC/DC converter output is 48V in the IBA. This voltage is further stepped down to an intermediate voltage of 12V by an isolated IBC. This voltage is further stepped down to the required low voltage using PoL. DOSA Quarter Brick DC/DC converters are offered in through-hole configurations only. FIGURE 3: REMOTE ON/OFF Remote ON/OFF – (I/P) Some advantages of the Quarter Brick Converter are: • • • • • Improved dynamic response Highest packaging density Improved converter efficiency Isolation near the load end Output voltage ripple below the required limit +3.3V ANA R U Remote ON/OFF Signal to DSC C 1 DC/DC POWER MODULES BASICS R Before discussing the design aspects of the Quarter Brick Converter, the following requirements should be understood: • • • • • • • • Input Capacitance Output Capacitance Remote ON/OFF Control Ripple and Noise Remote Sense Forced Air Cooling Overvoltage Overcurrent Input Capacitance For DC/DC converters with tight output regulation requirements, it is recommended to use an electrolytic capacitor of 1 µF/W output power at the input to the Quarter Brick Converter. In the Quarter Brick Converter designs, these capacitors are external to the converter. Output Capacitance To meet the dynamic current requirements and the output voltage regulations at the load end, additional electrolytic capacitors must be added. As a design guideline, in Quarter Brick Converter designs, 100 µF/A to 200 µF/A of output current can be added and an effective lower Equivalent Series Resistance (ESR) can be achieved by using a number of capacitors in parallel. © 2010 Microchip Technology Inc. C GND DIG_GND Ripple and Noise The output of a rectifier consists of a DC component and an AC component. The AC component, also known as ripple, is undesirable and causes pulsations in the rectifier output. Ripple is an artifact of the power converter switching and filtering action, and has a frequency of some integral multiple of the power converter operating switching frequency. Noise occurs at multiples of the power converter switching frequency, and is caused by a quick charge and discharge of the small parasitic capacitances in the power converter operations. Noise amplitude depends highly on load impedance, filter components and the measurement techniques. Remote Sense Remote sense can be used to compensate voltage drop in the set voltage when long traces/wires are used to connect the load. In applications where remote sensing is not required, the sense pins can be connected to the respective output pins. DS01335A-page 3 AN1335 Forced Air Cooling To remove heat from the high density board mount power supplies, forced air cooling is applied using a fan. Forced air cooling greatly reduces the required PCB size and heat sink. However, installation of a fan consumes additional power, causes acoustic noise and also the maintenance requirements are significant. In forced air cooling SMPS applications, reliability of the converter highly depends on the fan. A temperature sensing device is used to monitor the temperature and shuts down the converter when the Quarter Brick Converter exceeds the maximum operating temperature. Overvoltage Overvoltage protection is required to protect the load circuit from excessive rated voltage because of a malfunction from the converter’s internal circuit. This protection can be implemented by Latch mode or Cycle-by-Cycle mode. In Latch mode, the circuit will be in the OFF condition on the occurrence of overvoltage fault until the input voltage is cycled. The system automatically recovers in the Cycle-by-Cycle mode. If faults still exist in the system, the system is turned OFF and this cycle is repeated. Overcurrent Overcurrent protection prevents damaging the converter from short circuit or overload conditions. In Hiccup mode, the converter will be OFF when an overcurrent or short circuit occurs, and will recover in the specified time period. If the converter still sees the fault, it will turn OFF the converter again and this cycle repeats. In the Latch mode, the circuit is recovered only after recycling the input power. higher switching losses while the switch turns ON or OFF, which results in a reduction in the efficiency of the converter. Soft switching techniques are used to reduce the switching losses of the PWM converter by controlling the ON/OFF switching of the power devices. Soft switching can be done using the Zero Voltage Switching (ZVS) and Zero Current Switching (ZCS) techniques. These soft switching techniques have some design complexity and in turn, produce higher efficiency at high-power levels. Non-Isolated Forward Mode Buck Converter If the required output voltage is always less than the specified input voltage, the Buck Converter can be selected from the following three basic topologies: Buck, Boost and Buck Boost. The Buck topology can be implemented in the isolated and non-isolated versions. As per the bus converter specification requirement, isolated converter design is selected for this application. In the Forward mode Buck Converter, energy is transferred from the primary side to the secondary side when the primary side switch is turned ON. The output voltage can be controlled by varying the duty cycle with respect to the input voltage and load current. This is done with the feedback loop from the output that controls the duty cycle of the converter to maintain the regulated output. FIGURE 4: NON-ISOLATED FORWARD MODE BUCK CONVERTER Q1 L1 VOUT+ VIN+ D1 C1 TOPOLOGY SELECTION VOUT- VIN - The bus converter specifications are standardized, and are used or assembled as one of the components in the final system. The user must consider the end-system characteristics such as reliability, efficiency, foot prints and cost. There is no universally accepted topology for the bus converters. However, the following sections describe a few topologies that are commonly used for DC/DC converter applications with their pros and cons. A fundamental distinction among the PWM switching topologies is hard switching and soft switching/ resonant topologies. Typically, high frequency switching power converters reduce the size and weight of the converter by using small magnetics and filters. This in turn increases the power density of the converter. However, high frequency switching causes DS01335A-page 4 Isolated Forward Converter In the Forward Converter, the energy from the input to the output is transferred when the switch Q1 is ON. During this time, diode D1 is forward biased and diode D2 is reverse biased. The power flow is from D1 and L1 to output. During the switch Q1 OFF time, the transformer (T1) primary voltages reverse its polarity due to change in primary current. This also forces the secondary of T1 to reverse polarity. Now, the secondary diode, D2 is forward biased and freewheels the energy stored in the inductor during switch Q1 ON time. This simple topology can be used for power levels of 100W. Some of the commonly used variations in Forward © 2010 Microchip Technology Inc. AN1335 Converter topologies are active Reset Forward Converter, Two Transistor Forward or Double-ended Forward Converter. FIGURE 7: HALF-BRIDGE CONVERTER VIN+ D1 FIGURE 5: ISOLATED FORWARD CONVERTER D1 VIN+ L1 T1 L1 VOUT+ Q3 T1 VOUT+ C1 Q4 VOUTD2 D2 C1 VIN- Q1 VOUT- VIN- Push-Pull Converter The Push-Pull Converter is a two transistor topology that uses a tapped primary on the converter transformer T1. The switches Q1 and Q2 conduct their respective duty cycles and the current in the primary changes, resulting in a bipolar secondary current waveform. This converter is preferred in low input voltage applications because the voltage stress is twice the input voltage due to the tapped primary transformer. FIGURE 6: PUSH-PULL CONVERTER Full-Bridge Converter The Full-Bridge Converter is configured using the four switches: Q1, Q2, Q3 and Q4. The diagonal switches Q1, Q4 and Q2, Q3 are switched ON simultaneously. This provides full input voltage (VIN) across the primary winding of the transformer. During each half cycle of the converter, the diagonal switches Q1, Q4 and Q2, Q3 are turned ON, and the polarity of the transformer reverses in each half cycle. In the Full-Bridge Converter, at a given power compared to the HalfBridge Converter, the switch current and primary current will be half. This makes the Full-Bridge Converter suitable for high-power levels. FIGURE 8: FULL-BRIDGE CONVERTER VIN + Q1 D1 T1 - VIN D1 L1 VOUT + Q1 Q3 T1 Q2 Q4 + L1 VOUT + C1 C1 VOUT D2 VOUT D2 VIN - Q2 Half-Bridge Converter Half-Bridge converters are also known as two switch converters. Half the input voltage level is generated by the two input capacitors, C1 and C2. The transformer primary is switched alternatively between VIN+ and input return VIN- such that the transformer primary sees only half the input voltage (VIN/2). The input switches, Q1 and Q2, measure the maximum input voltage, VIN compared to 2 * VIN in the Push-Pull Converter. This allows the Half-Bridge Converter to use higher power levels. © 2010 Microchip Technology Inc. However, the diagonal switches are hard switched resulting in high turn ON and turn OFF switching losses. These losses increase with frequency, which in turn limits the frequency of the operation. To overcome these losses, the PSFB converter is introduced. In this topology, the switch turns ON after discharging the voltage across the switch. This eliminates the turn ON switching losses. DS01335A-page 5 AN1335 FIGURE 9: ZERO VOLTAGE SWITCHING (ZVS) FIGURE 10: VDS(t) FULL-BRIDGE CONVERTER WITH SYNCHRONOUS RECTIFICATIONS PWMH PWMH VDS ID(t) Q3 Q1 Q6 PWML ID TX TXVPRI ZVS t PWM PWML PWML t Q4 Q2 PWMH Q5 Synchronous Rectification In synchronous rectification, the secondary diodes, D1 and D2 are replaced with MOSFETs. This yields lower rectification losses because a MOSFET will have minimum DC losses compared to the Schottky rectifiers. The forward DC losses of a Schottky rectifier diode will be forward voltage drop multiplied by the forward current. The power dissipation by a conducting MOSFET will be RDS(ON) multiplied by the square of the forward current. The loss comparison will be significant at considerably higher current >15A and lower output voltages. This configuration involves complexity and cost to an extent because a gate drive circuit is required to control the synchronous MOSFET. The efficiency of this configuration can be further increased by designing the complex gate drive signals, which are discussed in the section “Digital Nonlinear Implementations”. Many topologies are available and one of them can be chosen depending on the given power level, efficiency of the converter, input voltage variations, output voltage levels, availability of the components, cost, reliability of the design, and good performance characteristics. With the discussed advantages for the topologies and efficiency considerations, the PSFB topology was selected for the Quarter Brick DC/DC Converter design. The operation, design and performance of this topology is discussed in following sections. TABLE 1: TOPOLOGY COMPARISON Topology No. of Switches in the Primary Stress Level of Primary Switches Power Levels (Typical) Forward converter 2 VIN 100W Push-Pull converter 2 2 * VIN 150W Half-Bridge converter 2 VIN Full-Bridge converter 4 VIN PSFB converter 4 VIN DS01335A-page 6 200W ~ 200W ~ 200W © 2010 Microchip Technology Inc. AN1335 PRIMARY SIDE CONTROL VS. SECONDARY SIDE CONTROL After selecting the topologies based on the merits for the given application, the next challenge faced by designers is to position the controller either on the primary or secondary side. The power converter demands the galvanic isolation between primary (input) and secondary (output load) due to safety reasons. There should not be any direct conductive path between the primary and secondary. Isolation is required when signals are crossing from the primary to the secondary and vice versa. The power path isolation will be given by the high frequency transformers. Gate drive signals can be routed through optocouplers or gate drive transformers. FIGURE 11: In the primary side controllers, the output feedback signal is transferred from the secondary to the primary using the optocouplers. These devices have limited bandwidth, poor accuracy, and tend to degrade over time and temperature. Again, the transfer of signals from the primary to the secondary or the secondary to the primary is dependant on the features demanded by the application. Figure 11, Figure 12 and Table 2 show the comparison between the primary side controller and the secondary side controller. The secondary side controller is selected in this application. SECONDARY SIDE CONTROL VOUT + VIN + 36V-76V 12V/17A Sync Rectifier PSFB MOSFET 200W VIN - VOUT Driver Current TX Drive TX Driver Drive TX Driver dsPIC® DSC Communication Remote Control OPTO VINOV VINUV 3.3V Reg NCP 1031 © 2010 Microchip Technology Inc. Auxiliary TX 3.8V 12V To Driver’s VCC DS01335A-page 7 AN1335 TABLE 2: PRIMARY SIDE CONTROL VS. SECONDARY SIDE CONTROL Primary Side dsPIC® DSC Control Secondary Side dsPIC® DSC Control Isolated feedback is required to regulate the output. A linear optocoupler can be used to achieve the regulation, which requires an auxiliary supply and an amplifier in the secondary. Isolated feedback is not required because the controller is on the secondary. Remote ON/OFF signal isolation is not required. Remote ON/OFF signal isolation is required. Isolation is required for communication signals. Isolation is not required for communication signals. Load sharing signal is transferred from the secondary to Load sharing isolation is not required because the the primary. controller is in the secondary. Overvoltage protection signal is transferred from the secondary to the primary. Isolation for overvoltage is not required because the controller is in the secondary. Frequency synchronization signal is transferred from the Isolation for frequency synchronization is not required secondary to the primary. because the controller is in the secondary. Input undervoltage and overvoltage can be measured without isolation. Isolation is required. However, in this application, the input undervoltage or overvoltage protection is provided by the NCP 1031 auxiliary converter controller. Gate drive design for the primary side switches is simple. Gate drive is transferred from the secondary to the primary either by using driver transformers or opto isolators. FIGURE 12: PRIMARY SIDE CONTROL VOUT+ VIN+ 36V-76V 12V/17A Sync Rectifier PSFB MOSFET 200W VIN- VOUT Current TX Drive TX Drive TX Driver1 Driver2 dsPIC® DSC I/P UV O/P OV To Driver’s ICs Remote ON/OFF Drive TX Linear OPTO Driver3 LM358 OPTO Communication +12V Reg 3.3V 3.8V Isolated 12V for Driver3 VIN+ NCP1031 Auxiliary TX VIN- DS01335A-page 8 © 2010 Microchip Technology Inc. AN1335 VOLTAGE MODE CONTROL (VMC) VS. CURRENT MODE CONTROL (CMC) FIGURE 14: The preference to implement VMC or CMC as the feedback control method is based on applicationspecific requirements. In VMC, change in load current will have effect on the output voltage before the feedback loop reacts and performs a duty cycle correction. In CMC, change in load current is sensed directly and corrects the loop before the outer voltage loop reacts. This cause and then react process in the VMC is slower to respond than in the CMC for highly varying load transients. The fundamental difference between VMC and CMC is that CMC requires accurate and high grade current sensing. In VMC, output voltage regulation is independent of the load current. Therefore, relatively low grade current sensing is enough for overload protection. This saves significant circuit complexity and power losses. TABLE 3: VMC AND CMC DIFFERENCES VMC CMC Single feedback loop. Dual feedback loop. Provides good noise margin. Poor noise immunity. Slope compensation required, instability at more than 50% duty cycles. Poor dynamic response. Good dynamic response. VOLTAGE MODE CONTROLLER (VMC) ID L IS S PWM1H + Comp - Ramp Generator EA + IO D C Ref L ID IS IO D S PWM1H C + Comp - EA + Ref HARDWARE DESIGN FOR THE ISOLATED QUARTER BRICK DC/DC CONVERTER The average Current mode control PSFB topology with secondary side controller was selected for this design. The digital Quarter Brick DC/DC Converter design is discussed in the following sections. High switching frequency and high voltage stress on the primary side transistors produce switching losses. PSFB transformer isolated buck converter attains zero voltage transition (ZVT) without increasing the MOSFET’s peak voltage stress. Slope compensation not required. IL IL Phase-Shifted Full-Bridge (PSFB) Converter Design Current measurement not Current measurement required for feedback. required. FIGURE 13: CURRENT MODE CONTROLLER (CMC) In Figure 15, MOSFET (Q1-Q4), body diodes (D1-D4) leakage output capacitance (COSS1-COSS4) inductance of the transformer are illustrated. Leakage inductance causes the full-bridge switching network to drive an effective inductive load, and results in ZVT on the primary side switching devices. The output voltage is controlled through a phase shift between the two half-bridge. Both halves of the bridge switch network operate with a 50% duty cycle and the phase difference between the half-bridge switch networks is controlled. A maximum duty cycle of 50% ensures that the gate drive transformer and gate drive circuit design will be simple. The ZVT is load related and at some minimum load, the ZVT will be lost. Linear output voltage control can be achieved by controlling the phase shift between the right leg and left leg of the bridge configuration. In ZVT, the switches are turned ON when the voltage seen by the switches are zero, resulting in no switch ON losses. Phase shift control of a Full-Bridge Converter can provide ZVT in the primary side which results in lower primary side switching losses and lower EMI losses. © 2010 Microchip Technology Inc. DS01335A-page 9 AN1335 Operation of the PSFB converter and detailed primary side waveforms with different time intervals are illustrated in Figure 15. FIGURE 15: PSFB CONVERTER WITH FULL WAVE SYNCHRONOUS RECTIFICATIONOPERATIONAL WAVEFORMS Q3 Q1 TX Q6 LL IPRI L0 VPRI C0 Q4 Q2 V0 Q5 Q1 Q2 Q3 Q4 VPRI IPRI t t0 DS01335A-page 10 t1 t2 t3 t4 © 2010 Microchip Technology Inc. AN1335 • Initial Conditions t0: Q1 = ON; Q4 = ON; Q2 = OFF; Q3 = OFF • Time interval t3 to t4: Q3 = ON; Q2 = ON; Q1 = OFF; Q4 = OFF; The PSFB converter operation is described with the power transfer from primary side to secondary side with the conduction of diagonal switches, Q1 and Q4. The primary side current (IPRI) was conducting through the switches, Q4 and Q1, but in this period, the full input voltage VIN is across the primary side of the transformer TX and VIN/N is across the secondary of the transformer. The slope of the current is determined by VIN, magnetizing inductance and the output inductance. In this time interval, both the diagonal switches Q3 and Q2 are ON and input voltage VIN is applied across the primary of transformer. The rate of rise of the current is determined by the input voltage VIN, magnetizing inductance and the output inductance. However, the current flows at negative value as opposed to zero. Now, the current flowing through the primary switches is the magnetizing current along with the reflected secondary current into the primary. • Time interval t0 to t1: Q1 =ON; Q4 = OFF; Q2 =OFF; Q3 = OFF Switch Q4 is turned OFF and Switch Q1 remains ON, the primary current continues to flow taking the Q4 switch output capacitor C4. This charges the capacitor C4 to VIN from 0V, at the same time the capacitor C3 of Switch Q3 is discharged because its source voltage rises to input voltage VIN. This transition puts Q3 with no drain to source voltage prior to turn ON and ZVS can be observed. Therefore there will not be any turn ON switching losses. During this transition period, the primary voltage of the transformer decreases from VIN to zero, and the primary no longer supplies power to the output. Simultaneously, the energy stored in the output inductor starts supplying the decaying primary power. • Time interval t1 to t2: Q1 = ON; Q3 = ON; Q4 = OFF; Q2 =OFF; D3 = ON The input voltage, the transformer turns ratio and output voltage determine the exact diagonal switch ON time. After the switch-on time period of the diagonal switches, Q3 is turned OFF at t4. One switching cycle is completed when the switch Q3 is turned OFF and the resonant transition to switch Q4 starts. In the PSFB converter, the left leg transition requires more time than the right leg transition to complete. The maximum transition time occurs for the left leg at minimum load current and maximum input voltage, while minimum transition time occurs for the right leg at maximum load current and minimum input voltage. To achieve ZVT for all the switches, the leakage inductor must store sufficient energy to charge and discharge the output capacitance of the switches in the allocated time. The energy stored in the inductor must be greater than the capacitive energy required for the transition. After Q3 output capacitance is charged to full input voltage VIN, the primary current free wheels through switch Q1 and body diode D3 of switch Q3. The current remains constant until the next transition occurs. Q3 can be turned ON any time after t1 and the current shares between the body diode D3 and the switch Q3 channel. HARDWARE DESIGN AND SELECTION OF COMPONENTS • Time interval t2 to t3: Q3 = ON; Q1 = OFF; Q4 = OFF; Q2 =OFF; Specifications At time t2, Q1 is turned OFF, the primary current continues to flow through the body diode, D1 of the switch Q1. The direction of the current flow increases the switch Q1 source to drain voltage, and voltage across the switch Q2 decreases from high to lower voltage. During this transition, the primary current decays to zero. ZVS of the left leg switches depending on the energy stored in the resonant inductor, conduction losses in the primary switches and the losses in the transformer winding. Because the left leg transition depends on leakage energy stored in the transformer, it may require an external series inductor if the stored leakage energy is not enough for ZVS. When Q2 is then turned ON in the next interval, voltage VIN is applied across the primary in the reverse direction. © 2010 Microchip Technology Inc. Selection of components for a quarter brick converter design is critical to achieve high efficiency and high density. • • • • • • • • • • • Input voltage: VIN = 36 VDC-76 VDC Output voltage: VO = 12V Rated output current: IORATED = 17A Maximum output current: IO= 20A Output power: PO = 200W Estimated efficiency: 95% Switching frequency of the converter: FSW = 150 kHz Switching period of the converter: TP = 1/150 kHz = 6.66 µs Chosen duty cycle: D = 43.4% Full duty cycle: DMAX = 2 * 43.4% = 86.8% Input power pin = 214.75W DS01335A-page 11 AN1335 EQUATION 1: EQUATION 4: TURN ON TIME Conduction losses of the MOSFET at 48V: 43.4 TurnOnTime = 6.66μs × ---------- = 2.89μs 100 P COND = I 2 SRMS × R DS ( ON ) HOT = 0.171W where: PSFB MOSFET Selection ISRMS = Switch rms current EQUATION 2: Conduction losses of all the four PSFB MOSFETs = 0.687W Input line current at 36V P IN I AVE = ------------------ = 5.96A V INMIN EQUATION 5: Maximum Line Current at 36V I MAX 1 P SW = --- × V IN × I SRMS × T F × F SW = 0.05W 2 I AVE 5.96 = -------------- = ------------- = 6.87A D MAX 0.868 where: TF = Fall time of the MOSFET = 5.7ns Line rms current at 36V Switching losses of all the four PSFB MOSFETs = 0.21W I RMS = I MAX × D = 6.40A Switch rms current at 36V I SRMS In the ZVT, MOSFETs have only turn OFF switching losses. = I MAX × D ---- = 4.53A 2 EQUATION 6: Because the maximum input voltage is 76 VDC, select a MOSFET voltage rating that is higher than 76V and the current rating higher than IMAX at 36 VDC. RDS(ON) HOT can be calculated either from the graphs provided in the data sheet or by using the empirical formula shown in Equation 3. = 0.126W RDS(ON) HOT = 0.02625E where: RDS(ON) at 25 = 0.015E Maximum junction temperature, TMAX = 125oC Ambient temperature, TAMB = 25oC DS01335A-page 12 where: For all the four PSFB MOSFETs = 0.504W Bias voltage to the gate drive, VDD = 12V MOSFET total gate charge QG = 70 ns RDS(ON) EMPIRICAL FORMULA RDS(ON) HOT = RDS(ON) @ 25 * [1+0.0075*(TMAX-TAMB) MOSFET GATE CHARGE LOSS MOSFETGateCh arg eLosses = Q G × F SW × V DD The device selected is Renesas HAT2173 (LFPAK), and has VDS 100V, ID 25A, RDS(ON) 0.015E. EQUATION 3: SWITCHING LOSSES OF MOSFET Synchronous MOSFET Selection ] The ability of the MOSFET channel to conduct current in the reverse direction makes it possible to use a MOSFET where a fast diode or Schottky diode is used. In the fast diodes, junction contact potential limits to reduce the forward voltage drop of diodes. Schottky diodes will have reduced junction potential compared to the fast diode. In the MOSFETs, the conduction losses will be RDS(ON) * I2RMS. The on-resistance can be decreased by using parallel MOSFETs; this will reduce the losses further significantly. © 2010 Microchip Technology Inc. AN1335 When full wave center tapped winding is used in the transformer secondary side, the MOSFET voltage stress is twice the output voltage, as shown in Equation 7. EQUATION 7: MOSFET VOLTAGE STRESS MOSFETVoltageStress = 2 × ( V O + V FET + V DROP ) The maximum transition cannot exceed one-fourth of the resonant period to gain the ZVT. EQUATION 10: TTRANSMAX MaximumTransitionTime, T TRANSM AX π = --- ⋅ ( L R ⋅ C R ) 2 = 2 × ( 12 + 0.6 + 0.2 ) where: The capacitive energy required to complete the transition, ECR is shown in Equation 11. = 25.6V Secondary MOSFET Drop, VFET = 0.6V Total Trace Drops, VDROP = 0.2V This is the minimum voltage stress, seen by the MOSFET when the lower input voltage is 36V. For the maximum input voltage of 76V, the stress is as shown in Equation 8. EQUATION 8: 25.6 MOSFET Voltage Stress @ 76V = 76 × ---------- = 54.04V 36 EQUATION 11: 1 2 E CR = --- × C R × ( V INMAX ) 2 where: VIN MAX = maximum input voltage The energy stored in the resonant inductor LR must be greater than the energy required to charge and discharge the COSS of the MOSFET and transformer capacitance CTX of the leg transition within the maximum transition time. The device selected is Renesas HAT2173 (LFPAK). The energy stored in the resonant inductor (LR), is as shown in Equation 12. Transformer Design EQUATION 12: DESIGN CONSIDERATIONS FOR RESONANT TANK CIRCUIT ELEMENTS Design of resonant tank is critical to achieve ZVT. Resonant capacitor (CR) and resonant Inductor (LR) forms resonant tank. A factor of 4/3 is multiplied to the Output Capacitance of MOSFET (COSS) to accommodate the increase in capacitance with voltage, and a factor of two is also multiplied because two output capacitances (COSS) will come in parallel in each resonant transition. 1 2 E LR = --- × L R × I PRI 2 The slope of the primary current during transition is as shown in Equation 13. EQUATION 13: I PRI VP ------ = ----------------LR T TRANS EQUATION 9: 4 TotalResonantCapaci tan ce, C R = --- × 2 × C OSS + C TX 3 8--= × C OSS + C TX 3 = 1.387nF where: COSS = Output capacitance of the MOSFET = 5.20E-10F CTX = Transformer capacitance (neglected) ENERGY STORED IN THE RESONANT INDUCTOR (LR) NS MaximumPrimaryCurrent, I PRI = ------- × I O NP = 6.8A where: IO = Output current NP = TX primary turns = 5 NS = TX secondary turns = 2 VP = Input voltage = 32.5V FSW = Converter switching frequency ResonantTankFrequency, F R 1 = ------------------------------2π L R × C R where: TP = Switching period = 1/FSW Resonant transition estimated TTRANS = 0.15 * TP LR = Transformer leakage inductance + Additional leakage inductance © 2010 Microchip Technology Inc. DS01335A-page 13 AN1335 EQUATION 14: TTRANS LR T TRANS = 2 × I PRI × -----VP Two transitions per period. Hence, multiplied with 2. EQUATION 15: LR T TRANS V P L R = ------------------ × ---------2 I PRI L R = 2.3μH The energy stored in the inductor, ELR must be greater than the capacitive energy, ECR, which is required for the transition to occur within the allocated transition time. EQUATION 16: 2 1 2 1 E LR > E CR = --- × L R × I PRIMIN > --- × C R × V INMAX 2 2 2 ( V IN ) I PRIMIN > C R × ---------------- = 0.88A LR The magnetic cross section area must be large to minimize the number of turns that are required for the given application. Ensure that the core covers the winding that is laid on the PCB. Such design types reduce the EMI, heat dissipation and allow small height cores. Copper losses can be reduced by selecting the round center leg core because this reduces the length of turns. The Planar Magnetics design procedure is the same as that of the wire wound magnetics design: 1. 2. 3. 4. 5. 6. 7. 8. Select the optimum core cross-section. Select the optimum core window height. Iterate turns versus duty cycle. Iterate the core loss. Iterate the copper loss (Cu). Evaluate the thermal methods. Estimate the temperature rise. What is the cost trade-off versus the number of layers. 9. Does the mechanical design fit the envelope and pad layout? 10. Fit within core window height. 11. Is the size sufficient for power loss and thermal solution? Magnetics Design Full-Bridge Planar Transformer Design Magnetics design also plays a crucial role in achieving high efficiency and density. In the Quarter Brick DC/DC Converter design, planar magnetics are used to gain high efficiency and density. The two considerations for secondary rectifications are Full Wave Center Tapped (FWCT) rectifier configuration and Full Wave Current Doubler rectifier configurations. It is observed that the FWCT rectifier makes optimum use of board space and efficiency goals. Preliminary testing has validated this conclusion. DESIGN OF PLANAR MAGNETICS Planar magnetics are becoming popular in the high density power supply designs where the winding height is the thickness of the PCB. Planar magnetics design can be constructed stand-alone with a stacked layer design or as a small multi-layer PCB or integrated into a multi-layer board of the power supply. The advantages of planar magnetics are: • • • • • • Low leakage inductance Very low profile Excellent repeatability of performance Economical assembly Mechanical integrity Superior thermal characteristics Planar E cores offer excellent thermal resistance. Under normal operating conditions, it is less than 50% as compared to the conventional wire wound magnetics with the same effective core volume, VE. This is caused by the improved surface to the volume ratio. This results in better cooling capability and can handle higher power densities, while the temperature is within the acceptable limits. DS01335A-page 14 A further optimization goal is to offer a broad operating frequency from 125 kHz to 200 kHz to provide wide latitude for customers to optimize efficiency. The input voltage range is 36 VDC-76 VDC nominal with extended VINMIN OF 32.5 VDC. Analysis of the transformer design begins with the given input parameters: • VIN = 36V • Frequency = 150 kHz • TP = 6.667 x 10-6 The intended output voltage was meant to supply a typical bus voltage for distributed power applications and the output voltage. VO = 12.00V and the maximum output load current, IO = 25A No substitute exists for the necessary work to perform calculations sufficient to evaluate a particular core size, turns, and core and copper losses. These must be iterated for each design. One of the design considerations is to maximize the duty cycle, but the limitation of resolution offered by integer turns will quickly lead to the turn ratio of NP = 5 and NS = 2. © 2010 Microchip Technology Inc. AN1335 In the design of the magnetics, users must select the minimum number of turns. There is a cost or penalty to placing real-world turns on a magnetic structure such as, resistance, voltage drop and power loss. Therefore, use the least number of integer turns possible. FIGURE 16: PLANAR TRANSFORMER 21.00 mm (0.83”) 14.90 mm (0.59”) Thereafter, a reasonable assessment for turn ratio, duty cycle, peak flux density, and core loss can be done until a satisfactory point is reached for the designer. The duty cycle (more than each half-period) to produce the desired output is as follows: Champs Technologies MCHP-045-V31-1 • TON = 2.89 µs • D = TON/TP = 0.434 16.90” (0.67 mm) Over a full period, the duty cycle is 86.8% at a VIN of 36 VDC. 5.90 mm (0.23”) • Secondary MOSFET drop, VFET SEC = 0.1V • Total trace drops, VDROP = 0.2V • Primary MOSFET drop, VFET PRI = 0.6V 9.80 mm (0.39”) In this design, the following regulation drops are used: EQUATION 17: N V O = ( V IN – V FETPRI ) × -------S – V FETSEC – V DROP × 2D N P = 12.03V The iteration method is followed again to select the core size from the available cores. The selected core has the following magnetic parameters: • AC = 0.45 cm2 • LE = 3.09 cm • VE = 1.57 cm3 This core shape is a tooled core and is available from the Champs Technologies. In general, a power material in the frequency range of interest must be considered. Materials such as 2M, 3H from Nicera™, the PC95 from TDK™, or the 3C96, 3C95 from Ferroxcube™ are the most recommended options. The peak-to-peak and rms flux densities arising from this core choice are shown in Equation 18. EQUATION 18: 8 ( V IN × t ON ) × 10 B PKPK = ------------------------------------------NP × AC 3 B PKPK = 4.624 × 10 Gauss t ON B RMS = 2----× TP ∫ O 8 2 ( V IN × t ON ) × 10 ------------------------------------------- dT 2 × NP × AC 3 B RMS = 2.153 × 10 Gauss © 2010 Microchip Technology Inc. DS01335A-page 15 AN1335 The power loss density is calculated using the parameters shown in Table 4. TABLE 4: Material 3C92 3C96 3F35 Note: FIT PARAMETERS TO CALCULATE THE POWER LOSS DENSITY f (kHz) Cm x y Ct1 Ct0 20-100 26.500 1.19 2.65 2.68E-04 5.43E-02 3.75 100-200 0.349 1.59 2.67 1.51E-04 3.05E-02 2.55 200-400 1.19E-04 2.24 2.66 2.08E-04 4.37E-02 3.29 20-100 5.120 1.34 2.66 5.48E-04 1.10E-01 6.56 100-200 8.27E-02 1.72 2.80 1.83E-04 3.66E-02 2.83 200-400 9.17E-05 2.22 2.46 2.33E-04 4.72E-02 3.39 400-1000 1.23E-08 2.95 2.94 1.38E-04 2.41E-02 2.03 Source – New ER Cores for Planar Converters, Ferroxcube™ Publication 939828800911, Sept. 2002. Core loss density can be approximated by the formula shown in Equation 19. The core constants are made available by Ferroxcube™. In this design: • • • • • • • • • Ct2 Temp = 50oC Frequency = 150000 Hz B = BRMS * 10-4 = 0.2153 Tesla x =1.72 y = 2.80 Ct2 = 1.83 * 10-4 Ct1 = 3.66 * 10-2 Ct0 = 2.83 Cm = 8.27 * 10-2 EQUATION 19: SECONDARY RMS CURRENT I SEC = I O × D I SEC = 16.47A Primary rms current is calculated as shown in Equation 21: EQUATION 21: PRIMARY RMS CURRENT NS I PRI = I O × 2D × ------NP CORE LOSS DENSITY Core Loss Density Pcore 2 x y C m × Freq × B × ⎛ C t0 – C t1 × Temp + C t2 × Temp ⎞ ⎝ ⎠ ------------------------------------------------------------------------------------------------------------------------------------------= 1000 3 P = 1.307 × 10 mW/Cm3 CoreLoss = P × V E × 10 EQUATION 20: –3 CoreLoss = 2.052W I PRI = 9.317A The DCR values are computed from the CAD drawings: • Secondary DCR: SecDCR = 0.0023E • Primary DCR: PriDCR = 0.025E Secondary copper loss is multiplied by two because it is a center tapped winding. EQUATION 22: Sec_Loss = 2 * I2SEC * Sec_DCR = 1.248W One of the benefits of using planar construction is the opportunity to utilize 2 oz., 3 oz., and 4 oz. copper weight, which results in very thin copper. The impact is that skin depth and proximity loss factors are usually considerably reduced versus using wire wound magnetic structures. The copper losses are calculated using DC Resistance (DCR). Pri_Loss = I2PRI* Pri_DCR = 2.17W Total_Loss = Sec_Loss + Pri_Loss + Core Loss Total_Loss = 5.466W The secondary rms current in each half of the center tapped winding is shown in Equation 20. DS01335A-page 16 © 2010 Microchip Technology Inc. AN1335 The stacking of the main transformer arrangement is shown in Table 5. TABLE 5: Planar Output Inductor Design layers The output inductor serves the following functions: • Stores the energy during the OFF period to keep the output current flowing continuously to the load. • Smooths out and average the output voltage ripple to an acceptable level. STACKING LAYERS FOR PLANAR TRANSFORMER Layers Cu Weight (Oz.) Winding Primary Sec1 Layer 1 Sec2 Sec2 2 Layer 2 2 Layer 3 Sec1 2 Layer 4 2 Layer 5 Primary 4 Layer 6 4 Layer 7 Sec1 2 Layer 8 2 Layer 9 Primary 2 Layer 10 3 Layer 11 Sec2 3 Layer 12 Layer 13 2 Primary 4 Layer 14 4 Layer 15 Sec1 2 Layer 16 2 Layer 17 Sec2 2 Layer 18 Turns 2 5 FIGURE 17: Q1 2 2 FULL-BRIDGE CONVERTER WITH CENTER TAPPED FULL WAVE SYNCHRONOUS RECTIFIER Q3 TX Q6 LL VPK2 IPRI L0 VPRI C0 Q2 Q4 © 2010 Microchip Technology Inc. V0 Q5 DS01335A-page 17 AN1335 FIGURE 18: PLANAR OUTPUT INDUCTOR • Switch turn ON time, TON = 2.89 μs • Total Switching period, TP = 6.667μs • Duty cycle, D = TON/TP = 0.434 18.35 mm (0.72”) Over a full period the duty cycle is 86.8% at VINMIN 36 VDC. 12.00 mm (0.47”) The duty cycle (more than each half-period) to produce the desired output is as follows: Champs Technologies MCHP1825-V31-1 EQUATION 23: V O = (V IN –V FETPRI N S ) × ------- – V –V × 2D FETSEC DROP N P 15.0 mm (0.59”) V O = 12.03V 5.40 mm (0.21”) 9.80 mm (0.39”) In the case of output inductor, consider the choice of inductance value at the maximum off time. This occurs in PWM regulated DC-DC converters at the maximum input voltage, VIN MAX = 76V, and the feedback loop adjusts the switch ON time accordingly. TONMIN = 1.415 µs The duty cycle is as follows: D_MIN = TONMIN/TP = 1.3689 µs The peak voltage at the transformer secondary is as shown in Equation 24. EQUATION 24: NS V PK2 = ( V INMAX – V FETPRI ) × ------- – V FETSEC – V DROP NP V PK2 = 28.26V Maximum output load current, IO = 25A. A ripple current of 25% of the total output current is considered in this design. EQUATION 25: I MIN = I O × 0.25 = 6.5A EQUATION 26: L OMIN OUTPUT INDUCTANCE (LOMIN) ( V PK2 – V O ) × T ONMIN = --------------------------------------------------------- = 3.54μH I MIN This core is also a tooled core as the main transformer, TX1. It is available from Champs Technologies as PN MCHP1825-V31-1. Materials such as 7H from Nicera™, the PC95 from TDK™, or the 3C94, 3C92 from Ferroxcube™ are the recommended choices. • • • • Core cross section, AC = 0.4 cm2 Core path length, LCORE = 3.09 cm Rated output current: IRATED = 17A Defined saturation current: ISAT = 20A The process of inductor design involves iterating the number of turns possible and solving for a core air gap. The air gap is checked for operating the flux below maximum rated flux in the core material at the two operating current values that is rated current and saturation current. In this design, if the 18 layers are available, these layers can be split into balanced integer turns. This is a practical method and the number of turns Nt = 6. In this design, a fringing flux factor assumption of 15% is done that is FFF = 1.15. The iterative process begins by calculating the air gap equation. The air gap is calculated using Equation 27. EQUATION 27: 2 In this design, the core window height and its adequacy in terms of accommodating the 18 layer PCB stack is to be assessed since the windings/turns for the inductor are also embedded. DS01335A-page 18 –8 ⎛ 0.4 × π × Nt × A C × 10 ⎞ L GAP = ⎜ --------------------------------------------------------------⎟ × FFF L OMIN ⎝ ⎠ = 0.058cm L GAP L GAPIN = ------------- = 0.023inch 2.54 © 2010 Microchip Technology Inc. AN1335 EQUATION 28: OPERATING FLUX DENSITY AT DEFINED SATURATION CURRENT 0.4 × π × Nt × I SAT B DC = ---------------------------------------------L GAP 3 B DC = 2.598 × 10 Gauss EQUATION 29: OPERATING FLUX DENSITY AT RATED CURRENT 0.4 × π × Nt × I RATED B RATED = ----------------------------------------------------L GAP 3 B RATED = 2.208 × 10 Gauss The BDC and BRATED values are conservative compared to the commercially rated devices. Typical BMAX values are 3000 Gauss at 100ºC. The required AL value is calculated, as shown in Equation 30. EQUATION 30: AL VALUE 9 L OMIN × 10 - = 98.32mH A L = -----------------------------2 Nt This is helpful for instructing the core manufacturer for gapping instructions. The inductor traces are designed using a CAD package and are integrated into the PCB layout package. The CAD package facilitates the calculation of trace resistance for each layer. The calculated DCR values DCRRATED = 3.5 * 10-3E. Copper loss is computed at the DC values of rated and saturation-defined currents, as shown in Equation 31. EQUATION 31: 2 Cu LossSAT = ( I SAT ) × DCR RATED Cu LossSAT = 1.4W EQUATION 32: 2 Cu LossRATED = ( I RATED ) × DCR RATED Cu LossRATED = 1.012W One of the design goals is to make it universal for other lower and higher power implementations of the digital converter and to keep the overall efficiency high. It fits comfortably with its footprint in the PCB. However, we consider that a smaller core and footprint optimization is quite possible. © 2010 Microchip Technology Inc. Planar Drive Transformer Design To drive each leg (high side and low side) of the gates, the high side/low side driver, or low side driver with isolated drive transformer is required. A minimum of 500 VDC isolation is required in the drive transformer from the high side to low side winding. Because the gate drive is derived from secondary side controller, primary to secondary 2500 VDC isolation is required. The following critical parameters must be controlled while designing the gate drive transformer: • Leakage inductance • Winding capacitance A high leakage inductance and capacitance causes an undesirable gate signal in the secondary, such as phase shift, timing error, overshoot and noise. Winding capacitance results when the design has a higher number of turns. Leakage inductance results when the turns are not laid uniformly. Because planar magnetics are used in this application, these parameters may not be a problem. Since the absolute number of turns required is low and the primary and secondary side high/low drive windings can be interleaved to minimize leakage without increasing the overall capacitance. Typical gate drive transformers are designed with ferrite cores to reduce cost and to operate them at high frequencies. Ferrite is a special material that comprises high electrical resistivity and can be magnetized quickly with minor hysteresis losses. Because of its high resistance, eddy currents are also minimal at high frequency. Selection of Core Materials and Core Selection of core material depends on the frequency of the operation. 3F3 from Ferroxcube™ is one of the best options for the operating frequencies below 500 kHz. The power loss levels of gate drive transformers is usually not a problem and thus Ferroxcube RM4/ILP is selected. The magnetic parameters of Ferroxcube RM4/ILP are as follows: • • • • AC = 0.113 cm2 Lm = 1.73 cm AL =1200 nH µEFF = 1140 One of the primary goals of the design is to embed all the magnetics as part of the overall PCB design of the main power stage. A small size core geometry is selected, that has sufficient window height to accommodate the overall PCB thickness and also gives reasonable window width to accommodate the PCB trace width that comprises the turns. The resulting “footprint” or core cut-out required of the RM4/ ILP was found to be acceptable. DS01335A-page 19 AN1335 We will iterate the primary turns to arrive at a suitable peak flux density and magnetizing current using the formula shown in Equation 33. The peak and RMS flux densities can be pushed higher. However, a reasonably low value of magnetizing current has been maintained such that the driver is not loaded much. EQUATION 33: EQUATION 37: 8 NP ( V IN × T ON ) × 10 = --------------------------------------------B PKPK × A C CALCULATION OF MAGNETIZING INDUCTANCE 2 In the application, VIN = 12V as set by the bias supply. The operating frequency for main power processing is selected as 150 kHz. The result is the gate drive transformer operates at the same frequency. The duty cycle is also determined by the power stage. The basic input parameters, TP and TON are set. Iterating for primary turns, NP = 10. The peak-to-peak flux density can be achieved as shown in Equation 34: LA L = N P × A L × 10 –9 –4 LA L = 1.2 × 10 Henry Or, 2 –8 0.4 × π × μ EFF × ( N P ) × A C × 10 L M = --------------------------------------------------------------------------------------Lm –5 L M = 9.57 × 10 Henry Conversely, the inductance minimum will be between ~70 µH. L MIN = 0.75L M Henry EQUATION 34: PEAK-TO-PEAK FLUX DENSITY –5 L MIN = 6.699 × 10 Henry 8 ( V IN × T ON ) × 10 B PKPK = --------------------------------------------NP × AC The magnetizing current is thus reasonable for this application, and is shown in Equation 38: 3 B PKPK = 3.069 × 10 Gauss EQUATION 38: The peak flux density is shown in Equation 35, which yields a volt-µs rating of (VIN * TON) = 37.7. This is well below the typical saturation curves for 3F3 of 3000 Gauss at 85ºC operational ambient temperature. However, potential saturation is not a design concern. EQUATION 35: PEAK FLUX DENSITY 8 ( V IN × T ON ) × 10 B PK = --------------------------------------------2 × NP × AC 3 B RMS = O Farad EQUATION 39: F R = 2.3MHz 8 2 ( V IN × T ON ) × 10 --------------------------------------------2 × NP × AC ∫ – 12 1 F R = ------------------------------------------------2 × π × ( LM × CD ) RMS FLUX DENSITY T ON Assuming the worst case, the distributed capacitance is shown as follows: Any ringing on the gate drive waveforms due to the transformer will possess a frequency of 2.3 MHz. The RMS flux density is calculated as shown in Equation 36. 2----× TP di = 0.362A C D = 50 × 10 B PK = 1.534 × 10 Gauss EQUATION 36: V IN × T ON di = ------------------------LM × DT 3 B RMS = 1.363 × 10 Gauss results in mW of core loss. DS01335A-page 20 © 2010 Microchip Technology Inc. AN1335 In this design, the selection of track width or trace width was fairly conservative. Given the RM4/ILP core window width of 2.03 mm (80 mils), and an allowable PCB width accommodated inside this core of 1.63 mm (64 mils), and a further conservative assumption of trace-to-trace clearance of 0.3 mm (12 mils), we can either place 2T/layer of 0.39 mm (15 mil) width or 3T/layer of 0.18 mm (7 mils) width. If 4 oz. copper was used per layer the 0.18mm trace width would result in too much “under-etch” in the fabrication. We had ~14 layers dictated by the power stage and the resulting PCB thickness of 3.5-3.8 mm could be easily accommodated by the RM4/ILP core window height. Hence, it is easier to select 2T/layer. This selection also allowed three opportunities for an interleave to occur between the primary and each secondary drive winding. A choice of 3T/layer may have resulted in an imbalance and with less opportunity for interleave. EQUATION 42: –6 T ON = 2.89 × 10 Sec TP T OFF = ------ – T ON 2 –7 T OFF = 4.433 × 10 Sec The core used on this part = E5.3/2.7/2-3C96 The core parameters are as follows: • LM = 1.25 cm • AC = 0.0263 cm2 • VE = 0.0333 cm3 The nominal current sense termination resistor value: RB = 10.0E. EQUATION 43: Current Sense Transformer Design I MAX V PKSECY = ------------ × R B NS The current sense transformer selected is a conventional stand-alone magnetic device. The decision was made earlier to have a 1:100 current transformation ratio. Therefore, it is difficult to implement this device as an embedded structure. Therefore, the rating is 0.1 V/amp. We repeat some aspects of the TX1 main transformer design such as switching frequency. EQUATION 44: 8 EQUATION 40: 3 F SW = 150 × 10 HZ 1 T p = ---------F SW –6 T P = 6.667 × 10 Sec The transformation ratio, NC = NS/NP = 100 Maximum rated current, IMAX = 10A Therefore, secondary RMS current is computed as shown in Equation 41: EQUATION 41: V PKSECY = 1V SECONDARY RMS CURRENT I RMSPRIM I RMSSECY = -----------------------NC I RMSSECY = 0.093A © 2010 Microchip Technology Inc. ( V PKSECY × T ON ) × 10 B PK = ----------------------------------------------------------NS × AC B PK = 109.886Gauss It is considered that the peak flux density is very low and it is fine. Usually, the current to voltage gain is this low in most switched mode converters. The current ramp signal at the current sense (CS) input for most analog controllers is <1V so always select a low value termination resistor. In this case, the voltage gain is conditioned with differential op amps prior to sending it to the input ADC of the dsPIC® DSC. It is helpful to know that higher current to voltage gains are possible simply by selecting higher value termination resistors. The only limitation will be a ceiling imposed by the saturation of the ferrite core. The volt-µs rating of the CH-1005 Champs Technologies is 58V-µs. In this design, if a termination impedance of 100Ω is selected, a 10V signal amplitude is gained. The current transformer reproduces the current wave shape until it is not saturated, that is as long as it is performing as a transformer. In this design, a maximum ON time of 5.8 µs can be permitted. DS01335A-page 21 AN1335 EQUATION 46: The rated maximum flux is shown in Equation 45. RMS FLUX DENSITY t ON EQUATION 45: –6 B RMS = 8 ( 58 × 10 ) × 10 B RATED = ------------------------------------------NS × AC 2----× TP ∫ O 8 2 ( V PKSECY × T ON ) × 10 ----------------------------------------------------------- dTp 2 × NS × AC B RMS = 51.159Gauss 3 B RATED = 2.205 × 10 Gauss The BPK is rated as 2200 Gauss peak for 100ºC operation unipolar excursion. The RMS flux density is calculated as shown in Equation 46. EQUATION 47: x 2 y C m × F × ( B RMS ) × ( C t0 – Ct1 × Temp + C t2 × Temp ) CoreLossDensity, P = -----------------------------------------------------------------------------------------------------------------------------------1000 P = 0.048 mW/cm3 where setting up core loss coefficients: Cm = 8.27 * 10-2 x = 1.72 y = 2.80 Temp = 30 Ct1 = 3.66 * 10-2 Ct2 = 1.83 *10-4 Ct0 = 2.83 F = 1.5 * 105 CoreLoss = P × V E × 10 –3 –6 CoreLoss = 1.592 × 10 W Core loss is about zero or negligible. Secondary SecDCR = 6.6E. DS01335A-page 22 © 2010 Microchip Technology Inc. AN1335 FIGURE 19: CURRENT TRANSFORMER 2 3.70 mm (0.146'') Secloss = ( I RMSSECY ) × SecDCR 2 Priloss = ( I RMSPRIM ) × PriDCR 1.85 mm (0.073'') 1 8 = 0.057W 100T 7 1T 3 = 0.173W 5.3 mm (0.21'') EQUATION 48: 7 + 8 TotalLoss = Secloss + Priloss + Coreloss = 0.23W where: 4.9 mm (0.19'') Secloss = Secondary copper losses Priloss = Primary copper losses 0.25 mm (0.010'') PriDCR = 0.002E SecDCR = 6.6E Total loss for this device at maximum ratings is less than 1/4W. Calculate the inductance value for the selected 3C96 material. 7.80 mm (0.31'') 5.3 mm (0.21'') EQUATION 49: 2 L AL = ( N S ) × A L × 10 L AL = 3 × 10 –3 –9 ~ 3mH 0.006 where: Ns = 100 0.15 6.8 mm (0.27'') AL = 300 nH EQUATION 50: L MIN = L AL × 0.75 = 2.25mH Minimum secondary inductance = 2.25 mH. X L = 2 × π × f × L MIN 3 = 2.12 × 10 E Effective termination impedance is as shown in Equation 51: Planar Auxiliary Power Supply Transformer Design The digital DC/DC converter requires auxiliary power supply. The dsPIC DSC requires 3.3V and the gate drivers require 12V. The dsPIC DSC must have power supplied to it prior to start-up of the power converter. The scheme to accomplish this is to utilize an analog converter for start-up and also for continuous operation. This avoids possible glitches or uncontrolled operation events during abnormal operation or unanticipated transient conditions. The analog controller requires a boot strap supply once it has gone through soft-start. EQUATION 51: X L × Rb X EFF = ------------------X L + Rb X EFF = 9.953E Deviation from ideal is < 0.1%. © 2010 Microchip Technology Inc. DS01335A-page 23 AN1335 The dsPIC DSC requires 3.3V. A linear regulator is inserted prior to 3.3V so that the headroom required at one output is 4V. The 3.3V output voltage before regulator V01 = 4V. • • • • • • EQUATION 57: 2 × I SCDC PeakSecondaryCurrent, ISCPK = -----------------------DS Load current, I3.3V = 0.3A 12V output voltage before regulator, V02 = 12V Load current, I12V = 0.4A Total output power = 6W Consider an overall efficiency of 80% Input power = 7.5W 2---------------× 0.1 = 3.33A 0.6 D ------S × I SCPK 3 = 1.489A SecondaryRMSCurrent, I SRMS = Consider minimum input voltage, VINMIN = 32V. The converter is designed to operate at a maximum duty cycle, D = 40%. The nominal operating frequency, FSW of the IC is 250 kHz. where: Short circuit current: ISCDC = 1A Secondary duty cycle: DS = 0.6 The turns ratio for 12V and 3.3V output is shown in Equation 58. EQUATION 52: 3 F SW = 250 × 10 Hz EQUATION 58: N P [ V IN – ( I PPK × R DS ( ON ) ) ] × D ------- ≥ -------------------------------------------------------------------------NS ( V OUT + V fD1 ) × ( 0.8 – D ) Total period, TP = 4 µs On period, TON = 1.6 µs NP ---------- = 2.60 N S12 EQUATION 53: InputPower AverageCurrent, IAVE = -----------------------------------------------------------MinimumInputVoltage NP ------------- = 7.828 N S 3.3 7.5W = ------------ = 0.234A 32V where: Voltage drop on the diode, VFD1 = 0.7V RDS(ON) = 4E Peak current of a Discontinuous mode Flyback Converter, IPPK is shown in Equation 54. A quick check of the available standard core structures indicates that there was a distinct possibility to use a standard size RM-4 core. EQUATION 54: I AVE I PPK = 2 × ----------- = 1.17A D Primary rms current, IRMSPRIM is shown in Equation 55: An important feature of this core for this design is, it consists of a core window with nominal 4.3 mm that clears the 4.0 mm PCB thickness. The overall height of this core is 7.8 mm so it is <10 mm height of the DC/DC Converter mechanical height. The RM-4 core parameters are: EQUATION 55: T ON - = 0.427A I RMSPRI = I PPK × -------------3 × TP • • • • AE = 0.145 cm2 ICORE = 1.73 cm µ = 2000 VE = 0.25 cm3 EQUATION 56: V INMIN × D PrimaryInduc tan ceL P = ----------------------------F SW × I PPK 32 × 0.4 = ---------------------------------- = 43.6μH 250000 × 1.17 DS01335A-page 24 © 2010 Microchip Technology Inc. AN1335 FIGURE 20: AUXILIARY PLANAR TRANSFORMER EQUATION 61: 9 L P × 10 A L = --------------------2( N PRI ) 15.51 mm (0.61'') 2 1 5 A L = 164.063nH 6 The flux density is calculated as shown in Equation 62. 11.51 mm (0.45'') EQUATION 62: 7 4 0.4 × π × N PRI × I PPK B PK = ----------------------------------------------------L GAP 8 3 3 B PK = 1.959 × 10 Gauss Inner Layer Top Points 1 * BPK is lesser than BSAT limitation of 3000 Gauss at 85ºC. The required maximum output power for DCM operation, factoring in efficiency is shown in Equation 63. 5 * 6T 16T 6 2 3 * 7 * 6T EQUATION 63: 2T 4 1 2 P O = --- × L P × ( I PPK ) × F SW 2 8 The footprint (length x width) of the device is not greater than that of a stand-alone magnetic device. The footprint shown above has been further reduced in the final implementation and the entire bias converter has been implemented as part of the embedded design. P O = 7.46W The peak AC flux density is calculated as shown in Equation 64: EQUATION 64: EQUATION 59: 8 ( V IN × T ON ) × 10 B PKAC = --------------------------------------------N PRI × A E 8 V INMIN × T ON × 10 N PRI = -------------------------------------------------- = 16T B MAX × A E 3 B PKAC = 2.48 × 10 Gauss where: BMAX = 2200Gauss The required center post air gap based on the formula is shown in Equation 60: The RMS flux density is calculated as shown in Equation 65. EQUATION 65: EQUATION 60: 2 L GAP T ON –8 0.4 × π × ( N PRI ) × A E × 10 = ------------------------------------------------------------------------- × FFF LP B RMS = 1----× TP ∫ O 8 2 ( V IN × T ON ) × 10 -------------------------------------------- dTp 2 × N PRI × A E B RMS = 771.454Gauss L GAP = 0.012cm L GAP L GAPIN = ------------2.54 –3 L GAPIN = 4.591 × 10 in The core loss equation parameters are used for Ferroxcube “3C92” material at 40ºC rise in temperature. The AL value is calculated as shown in Equation 61. © 2010 Microchip Technology Inc. DS01335A-page 25 AN1335 EQUATION 66: A calculated core loss value of 76 mW is acceptable and a good reason to use ferrite for the core material. B RMS B = --------------10000 1 f = -----TP The CAD package is used in the PCB trace design to calculate the trace DCR for the primary and secondary DC resistance. • DCRSEC = 0.023E • DCRPRI = 0.088E 5 f = 2.5 × 10 Hz ° EQUATION 68: Temp = 40 C The operating coefficients are: CM = 9.17 x 10-5 EQUATION 67: 2 opperLoss = ⎛ I 2 × DCR PRI⎞ + ⎛ I RMSSEC × DCR SEC⎞ ⎝ RMSPRI ⎠ ⎝ ⎠ CORE LOSS DENSITY FORMULA CopperLoss = 0.067W 2 x y ( C t0 – C t1 ) × Temp + C t2 × Temp P = C M × f × B × ------------------------------------------------------------------------------------1000 P = 303.063 mW/cm3 The overall loss is shown in Equation 69. where: EQUATION 69: Ct2 = 2.33 * 10-4 Ct1 = 4.72 * 10-2 TotalLoss = CuLoss + CoreLoss Ct0 = 3.39 TotalLoss = 0.142 x = 2.22 y = 2.46 CoreLoss = P × V OL × 10 CoreLoss = 0.076W –3 The only efficiency penalty in using a digital controller is the bias supply converted efficiency of 80%. All converters will share approximately the same FET driver loss. The only further penalty is the footprint or space occupied by the bias supply within the available outline package of the converter itself. The main advantage as discussed at the outset is that the controller is “always on”, that is, it supplies power in a controlled fashion and rides out abnormalities and transients that might at the least require a hiccup start-up for an analog controller. DS01335A-page 26 © 2010 Microchip Technology Inc. AN1335 DESIGNING A DIGITAL QUARTER BRICK CONVERTER The Quarter Brick DC/DC Converter was designed using the dsPIC33FJ16GS502. The design analysis is described in the following sections. FIGURE 21: Sensing Element REAL WORLD SIGNAL CHAIN: DIGITAL POWER SUPPLY Scaling Filter What is a Digitally Controlled Power Supply? A digital power supply can be broadly divided into power control and power management. Power control is relatively a new trend when compared to power management. Power management is data communication, monitoring, data logging, power supply protection, and sequencing of the outputs. This is not real time because the switching frequencies of the converters are higher than the power management functions. Power control is defined as the flow of power in the converter and it is controlled from one PWM cycle to another PWM cycle. Power control is performed with both the DSCs and analog controllers without much variation in the design. Advantages of DSC In modern SMPS applications, power conversion is only part of the total system solution. In addition, many other requirements and features are required to make the system more reliable. These features can be realized using a DSC and are as follows: • Improved level of portability to other converter topologies • Adaptive and predictive control mechanism to achieve high efficiency and improved dynamic response • Software implementation of the protections to reduce the component count • Improved scalability • Active load balancing in the parallel connected systems • Improved overall system reliability and stability • System performance monitoring capability • Real time algorithms for the regulation of power converters • Less susceptibility to parameter variations from thermal effects and aging ADC CMP Load Power Converter Analog Hardware DSC Core PWM Digital Signal Controller (DSC) DIGITAL PHASE-SHIFTED FULLBRIDGE (PSFB) DESIGN In the digital power supply design, the power train is same as the analog power converter design. The difference exists in the way it is controlled in the digital domain. The analog signals such as voltage and current are digitized by using the ADC, and fed to the DSC. These feedback signals are processed with the digital compensator and modulate the PWM gate drive to get the desired control on the output. Few critical peripherals that are used in digital power supply are listed below: • PWM generator • ADC • Analog comparator PWM Generator The PWM generator must have the ability to generate high operating frequencies with good resolution, dynamically control PWM parameters such as duty cycle, period, and phase, and to synchronously control all PWMs, fault handling capability, and CPU load staggering to execute multiple control loops. The PWM resolution determines the smallest correction to be done on the PWM time base. EQUATION 70: PWMClockFrequency PWMResolution = ---------------------------------------------------------------DesiredPWMFrequency EQUATION 71: PWMClockFrequency BitResolution = log 2 ⎛ ----------------------------------------------------------------⎞ ⎝ DesiredPWMFrequency⎠ © 2010 Microchip Technology Inc. DS01335A-page 27 AN1335 EXAMPLE 1: EQUATION 72: FullScaleVoltageADCResolution = ---------------------------------------------n 2 where: PWM Clock Frequency = 60 MHz Desired PWM Frequency = 500 kHz PWM Resolution = 120 = One part in 120 n = Number of bits in the ADC Bit Resolution = log2 (120) ~ 7 bits EXAMPLE 3: EXAMPLE 2: Example A: PWM Clock Frequency = 1000 MHz ADC full voltage = 3.3V Desired PWM Frequency = 500 kHz PWM Resolution = 2000 = One part in 2000 Bit Resolution = log2 (2000) ~ 11 bits A resolution of 11 bits indicates that the user can have 2048 different steps from zero to full power of the converter. This gives finer granularity in control of the duty cycle when compared to the seven bits resolution where only 128 steps are available for control. Analog-to-Digital Converter (ADC) All the real world feedback signals are continuous signals, and should be digitized to process in the DSC. A built-in ADC performs this process. ADC requires a voltage signal that is to be provided as an input. The input signals are scaled down to the ADC reference voltage. These voltages are typically 3.3V and 5V. FIGURE 22: ANALOG-TO-DIGITAL CONVERTER (ADC) AN1 Inputs MUX SAR Core Data Format ADC Result Buffers ANx Sample and Hold Circuit In digital SMPS applications, higher bit resolutions and higher speed are the two characteristics that determine the ADC selection. The ADC resolution indicates the number of discrete values it can produce over the range of analog values, hence the resolution is expressed in bits. DS01335A-page 28 CALCULATING THE ADC RESOLUTION Number of bits in an ADC = 10 Therefore, ADC resolution = 3.22mV Another parameter to be considered is the sample and conversion time (time taken by ADC to sample an analog signal and to deliver the equivalent digital value). Usually, the conversion time is specified in million samples per second (Msps). For example, if the conversion time is specified as 2 Msps, the ADC can convert two million samples in one second. Hence, the sample and conversion time is 0.5 µs. The conversion speed plays an important role to replicate the sampled signal. As per Nyquist criterion, the sampling frequency must be greater than twice the bandwidth of the input signal (Nyquist frequency). As a guideline in SMPS applications, sampling of the analog signal at a frequency greater than 10x of the signal bandwidth is required to maintain fidelity. Analog Comparator Most of the DSCs consists of an analog comparator as a built-in peripheral which enhances the performance of SMPS applications. Analog comparator can be used in cycle-by-cycle control method to improve the response time of the converter and also in the fault protection applications. ADC and PWM Resolution in SMPS Applications Usually, analog controllers provide fine resolution to position the output voltage. The output voltage can be adjusted to any arbitrary value, and is only limited by loop gain and noise levels. However, a DSC consists of a finite set of discrete levels, because the quantizing elements, ADC and PWM generator exist in the digital control loop. Therefore, the quantization of ADC and PWM generator is critical to both static and dynamic performance of switched mode power supplies. © 2010 Microchip Technology Inc. AN1335 The ADC resolution must be lower than the permitted output voltage variation to achieve the specified output voltage regulation. The required ADC resolution is shown in Equation 73. EQUATION 73: ⎛ V MAX A ⁄ D V o ⎞ N A ⁄ D = Int log 2 ⎜ ----------------------- × ----------⎟ ΔV o⎠ ⎝ V REF where: VMAX A/D = ADC full range voltage in this application VREF = Reference voltage The digital PWM produces an integer number of duty values (it produces a discrete set of output voltage values). If the desired output value does not belong to any of these discrete values, the feedback controller switches among two or more discrete values of the duty ratio. In digital control system, this is called as limit cycle and it is not desirable. Limit cycling can be avoided by selecting the change in output voltage caused by one LSB change in the duty ratio has to be smaller than the analog equivalent of the LSB of ADC. For a buck type forward regulator, NPWM is shown in Equation 74. EQUATION 74: NPWM > = NA/D + log2 NA/D = Number of bits in ADC VO = Signal to be measured (output voltage) ΔVO = Allowed output voltage variation Int [ ] = Denotes taking the upper rounded integer EXAMPLE 4: ADC Resolution Vref VMAX A/D * D where: NPWM = Number of bits in a PWM controller D = Duty ratio To generalize, NPWM must be minimum of one bit more than NA /D. VMAX A/D = 3.3V Note: VO = 12V Δ VO = 1% of 12V = 120 mV VREF = 2.6V which is 80% of the ADC full range voltage To have a stable output, that is without limit cycling, the down stream quantizer of the ADC should have higher resolution. NA/D = 7, (therefore, a 7-bit ADC can be used) ADC resolution can also be expressed as follows: ADC LSB << (VREF/VO) * ΔVO TABLE 6: SWITCHING FREQUENCIES OF THE CONVERTER Signal Name Description Type of Signal dsPIC® DSC Resource Frequency of Operation PWM1H,PWM1L Left Leg Gate Drive PWM Output PWM1H,PWM1L 150 kHz PWM2H,PWM2L Right Leg Gate Drive PWM Output PWM2H,PWM2L 150 kHz Synchronous Rectifier Gate Drive PWM Output PWM3H,PWM3L 150 kHz — — 75 kHz PWM3H,PWM3L — Control Loop Frequency © 2010 Microchip Technology Inc. DS01335A-page 29 AN1335 TABLE 7: Pin DSC PERIPHERALS MAPPED TO PSFB CONVERTER Peripheral Description 1 AN2 Load share 2 AN3 Temp 3 CMP2C Output overvoltage 4 RP10 TX secondary voltage 5 VSS Ground 6 CMP4A TX overcurrent 7 RP2 EXT SYNCI1 8 PGD2 Programming 9 PGC2 Programming 10 VDD Bias supply +ve 11 RB8 COM1 12 RB15 COM2 13 RB5 Remote ON/OFF 14 SCL1 COM4 15 SDA1 COM3 16 VSS Ground 17 VDDcore VDD core 18 PWM3H Sync gate drive 19 PWM3L Sync gate drive 20 PWM2H PSFB gate drive 21 PWM2L PSFB gate drive 22 PWM1H PSFB gate drive 23 PWM1L PSFB gate drive 24 AVSS Ground 25 AVDD Bias supply +ve 26 MCLR Master clear 27 AN0 TX current 28 AN1 12V output DS01335A-page 30 © 2010 Microchip Technology Inc. AN1335 FIGURE 23: dsPIC® DSC RESOURCES FOR THE QUARTER BRICK CONVERTER Full-Bridge Converter + Output Voltage 36 VDC – 76 VDC Synchronous Rectifier + 12 VDC /17A CT - - Drive TX Drive IC Drive IC Drive TX Drive IC PWM1 PWM2 AN1 Remote ON/OFF Opto Isolator AN0 dsPIC33FJ16GS502 RB5 RB8 PWM3 RB15 SCL1 SDA1 RP2 AN3 AN2 External Communication External Over Sync Temp © 2010 Microchip Technology Inc. Load Share DS01335A-page 31 AN1335 DIGITAL CONTROL SYSTEM DESIGN Digital Average Current Mode Control Technique Digital control system design is a process of selecting the difference equation or Z-domain transfer function for the controller to achieve good closed loop response. Parameters such as settling time, output overshoot, rise time, control loop frequency and bandwidth must be considered to achieve acceptable performance. Digital current mode control is a new approach for improving the dynamic performance of high frequency switched mode PWM converters, and is used in this design. In this method, DSC performs the entire control strategy in software. The current mode control (CMC) strategy consists of two control loops. The inner current loop subtracts a scaled version of the inductor current from the current reference. The current error is further processed with the PID or PI compensator and the result is appropriately converted into duty or phase. Any dynamic changes in the output load current directly modifies the duty or phase of the converter. The outer loop subtracts the scaled output voltage from a reference and the error is processed using the PID or PI compensator. The output of the voltage loop compensator provides the current reference for the inner loop. Current and voltage compensators allow tuning of the inner and outer loops to ensure converter stability and to achieve the desired transient response. The denominator polynomial of transfer function provides the roots of the equation. These roots are the poles of the transfer function. This equation is called the characteristic equation. The nature of roots of the characteristic equation provides an indication of the time response. The system stability can be determined by finding the roots of the characteristic equation and its location. The system is considered to be stable if the roots of the characteristic equation are located in left half of the ‘S’ plane. This causes the output response due to bounded input to decrease to zero as the time approaches infinity. In the quarter brick converter design, the controller is designed in the continuous time domain and then converted to an equivalent digital controller. This approach is called digital re-design approach or digital design through emulation. FIGURE 24: AVERAGE CURRENT MODE CONTROL DCR Compensation Voltage Loop Compensation VO* + Current Loop Compensation IL * VERROR - VO + + PI Control P Control - Phase/Duty VO Plant + IL VO Decouple Compensation ITX ADC Sensor ADC Sensor Digital Signal Controller (DSC) DS01335A-page 32 © 2010 Microchip Technology Inc. AN1335 Deriving the Characteristic Equation for the Current Mode Control (CMC) Let us take a simple buck converter to derive the characteristic equation. FIGURE 25: BUCK CONVERTER IL VL D * VIN LM DCR VX Buck Inductor ESR VIN IC Output Capacitor IO VO Load C Based on Figure 25, and applying Kirchhoff's laws results in the expressions and equations shown in Equation 75. EQUATION 75: (A) IC = IL – IO (B) V O = D × V IN – V L (C) VL VL VL VL I L = ------ = ------------ = ---------- = -----XL 2πfL JωL sL (D) VO IC IC Ic = I C × X c = ------------- = ----------- = -----2πfC JωC sC The current compensator proportional gain is denoted as RA, and it has a dimension of resistance. The value of RA can be determined from the system characteristic equation. Higher value of RA implies higher current loop bandwidth. With the current mode control, the ‘D’ term performance in the voltage PID can be achieved. EQUATION 76: VX = VO + VL The current reference (IL*) is generated using the outer voltage loop. [IL* = (VO* - VO) * G] (because current loop performs the function of differential gain in the voltage loop, the outer voltage loop will have only proportional and integral gain). From the physical capacitor system, IC = IL - IO. In the equation, IO is made as constant and analyzed the relation between VO and VO*. Therefore, IL = SCVO. EQUATION 77: (F) KI ( I L )∗ = ( V O∗ – V O ) × ⎛ K P + -----⎞ ⎝ s⎠ KI ( Ra + sL ) I L × ------------------------ = [ ( V O∗ ) – V O ] × K P + ⎛ -----⎞ ⎝ s⎠ Ra The Equation 77 is rearranged to find VO*/VO and is shown in Equation 78. EQUATION 78: KI K P × R A + ⎛⎝ -----⎞⎠ V O∗ S ---------- = ---------------------------------------------------------------------------------------------------------VO KI 2 s LC + ( sC × R A ) + ( K P × R A ) + ⎛ -----⎞ × R A ⎝ s⎠ V X = V O + sLI L = R A × [ ( I L∗ ) – I L ] + [ V X – sLI L ] (E) [ R A × ( I L∗ ) ] I L = -----------------------------R A + sL © 2010 Microchip Technology Inc. DS01335A-page 33 AN1335 The denominator [s2LC + sCRa + KPRa + (KI/s)Ra] denotes the characteristic equation. The denominator should have three roots known as three poles or three bandwidths, f1 > f2 > f3 (units of Hz) of the controller. These roots correspond to current loop bandwidth (f1), proportional voltage loop bandwidth (f2) and integral voltage loop bandwidth (f3). These roots should be selected based on the system specifications. f1, f2 and f3 should be separated with a factor minimum of three between them. This ensures that any parameter variation (L and C) due to manufacturing tolerance or inductor saturation will not affect the stability of the system. EQUATION 81: The f3 determines the settling time (TS), that is the output voltage of the converter takes to settle within 98% of VO* for a step change in load. Ts should be selected less than the specification settling time. Substituting the actual design parameters used in the PSFB converter to have the KP, KI, RA gains. TS = 4/2πf3 The f2 determines the ability of the controller to track changes in VO*. If VO* varies, VO can track VO* variations up to a frequency f2 Hz. The f1 exists only to make the system non-oscillatory or resonant at frequencies greater than f2. The gains KP, KI and RA can be determined once f1, f2 and f3 are selected. The characteristic equation: s3LC + s2CRa + s KP Ra + KI Ra = 0 is a cubic equation. Because ‘s’ is -2πf1(ω1), -2πf2 (ω2) and -2πf3 (ω3), which are the roots of the characteristic equation and should make the equation equal to zero after substituting for ‘s’. The three unknown coefficients KP, KI and RA can be obtained by solving the following three equations shown in Equation 79: EQUATION 79: 2 3 2 3 ω 1 CR A + ω 1 K P R A + K I R A = – ω 1 LC ω 2 CR A + ω 2 K P R A + K I R A = – ω 2 LC 2 3 ω 3 CR A + ω 3 K P R A + K I R A = – ω 3 LC Y1 –1 Y = Y2 = A × B Y3 Y1 = C RA and RA = Y1/C Y2 = KP RA and KP = Y2/RA Y3 = KI RA and KI = Y3/RA Finding the Gains • Transformer turns ratio = 5:2 • Primary input voltage, VIN = 76V • Nominal primary input voltage, VNOM = 48V The maximum primary input current is selected as 9.75A and is reflected to the secondary because the controller exists on the secondary side of the isolation barrier. The base value of the current INBASE is 24.38A and the base value of the voltage VNBASE is 14.2V. All the voltage and current quantities are referenced with the base values INBASE and VNBASE. Transformer secondary voltage is: • VINS = VIN/turns ratio = 30.4V • Output inductor L = 3.4e-6 Henry • DC resistance of the inductor and tracks is considered as DCR = 0.05E • Output capacitance, C = 4576e-6F (4400 µF external to converter) • Equivalent series resistance of the capacitor, ESR = 0.0012E • Switching frequency of the converter, FSW = 150000 Hz • Control loop frequency TS is 1/2 of the FSW that is: 1 T S = ----------------F SW ⁄ 2 This can be solved by using the matrix method shown in Equation 80. • Integral voltage BW, f3 = -1000 * 2 * π • Proportional voltage BW, f2 = -2000 * 2 * π • Proportional current loop BW, f1 = -4000 * 2 * π EQUATION 80: The characteristic equation is solved using the above three bandwidths. ω1 2 ω1 1 ω2 2 ω2 ω3 2 ω3 3 – ω 1 LC CRa 3 1 × K P R A = – ω 2 LC KI RA 3 1 – ω 3 LC • RA = 0.1495 • KP = 57.5037 • KI = 2.0646e + 005 The matrix shown in Equation 80 is made equivalent to A * Y = B for simplicity purpose. DS01335A-page 34 © 2010 Microchip Technology Inc. AN1335 FIGURE 26: CONTROL LOOP COMPENSATOR DESIGN BLOCK DIAGRAM Inner Current Loop Compensator Outer Voltage Loop Compensator IL * DCR + VERROR + IREF (IL*) IERROR + Compensator VO* VO Scaling The gains calculated previously are based on real units (volts, amps, and so on). The dsPIC DSC consists of a fixed point processor and the values in the processor comprise linear relationship with the actual physical quantities they represent. The gains calculated are in real units, and cannot be directly applied to these scaled values (representation of physical quantities). Therefore, for the consistency these gains must be scaled. The scaling feedback section and the prescalar section provide general concepts of scaling. The basic idea behind scaling is the quantities that are to be added or subtracted should have the same scale. Scaling does not affect the structure of the control system block diagram. Scaling only affects the software representation of various quantities used in the software. Scaling Feedback To properly scale the PID gains, it is imperative to understand the feedback gain calculation. The feedback can be represented in various formats. Fractional format (Q15) is a very convenient representation. Fractional format allows easy migration of code from one design to another with different ratings where most of the changes that exist only in the coefficients and are defined in the header file. To use the available 16 bits in the processor, the Q15 format is most convenient as it allows signed operations and full utilization of the available bits (maximum resolution). Other formats can also be used, but resolution is lost in the process. Q15 allows using the fractional multiply MAC and MPY operation of the dsPIC DSC effectively. The feedback signal (typically voltage or current) is usually from a 10-bit ADC. Based on the potential divider or amplifier in the feedback circuitry, actual voltage and current is scaled. © 2010 Microchip Technology Inc. Compensator VX VL Phase/Duty + IL + VO Typically, the feedback 10-bit value (0 -1023) is brought to ±32767 range by multiplying with 32. This format is also known as Q15 format: Q15(m) where -1<m<1 and is defined as (int) (m * 32767). These formulae will have some error as 215 = 32768 is required, but due to finite resolution of 15 bits, only ±32767 is used. From a control perspective, for most systems these hardly introduce any significant error. In this format, +32767 correspond to +3.3V and 0 corresponds to 0V. Prescalar As most physical quantities are represented as Q15 format for easy multiplication with gains, the gains must also be represented in fractional format. If the value of gain (G * VNBASE/INBASE) is between -1 and +1, it can be easily represented as fractional format. Multiplication can then be performed using fractional multiply functions such as MAC or using builtin_mul functions and shifting appropriately. For example, z = (__builtin_mulss(x,y) >> 15) results in z = Q15(fx,fy), where all x, y, and z are in Q15 format (fx and fy are the fractions that are represented by x and y). In many cases, the gain terms are greater than unity. Because 16-bit fixed point is a limitation, a prescalar may be used to bring the gain term within the ± range. In this application, voltage loop proportional gain KP value is higher than one. Therefore, it is normalized using the defined current, voltage base values with the pre scalar 32. For simplifying the calculations, the voltage integral gain (KI) is also scaled with 32, that means if a prescalar is used for P term in a control block, it must also be used for the ‘I’ and ‘D’ term in the control block since all the terms are added together. To prevent the number overflows, PID output and ‘I’ output must be saturated to ±32767. The saturation limits for the PID output must be set at 1/32 of the original ±32767 to account for the prescalar. Therefore, saturation limits are set at ±1023. Finally, after saturation, the output must be post scaled by five to bring it to proper scale again. DS01335A-page 35 AN1335 Gain Scaling LOAD SHARING The voltage compensator input is in voltage dimensions and the output is in current dimensions, the voltage loop coefficients dimensions will be in mho (Siemens). In the traditional analog controller, regulation of the converter is achieved by a simple PWM controller, and load sharing of the converter is achieved by an additional load sharing controller/equivalent amplifier circuit. Recently, high end systems are calling for logging of converter parameters, which requires a microcontroller to communicate to the external world. Therefore, each converter needs a PWM controller, a load sharing controller, and a Microcontroller to meet the desired specifications. New value voltage loop proportional gain KP after normalizing and scaling will be (KP * VNBASE)/(INBASE * prescalar) that is 1.04. New value voltage loop integral gain, KI after normalizing and scaling will be KI * TS * VNBASE/ (INBASE * prescalar) = 0.0501. The current compensator input is in current dimensions and the output is in voltage dimensions, the current loop coefficients dimensions will be in Ω. New value current loop Integral gain, RA after normalizing is [(RA/VINS) * INBASE] = 0.1495. A few more contributors for the Phase/Duty control, are voltage decouple term and DCR compensation term. These are discussed below. Because at steady state (VL = 0), the average output of switching action will be equal to VO. A contribution of VO can be applied towards VX (the desired voltage at primary of the transformer). VO information is available in the software, so the voltage decouple term can be easily calculated. This will improve the dynamic performance and make the design of control system easier. PI output performs only small changes to correct for load and line variations and most of the variation in PHASE/DUTY is contributed by VO. The voltage decouple term after scaling will be VNBASE/VINS. The other parameters that need to be addressed are the resistance drops in the traces and magnetic winding resistance drop which may cause the current loop to function less than ideal. The dimension of gain of the current loop is in ohms. The physical resistance may interfere with the control action. If this resistance is known and measured during the design stage, then this resistance drop in the software can be compensated. The DC resistance compensation term after scaling will be (DCR/VINS) * INBASE. The input quantity should be in fractional format (this must be ensured in code). Then, the output current quantity will automatically be in the correct fractional quantity. This essentially solves the objective of scaling. The same logic applies to any control block. In the recent past, cost of the DSCs has reduced drastically and are highly attractive for use by power supply designers in their applications. Digital controllers are immune to component variations and have the ability to execute sophisticated nonlinear control algorithms, which are not common or unknown in analog controlled power systems. Apart from closing the control loop digitally, the DSC can perform fault management and communicate with the external applications which is becoming more and more significant in server applications. Digitally controlled power systems also offer advantages where very high precision, flexibility and intelligence are required. For the overcurrent protection or short circuit protection of the converters, load current or load equivalent current will be measured and the same will be used for the load sharing between the converters. Therefore, an additional circuitry/additional controller is not required in the case of a digitally controlled power supply compared to its analog counterpart for load sharing. This reduces overall cost as the component count is lower and easier to implement by adding a few lines of code to the stand-alone converter design. Digital Load Sharing Implementation Basic operation of the analog and digital load sharing concept is the same; however, implementation is completely different. In the digital implementation, the ADC will sample the continuous signals of output voltage and output current. The sampling frequency of the output voltage and output current signal is user configurable. The PID compensator design calculations are performed in the Interrupt Service Routine (ISR) and are updated based on the control loop frequency. By considering the input and output units and scale of each block to be implemented in software, the proper scaled values can be arrived. DS01335A-page 36 © 2010 Microchip Technology Inc. AN1335 In the dual load sharing implementation, for additional current, error information is added and this combined data will be given to the PWM module to generate appropriate phase/duty cycle. The PID compensator design will be same as the standalone individual converter. The load sharing compensator depends on the expected dynamic performance and this depends on the bandwidth of the current feedback. The current loop compensator forces the steady state error, (δIL) between individual converter currents IL1, IL2 and average current (IAVE) to zero. load sharing will be done with the single controller and this results in fewer components, less complexity and increased reliability. Poor noise immunity is a disadvantage of this design. Load sharing loop be 2πfL = 0.0021 gain, IKP will Load sharing loop integral gain, IKI will be 2πf5 IKI = 0.3356, where f5 (25 Hz) is the zero of the PI. New value voltage loop proportional gain, IKI after normalizing and scaling w ill be as shown below: Typically, temperature is a criteria for stress on the components and the junction temperature bandwidth is around 5 ms (about 30 Hz). Therefore, it is sufficient to use ~500 Hz bandwidth current data and the current share loop can have a bandwidth of ~100 Hz. Here, the DSC allows output voltage regulation by designing the voltage/current loop compensator and load current sharing by load current loop compensator design. Effectively, both the output voltage regulation and the FIGURE 27: proportional IKP * INBASE/VNBASE * prescaler2 * 1.25 = 0.0734 New value voltage loop proportional gain, IKI after normalizing and scaling will be as follows: IKI * INBASE/VNBASE * prescaler2 * TSLOADSHARE = 0.0092 In this application, the load sharing sampling time (TS LOADSHARE) is selected as 1 kHz. SINGLE WIRE LOAD SHARE COMPENSATOR DESIGN BLOCK DIAGRAM Outer Voltage Loop Compensator (IL * DCR) Inner Current Loop Compensator VERROR + VO* + IREF(IL*) + IERROR Compensator VX Phase/Duty + - VO + VL Compensator + IL1 VO Load Share Loop Compensator Load Share IL1 Converter 1 δ IL Compensator IAVEBLOCK (IL1 + IL2)/2 Load Share Loop Compensator Converter 2 δ IL Compensator IL2 Load Share Outer Voltage Loop Compensator Inner Current Loop Compensator + I REF(I L*) V ERROR + Compensator VO * (IL * DCR) IL2 - + - + IERROR Compensator VL Vx + Phase/Duty + VO VO © 2010 Microchip Technology Inc. DS01335A-page 37 AN1335 MATLAB MODELING The disturbance rejection plot is defined as: I(S)/VO(S). The .m file is used to generate the coefficients that are used in the MATLAB model (.mdl). This file also generates the scaled values to be used in the software. The generated values are in fractional format. In software, the coefficients must be represented as Q15(x), where ‘x’ is a fractional value. For more detailed calculations, refer to the MATLAB (.m) file in the PSFB_MATLAB file. For the MATLAB Simulink block diagram, refer to the MATLAB (.mdl) file. The transfer function IO(S)/VO(S) (with VO*(S) = 0) is called as dynamic stiffness or disturbance rejection. This plot explains us for a unit amplitude distortion in VO, the amount of load needed as a function of frequency. The system needs to be as robust as possible so that the output does not change under load. The higher this absolute figure of merit, the stiffer (better) the power supply output will be. The minimum is 35 db in this application, which will correlate to 56A (20logI = 35 dB) at approximately 1300 Hz of load producing 1.0V ripple on the output voltage. The following Bode plots (Figure 29 through Figure 31) are generated from the MATLAB (.m) file. Each plot is used to describe the behavior of the system. FIGURE 28: MATLAB® DIGITAL IMPLEMENTATION FOR THE PSFB CONVERTER (FROM MATLAB FILE) Quantizer2 Quantizer1 VO IL 12 VO * Phifactor VO* VIN Control System Zero-Order Hold2 Zero-Order Hold1 Phifactor VIN_PHIFACTOR LC Voltage 1 IL1 VIN IL1 VO1 VO2 ZVT Modulation L1 13.6 iLoad C VIN Scope1 Pulse Generator DS01335A-page 38 © 2010 Microchip Technology Inc. AN1335 FIGURE 29: DISTURBANCE REJECTION PLOT © 2010 Microchip Technology Inc. DS01335A-page 39 AN1335 The loop gain voltage plot illustrated in Figure 30 is used to calculate the phase and gain margin. In the plot, the phase margin (difference between 180º and the phase angle where the gain curve crosses 0 db) is 50º. To prevent the system from being conditionally unstable, it is imperative that the gain plot drops below 0 db when the phase reaches 180º. FIGURE 30: DS01335A-page 40 The blue curve is for the analog implementation and the green curve is for the digital implementation. It is generally recommended to have a phase margin of at least 40º to allow for parameter variations. The gain margin is the difference between gain curve at 0 db and where the phase curve hits 180º. The gain margin (where the green line on the phase plot reaches 180º) is -20 db. LOOP GAIN PLOT © 2010 Microchip Technology Inc. AN1335 Figure 31 illustrates the closed loop Bode plot. The point where the gain crosses -3 db or -45º in phase is usually denoted as the bandwidth. In this system, the bandwidth of the voltage loop is approximately 2700 Hz (17000 rad/s), which is closely matched by the Bode plot. FIGURE 31: CLOSED LOOP PLOT © 2010 Microchip Technology Inc. DS01335A-page 41 AN1335 SOFTWARE IMPLEMENTATION The Quarter Brick DC/DC Converter is controlled using the dsPIC33FJ16GS502 device. This device controls the power flow in the converter, fault protection, soft start, remote ON/OFF functionality, external communication, adaptive control for the synchronous MOSFET’s and single wire load sharing. Note: For more information on this device, refer to the “dsPIC33FJ06GS101/X02 and dsPIC33FJ16GSX02/X04 Data Sheet” (DS70318). For information on the peripherals, refer to Section 43. “High-Speed PWM” (DS70323), Section 44. “High-Speed 10Bit Analog-to-Digital Converter (ADC)” (DS70321), and Section 45. “HighSpeed Analog Comparator” (DS70296) in the “dsPIC33F/PIC24H Family Reference Manual”. These documents are available from the MIcrochip website (www.microchip.com). Init_CMC.c Functions present in this file are: init_PSFBDrive () Configure the primary MOSFET’s PWM module. init_SYNCRECTDrive () Configure the synchronous MOSFET’s PWM module. init_ADC() Configure the ADC module. InitRemoteON_OFF() Configure the System state for remote ON/OFF functionality. init_Timer1() Configure Timer1. Variables_CMC.c Description of Software Functional Blocks Declarations and Initialization of all the global variables. The source files and header files describe the functions used in the software. Compensator_CMC.c Source Files DigitalCompensator(void) Main_CMC.c Function to execute the voltage PI compensator and current P compensator. LoadshareCompensator(void) Functions present in this file are: main() Configures the operating frequency of the device. Function to execute compensator. the load share PI delay.s Configures the auxiliary clock module. Calls functions for configuring GPIO, ADC and PWM modules. _Delay to get ms delay. _Delay_Us to get µs delay. Checks for fault status. ADCP1Interrupt() Read values of currents and voltages. Check for any fault condition. If fault does not exist, execute the control loop. If fault exists, disable PWM outputs. INT1Interrupt() Remote ON/OFF functionality. T1Interrupt() Averaging the PID output. Over current limit selection. Over temperature fault. DS01335A-page 42 © 2010 Microchip Technology Inc. AN1335 Header Files dsp.h Define_CMC.h Standard library file for all DSP related operations. This file has all the global function prototype definitions and global parameter definitions. delay.h This is the file where all the modifications must be done based on the requirements of hardware components, power level, control loop bandwidth and other parameters. They are given below for reference. Presentable delay definition in ms and µs. Variables_CMC.h Supporting file for Variables_CMC.c and contains all the external global definitions. FIGURE 32: SOFTWARE FLOW CONTROL CMC WITH LOAD SHARING Wait for A/D Interrupt Initialization Soft Start Reset Voltage PI Compensator VO IREF Phase + ΔPhase PWM Phase Load Sharing PI Compensator ΔPhase ISHARE © 2010 Microchip Technology Inc. Current P Compensator IPSFB IPSFB DS01335A-page 43 AN1335 Digital Nonlinear Implementations DSCs allow implementing customized configurations to gain performance improvements of the SMPS. Adaptive Control to Improve the Efficiency Achieving ultra high efficiency specifications in power supply designs require unique configuration of PWM. This can be achieved by using external hardware or with software in digital controllers. In the PSFB converter, the software is designed to get the efficiency benefit at higher specified input voltages. Most of the DC/DC converters (part of AC/DC converter/Brick DC/DC converter) are designed using the isolation transformer for user safety and is also imposed by regulatory bodies. These power supplies are designed primary with push-pull, half-bridge, fullbridge and PSFB, in the secondary with synchronous MOSFET configurations to gain high efficiency. To avoid cross conduction, there will be a defined dead band and during this period neither of the synchronous MOSFET’s conduct so, the current will take the path of MOSFET body diode. These MOSFET body diodes has high forward drop compared to the RDS(ON) of the MOSFET, that is, VF * I >> IRMS2 * RDS(ON). Therefore, the losses are higher and the efficiency is less. DS01335A-page 44 © 2010 Microchip Technology Inc. AN1335 FIGURE 33: FULL-BRIDGE CONVERTER WITH CONVENTIONAL SYNCHRONOUS MOSFET GATE DRIVES Q1 Q3 Q6 TX LO TXVPRI Q2 Q4 CO Q5 Q1 / Q5 Q2 /Q 6 Q3 Q4 VPRI t During this period, there will be circulating currents in the Primary side MOSFET’s. These circulating currents are prominent at higher input voltages © 2010 Microchip Technology Inc. DS01335A-page 45 AN1335 These problems can be overcome by unique configuration of PWM gate drive of the synchronous MOSFETs. To control the output voltage of the converter with variation of input voltage, the duty cycle/phase is controlled. At high input voltages, the energy transfer from primary side to secondary side will be in small portions of the total period (zero states will exist). Due to the presence of inductors in secondary side of the converter, current continues to flow through the transformer coils through the MOSFET’s channel or through MOSFET body diodes. Due to reflection of current from secondary to primary, there will be a circulating current during the zero states in the primary and particularly this will be predominant at higher input voltage than the nominal input voltages of the input voltage range. Losses occurring during zero state of the primary side of the transformer can be avoided by overlapping the PWM gate drive of the synchronous MOSFETs. This method solves the problems which cause losses during zero states of the transformer. DS01335A-page 46 © 2010 Microchip Technology Inc. AN1335 FIGURE 34: FULL-BRIDGE CONVERTER WITH OVERLAP OF SYNCHRONOUS MOSFET GATE DRIVES Q1 Q3 Q6 TX L LO TXVPRI Q4 Q2 CO Q5 Q1 Q2 Q3 Q4 VPRI Q5 Q6 t Zero States © 2010 Microchip Technology Inc. DS01335A-page 47 AN1335 MOSFET body diode conduction in the primary side of the transformer is stopped so there are no reflected currents from the secondary side. The secondary side coils conduct in a way that there are no circulating currents in the primary side, effectively cancellation of currents. If a center tapped configuration is used in the secondary side of the transformer, the two coils cancel the flux and no flux is linked to the primary side because of the cancellation of currents. In case of “synchronous current doubler configuration” in the secondary side, both the synchronous MOSFETS are ON and the current does not pass in secondary side coil of the transformer, and therefore there is no reflected current in the primary side of the converter. This drastically reduces the circulating current losses in primary side body diodes of the MOSFETs. • In the case of center tapped transformer secondary configuration, instead of one synchronous MOSFET and one coil of the center tapped transformer, two synchronous MOSFETS and two transformer coils conduct simultaneously. Therefore, the secondary current will have only half the effective resistance, and the losses are reduced by half compared to when only one synchronous MOSFET is ON. • In the conventional switching methodology, intentional dead time is introduced between the two synchronous MOSFETS and typically this may be 10% of switching period based on the designs. During this dead time, the high secondary current flows through the high forward drop body MOSFET and cause losses. By configuring the overlap of the PWM gate drive of the synchronous MOSFET, the high secondary currents flow through the channel of the MOSFET. In this instance there will be only RDS(ON) losses that are very less compared to the losses incurred by the MOSFET body diodes in the dead time. Overcurrent Protection Implementation A current transformer is located in the primary side of the converter and the output of the current transformer also varies with the line conditions. To have the specific current limit across the line voltages, the compensator final output is averaged over a period of 10 ms. The compensator final output provides the line voltage variation data. This data is used as a modifier to change the current limit setting. DS01335A-page 48 PRINTED CIRCUIT BOARD (PCB) In the Quarter Brick DC/DC Converter design, an 18layer PCB is used to achieve the standard quarter brick dimensions. The PCB tracks routing is a challenging task in the quarter brick converter design. The PCB layers are described in Table 8. TABLE 8: PCB Layer Stacking of PCB Layers PCB Layer Description 1 Top layer traces, magnetic winding and component assembly. 2 Analog GND, magnetics and primary, and secondary side Cu pours. 3 4 5 6 Analog GND, +3.3V, magnetics and primary, and secondary side Cu pours. 7 Analog GND, gate drive traces, magnetics and primary, and secondary side Cu pours. 8 Analog GND, magnetics and primary, and secondary side Cu pours. 9 10 11 Analog GND, DIG GND, magnetics and primary, and secondary side Cu pours. 12 Analog GND, DIG GND, gate drive traces, magnetics and primary, and secondary side Cu pours. 13 Analog GND, DIG GND, magnetics and primary, and secondary side Cu pours. 14 Analog GND, DIG GND, gate drive traces, magnetics and primary, and secondary side Cu pours. 15 Analog GND, DIG GND, magnetics and primary, and secondary side Cu pours. 16 Analog GND, DIG GND, magnetics and primary, and secondary side Cu pours. 17 Digital GND and signal traces, magnetics and primary, and secondary side Cu pours. 18 Bottom layer traces, magnetic winding and component assembly. © 2010 Microchip Technology Inc. AN1335 LABORATORY TEST RESULTS AND CIRCUIT SCHEMATICS The Laboratory test results provide an overview of the quarter brick PSFB electrical specifications as well as the scope plots from initial test results. The test results are illustrated in Figure 35 to Figure 65. FIGURE 35: OUTPUT VOLTAGE RIPPLE: 8.5A AT 75V FIGURE 36: OUTPUT VOLTAGE RIPPLE: 17A AT 75V © 2010 Microchip Technology Inc. DS01335A-page 49 AN1335 FIGURE 37: OUTPUT VOLTAGE RIPPLE: 0A AT 75V FIGURE 38: OUTPUT VOLTAGE RIPPLE: 8.5A AT 48V DS01335A-page 50 © 2010 Microchip Technology Inc. AN1335 FIGURE 39: OUTPUT VOLTAGE RIPPLE: 17A AT 48V FIGURE 40: OUTPUT VOLTAGE RIPPLE: 0A AT 75V © 2010 Microchip Technology Inc. DS01335A-page 51 AN1335 FIGURE 41: OUTPUT VOLTAGE RIPPLE: 8.5A AT 36V FIGURE 42: OUTPUT VOLTAGE RIPPLE: 17A AT 36V DS01335A-page 52 © 2010 Microchip Technology Inc. AN1335 FIGURE 43: OUTPUT VOLTAGE TRANSIENT: 4.25A, 12.75A AT 48V FIGURE 44: OUTPUT VOLTAGE TRANSIENT: 4.25A, 12.75A AT 75V © 2010 Microchip Technology Inc. DS01335A-page 53 AN1335 FIGURE 45: OUTPUT VOLTAGE TRANSIENT: 4.25A, 12.75A AT 36V FIGURE 46: START-UP TIME: 8.5A AT 53V DS01335A-page 54 © 2010 Microchip Technology Inc. AN1335 FIGURE 47: OUTPUT VOLTAGE RAMP UP TIME: 17A AT 53V FIGURE 48: OUTPUT VOLTAGE RIPPLE: 8.5A AT 53V © 2010 Microchip Technology Inc. DS01335A-page 55 AN1335 FIGURE 49: OUTPUT VOLTAGE OVERSHOOT: 8.5A AT 53V FIGURE 50: REMOTE ON/OFF, OUTPUT VOLTAGE RISE TIME: 17A AT 53V DS01335A-page 56 © 2010 Microchip Technology Inc. AN1335 FIGURE 51: REMOTE ON/OFF, OUTPUT VOLTAGE FALL TIME: 17A AT 53V FIGURE 52: REMOTE ON/OFF, OUTPUT VOLTAGE FALL TIME: 0A AT 53V © 2010 Microchip Technology Inc. DS01335A-page 57 AN1335 FIGURE 53: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 48V/8.5A FIGURE 54: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 48V/17A DS01335A-page 58 © 2010 Microchip Technology Inc. AN1335 FIGURE 55: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 48V/0A FIGURE 56: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 76V/8.5A © 2010 Microchip Technology Inc. DS01335A-page 59 AN1335 FIGURE 57: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 76V/17A FIGURE 58: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 76V/0A DS01335A-page 60 © 2010 Microchip Technology Inc. AN1335 FIGURE 59: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 36V/8.5A FIGURE 60: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 36V/17A © 2010 Microchip Technology Inc. DS01335A-page 61 AN1335 FIGURE 61: PRIMARY TX AND SYNCHRONOUS FET GATE WAVEFORMS: 36V/0A FIGURE 62: SYNCHRONOUS MOSFET GATE AND DRAIN WAVEFORM: 48V/17A DS01335A-page 62 © 2010 Microchip Technology Inc. AN1335 FIGURE 63: SYNCHRONOUS MOSFET GATE AND DRAIN WAVEFORM: 76V/17A FIGURE 64: PRIMARY MOSFET GATE AND DRAIN WAVEFORM: 48V/17A © 2010 Microchip Technology Inc. DS01335A-page 63 AN1335 FIGURE 65: DS01335A-page 64 PRIMARY MOSFET GATE AND DRAIN WAVEFORM: 76V/17A © 2010 Microchip Technology Inc. AN1335 FIGURE 66: LOOP GAIN PLOT: 36V AND 12V/9A Phase Margin: 61.430 Gain Margin:-7.53 dB Crossover frequency: 2.17 kHz © 2010 Microchip Technology Inc. DS01335A-page 65 AN1335 FIGURE 67: LOOP GAIN PLOT: 48V AND 12V/9A Phase Margin: 59.800 Gain Margin:-6.508 dB Crossover frequency: 2.67 kHz DS01335A-page 66 © 2010 Microchip Technology Inc. AN1335 FIGURE 68: LOOP GAIN PLOT: 76V AND 12V/9A Phase Margin: 53.080 Gain Margin:-3.60 dB Crossover frequency: 3.57 kHz © 2010 Microchip Technology Inc. DS01335A-page 67 AN1335 CONCLUSION REFERENCES This application note presents the design of a PSFB Quarter Brick DC/DC Converter through the average current mode control using a Microchip dsPIC “GS” family Digital Signal Controller (DSC). Various nonlinear techniques implemented in this design explore the benefits of DSCs in Switched Mode Power Converter applications. The following resources are available from Microchip Technology Inc., and describe the use of dsPIC DSC devices for power conversion applications: Microchip has various resources to assist you in developing this integrated application. For more details on the PSFB Quarter Brick DC/DC Converter Reference Design using a dsPIC DSC, please contact your local Microchip sales office. DS01335A-page 68 • “dsPIC33FJ06GS101/X02 and dsPIC33FJ16GSX02/X04 Data Sheet” (DS70318) • Dedicated Switch Mode Power Supply (SMPS) Web site: http://www.microchip.com/SMPS In addition, the following resource was used in the development of this application note: “Design and Implementation of a Digital PWM Controller for a High-Frequency Switching DC-DC Power Converter”. Aleksandar Prodic, Dragan Maksimovic and Robert W. Erickson © 2010 Microchip Technology Inc. AN1335 APPENDIX A: SOURCE CODE Software License Agreement The software supplied herewith by Microchip Technology Incorporated (the “Company”) is intended and supplied to you, the Company’s customer, for use solely and exclusively with products manufactured by the Company. The software is owned by the Company and/or its supplier, and is protected under applicable copyright laws. All rights are reserved. Any use in violation of the foregoing restrictions may subject the user to criminal sanctions under applicable laws, as well as to civil liability for the breach of the terms and conditions of this license. THIS SOFTWARE IS PROVIDED IN AN “AS IS” CONDITION. NO WARRANTIES, WHETHER EXPRESS, IMPLIED OR STATUTORY, INCLUDING, BUT NOT LIMITED TO, IMPLIED WARRANTIES OF MERCHANTABILITY AND FITNESS FOR A PARTICULAR PURPOSE APPLY TO THIS SOFTWARE. THE COMPANY SHALL NOT, IN ANY CIRCUMSTANCES, BE LIABLE FOR SPECIAL, INCIDENTAL OR CONSEQUENTIAL DAMAGES, FOR ANY REASON WHATSOEVER. All of the software covered in this application note is available as a single WinZip archive file. This archive can be downloaded from the Microchip corporate Web site at: www.microchip.com © 2010 Microchip Technology Inc. DS01335A-page 69 FIGURE B-1: PSFB QUARTER BRICK DC/DC CONVERTER BOARD LAYOUT AND SCHEMATICS PSFB Quarter Brick DC/DC Converter Board Layout (Bottom View) Output Inductor (L2) Main Transformer (TX1) DT1 VO +ve VIN +ve U5 Q2 Q5 Remote ON/OFF © 2010 Microchip Technology Inc. VO -ve VIN -ve Q6 3.3V Regulator (U8) Auxiliary Transformer (TX3) U7 Auxiliary Controller (U9) DT2 dsPIC33FJ16GS502 (U1) Note: This view lists a few key components. Refer to the Bottom Silk drawing in Figure B-2, which lists all board components. AN1335 DS01335A-page 70 APPENDIX B: © 2010 Microchip Technology Inc. FIGURE B-2: PSFB Quarter Brick DC/DC Converter Board Layout (Bottom Silk) C47 C10 DT1 Q1 L2 C11 TX1 R5 C9 R1 C1 D1 R47 C40 DS01335A-page 71 C24 D15 D16 R59 C37 R61 DT2 AN1335 RS7 C23 C27 TX3 C41 R40 C25 C36 R43 C20 U9 C3 R62 R80 C14 R6 C31 D18 R58 L4 U1 R55 C32 L3 U8 R54 R76 C26 R48 C30 R77 C46 C13 U7 R74 C22 R60 C39 C28 C35 R81 C48 R46 C21 D2 R33 R41 U5 R35 C19 R73 Q5 R2 R44 C29 Q2 Q6 C7 PSFB Quarter Brick DC/DC Converter Board Layout (Top View) Q3 TX1 Q14 Q13 AN1335 DS01335A-page 72 FIGURE B-3: L2 DT1 Q4 J4 TX2 U4 © 2010 Microchip Technology Inc. U6 DT2 Note: TX3 U3 U2 This view lists a few key components. Refer to the Top Silk drawing in Figure B-4, which lists all board components. J1 © 2010 Microchip Technology Inc. FIGURE B-4: PSFB Quarter Brick DC/DC Converter Board Layout (Top Silk) C4 M DT1 Q3 C5 L2 C6 C8 R3 R11 R50 7 6 C18 R75 C42 C44 D7 R52 R39 R12 TX2 D8 R8 D4 R13 R4 15 14 R10 J4 R51 Q13 Q14 R49 Q4 D3 C2 R7 TX1 R28 C15 C12 R18 R15 R14 R20 R19 R16 R78 R79 C43 R69 R64 U6 R65 R66 DS01335A-page 73 J1 C38 R63 R68 AN1335 D14 R17 C45 C16 TX3 C33 R56 R22 U2 R21 C34 DT2 D17 R23 R53 C17 U4 U3 PSFB Quarter Brick DC/DC Converter Board Dimensions AN1335 DS01335A-page 74 FIGURE B-5: VO+ VIN+ ON/OFF VIN- VO- © 2010 Microchip Technology Inc. © 2010 Microchip Technology Inc. FIGURE B-6: PSFB Quarter Brick DC/DC Converter Schematic (Sheet 1 of 4) +12V R23 10 U2 1 2 3 4 PWM1H PWM1L R75 10K NC IN/A GND IN/B 8 NC1 7 OUT/A 6 VDD 5 OUT/B +12V C42 .1 µF R28 10 DT1 1 5 4 MCP1404-SO8 R74 10K GATE1 U3 1 2 3 4 PWM2H 6 7 S1 PWM2L NC IN/A GND IN/B R77 10K C16 2.2 µF 8 8 7 6 5 NC1 OUT/A VDD OUT/B DT2 1 5 4 6 7 MCP1404-SO8 GATE3 S3 C17 2.2 µF R76 10K GATE2 C44 .1 µF 8 GATE4 ANA_GND ANA_GND SECY side components. Require 2250 VDC isolation. SECY side components. Require 2250 VDC isolation. +12V R39 10 1 2 3 4 PWM3H PWM3L R79 10K NC IN/A GND IN/B 8 7 6 5 NC1 OUT/A VDD OUT/B GATE5 TEMP U5 5 NC1 NC VSS 4 VDD VOUT MCP9700-SC70 GATE6 MCP1404-SO8 +3.3V_ANA C19 2.2 µF ANA_GND C18 2.2 µF ANA_GND ANA_GND AN1335 DS01335A-page 75 R78 10K U4 1 2 3 PSFB Quarter Brick DC/DC Converter Schematic (Sheet 2 of 4) Pin 1 AN1335 4.7 S1 R3 GATE1 5 4.7 GATE6 R11 HAT2173H 10K 4 CT 5 5 3 2 1 R7 10K Pin 5 9 4.7 VO+ GATE3 S3 C2 L2a 11 C4 3.5 µH 22 µF C6 C7 C8 22 µF 22 µF 22 µF C5 22 5F C9 22 µF C10 C11 22 µF 22 µF R2 R4 4.7 8 7 5 VAux eSMP D18 TX2 Pin 4 L2b C45 47 µF R13 10K 3 2 1 GATE4 4 HAT2173H ANA_GND R12 5 5 Q6 DIG_GND 4.7 GATE5 4 3 2 1 Pin 3 R73 0.0 1 3 10K INPUT VOLTAGEGATE2 VO- 1 2 3 R8 1 2 3 R6 10K 4.7 D4 BAT54-XV 1 2 Q4 HAT2173H 4 D2 BAT54-XZ 2 1 Q2 HAT2173H 5 2.2 µF 4 2.2 µF R10 Q13 L1 C1 HAT2173H 7 4 1 TX1 3 2 1 R1 1 BAT54-XV 1 2 3 R5 10K BAT54-XV D3 2 1 2 3 1 Q3 HAT2173H 4 D1 2 5 Q1 HAT2173H 5 Q5 INPUT VOLTAGE+ 4 DS01335A-page 76 FIGURE B-7: Q14 HAT2173H R80 Pin 2 470 pF C46 150K CT REMOTE ON/OFF-I/P D7 BAS40 3 3 R81 4.99K D8 BAS40 +3.3V_ANA 1 8 R69 LOAD SHARE 47 C43 .1 µF U6B MCP6022-TSSOP + 7 - 5 R63 620 ANA_GND R15 100 R68 2K C38 470 pF R16 620 8 3 R14 5.6 ANA_GND 2.2 µF 1 2 100 6 4 R66 2 R18 2K C12 470 pF + 2 +3.3V_ANA U6A MCP6022-TSSOP R20 47 1 TX CURRENT C14 - .1 µF © 2010 Microchip Technology Inc. 4 ANA_GND R21 4.7K TX OVER CURRENT ANA_GND ANA_GND ANA_GND R64 R65 2K 620 ANA_GND ANA_GND R17 620 C15 470 pF R22 4.7K R19 2K ANA_GND ANA_GND ANA_GND ANA_GND DIG_GND VSecy C13 PSFB Quarter Brick DC/DC Converter Schematic (Sheet 3 of 4) R33 10 VO+ +3.3V_DIG REMOTE+ MCLR R48 4.7K R44 5.23K 1 2.2uF C22 2.2 +3.3V_ANA 2 OUTPUT FEEDBACK R46 1.5K C21 2200pF 2200 7.5K OUTPUT OVERVOLTAGE C20 TX OVER CURRENT EXTSYNCI1 1 2 3 4 5 6 7 AN2 AN3 CMP2C RP10 VSS CMP4a RP2 470pF 8 9 10 11 12 13 14 dsPIC33FJ16GS502-QFN28 DIG_GND VSS PGD 5 PGC N/C ICSP_6_HDR PWM2L PWM2H PWM3L PWM3H VDDCORE VSS SDA1 21 20 19 18 17 16 15 PWM2L PWM2H PWM3L PWM3H C23 2.2 µF PGD2 PGC2 DIG_GND VDD 6 GND LOAD SHARE VSecy OUTPUT OVERVOLTAGE TEMP R41 AN1 AN0 MCLR AVdd AVss PWM1L PWM1H U1 VO+ PGC2 28 27 26 25 24 23 22 R35 10 DIG_GND PWM1L PWM1H MCLR TX CURRENT OUTPUT FEEDBACK J1 MCLR 29 VO- 1K 4 PGD2 REMOTE- R43 3 C26 .1 µF ANA_GND PGD2 PGC2 Vdd RB8 RB15 RB5 SCL1 © 2010 Microchip Technology Inc. FIGURE B-8: DIG_GND COM3 COM4 REMOTE ON/OFF COM2 COM1 +3.3V_DIG REMOTE ON/OFF-I/P C24 2.2 µF R62 DIG_GND +3.3V_ANA 1.6K 4 1 +3.3V_DIG REMOTE ON/OFF 3 2 U7 2801 R49 R47 C25 470 pF 6.8K DIG_GND R50 4.99K COM3 DIG_GND COM4 J4.1 R51 220 DS01335A-page 77 R52 COM1 14 15 COM2 EXTSYNCI1 12 13 DIG_GND 10 11 8 9 6 7 220 LOAD SHARE REMOTE+ REMOTE- AN1335 PRI side components. Require 2250 VDC Isolation. 4.99K PSFB Quarter Brick DC/DC Converter Schematic (Sheet 4 of 4) AN1335 DS01335A-page 78 FIGURE B-9: VAux +12V D15 2 TX3 5 eSMP L4 C41 XPL2010 C30 47 µF 47 µF INPUT VOLTAGE + C33 2200 pF 6 7 4 8 R56 1K 1 MB 2 U8 2 D17 MURA110 eSMP C3 15uF 6 VIN VEN 47 2 4 7 D16 1 R57 +3.3V_DIG ANA_GND D14 GND GND1 R53 1 3 VOUT VOUT-SEN BYPASS LP3961-LLP6 10000 pF 1 +3.3V_ANA L3 1 3 5 XPL2010 C40 C29 C39 15 µF 15uF BAT54-XV C31 DIG_GND R54 10000 pF R58 10 R59 4.7K R60 1.6K 45.3K 8 7 6 5 VDRAIN VSS VCC CT UV FB OV COMP 1 2 3 4 +3.3V_ANA C35 NCP1031-SO8 U9 C32 10000 pF R55 C36 680pF C34 34K 2.2 µF C27 680 pF C37 10000 pF R40 0.0 R61 10K DIG_GND INPUT VOLTAGE - 2.2uF ANA_GND ANA_GND C28 15 µF © 2010 Microchip Technology Inc. AN1335 TABLE B-1: PSFB Quarter Brick DC/DC Converter Pin Out Details Pin Number Pin Designation Function 1 VIN+ Input Voltage Plus 2 Remote ON/OFF Remote ON/OFF 3 VIN- Input Voltage Minus 4 V0- Output Voltage Minus 5 V0+ Output Voltage Plus J4-6 Remote+ Remote Sense Plus J4-7 Remote- Remote Sense Minus J4-8 Load Share J4-9 NC J4-10 COM 4 J4-11 COM 3 J4-12 EXTSYNCI 1 J4-13 DIG_GND Single Wire Load Share Not Connected Serial Clock Input/Output Serial Data Input/Output External Synchronization Signal Digital Ground J4-14 COM 1 PORTB - 8 J4-15 COM 2 PORTB - 15 J1-1 MCLR Master Clear J1-2 +3.3V J1-3 DIG_GND J1-4 PGD2 Data I/O Pin for Programming/Debugging J1-5 PGC2 Clock Input Pin for Programming/Debugging © 2010 Microchip Technology Inc. Supply Digital Ground DS01335A-page 79 BASE BOARD SCHEMATIC P1 5 4 1 N/C VO- 5 TP7 TP8 White White Fan circuitry U2 7 1 8 C17 0.1 µF L1 D2 50SQ100 ON/OFF 3 FB GND R4 139K 5 SGND C15 68 µF LX 6 R3 1M BST VIN VD 220 µH C16 15 µF 4 2 MAX5035 J6 1 2 Fan Header 3 C18 0.1 µF 4 © 2010 Microchip Technology Inc. J3 Auxiliary Fan Input Input Power Selection for Fan S1 1 1 2 J7 3 1 3 4 5 2 2 4 C14 22 µF Black TP2 White C13 22 µF 2 J2 GND REMOTE- COM4 COM3 COM2 C12 100 µF 1 20 Black 14 COM1 11 TP3 VI- 13 3 C11 47 µF 3 ON/OFF C9 2200 µF 2 9 10 +3.3V MCLR 6 7 8 TP6 C8 2200 µF REMOTE+ 2 D1 3.3V TP1 4 DC-DC LOADSHARE TP5 VO+ 19 C7 5 4 PGC Orange C6 17 C5 Red 18 C4 EXTSYNCI1 C3 16 C2 1 3 J1 P2 2 2 C1 180 µF/100V 12 1 2.2 µF/100V C2-C7 VI+ PGD U1 1 GND Red TP4 15 FIGURE C-1: BASE BOARD SCHEMATIC AND LAYOUT AN1335 DS01335A-page 80 APPENDIX C: 6 J4 MCLR J5 1 1 2 VDD 2 3 VSS 3 4 PGD 4 PGC N.C. RJ-11 5 6 PMBUS TP9 White © 2010 Microchip Technology Inc. FIGURE C-2: BASE BOARD LAYOUT (TOP VIEW) R1 C7 C8 C5 C4 C3 C2 R5 D1 C10 C12 C14 C18 D3 U5 D5 C17 R3 C19 R4 AN1335 DS01335A-page 81 AN1335 BASE BOARD LAYOUT (BOTTOM VIEW) DS01335A-page 82 FIGURE C-3: © 2010 Microchip Technology Inc. AN1335 C.1 Efficiency Improvement Proposals The following proposals can be implemented to improve the efficiency of the converter. 1. 2. 3. 4. 5. Improving the rise and fall times of the MOSFETs. Investigating the feasibility of using a single gate drive transformer in the Full-Bridge reference design. Investigating the feasibility of using high-side and low-side drivers. Using 3+3 synchronous MOSFETs in the secondary rectifications. Investigating the feasibility of using fractional turns in the main transformer. In the present design, some of the layers were made using 2 oz. copper. As an improvement, these layers could be made using 4 oz. copper. © 2010 Microchip Technology Inc. DS01335A-page 83 AN1335 APPENDIX D: PSFB QUARTER BRICK DC/DC REFERENCE DESIGN DEMONSTRATION This appendix guides the user through the evaluation process to test the Quarter Brick DC/DC Converter. The Phase-Shifted Full-Bridge Quarter Brick DC/DC Converter Reference Design is a 200W output isolated converter with 36V-76V DC input and produces 12V DC output voltage. D.1 Tests Performed on the Quarter Brick DC/DC Converter • Input characteristics - Input undervoltage/overvoltage - No load power - Input power when remote ON/OFF is active • Output characteristics - Line regulation - Load regulation - Output voltage ramp-up time - Start-up time - Remote ON/ OFF start-up time - Remote ON/OFF shutdown fall time - Output overcurrent threshold - Output voltage ripple and noise - Load transient response • Efficiency of the converter FIGURE D-1: D.2 Test Equipment Required • DC source 30 VDC-100 VDC @ 8A (programmable DC power supply, 62012P-600-8 from Chroma or equivalent) • DC electronic load (DC electronic load 6314/ 63103 from Chroma or equivalent) • Digital multimeters (six and one-half digit multimeter, 34401A from Agilent or equivalent) • Oscilloscope (mixed-signal oscilloscope, MSO7054A from Agilent or equivalent) • Differential probe (high-voltage differential probe, P5200 from Tektronix or equivalent) D.3 Test Setup Description The Quarter Brick DC/DC Converter is assembled on the base board for evaluation purposes. The location of the Quarter Brick DC/DC Converter and its associated components used for testing are illustrated in Figure D-1. QUARTER BRICK DC/DC CONVERTER CONNECTED TO THE BASE BOARD IN THE REFERENCE DESIGN ENCLOSURE Quarter Brick DC/DC Converter DS01335A-page 84 © 2010 Microchip Technology Inc. AN1335 FIGURE D-2: FRONT VIEW OF THE QUARTER BRICK DC/DC CONVERTER REFERENCE DESIGN Note: The check mark on the front of the enclosure identifies the reference design model. FB = Full-Bridge Quarter Brick DC/DC Converter (to be discussed in a future application note) PSFB = Phase-Shifted Full-Bridge DC/DC Converter Use the following procedure to connect the DC load and source. 1. Connect the DC source +ve terminal and -ve terminals to the + and – input terminals (INPUT 3676V) of the connector, as illustrated in Figure D-3. FIGURE D-3: LEFT SIDE VIEW OF THE QUARTER BRICK DC/DC CONVERTER REFERENCE DESIGN © 2010 Microchip Technology Inc. 2. Connect the DC load +ve terminal and –ve terminals to the + and – output terminals (OUTPUT 12V) of the converter, as illustrated in Figure D-4. Note: The PROGRAM/DEBUG socket is used to program the converter with software. FIGURE D-4: RIGHT SIDE VIEW OF THE QUARTER BRICK DC/DC CONVERTER REFERENCE DESIGN DS01335A-page 85 AN1335 Use the following procedure to prepare the reference design for testing. 1. Connect the DMM +ve terminal and –ve terminals to the +ve and –ve terminals of the input current measurement resistor, as illustrated in Figure D-5. The current measurement resistor used to measure the input current is 10 mE. For example, if the measured voltage across the resistor is 60 mV, the input current will be 6A. FIGURE D-5: 2. INPUT CURRENT MEASUREMENT Connect the DMM +ve terminal and –ve terminals to the +ve and –ve terminals of the output current measurement resistor, as illustrated in Figure D-6. The current measurement resistor used to measure the output current is 5 mE. For example, if the measured voltage across the resistor is 85 mV, then the output current will be 17A. FIGURE D-6: DS01335A-page 86 OUTPUT CURRENT MEASUREMENT © 2010 Microchip Technology Inc. AN1335 3. Connect the DMM for input voltage measurement, as illustrated in Figure D-7. FIGURE D-7: 4. INPUT VOLTAGE MEASUREMENT Connect the DMM for output voltage measurement, as illustrated in Figure D-8. FIGURE D-8: OUTPUT VOLTAGE MEASUREMENT © 2010 Microchip Technology Inc. DS01335A-page 87 AN1335 5. Connect the oscilloscope probe for output voltage (in DC coupling) and ripple and noise (In AC coupling) measurement, as illustrated in Figure D-9. FIGURE D-9: 6. OUTPUT VOLTAGE MEASUREMENT Connect the oscilloscope probe for remote ON/ OFF testing, as illustrated in Figure D-10. FIGURE D-10: CONNECTING THE OSCILLOSCOPE PROBE FOR REMOTE ON/OFF TESTING Remote ON/OFF Pin Note: Differential probe must be used to monitor the remote ON/OFF signal. DS01335A-page 88 © 2010 Microchip Technology Inc. AN1335 7. Connect the oscilloscope probe for start-up time, as illustrated in the Figure D-11. FIGURE D-11: CONNECTING THE OSCILLOSCOPE PROBE FOR START-UP TIME Note: Differential probe must be used to monitor the input voltage. Instructions to connect two quarter brick converters for parallel operation 1. 2. 3. For N+1 system operations connect COMM2-3 (load share) of converter 1 to converter2 COMM2-3(load share). Connect a common DC source to the Converter1 and Converter2 input terminals as illustrated in Figure D-3. Connect a Common DC electronic load to the Converter1 and Converter2 output terminal as illustrated in Figure D-4. © 2010 Microchip Technology Inc. DS01335A-page 89 AN1335 D.4 Forced Air Cooling a) The Quarter Brick DC/DC Converter is designed to work with forced air cooling, which is provided by the fans illustrated in Figure D-2. Ensure that the fans are circulating air into the enclosure after providing the DC input supply at the + and – input terminals (INPUT 36-76V) of the connector, as illustrated in Figure D-3. D.5 b) Powering Up the Quarter Brick DC/DC Converter c) Before powering up the converter, ensure that polarity of the input source and DC load are connected as per the guidelines described in the section “Test Setup Description”. Typically, the unit may enter into the regulation range at around 35 VDC, undervoltage lockout at approximately 33.5 VDC, and overvoltage lockout at approximately 81 VDC. Use the following procedure to power up the reference design. 1. 2. 3. Turn the DC source ON and measure the input voltage with DMM, as illustrated in Figure D-7. This voltage should be in the range of 36 VDC76 VDC. Check to see that the fans are circulating air into the enclosure. Ensure that the connected DC load is in the range of 0A-17A. The output load current measurement resistor provides a value in the range of 0 mV-85 mV when measuring with DMM, as illustrated in Figure D-6. Ensure that the output voltage read by DMM (see Figure D-8) is in the range of 11.88 VDC to 12.12 VDC. D.6 Test Procedure The following two sections provide detailed procedures for each test. D.6.1 1. INPUT CHARACTERISTICS Input undervoltage/overvoltage. The Quarter Brick DC/DC Converter is rated to operate with regulation between the input voltage ranges 36 VDC-76 VDC. The converter features input undervoltage and overvoltage protection. This feature will not allow the converter to start-up unless the input voltage exceeds the turn-on voltage threshold and shuts down the converter when the input voltage exceeds the overvoltage threshold. DS01335A-page 90 Set the DC load at 8.5A and increment the input voltage from 33 VDC (read the input voltage with DMM illustrated in Figure D-7) to the voltage where output voltage is in the regulation range of 11.88 VDC to 12.12 VDC. Read the output voltage with DMM illustrated in Figure D-8. Start decrementing the input voltage and observe at what input voltage the converter shuts OFF. This input voltage point will be the input undervoltage threshold. Start incrementing the voltage from 76 VDC input and observe at what input voltage converter shuts OFF. This input voltage point will be the input overvoltage threshold. 2. No load power. a) Set the input voltage at 53 VDC and disconnect or turn OFF the load from the converter and record the input power. This value will be the product of input voltage and input current measured using the DMM illustrated in Figure D-5 and Figure D-7. 3. Input power when remote ON/OFF is active. Remote ON/OFF will be used to turn OFF the converter by applying a 3.3 VDC signal on the pin illustrated in Figure D-10. A high signal (3.3 VDC) will turn OFF the converter and there is no output. When a high signal is sensed by the dsPIC DSC, all of the PWM generators are shutdown. When the dsPIC DSC detects a low remote ON/OFF signal, the converter will be turned ON. a) b) Turn ON the converter with 53 VDC input at 8.5A output load. Connect an oscilloscope voltage probe to measure the output voltage and a differential voltage probe to measure the external 3.3 VDC supply, as illustrated in Figure D-10. Turn ON the external 3.3 VDC supply and the system will shut down (there will be no voltage at the output of the converter). Record the input voltage and input current to calculate the input power. © 2010 Microchip Technology Inc. AN1335 D.6.2 1. OUTPUT CHARACTERISTICS 7. The output overcurrent limit will protect the unit from excessive loading than the rated load current. Increment the output load beyond the rated 17A, the converter enters into Hiccup mode for a few milliseconds. If overcurrent persists, the converter enters into Latch mode. Line regulation. Change the input DC voltage from 36 VDC to 76 VDC to the converter and record the output voltage. The output voltage deviation should be in the range of 11.88 VDC to 12.12 VDC. 2. Load regulation. Change the output load from 0A to 17A at various input voltages in the range of 36 VDC to 76 VDC and record the output voltage variations. The output voltage deviation should be in the range of 11.88 VDC to 12.12 VDC. 3. Set the input voltage at various points in the specified range 36 VDC to 76 VDC and increment the load at the output insteps. To monitor the output voltage, connect the voltage probe, as illustrated in Figure D-9. 8. Output voltage ramp-up time. Start-up time. This is the time when the input voltage applied to the converter (in the range of 36 VDC-76 VDC) when the output voltage reaches 90% of the rated 12V output voltage. Connect the voltage differential probe at the input voltage terminals and the voltage probe at the output to the oscilloscope, as illustrated in Figure D-11. 5. 9. Output voltage ripple and noise. Measure the AC component on the output voltage of the converter by connecting the oscilloscope output voltage probe, as illustrated in Figure D-9. Read the output voltage by configuring the oscilloscope in the AC couple mode. The output ripple is measured in terms of peak-to-peak voltage. Remote ON/OFF start-up time. Remote ON/OFF will be used to disable/enable the converter by applying or removing a 3.3 VDC signal on the Remote ON/OFF pin, as illustrated in Figure D-10. Applying 3.3 VDC on the remote ON/ OFF pin turns the converter OFF. Remote ON/OFF start-up time is the time duration from when the remote ON/OFF is disabled, to when the output voltage rises to 90% of the rated output voltage. 6. Load transient response. Observe the variation on the DC output voltage while step changing the output load from 25% to the 75% of the rated output load 17A. The parameters to be measured are peak-to-peak output voltage variation and load transient recovery time. Configure the oscilloscope in AC couple mode and connect the oscilloscope output voltage probe as illustrated in Figure D-9 to measure the peak-to-peak output voltage variation and load transient recovery time. Turn ON the converter with the specified input voltage in the range of 36 VDC to 76 VDC and observe the DC output voltage raise time. Ramp-up time is the time taken to reach output voltage from 10% to 90% of the rated output voltage. Ramp-up time can be measured by connecting the oscilloscope voltage probe, as illustrated in Figure D-9. 4. Output overcurrent threshold. Remote ON/OFF shut down fall time. Removing the 3.3 VDC signal on the remote ON/ OFF pin, turns the converter ON. The remote ON/ OFF fall time is the time duration from when the remote ON/OFF signal is enabled, to when the output voltage falls to 10% of the rated output voltage. D.7 Efficiency is the ratio of output power to the input power: Efficiency (%) = Output Power / Input Power * 100 = [(Output voltage * Output current) / (Input Voltage * Input Current)] * 100 Use the following procedure to measure the efficiency of the converter. 1. 2. 3. 4. © 2010 Microchip Technology Inc. Efficiency of the Quarter Brick DC/ DC Converter Connect the DMM +ve terminal and –ve terminals to the +ve and –ve of the input current measurement resistor, as illustrated in Figure D-5. Connect the DMM +ve terminal and –ve terminals to the +ve and –ve of the output current measurement resistor, as illustrated in Figure D-6. Connect the DMM for input voltage measurement, as illustrated in Figure D-7. Connect the DMM for output voltage measurement, as illustrated in Figure D-8. DS01335A-page 91 AN1335 D.8 COMM 1 and COMM 2 Connectivity The COMM 1 and COMM 2 signal connectors, pin termination, and functionality are described in Table D-1. The pin sequence is illustrated in Figure D-12. TABLE D-1: Pin PIN, PERIPHERAL AND FUNCTIONALITY TABLE Peripheral COMM 1 - 1 RB8 Functionality Remappable I/O COMM 1 - 2 — COMM 1 - 3 VSS COMM 1 - 4 SDA1 Synchronous serial data input/output for I2C1. COMM 1 - 5 SCL1 Synchronous serial clock input/output for I2C1. COMM 1 - 6 RB15 Remappable I/O. COMM 2 - 1 — COMM 2 - 2 — COMM 2 - 3 AN2 COMM 2 - 4 RP2/SYNCI1 No connect. DIG_GND Remote Sense -ve. Remote Sense +ve. Load share. External synchronization signal to PWM master time base. FIGURE D-12: COMM 1 AND COMM 2 SIGNAL CONNECTORS Pin 6 Pin 1 Pin 4 Pin 1 Note 1: For N+1 system operations connect COMM2-3 (load share) of Converter 1 to Converter2 COMM2-3(load share). 2: Connect a common DC source to the Converter1 and Converter2 input terminals as illustrated in Figure D-3. 3: Connect a Common DC electronic load to the Converter1 and Converter2 output terminal as illustrated in Figure D-4. DS01335A-page 92 © 2010 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART, PIC32 logo, rfPIC and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor, MXDEV, MXLAB, SEEVAL and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, HI-TIDE, In-Circuit Serial Programming, ICSP, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, mTouch, Octopus, Omniscient Code Generation, PICC, PICC-18, PICDEM, PICDEM.net, PICkit, PICtail, REAL ICE, rfLAB, Select Mode, Total Endurance, TSHARC, UniWinDriver, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2010, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. ISBN: 978-1-60932-409-4 Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. © 2010 Microchip Technology Inc. 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