LT3957 - Boost, Flyback, SEPIC and Inverting Converter with 5A, 40V Switch

LT3957
Boost, Flyback, SEPIC and
Inverting Converter
with 5A, 40V Switch
DESCRIPTION
FEATURES
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Wide Input Voltage Range: 3V to 40V
Single Feedback Pin for Positive or Negative
Output Voltage
Internal 5A/40V Power Switch
Current Mode Control Provides Excellent Transient
Response
Programmable Operating Frequency (100kHz to
1MHz) with One External Resistor
Synchronizable to an External Clock
Low Shutdown Current < 1μA
Internal 5.2V Low Dropout Voltage Regulator
Programmable Input Undervoltage Lockout with
Hysteresis
Programmable Soft-Start
Thermally Enhanced QFN (5mm × 6mm) Package
The LT®3957 is a wide input range, current mode DC/DC
converter which is capable of generating either positive
or negative output voltages. It can be configured as either
a boost, flyback, SEPIC or inverting converter. It features
an internal low side N-channel power MOSFET rated for
40V at 5A and driven from an internal regulated 5.2V
supply. The fixed frequency, current-mode architecture
results in stable operation over a wide range of supply
and output voltages.
The operating frequency of LT3957 can be set with an
external resistor over a 100kHz to 1MHz range, and can
be synchronized to an external clock using the SYNC pin.
A minimum operating supply voltage of 3V, and a low
shutdown quiescent current of less than 1μA, make the
LT3957 ideally suited for battery-powered systems.
The LT3957 features soft-start and frequency foldback
functions to limit inductor current during start-up.
APPLICATIONS
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Automotive
Telecom
Industrial
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents pending.
TYPICAL APPLICATION
High Efficiency Output Boost Converter
Efficiency vs Output Current
VIN
4.5V TO 16V
10μF
200k
VIN
10μF
s2
SW
GND
EN/UVLO
95.3k
LT3957
SGND
SENSE1
SYNC
SENSE2
226k
VOUT
24V
600mA
100
VIN = 12V
95
EFFICIENCY (%)
10μH
90
85
80
FBX
RT
41.2k
300kHz
SS
VC
0.33μF
INTVCC
6.8k
15.8k
4.7μF
75
70
0
22nF
100
600
400
500
200
300
OUTPUT CURRENT (mA)
3957 TA01b
3957 TA01a
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LT3957
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
VC
FBX
SS
RT
SYNC
NC
TOP VIEW
NC
VIN, EN/UVLO (Note 5), SW ......................................40V
INTVCC ......................................................VIN + 0.3V, 8V
SYNC ..........................................................................8V
VC, SS .........................................................................3V
RT ............................................................................................... 1.5V
SENSE1, SGND .................. Internally Connected to GND
SENSE2..................................................................±0.3V
FBX ................................................................. –6V to 6V
Operating Junction Temperature Range
(Note 2).................................................. –40°C to 125°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range .................. –65°C to 125°C
36 35 34 33 32 31 30
NC 1
28 INTVCC
27 VIN
NC 2
SENSE2 3
SGND
37
SGND 4
25 EN/UVLO
24 SGND
23 SGND
SENSE1 6
SW
38
SW 8
SW 9
21 SW
20 SW
NC 10
GND
GND
GND
GND
GND
GND
12 13 14 15 16 17
UHE PACKAGE
36-LEAD (5mm s 6mm) PLASTIC QFN
TJMAX = 125°C, θJA = 42°C/W, θJC = 3°C/W
EXPOSED PAD (PIN 37) IS SGND, MUST BE SOLDERED TO SGND PLANE
EXPOSED PAD (PIN 38) IS SW, MUST BE SOLDERED TO SW PLANE
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3957EUHE#PBF
LT3957EUHE#TRPBF
3957
36-Lead (5mm × 6mm) Plastic QFN
–40°C to 125°C
LT3957IUHE#PBF
LT3957IUHE#TRPBF
3957
36-Lead (5mm × 6mm) Plastic QFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3957f
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LT3957
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA ≈ TJ = 25°C. VIN = 24V, EN/UVLO = 24V, SENSE2 = 0V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
VIN Operating Range
TYP
3
MAX
UNITS
40
V
VIN Shutdown IQ
EN/UVLO = 0V
EN/UVLO = 1.15V
0.1
1
6
μA
μA
VIN Operating IQ
VC = 0.3V, RT = 41.2k
1.7
2.3
mA
VIN Operating IQ with Internal LDO Disabled
VC = 0.3V, RT = 41.2k, INTVCC = 5.5V
350
400
μA
5.9
6.8
A
l
SW Pin Current Limit
5
SW Pin On Voltage
ISW = 3A
100
mV
SENSE2 Input Bias Current
Current Out of Pin
–65
μA
Error Amplifier
FBX Regulation Voltage (VFBX(REG))
FBX > 0V (Note 3)
FBX < 0V (Note 3)
FBX Overvoltage Lockout
FBX > 0V (Note 4)
FBX < 0V (Note 4)
FBX Pin Input Current
FBX = 1.6V (Note 3)
FBX = –0.8V (Note 3)
l
l
1.569
–0.816
1.6
–0.800
1.631
–0.784
V
V
6
7
8
11
10
14
%
%
70
100
10
nA
nA
–10
Transconductance gm (ΔIVC /ΔFBX)
(Note 3)
230
μS
VC Output Impedance
(Note 3)
5
MΩ
VFBX Line Regulation (ΔVFBX /[ΔVIN • VFBX(REG)])
FBX > 0V, 3V < VIN < 40V (Notes 3, 6)
FBX < 0V, 3V < VIN < 40V (Notes 3, 6)
0.04
0.03
VC Current Mode Gain (ΔVVC /ΔVSENSE)
0.06
0.06
%/V
%/V
10
V/V
VC Source Current
VC = 1.5V, FBX = 0V, Current Out of Pin
–15
μA
VC Sink Current
FBX = 1.7V
FBX = –0.85V
12
11
μA
μA
Oscillator
Switching Frequency
RT = 140k to SGND, FBX = 1.6V, VC = 1.5V
RT = 41.2k to SGND, FBX = 1.6V, VC = 1.5V
RT = 10.5k to SGND, FBX = 1.6V, VC = 1.5V
RT Voltage
FBX = 1.6V
80
270
850
100
300
1000
120
330
1200
1.2
kHz
kHz
kHz
V
SW Minimum Off-Time
220
275
ns
SW Minimum On-Time
240
320
ns
SYNC Input Low
0.4
SYNC Input High
SS Pull-Up Current
1.5
SS = 0V, Current Out of Pin
–10
μA
Low Dropout Regulator
l
INTVCC Regulation Voltage
INTVCC Undervoltage Lockout Threshold
Falling INTVCC
UVLO Hysteresis
5
5.2
5.45
V
2.6
2.7
0.15
2.85
V
V
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LT3957
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA ≈ TJ = 25°C. VIN = 24V, EN/UVLO = 24V, SENSE2 = 0V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
INTVCC Current Limit
VIN = 40V
VIN = 15V
32
40
95
55
INTVCC Load Regulation (ΔVINTVCC / VINTVCC)
0 < IINTVCC < 20mA, VIN = 8V
–1
–0.5
UNITS
mA
mA
%
INTVCC Line Regulation (ΔVINTVCC / [ΔVIN • VINTVCC]) 6V < VIN < 40V
0.02
Dropout Voltage (VIN – VINTVCC)
VIN = 5V, IINTVCC = 20mA, VC = 0V
450
mV
INTVCC Current in Shutdown
EN/UVLO = 0V, INTVCC = 6V
17
μA
INTVCC Voltage to Bypass Internal LDO
0.05
%/V
5.5
V
Logic Inputs
l
VIN = INTVCC = 6V
EN/UVLO Threshold Voltage Falling
1.17
EN/UVLO Voltage Hysteresis
IVIN Drops Below 1μA
EN/UVLO Pin Bias Current Low
EN/UVLO = 1.15V
EN/UVLO Pin Bias Current High
EN/UVLO = 1.33V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3957E is guaranteed to meet performance specifications
from the 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LT3957I is guaranteed over the full –40°C to 125°C operating junction
temperature range.
1.7
Positive Feedback Voltage
vs Temperature, VIN
VIN = 8V
1590
VIN = INTVCC = 3V,
SHDN/UVLO = 1.33V
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3957 G01
VIN = INTVCC = 3V
SHDN/UVLO = 1.33V
–792
–794
VIN = 8V
–796
–798
–800
2
2.5
μA
20
100
nA
1.8
QUIESCENT CURRENT (mA)
REGULATED FEEDBACK VOLTAGE (mV)
REGULATED FEEDBACK VOLTAGE (mV)
VIN = 24V
–790
V
Quiescent Current
vs Temperature, VIN
–788
VIN = 40V
0.4
TA ≈ TJ = 25°C, unless otherwise noted.
Negative Feedback Voltage
vs Temperature, VIN
1605
1580
–50
V
mV
Note 3: The LT3957 is tested in a feedback loop which servos VFBX to the
reference voltages (1.6V and –0.8V) with the VC pin forced to 1.3V.
Note 4: FBX overvoltage lockout is measured at VFBX(OVERVOLTAGE) relative
to regulated VFBX(REG).
Note 5: For 3V ≤ VIN < 6V, the EN/UVLO pin must not exceed VIN.
Note 6: EN/UVLO = 1.33V when VIN = 3V.
TYPICAL PERFORMANCE CHARACTERISTICS
1585
1.27
20
EN/UVLO Input Low Voltage
1600
1.22
VIN = 24V
VIN = 40V
1.7
VIN = 40V
VIN = 24V
1.6
VIN = INTVCC = 3V
1.5
–802
–804
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3957 G02
1.4
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3957 G03
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LT3957
TYPICAL PERFORMANCE CHARACTERISTICS
Dynamic Quiescent Current
vs Switching Frequency
TA ≈ TJ = 25°C, unless otherwise noted.
Normalized Switching
Frequency vs FBX
RT vs Switching Frequency
120
1000
12
NORMALIZED FREQUENCY (%)
10
RT (kΩ)
IQ(mA)
8
6
100
4
2
10
0
100 200 300 400 500 600 700 800 900 1000
SWITCHING FREQUENCY (kHz)
40
20
310
305
300
295
290
285
0
0.4
0.8
FBX VOLTAGE (V)
1.2
1.6
SW Pin Current Limit
vs Duty Cycle
6.6
6.6
6.4
6.4
SW PIN CURRENT LIMIT (A)
RT = 41.2k
315
–0.4
3957 G06
SW Pin Current Limit
vs Temperature
SW PIN CURRENT LIMIT (A)
SWITCHING FREQUENCY (kHz)
60
3957 G05
Switching Frequency
vs Temperature
320
80
0
–0.8
0 100 200 300 400 500 600 700 800 900 1000
SWITCHING FREQUENCY (kHz)
3957 G04
325
100
6.2
6.0
5.8
5.6
6.2
6.0
5.8
5.6
280
275
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
5.4
–50
125
5.4
–25
50
25
0
75
TEMPERATURE (°C)
100
EN/UVLO Threshold
vs Temperature
1.24
EN/UVLO FALLING
40
2.4
30
2.2
20
10
0
50
25
0
75
TEMPERATURE (°C)
100
125
3957 G10
80
100
2.0
1.8
1.20
–25
40
60
DUTY CYCLE (%)
EN/UVLO Hysteresis Current
vs Temperature
IEN/UVLO (μA)
EN/UVLO CURRENT (μA)
EN/UVLO VOLTAGE (V)
1.26
EN/UVLO RISING
20
3957 G09
EN/UVLO Current vs Voltage
1.28
1.18
–50
0
3957 G08
3957 G07
1.22
125
0
10
20
30
EN/UVLO VOLTAGE (V)
40
3957 G11
1.6
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3957 G12
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LT3957
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC Minimum Output
Current Limit vs VIN
INTVCC vs Temperature
90
5.4
5.2
5.1
VIN = 6V
5.2
70
INTVCC VOLTAGE (V)
INTVCC CURRENT (mA)
5.3
INTVCC Load Regulation
5.3
TJ = 125°C
INTVCC = 3V
80
INTVCC (V)
TA ≈ TJ = 25°C, unless otherwise noted.
60
50
40
30
20
5.1
5.0
4.9
10
5.0
–50
50
25
0
75
TEMPERATURE (°C)
–25
100
0
125
1
10
VIN (V)
4.8
100
0
40
30
20
INTVCC LOAD (mA)
3957 G14
3957 G13
700
5.30
VIN = 5V
50
125°C
45
5.15
40
75°C
500
ON-RESISTANCE (mΩ)
DROPOUT VOLTAGE (mV)
5.20
25°C
400
0°C
300
–40°C
200
35
30
25
20
15
10
100
5
0
5.10
0
5
10
15
20 25
VIN (V)
30
35
5
0
40
10
15
20
0
–50
–25
0
25
50
28.2
100
125
3957 G18
SEPIC FBX Frequency Foldback
Waveforms During Overcurrent
SEPIC Typical Start-Up
Waveforms
Internal Switch On-Resistance
vs INTVCC
75
TEMPERATURE (°C)
INTVCC LOAD (mA)
3957 G16
3957 G17
VIN = 12V
VIN = 12V
VOUT
10V/DIV
28.0
ON-RESISTANCE (mΩ)
60
Internal Switch On-Resistance
vs Temperature
600
5.25
50
3957 G15
INTVCC Dropout Voltage
vs Current, Temperature
INTVCC Line Regulation
INTVCC VOLTAGE (V)
10
27.8
VOUT
5V/DIV
27.6
VSW
20V/DIV
27.4
27.2
IL1A + IL1B
5A/DIV
IL1A + IL1B
2A/DIV
27.0
26.8
5ms/DIV
26.6
3
4
5
6
7
8
3957 G20
SEE TYPICAL APPLICATION: 5V TO 16V INPUT,
12V OUTPUT SEPIC CONVERTER
50μs/DIV
3957 G21
SEE TYPICAL APPLICATION: 5V TO 16V INPUT,
12V OUTPUT SEPIC CONVERTER
INTVCC (V)
3957 G19
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LT3957
PIN FUNCTIONS
NC (Pins 1, 2, 10, 35, 36): No Internal Connection. Leave
these pins open or connect them to the adjacent pins.
SENSE2 (Pin 3): The Current Sense Input for the Control
Loop. Connect this pin to SENSE1 pin directly or through
a low pass filter (connect this pin to SENSE1 pin through
a resistor, and to SGND through a capacitor).
SGND (Pins 4, 23, 24, Exposed Pad Pin 37): Signal
Ground. All small-signal components should connect to
this ground. SGND is connected to GND inside the IC to
ensure Kelvin connection for the internal switch current
sensing. Do not connect SGND and GND externally.
SENSE1 (Pin 6): The Current Sense Output of the Internal N-channel MOSFET. Connect this pin to SENSE2 pin
directly or through a low pass filter (connect this pin to
SENSE1 pin through a resistor, then connect SENSE2 to
SGND through a capacitor).
SW (Pins 8, 9, 20, 21, Exposed Pad Pin 38): Drain of
Internal Power N-channel MOSFET.
GND (Pins 12, 13, 14, 15, 16, 17): Ground. These pins
connect to the source terminal of internal power N-channel
MOSFET through an internal sense resistor. GND is connected to SGND inside the IC to ensure Kelvin connection
for the internal switch current sensing. Do not connect
GND and SGND externally.
EN/UVLO (Pin 25): Shutdown and Undervoltage Detect
Pin. An accurate 1.22V (nominal) falling threshold with
externally programmable hysteresis detects when power
is okay to enable switching. Rising hysteresis is generated
by the external resistor divider and an accurate internal
2μA pull-down current. An undervoltage condition resets
soft-start. Tie to 0.4V, or less, to disable the device and
reduce VIN quiescent current below 1μA.
VIN (Pin 27): Input Supply Pin. The VIN pin can be locally
bypassed with a capacitor to GND (not SGND).
INTVCC (Pin 28): Regulated Supply for Internal Loads
and Gate Driver. Supplied from VIN and regulated to
5.2V (typical). INTVCC must be bypassed to SGND with a
minimum of 4.7μF capacitor placed close to pin. INTVCC
can be connected directly to VIN, if VIN is less than 8V.
INTVCC can also be connected to a power supply whose
voltage is higher than 5.5V, and lower than VIN, provided
that supply does not exceed 8V.
VC (Pin 30): Error Amplifier Compensation Pin. Used to
stabilize the voltage loop with an external RC network.
Place compensation components between the VC pin
and SGND.
FBX (Pin 31): Positive and Negative Feedback Pin. Receives the feedback voltage from the external resistor
divider between the output and SGND. Also modulates the
switching frequency during start-up and fault conditions
when FBX is close to SGND.
SS (Pin 32): Soft-Start Pin. This pin modulates compensation pin voltage (VC) clamp. The soft-start interval is
set with an external capacitor between SS pin and SGND.
The pin has a 10μA (typical) pull-up current source to
an internal 2.5V rail. The soft-start pin is reset to SGND
by an undervoltage condition at EN/UVLO, an INTVCC
undervoltage or overvoltage condition or an internal
thermal lockout.
RT (Pin 33): Switching Frequency Adjustment Pin. Set
the frequency using a resistor to SGND. Do not leave this
pin open.
SYNC (Pin 34): Frequency Synchronization Pin. Used to
synchronize the switching frequency to an outside clock.
If this feature is used, an RT resistor should be chosen
to program a switching frequency 20% slower than the
SYNC pulse frequency. Tie the SYNC pin to SGND if this
feature is not used. SYNC is bypassed when FBX is close
to SGND.
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LT3957
BLOCK DIAGRAM
CDC
L1
R4
R3
L2
EN/UVLO
A10
IS1
2μA
IS2
10μA
–
+
VIN
27
COUT
•
SW
8, 9, 20,
21, 38
1.22V
INTERNAL
REGULATOR
AND UVLO
CURRENT
LIMIT
UVLO
M2
5.2V LDO
A8
Q3
2.5V
VOUT
CIN
25
2.5V
D1
•
VIN
28
G4
INTVCC
2.7V
1.72V
–
+
CVCC
IS3
A11
TLO
165˚C
DRIVER
G6
–0.88V
–
+
A12
VC
Q2
1.6V
SR1
–
+A7
G5
R
G2
O
M1
S
6
PWM
COMPARATOR
+
A1
–
RSENSE
VISENSE
SLOPE
–
A6
+
–0.8V
RAMP
+
A2
–
1.28V
RAMP
GENERATOR
–
+A3
1.2V
+
+
–
31
FBX
30
VC
32
SS
34
SYNC
+
–
A4
CC2
RC
CSS
3
SENSE2
Q1
FREQ
PROG
33
RT
SGND
4, 23,
24, 37
R2
VOUT
12, 13, 14,
15, 16, 17
100kHz-1MHz
OSCILLATOR
G1
FREQUENCY
FOLDBACK
A5
GND
48mV
SENSE
SENSE1
3957 F01
RT
R1
CC1
Figure 1. LT3957 Block Diagram Working as a SEPIC Converter
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LT3957
APPLICATIONS INFORMATION
Main Control Loop
The LT3957 uses a fixed frequency, current mode control
scheme to provide excellent line and load regulation. Operation can be best understood by referring to the Block
Diagram in Figure 1.
The start of each oscillator cycle sets the SR latch (SR1)
and turns on the internal power MOSFET switch M1 through
driver G2. The switch current flows through the internal
current sensing resistor RSENSE and generates a voltage
proportional to the switch current. This current sense
voltage VISENSE (amplified by A5) is added to a stabilizing
slope compensation ramp and the resulting sum (SLOPE)
is fed into the positive terminal of the PWM comparator A7.
When SLOPE exceeds the level at the negative input of A7
(VC pin), SR1 is reset, turning off the power switch. The
level at the negative input of A7 is set by the error amplifier
A1 (or A2) and is an amplified version of the difference
between the feedback voltage (FBX pin) and the reference
voltage (1.6V or –0.8V, depending on the configuration).
In this manner, the error amplifier sets the correct peak
switch current level to keep the output in regulation.
The LT3957 has a switch current limit function. The current
sense voltage is input to the current limit comparator A6.
If the SENSE2 pin voltage is higher than the sense current
limit threshold VSENSE(MAX) (48mV, typical), A6 will reset
SR1 and turn off M1 immediately.
The LT3957 is capable of generating either positive or
negative output voltage with a single FBX pin. It can be
configured as a boost, flyback or SEPIC converter to generate positive output voltage, or as an inverting converter
to generate negative output voltage. When configured as
a SEPIC converter, as shown in Figure 1, the FBX pin is
pulled up to the internal bias voltage of 1.6V by a voltage divider (R1 and R2) connected from VOUT to SGND.
Comparator A2 becomes inactive and comparator A1
performs the inverting amplification from FBX to VC.
When the LT3957 is in an inverting configuration, the
FBX pin is pulled down to –0.8V by a voltage divider
connected from VOUT to SGND. Comparator A1 becomes
inactive and comparator A2 performs the noninverting
amplification from FBX to VC.
The LT3957 has overvoltage protection functions to
protect the converter from excessive output voltage
overshoot during start-up or recovery from a short-circuit
condition. An overvoltage comparator A11 (with 20mV
hysteresis) senses when the FBX pin voltage exceeds the
positive regulated voltage (1.6V) by 8% and provides a
reset pulse. Similarly, an overvoltage comparator A12
(with 10mV hysteresis) senses when the FBX pin voltage
exceeds the negative regulated voltage (–0.8V) by 11%
and provides a reset pulse. Both reset pulses are sent to
the main RS latch (SR1) through G6 and G5. The power
MOSFET switch M1 is actively held off for the duration of
an output overvoltage condition.
Programming Turn-On and Turn-Off Thresholds with
the EN/UVLO Pin
The EN/UVLO pin controls whether the LT3957 is enabled
or is in shutdown state. A micropower 1.22V reference,
a comparator A10 and a controllable current source IS1
allow the user to accurately program the supply voltage
at which the IC turns on and off. The falling value can be
accurately set by the resistor dividers R3 and R4. When
EN/UVLO is above 0.4V, and below the 1.22V threshold,
the small pull-down current source IS1 (typical 2μA) is
active.
The purpose of this current is to allow the user to program
the rising hysteresis. The Block Diagram of the comparator
and the external resistors is shown in Figure 1. The typical
falling threshold voltage and rising threshold voltage can
be calculated by the following equations:
(R3 + R4)
R4
VVIN,RISING = 2µA • R3 + VIN,FALLING
VVIN,FALLING = 1.22 •
For applications where the EN/UVLO pin is only used as
a logic input, the EN/UVLO pin can be connected directly
to the input voltage VIN for always-on operation.
3957f
9
LT3957
APPLICATIONS INFORMATION
INTVCC Regulator Bypassing and Operation
An internal, low dropout (LDO) voltage regulator produces
the 5.2V INTVCC supply which powers the gate driver, as
shown in Figure 1. The LT3957 contains an undervoltage
lockout comparator A8 for the INTVCC supply. The INTVCC
undervoltage (UV) threshold is 2.7V (typical), with 0.1V
hysteresis, to ensure that the internal MOSFET has sufficient gate drive voltage before turning on. When INTVCC
is below the UV threshold, the internal power switch will
be turned off and the soft-start operation will be triggered.
The logic circuitry within the LT3957 is also powered from
the internal INTVCC supply.
The INTVCC regulator must be bypassed to SGND immediately adjacent to the IC pins with a minimum of 4.7μF
ceramic capacitor. Good bypassing is necessary to supply
the high transient currents required by the MOSFET gate
driver.
In an actual application, most of the IC supply current is
used to drive the gate capacitance of the internal power
MOSFET. The on-chip power dissipation can be significant
when the internal power MOSFET is being driven at a high
frequency and the VIN voltage is high.
An effective approach to reduce the power consumption of
the internal LDO for gate drive and to improve the efficiency
is to tie the INTVCC pin to an external voltage source high
enough to turn off the internal LDO regulator.
In SEPIC or flyback applications, the INTVCC pin can be
connected to the output voltage VOUT through a blocking
diode, as shown in Figure 2, if VOUT meets the following
conditions:
1. VOUT < VIN (pin voltage)
2. VOUT < 8V
A resistor RVCC can be connected, as shown in Figure 2, to
limit the inrush current from VOUT. Regardless of whether
or not the INTVCC pin is connected to an external voltage
source, it is always necessary to have the driver circuitry
bypassed with a 4.7μF low ESR ceramic capacitor to ground
immediately adjacent to the INTVCC and SGND pins.
If LT3957 operates at a low VIN and high switching frequency, the voltage drop across the drain and the source of
the LDO PMOS (M2 in Figure 1) could push INTVCC to be
below the UV threshold. To prevent this from happening,
the INTVCC pin can be shorted directly to the VIN pin. VIN
must not exceed the INTVCC Absolute Maximum Rating
(8V). In this condition, the internal LDO will be turned off
and the gate driver will be powered directly from VIN. It is
recommended that INTVCC pin be shorted to the VIN pin if
VIN is lower than 3.5V at 1MHz switching frequency, or VIN
is lower than 3.2V at 100kHz switching frequency. With
the INTVCC pin shorted to VIN, however, a small current
(around 16μA) will load the INTVCC in shutdown mode.
DVCC
INTVCC
LT3957
RVCC
VOUT
CVCC
4.7μF
SGND
3957 F02
Figure 2. Connecting INTVCC to VOUT
3957f
10
LT3957
APPLICATIONS INFORMATION
Operating Frequency and Synchronization
Duty Cycle Consideration
The choice of operating frequency may be determined by
on-chip power dissipation (a low switching frequency may
be required to ensure IC junction temperature does not
exceed 125°C), otherwise it is a trade-off between efficiency
and component size. Low frequency operation improves
efficiency by reducing gate drive current and MOSFET
and diode switching losses. However, lower frequency
operation requires a physically larger inductor. Switching
frequency also has implications for loop compensation.
The LT3957 uses a constant-frequency architecture that
can be programmed over a 100kHz to 1000kHz range
with a single external resistor from the RT pin to SGND,
as shown in Figure 1. A table for selecting the value of RT
for a given operating frequency is shown in Table 1.
Switching duty cycle is a key variable defining converter operation. As such, its limits must be considered.
Minimum on-time is the smallest time duration that the
LT3957 is capable of turning on the power MOSFET. This
time is typically about 240ns (see Minimum On-Time in
the Electrical Characteristics table). In each switching
cycle, the LT3957 keeps the power switch off for at least
220ns (typical) (see Minimum Off-Time in the Electrical
Characteristics table).
Table 1. Timing Resistor (RT ) Value
Maximum duty cycle = 1 – (minimum off-time • frequency)
SWITCHING FREQUENCY (kHz)
RT (kΩ)
100
140
200
63.4
300
41.2
400
30.9
500
24.3
600
19.6
700
16.5
800
14
900
12.1
1000
10.5
The operating frequency of the LT3957 can be synchronized
to an external clock source. By providing a digital clock
signal into the SYNC pin, the LT3957 will operate at the
SYNC clock frequency. The LT3957 detects the rising edge
of each clock cycle. If this feature is used, an RT resistor
should be chosen to program a switching frequency 20%
slower than SYNC pulse frequency. It is recommended that
the SYNC pin has a minimum pulse width of 200ns. Tie
the SYNC pin to SGND if this feature is not used.
The minimum on-time, minimum off-time and the switching
frequency define the minimum and maximum switching
duty cycles a converter is able to generate:
Minimum duty cycle = minimum on-time • frequency
Programming the Output Voltage
The output voltage VOUT is set by a resistor divider, as
shown in Figure 1. The positive and negative VOUT are set
by the following equations:
⎛ R2 ⎞
VOUT,POSITIVE = 1.6V • ⎜ 1+ ⎟
⎝ R1⎠
⎛ R2 ⎞
VOUT,NEGATIVE = –0.8V • ⎜ 1+ ⎟
⎝ R1⎠
The resistors R1 and R2 are typically chosen so that
the error caused by the current flowing into the FBX pin
during normal operation is less than 1% (this translates
to a maximum value of R1 at about 158k).
3957f
11
LT3957
APPLICATIONS INFORMATION
Soft-Start
FBX Frequency Foldback
The LT3957 contains several features to limit peak switch
currents and output voltage (VOUT) overshoot during
start-up or recovery from a fault condition. The primary
purpose of these features is to prevent damage to external
components or the load.
When VOUT is very low during start-up, or an output shortcircuit on a SEPIC, an inverting, or a flyback converter, the
switching regulator must operate at low duty cycles to keep
the power switch current below the current limit, since
the inductor current decay rate is very low during switch
off time. The minimum on-time limitation may prevent the
switcher from attaining a sufficiently low duty cycle at the
programmed switching frequency. So, the switch current
may keep increasing through each switch cycle, exceeding the programmed current limit. To prevent the switch
peak currents from exceeding the programmed value, the
LT3957 contains a frequency foldback function to reduce
the switching frequency when the FBX voltage is low (see
the Normalized Switching Frequency vs FBX graph in the
Typical Performance Characteristics section).
High peak switch currents during start-up may occur in
switching regulators. Since VOUT is far from its final value,
the feedback loop is saturated and the regulator tries to
charge the output capacitor as quickly as possible, resulting
in large peak currents. A large surge current may cause
inductor saturation or power switch failure.
The LT3957 addresses this mechanism with the SS pin.
As shown in Figure 1, the SS pin reduces the power
MOSFET current by pulling down the VC pin through
Q2. In this way the SS allows the output capacitor to
charge gradually toward its final value while limiting the
start-up peak currents. The typical start-up waveforms
are shown in the Typical Performance Characteristics
section. The inductor current IL slewing rate is limited by
the soft-start function.
Besides start-up (with EN/UVLO), soft-start can also be
triggered by the following faults:
1. INTVCC < 2.85V
2. Thermal lockout (TLO > 165°C)
Any of these three faults will cause the LT3957 to stop
switching immediately. The SS pin will be discharged by
Q3. When all faults are cleared and the SS pin has been
discharged below 0.2V, a 10μA current source IS2 starts
charging the SS pin, initiating a soft-start operation.
The soft-start interval is set by the soft-start capacitor
selection according to the equation:
TSS = CSS •
1.25V
10µA
During frequency foldback, external clock synchronization is disabled to prevent interference with frequency
reducing operation.
Loop Compensation
Loop compensation determines the stability and transient
performance. The LT3957 uses current mode control to
regulate the output which simplifies loop compensation.
The optimum values depend on the converter topology, the
component values and the operating conditions (including
the input voltage, load current, etc.). To compensate the
feedback loop of the LT3957, a series resistor-capacitor
network is usually connected from the VC pin to SGND.
Figure 1 shows the typical VC compensation network.
For most applications, the capacitor should be in the
range of 470pF to 22nF, and the resistor should be in the
range of 5k to 50k. A small capacitor is often connected
in parallel with the RC compensation network to attenuate the VC voltage ripple induced from the output voltage
ripple through the internal error amplifier. The parallel
capacitor usually ranges in value from 10pF to 100pF. A
practical approach to design the compensation network
is to start with one of the circuits in this data sheet that
is similar to your application, and tune the compensation
network to optimize the performance. Stability should
then be checked across all operating conditions, including
load current, input voltage and temperature. Application
Note 76 is a good reference on loop compensation.
3957f
12
LT3957
APPLICATIONS INFORMATION
The Internal Power Switch Current
On-Chip Power Dissipation and Thermal Lockout (TLO)
For control and protection, the LT3957 measures the
internal power MOSFET current by using a sense resistor
(RSENSE) between GND and the MOSFET source. Figure 3
shows a typical waveform of the internal switch current
(ISW).
The on-chip power dissipation of LT3957 can be estimated
using the following equation:
Due to the current limit (minimum 5A) of the internal power
switch, the LT3957 should be used in the applications
that the switch peak current ISW(PEAK) during steady state
normal operation is lower than 5A by a sufficient margin
(10% or higher is recommended).
The LT3957 switching controller incorporates 100ns
timing interval to blank the ringing on the current sense
signal across RSENSE immediately after M1 is turned on.
This ringing is caused by the parasitic inductance and
capacitance of the PCB trace, the sense resistor, the diode,
and the MOSFET. The 100ns timing interval is adequate
for most of the LT3957 applications. In the applications
that have very large and long ringing on the current sense
signal, a small RC filter can be added to filter out the excess
ringing. Figure 4 shows the RC filter on the SENSE1 and
SENSE2 pins. It is usually sufficient to choose 22Ω for
RFLT and 2.2nF to 10nF for CFLT. Keep RFLT’s resistance
low. Remember that there is 65μA (typical) flowing out of
the SENSE2 pin. Adding RFLT will affect the internal power
switch current limit threshold:
PIC ≈ I2SW • D • RDS(ON) + V2PEAK • ISW • ƒ • 200pF/A +
VIN • (1.6mA + ƒ • 10nC)
where RDS(ON) is the internal switch on-resistance which
can be obtained from the Typical Performance Characteristics section. VSW(PEAK) is the peak switch off-state voltage.
The maximum power dissipation PIC(MAX) can be obtained
by comparing PIC across all the VIN range at the maximum
output current . The highest junction temperature can be
estimated using the following equation:
TJ(MAX) ≈ TA + PIC(MAX) • 42°C/W
It is recommended to measure the IC temperature in steady
state to verify that the junction temperature limit is not
exceeded. A low switching frequency may be required to
ensure TJ(MAX) does not exceed 125°C.
If LT3957 die temperature reaches thermal lockout
threshold at 165°C (typical), the IC will initiate several
protective actions. The power switch will be turned off.
A soft-start operation will be triggered. The IC will be enabled again when the junction temperature has dropped
by 5°C (nominal).
LT3957
SENSE1
⎛ 65µA • RFLT ⎞
ISW _ILIM = ⎜ 1−
⎟ • 5A
⎝
48mV ⎠
RFLT
SENSE2
CFLT
SGND
ISW
3957 F04
$ISW
Figure 4. The RC Filter on SENSE1 Pin and SENSE2 Pin
ISW(PEAK)
t
DTS
TS
3957 F03
Figure 3. The Switch Current During a Switching Cycle
3957f
13
LT3957
APPLICATIONS INFORMATION
APPLICATION CIRCUITS
The LT3957 can be configured as different topologies. The
first topology to be analyzed will be the boost converter,
followed by the flyback, SEPIC and inverting converters.
Boost Converter: Switch Duty Cycle and Frequency
The LT3957 can be configured as a boost converter for
the applications where the converter output voltage is
higher than the input voltage. Remember that boost converters are not short-circuit protected. Under a shorted
output condition, the inductor current is limited only by
the input supply capability. For applications requiring a
step-up converter that is short-circuit protected, please
refer to the Applications Information section covering
SEPIC converters.
The conversion ratio as a function of duty cycle is
VOUT
1
=
VIN 1− D
in continuous conduction mode (CCM).
For a boost converter operating in CCM, the duty cycle
of the main switch can be calculated based on the output
voltage (VOUT) and the input voltage (VIN). The maximum
duty cycle (DMAX) occurs when the converter has the
minimum input voltage:
DMAX =
VOUT − VIN(MIN)
VOUT
Discontinuous conduction mode (DCM) provides higher
conversion ratios at a given frequency at the cost of reduced
efficiencies and higher switching currents.
Boost Converter: Maximum Output Current Capability
and Inductor Selection
For the boost topology, the maximum average inductor
current is:
I L(MAX) = IO(MAX) •
1
1− DMAX
Due to the current limit of its internal power switch, the
LT3957 should be used in a boost converter whose maximum output current (IO(MAX)) is less than the maximum
output current capability by a sufficient margin (10% or
higher is recommended):
I O(MAX) <
VIN(MIN)
VOUT
• ( 5A − 0.5 • ΔISW )
The inductor ripple current ΔISW has a direct effect on the
choice of the inductor value and the converter’s maximum
output current capability. Choosing smaller values of
ΔISW increases output current capability, but requires
large inductances and reduces the current loop gain (the
converter will approach voltage mode). Accepting larger
values of ΔISW provides fast transient response and
allows the use of low inductances, but results in higher
input current ripple and greater core losses, and reduces
output current capability.
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the inductor,
the inductor value of the boost converter can be determined
using the following equation:
L=
VIN(MIN)
ΔISW • ƒ
• DMAX
The peak inductor current is the switch current limit (5.9A
typical), and the RMS inductor current is approximately
equal to IL(MAX). The user should choose the inductors
having sufficient saturation and RMS current ratings.
Boost Converter: Output Diode Selection
To maximize efficiency, a fast switching diode with low
forward drop and low reverse leakage is desirable. The
peak reverse voltage that the diode must withstand is
equal to the regulator output voltage plus any additional
ringing across its anode-to-cathode during the on-time.
The average forward current in normal operation is equal
to the output current.
It is recommended that the peak repetitive reverse voltage
rating VRRM is higher than VOUT by a safety margin (a 10V
safety margin is usually sufficient).
3957f
14
LT3957
APPLICATIONS INFORMATION
The power dissipated by the diode is:
tON
tOFF
$VCOUT
PD = IO(MAX) • VD
where VD is diode’s forward voltage drop, and the diode
junction temperature is:
TJ = TA + PD • RθJA
The RθJA to be used in this equation normally includes the
RθJC for the device plus the thermal resistance from the board
to the ambient temperature in the enclosure. TJ must not
exceed the diode maximum junction temperature rating.
Boost Converter: Output Capacitor Selection
Contributions of ESR (equivalent series resistance), ESL
(equivalent series inductance) and the bulk capacitance
must be considered when choosing the correct output
capacitors for a given output ripple voltage. The effect of
these three parameters (ESR, ESL and bulk C) on the output
voltage ripple waveform for a typical boost converter is
illustrated in Figure 5.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step ΔVESR and the charging/discharging ΔVCOUT. For the purpose of simplicity, we will choose
2% for the maximum output ripple, to be divided equally
between ΔVESR and ΔVCOUT. This percentage ripple will
change, depending on the requirements of the application,
and the following equations can easily be modified. For a
1% contribution to the total ripple voltage, the ESR of the
output capacitor can be determined using the following
equation:
ESRCOUT ≤
0.01• VOUT
ID(PEAK)
For the bulk C component, which also contributes 1% to
the total ripple:
COUT ≥
VOUT
(AC)
$VESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
3957 F05
Figure 5. The Output Ripple Waveform of a Boost Converter
The output capacitor in a boost regulator experiences high
RMS ripple currents, as shown in Figure 5. The RMS ripple
current rating of the output capacitor can be determined
using the following equation:
IRMS(COUT) ≥IO(MAX) •
DMAX
1− DMAX
Multiple capacitors are often paralleled to meet ESR requirements. Typically, once the ESR requirement is satisfied, the
capacitance is adequate for filtering and has the required
RMS current rating. Additional ceramic capacitors in parallel are commonly used to reduce the effect of parasitic
inductance in the output capacitor, which reduces high
frequency switching noise on the converter output.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical
than the output capacitor, due to the fact that the inductor
is in series with the input, and the input current waveform is continuous. The input voltage source impedance
determines the size of the input capacitor, which is typically in the range of 1μF to 100μF. A low ESR capacitor
is recommended, although it is not as critical as for the
output capacitor.
The RMS input capacitor ripple current for a boost converter is:
IRMS(CIN) = 0.3 • ΔIL
IO(MAX)
0.01• VOUT • ƒ
3957f
15
LT3957
APPLICATIONS INFORMATION
FLYBACK CONVERTER APPLICATIONS
The LT3957 can be configured as a flyback converter for the
applications where the converters have multiple outputs,
high output voltages or isolated outputs. Due to the 40V
rating of the internal power switch, LT3797 should be used
in low input voltage flyback converters. Figure 6 shows a
simplified flyback converter.
The flyback converter has a very low parts count for multiple outputs, and with prudent selection of turns ratio, can
have high output/input voltage conversion ratios with a
desirable duty cycle. However, it has low efficiency due to
the high peak currents, high peak voltages and consequent
power loss. The flyback converter is commonly used for
an output power of less than 50W.
The flyback converter can be designed to operate either
in continuous or discontinuous mode. Compared to continuous mode, discontinuous mode has the advantage of
smaller transformer inductances and easy loop compensation, and the disadvantage of higher peak-to-average
current and lower efficiency.
SUGGESTED
RCD SNUBBER
VIN
–
+
CIN
VSN
+
CSN
RSN
ID
LP
LS
+
+
VOUT
COUT
–
DSN
ISW
ISW(MAX)
ID
ID(MAX)
DTS
D2TS
TS
t
D3TS
3957 F07
The flyback converter conversion ratio in the discontinuous mode operation is:
VOUT NS D
=
•
VIN NP D2
ISW
SW
According to Figure 6, the peak SW voltage is:
LT3957
GND
VSW
Figure 7. Waveforms of the Flyback Converter
in Discontinuous Mode Operation
D
NP:NS
Figure 7 shows the waveforms of the flyback converter
in discontinuous mode operation. During each switching
period TS, three subintervals occur: DTS, D2TS, D3TS.
During DTS, M is on, and D is reverse-biased. During
D2TS, M is off, and LS is conducting current. Both LP and
LS currents are zero during D3TS.
3957 F06
Figure 6. A Simplified Flyback Converter
Flyback Converter: Switch Duty Cycle and Turns Ratio
The flyback converter conversion ratio in the continuous
mode operation is:
VOUT NS D
=
•
VIN NP 1− D
VSW(PEAK) = VIN(MAX) + VSN
where VSN is the snubber capacitor voltage. A smaller VSN
results in a larger snubber loss. A reasonable VSN is 1.5
to 2 times of the reflected output voltage:
VSN = k •
VOUT • NP
NS
k = 1.5 ~ 2
where NS/NP is the second to primary turns ratio. D is
duty cycle.
3957f
16
LT3957
APPLICATIONS INFORMATION
According to the Absolute Maximum Ratings table, the SW
voltage Absolute Maximum value is 40V. Therefore, the
maximum primary to secondary turns ratio (for both the
continuous and the discontinuous operation) should be.
NP 40V − VIN(MAX)
≤
NS
k • VOUT
According to the preceding equations, the user has relative
freedom in selecting the switch duty cycle or turns ratio to
suit a given application. The selections of the duty cycle
and the turns ratio are somewhat iterative processes, due
to the number of variables involved. The user can choose
either a duty cycle or a turns ratio as the start point. The
following trade-offs should be considered when selecting the switch duty cycle or turns ratio, to optimize the
converter performance. A higher duty cycle affects the
flyback converter in the following aspects:
• Lower MOSFET RMS current ISW(RMS), but higher
MOSFET VSW peak voltage
• Lower diode peak reverse voltage, but higher diode
RMS current ID(RMS)
• Higher transformer turns ratio (NP/NS)
It is recommended to choose a duty cycle between 20%
and 80%.
output current capability by a sufficient margin (10% or
higher is recommended):
IO(MAX) <
VIN(MIN)
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the primary
winding, the primary winding inductance can be calculated
using the following equation:
L=
VIN(MIN)
ΔISW • ƒ
The maximum output current capability and transformer
design for continuous conduction mode (CCM) is chosen
as presented here.
ILS(RMS) ≈
DMAX =
S
⎛N ⎞
VOUT • ⎜ P ⎟ + VIN(MIN)
⎝ NS ⎠
Due to the current limit of its internal power switch, the
LT3957 should be used in a flyback converter whose maximum output current (IO(MAX)) is less than the maximum
• DMAX
The primary winding peak current is the switch current
limit (typical 5.9A). The primary and secondary maximum
RMS currents are:
ILP(RMS) ≈
⎛N ⎞
VOUT • ⎜ P ⎟
⎝N ⎠
• DMAX • ( 5A − 0.5 • ΔISW )
The transformer ripple current ΔISW has a direct effect on
the design/choice of the transformer and the converter’s
output current capability. Choosing smaller values of
ΔISW increases the output current capability, but requires
large primary and secondary inductances and reduce the
current loop gain (the converter will approach voltage
mode). Accepting larger values of ΔISW allows the use
of low primary and secondary inductances, but results
in higher input current ripple, greater core losses, and
reduces the output current capability.
Flyback Converter: Maximum Output Current
Capability and Transformer Design
The maximum duty cycle (DMAX) occurs when the converter
has the minimum VIN:
VOUT
POUT(MAX)
DMAX • VIN(MIN) • η
I OUT(MAX)
1− DMAX
where η is the converter efficiency.
Based on the preceding equations, the user should design/choose the transformer having sufficient saturation
and RMS current ratings.
Flyback Converter: Snubber Design
Transformer leakage inductance (on either the primary or
secondary) causes a voltage spike to occur after the MOSFET turn-off. This is increasingly prominent at higher load
currents, where more stored energy must be dissipated.
3957f
17
LT3957
APPLICATIONS INFORMATION
In some cases a snubber circuit will be required to avoid
overvoltage breakdown at the MOSFET’s drain node. There
are different snubber circuits (such as RC snubber, RCD
snubber, Zener clamp, etc.), and Application Note 19 is a
good reference on snubber design. An RC snubber circuit
can be connected between SW and GND to damp the
ringing on SW pins. The snubber resistor values should
be close to the impedance of the parasitic resonance. The
snubber capacitor value should be larger than the circuit
parasitic capacitance, but be small enough to keep the
snubber resistor power dissipation low.
Approximate the required peak repetitive reverse voltage
rating VRRM using:
If the RC snubber is insufficient to prevent SW pins overvoltage, the RCD snubber can be used to limit the peak
voltage on the SW pins, which is shown in Figure 6.
The RθJA to be used in this equation normally includes the
RθJC for the device, plus the thermal resistance from the board
to the ambient temperature in the enclosure. TJ must not
exceed the diode maximum junction temperature rating.
The snubber resistor value (RSN) can be calculated by the
following equation:
RSN = 2 •
V 2 SN − VSN • VOUT •
NP
NS
I2 SW(PEAK) • L LK • ƒ
LLK is the leakage inductance of the primary winding,
which is usually specified in the transformer characteristics. LLK can be obtained by measuring the primary
inductance with the secondary windings shorted. The
snubber capacitor value (CSN) can be determined using
the following equation:
VSN
CCN =
ΔVSN • RSN • ƒ
where ΔVSN is the voltage ripple across CSN. A reasonable
ΔVSN is 5% to 10% of VSN. The reverse voltage rating of
DSN should be higher than the sum of VSN and VIN(MAX).
A Zener clamp can also be connected between SW and
GND to ensure SW voltage does not exceed 40V.
Flyback Converter: Output Diode Selection
The output diode in a flyback converter is subject to large
RMS current and peak reverse voltage stresses. A fast
switching diode with a low forward drop and a low reverse
leakage is desired. Schottky diodes are recommended if
the output voltage is below 100V.
VRRM >
NS
•V
+V
NP IN(MAX) OUT
The power dissipated by the diode is:
PD = IO(MAX) • VD
and the diode junction temperature is:
TJ = TA + PD • RθJA
Flyback Converter: Output Capacitor Selection
The output capacitor of the flyback converter has a similar
operation condition as that of the boost converter. Refer to
the Boost Converter: Output Capacitor Selection section
for the calculation of COUT and ESRCOUT.
The RMS ripple current rating of the output capacitors
in continuous operation can be determined using the
following equation:
DMAX
1− DMAX
IRMS(COUT),CONTINUOUS ≈ IO(MAX) •
Flyback Converter: Input Capacitor Selection
The input capacitor in a flyback converter is subject to
a large RMS current due to the discontinuous primary
current. To prevent large voltage transients, use a low
ESR input capacitor sized for the maximum RMS current.
The RMS ripple current rating of the input capacitors in
continuous operation can be determined using the following equation:
IRMS(CIN),CONTINUOUS ≈
POUT(MAX)
VIN(MIN) • η
•
1− DMAX
DMAX
3957f
18
LT3957
APPLICATIONS INFORMATION
SEPIC CONVERTER APPLICATIONS
The LT3957 can be configured as a SEPIC (single-ended
primary inductance converter), as shown in Figure 1. This
topology allows for the input to be higher, equal, or lower
than the desired output voltage. The conversion ratio as
a function of duty cycle is:
VOUT + VD
D
=
VIN
1− D
in continuous conduction mode (CCM).
In a SEPIC converter, no DC path exists between the input
and output. This is an advantage over the boost converter
for applications requiring the output to be disconnected
from the input source when the circuit is in shutdown.
Compared to the flyback converter, the SEPIC converter
has the advantage that both the power MOSFET and the
output diode voltages are clamped by the capacitors (CIN,
CDC and COUT), therefore, there is less voltage ringing
across the power MOSFET and the output diodes. The
SEPIC converter requires much smaller input capacitors
than those of the flyback converter. This is due to the fact
that, in the SEPIC converter, the current through inductor
L1 (which is series with the input) is continuous.
SEPIC Converter: Switch Duty Cycle and Frequency
For a SEPIC converter operating in CCM, the duty cycle
of the main switch can be calculated based on the output
voltage (VOUT), the input voltage (VIN) and the diode
forward voltage (VD).
The maximum duty cycle (DMAX) occurs when the converter
has the minimum input voltage:
DMAX =
VOUT + VD
VIN(MIN) + VOUT + VD
SEPIC Converter: The Maximum Output Current
Capability and Inductor Selection
As shown in Figure 1, the SEPIC converter contains two
inductors: L1 and L2. L1 and L2 can be independent, but
can also be wound on the same core, since identical voltages are applied to L1 and L2 throughout the switching
cycle.
For the SEPIC topology, the current through L1 is the
converter input current. Based on the fact that, ideally, the
output power is equal to the input power, the maximum
average inductor currents of L1 and L2 are:
IL1(MAX) =IIN(MAX) =IO(MAX) •
DMAX
1− DMAX
IL2(MAX) =IO(MAX)
Due to the current limit of its internal power switch,
the LT3957 should be used in a SEPIC converter whose
maximum output current (IO(MAX)) is less than the output
current capability by a sufficient margin (10% or higher
is recommended):
IO(MAX) < (1− DMAX ) • ( 5A − 0.5 • ΔISW )
The inductor ripple currents ΔIL1 and ΔIL2 are identical:
ΔIL1 = ΔIL2 = 0.5 • ΔISW
The inductor ripple current ΔISW has a direct effect on the
choice of the inductor value and the converter’s maximum
output current capability. Choosing smaller values of ΔISW
requires large inductances and reduces the current loop
gain (the converter will approach voltage mode). Accepting
larger values of ΔISW allows the use of low inductances,
but results in higher input current ripple and greater core
losses and reduces output current capability.
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the inductor, the inductor value (L1 and L2 are independent) of the
SEPIC converter can be determined using the following
equation:
L1 = L2 =
VIN(MIN)
0.5 • ΔISW • ƒ
• DMAX
For most SEPIC applications, the equal inductor values
will fall in the range of 1μH to 100μH.
3957f
19
LT3957
APPLICATIONS INFORMATION
By making L1 = L2, and winding them on the same core, the
value of inductance in the preceding equation is replaced
by 2L, due to mutual inductance:
L=
VIN(MIN)
ΔISW • ƒ
• DMAX
This maintains the same ripple current and energy storage
in the inductors. The peak inductor currents are:
IL1(PEAK) = IL1(MAX) + 0.5 • ΔIL1
IL2(PEAK) = IL2(MAX) + 0.5 • ΔIL2
The maximum RMS inductor currents are approximately
equal to the maximum average inductor currents.
Based on the preceding equations, the user should choose
the inductors having sufficient saturation and RMS current ratings.
SEPIC Converter: Output and Input Capacitor Selection
The selections of the output and input capacitors of the
SEPIC converter are similar to those of the boost converter.
Please refer to the Boost Converter: Output Capacitor
Selection and Boost Converter: Input Capacitor Selection
sections.
SEPIC Converter: Selecting the DC Coupling Capacitor
The DC voltage rating of the DC coupling capacitor (CDC,
as shown in Figure 1) should be larger than the maximum
input voltage:
VCDC > VIN(MAX)
CDC has nearly a rectangular current waveform. During
the switch off-time, the current through CDC is IIN, while
approximately –IO flows during the on-time. The RMS
rating of the coupling capacitor is determined by the following equation:
SEPIC Converter: Output Diode Selection
To maximize efficiency, a fast switching diode with a low
forward drop and low reverse leakage is desirable. The
average forward current in normal operation is equal to
the output current.
It is recommended that the peak repetitive reverse voltage
rating VRRM is higher than VOUT + VIN(MAX) by a safety
margin (a 10V safety margin is usually sufficient).
The power dissipated by the diode is:
PD = IO(MAX) • VD
where VD is diode’s forward voltage drop, and the diode
junction temperature is:
TJ = TA + PD • RθJA
The RθJA used in this equation normally includes the RθJC
for the device, plus the thermal resistance from the board,
to the ambient temperature in the enclosure. TJ must not
exceed the diode maximum junction temperature rating.
VOUT + VD
VIN(MIN)
IRMS(CDC) >IO(MAX) •
A low ESR and ESL, X5R or X7R ceramic capacitor works
well for CDC.
INVERTING CONVERTER APPLICATIONS
The LT3957 can be configured as a dual-inductor inverting
topology, as shown in Figure 8. The VOUT to VIN ratio is:
VOUT − VD
D
=−
VIN
1− D
in continuous conduction mode (CCM).
CDC
L1
VIN
+
L2
–
+
–
CIN
COUT
SW
LT3957
GND
VOUT
+
D1
+
3757 F10
Figure 8. A Simplified Inverting Converter
3957f
20
LT3957
APPLICATIONS INFORMATION
Inverting Converter: Switch Duty Cycle and Frequency
Inverting Converter: Selecting the DC Coupling Capacitor
For an inverting converter operating in CCM, the duty cycle
of the main switch can be calculated based on the negative
output voltage (VOUT) and the input voltage (VIN).
The DC voltage rating of the DC coupling capacitor
(CDC, as shown in Figure 10) should be larger than the
maximum input voltage minus the output voltage (negative voltage):
The maximum duty cycle (DMAX) occurs when the converter
has the minimum input voltage:
DMAX =
VOUT − VD
VOUT − VD − VIN(MIN)
Inverting Converter: Output Diode and Input Capacitor
Selections
The selections of the inductor, output diode and input
capacitor of an inverting converter are similar to those
of the SEPIC converter. Please refer to the corresponding
SEPIC converter sections.
Inverting Converter: Output Capacitor Selection
The inverting converter requires much smaller output
capacitors than those of the boost, flyback and SEPIC
converters for similar output ripples. This is due to the fact
that, in the inverting converter, the inductor L2 is in series
with the output, and the ripple current flowing through the
output capacitors are continuous. The output ripple voltage
is produced by the ripple current of L2 flowing through the
ESR and bulk capacitance of the output capacitor:
⎛
⎞
1
ΔVOUT(P – P) = ΔIL2 • ⎜ ESRCOUT +
8 • ƒ • COUT ⎟⎠
⎝
After specifying the maximum output ripple, the user can
select the output capacitors according to the preceding
equation.
The ESR can be minimized by using high quality X5R or
X7R dielectric ceramic capacitors. In many applications,
ceramic capacitors are sufficient to limit the output voltage ripple.
The RMS ripple current rating of the output capacitor
needs to be greater than:
IRMS(COUT) > 0.3 • ΔIL2
VCDC > VIN(MAX) – VOUT
CDC has nearly a rectangular current waveform. During
the switch off-time, the current through CDC is IIN, while
approximately –IO flows during the on-time. The RMS
rating of the coupling capacitor is determined by the following equation:
IRMS(CDC) >IO(MAX) •
DMAX
1− DMAX
A low ESR and ESL, X5R or X7R ceramic capacitor works
well for CDC.
Board Layout
The high power and high speed operation of the LT3957
demands careful attention to board layout and component
placement. Careful attention must be paid to the internal
power dissipation of the LT3957 at high input voltages,
high switching frequencies, and high internal power switch
currents to ensure that a junction temperature of 125°C is
not exceeded. This is especially important when operating
at high ambient temperatures. Exposed pads on the bottom of the package are SGND and SW terminals of the IC,
and must be soldered to a SGND ground plane and a SW
plane respectively. It is recommended that multiple vias
in the printed circuit board be used to conduct heat away
from the IC and into the copper planes with as much as
area as possible.
To prevent radiation and high frequency resonance
problems, proper layout of the components connected
to the IC is essential, especially the power paths with
higher di/dt. The following high di/dt loops of different
topologies should be kept as tight as possible to reduce
inductive ringing:
• In boost configuration, the high di/dt loop contains the
output capacitor, the internal power MOSFET and the
Schottky diode.
3957f
21
LT3957
APPLICATIONS INFORMATION
• In flyback configuration, the high di/dt primary loop
contains the input capacitor, the primary winding, the
internal power MOSFET. The high di/dt secondary loop
contains the output capacitor, the secondary winding
and the output diode.
• In SEPIC configuration, the high di/dt loop contains
the internal power MOSFET, output capacitor, Schottky
diode and the coupling capacitor.
• In inverting configuration, the high di/dt loop contains
internal power MOSFET, Schottky diode and the coupling
capacitor.
Check the stress on the internal power MOSFET by measuring the SW-to-GND voltage directly across the IC terminals.
Make sure the inductive ringing does not exceed the
maximum rating of the internal power MOSFET (40V).
The small-signal components should be placed away from
high frequency switching nodes. For optimum load regulation and true remote sensing, the top of the output voltage
sensing resistor divider should connect independently to
the top of the output capacitor (Kelvin connection), staying
away from any high dV/dt traces. Place the divider resistors near the LT3957 in order to keep the high impedance
FBX node short.
Figure 9 shows the suggested layout of the 4.5V to16V
input, 24V output boost converter in the Typical Application section.
R1
VIA TO VOUT
R2
CSS
RT
RC
CC
36 35 34 33 32 31 30
1
28
2
27
3
R3
37
4
25
24
6
CVCC
LT3957
R4
23
38
8
21
9
20
10
12 13 14 15 16 17
L1
COUT
COUT
D1
CIN
GND
VOUT
VIA TO VOUT
VIN
3958 F09
VIAS TO SGND GROUND PLANE
VIAS TO SW PLANE
Figure 9. Suggested Layout of the 4.5V to 16V Input. 24V Output Boost Converter in the Typical Application Section
3957f
22
LT3957
APPLICATIONS INFORMATION
Recommended Component Manufacturers
Some of the recommended component manufacturers
are listed in Table 2.
Table 2. Recommended Component Manufacturers
VENDOR
COMPONENTS
WEB ADDRESS
Capacitors
avx.com
Inductors,
Transformers
bhelectronics.com
Coilcraft
Inductors
coilcraft.com
Cooper Bussmann
AVX
BH Electronics
Inductors
bussmann.com
Diodes, Inc
Diodes
diodes.com
General Semiconductor
Diodes
generalsemiconductor.
com
International Rectifier
Diodes
irf.com
Kemet
Tantalum Capacitors
kemet.com
Toroid Cores
mag-inc.com
Microsemi
Diodes
microsemi.com
Murata-Erie
Inductors, Capacitors
murata.co.jp
Capacitors
nichicon.com
Magnetics Inc
Nichicon
On Semiconductor
Diodes
onsemi.com
Panasonic
Capacitors
panasonic.com
Pulse
Inductors
pulseeng.com
Sanyo
Capacitors
sanyo.co.jp
Sumida
Inductors
sumida.com
Taiyo Yuden
Capacitors
t-yuden.com
Capacitors, Inductors
component.tdk.com
Thermalloy
Heat Sinks
aavidthermalloy.com
Tokin
Capacitors
nec-tokinamerica.com
Toko
Inductors
tokoam.com
United Chemi-Con
Capacitors
chemi-com.com
TDK
Vishay
Inductors
vishay.com
Würth Elektronik
Inductors
we-online.com
Capacitors
vishay.com
Small-Signal Discretes
zetex.com
Vishay/Sprague
Zetex
3957f
23
LT3957
TYPICAL APPLICATIONS
4.5V to 16V Input, 24V Output Boost Converter
VIN
4.5V TO 16V
L1
10μH
CIN
10μF
25V
X5R
R3
200k
VIN
D1
GND
EN/UVLO
R4
95.3k
COUT
10μF
50V
X5R
s2
SW
VOUT
24V
600mA
LT3957
SGND
SENSE1
SYNC
SENSE2
R2
226k
FBX
RT
RT
41.2k
300kHz
SS
VC
CSS
0.33μF
R1
15.8k
INTVCC
RC
6.8k
CC
22nF
CVCC
4.7μF
10V
X5R
3957 TA02a
CIN: MURATA GRM31ER61H106KA12
COUT: TAIYO YUDEN UMK325BJ106MM
D1: VISHAY SILICONIX 10BQ040
L1: VISHAY SILICONIX IHLP-5050CE-1
Efficiency vs Output Current
100
VIN = 12V
EFFICIENCY (%)
95
90
85
80
75
70
0
100
600
400
500
200
300
OUTPUT CURRENT (mA)
3957 TA02b
3957f
24
LT3957
TYPICAL APPLICATIONS
5V to 16V Input, 12V Output SEPIC Converter
CDC
4.7μF, 25V
X5R
D1
VOUT
12V
COUT 1A
22μF
16V
X5R
s2
•
L1A
VIN
5V TO 16V
CIN
4.7μF
25V
X5R
VIN
200k
L1B
SW
GND
EN/UVLO
82.5k
•
LT3957
SGND
SENSE1
SYNC
SENSE2
105k
FBX
RT
41.2k
300kHz
SS
INTVCC
VC
0.47μF
10k
10nF
15.8k
CVCC
4.7μF
10V
X5R
3957 TA03a
CIN, CDC: MURATA GRM21BR61E475KA12L
COUT: MURATA GRM32ER61C226KE20
D1: VISHAY SILICONIX 30BQ040
L1A, L1B: COILTRONICS DRQ127-100
Efficiency vs Output Current
100
Load Step Waveforms
VIN = 12V
VIN = 12V
95
EFFICIENCY (%)
90
VOUT
0.5V/DIV
(AC)
85
80
75
IOUT 0.8A
0.5A/DIV
0.2A
70
65
60
500μs/DIV
55
3957 TA03c
50
0
800
200
400
600
OUTPUT CURRENT (mA)
1000
3957 TA03b
Frequency Foldback Waveforms
When Output Short-Circuit
Start-Up Waveforms
VIN = 12V
VIN = 12V
VOUT
10V/DIV
VOUT
5V/DIV
VSW
20V/DIV
IL1A + IL1B
5A/DIV
IL1A + IL1B
2A/DIV
5ms/DIV
3957 TA03d
50μs/DIV
3957 TA03e
3957f
25
LT3957
TYPICAL APPLICATIONS
5V to 16V Input, –12V Output Inverting Converter
CDC
4.7μF, 50V
X7R
•
L1B
CIN
4.7μF
25V
X5R
VIN
200k
D1
SW
GND
EN/UVLO
82.5k
VOUT
–12V
COUT 1A
22μF
16V
X5R
s2
•
L1A
VIN
5V TO 16V
LT3957
SGND
SENSE1
SYNC
SENSE2
105k
FBX
RT
41.2k
300kHz
SS
INTVCC
VC
0.47μF
7.5k
CVCC
4.7μF
10V
X5R
10k
10nF
3957 TA04a
CIN: MURATA GRM21BR61E475KA12L
CDC: TAIYO YUDEN UMK316BJ475KL
COUT: MURATA GRM32ER61C226KE20
D1: VISHAY SILICONIX 30BQ040
L1A, L1B: COILTRONICS DRQ127-100
Efficiency vs Output Current
100
Load Step Waveforms
VIN = 12V
VIN = 12V
95
EFFICIENCY (%)
90
85
VOUT
1V/DIV
(AC)
80
75
IOUT 0.8A
0.5A/DIV
0.2A
70
65
60
500μs/DIV
55
3957 TA04c
50
0
800
200
400
600
OUTPUT CURRENT (mA)
1000
3957 TA04b
Frequency Foldback Waveforms
When Output Short-Circuit
Start-Up Waveforms
VIN = 12V
VOUT
5V/DIV
VOUT
10V/DIV
VIN = 12V
VSW
20V/DIV
IL1A + IL1B
2A/DIV
IL1A + IL1B
5A/DIV
5ms/DIV
3957 TA04d
50μs/DIV
3957 TA04e
3957f
26
LT3957
PACKAGE DESCRIPTION
UHE Package
Variation: UHE28MA
36-Lead Plastic QFN (5mm × 6mm)
(Reference LTC DWG # 05-08-1836 Rev B)
28
27
25
24
23
21
20
0.70 p0.05
30
31
5.50 p 0.05
4.10 p 0.05
1.50 REF
16
3.00 p 0.05
3.00 p 0.05
32
33
17
1.53
p 0.05
1.88
p 0.05
0.12
p 0.05
15
14
PACKAGE OUTLINE
13
0.48 p 0.05
34
12
35
36
1
2
3
4
6
0.50 BSC
8
9
0.25 p0.05
10
2.00 REF
5.10 p 0.05
6.50 p 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
5.00 p 0.10
0.75 p 0.05
R = 0.10
TYP
PIN 1
TOP MARK
(NOTE 6)
30
31
32
1.50 REF
33 34 35
28
27
2.00 REF
25
24
6.00 p 0.10
1
1.88 p 0.10
3.00 p 0.10
0.12
p 0.10
20
2
3
4
6
23
21
36
PIN 1 NOTCH
R = 0.30 OR
0.35 s 45o
CHAMFER
1.53 p 0.10
0.48 p 0.10
3.00 p 0.10
8 R = 0.125
TYP
9
10
0.40 p 0.10
0.200 REF
0.00 – 0.05
17 16 15
0.25 p 0.05
0.50 BSC
14 13 12
(UHE28MA) QFN 0409 REV B
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
3957f
27
LT3957
TYPICAL APPLICATIONS
4V to 6V Input, 180V Output Flyback Converter
DANGER! HIGH VOLTAGE!
T1
1:10
VIN
4V TO 6V
D1
VOUT
180V
15mA
•
CIN
100μF
6.3V
s2
•
COUT
68nF
s2
D2
220pF
22Ω
75k
SW
GND
EN/UVLO
FBX
VIN
37.4k
LT3957
1.80M
SENSE1
22Ω
SGND
15.8k
SENSE2
SYNC
10nF
VC INTVCC
RT SS
140k
100kHz
0.1μF
10k
100pF
4.7μF
10V
X5R
10nF
3957 TA05
T1: TDK DCT15EFD-U44S003
CIN: GRM31CR60J107ME39L
COUT: GRM43QR72J683KW01L
D1: VISHAY SILICONIX GSD2004S DUAL DIODE CONNECTED IN SERIES
D2: DIODES MMSZ5258B
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3958
High Input Voltage, Boost, Flyback, SEPIC and
Inverting Converter with 3.5A/80V Switch
5V ≤ VIN < 80V, Current Mode Control, 100kHz to 1MHz Programmable
Operation Frequency, 5mm × 6mm QFN-36 Package
LT3580
Boost/Inverting DC/DC Converter with 2A Switch, Soft- 2.5V ≤ VIN ≤ 32V, Current Mode Control, 200kHz to 2.5MHz, 3mm × 3mm
Start and Synchronization
DFN-8, MSOP-8E
LT3573
Isolated Flyback Converter with 1.25A/60V Integrated
Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or "Third Winding" Required, Up to 7W,
MSOP-16E
LT3574
Isolated Flyback Converter with 0.65A/60V Integrated
Switch
3V ≤ VIN ≤ 40V, No Opto-Isolator or “Third Winding” Required, Up to 3W,
MSOP-16
LT3757
Boost, Flyback, SEPIC and Inverting Controller
2.9V ≤ VIN ≤ 40V, Current Mode Control, 100kHz to 1MHz Programmable
Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Package
LT3758
Boost, Flyback, SEPIC and Inverting Controller
5.5V ≤ VIN ≤ 100V, Current Mode Control, 100kHz to 1MHz Programmable
Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Package
LTC1871/LTC1871-1/ Wide Input Range, No RSENSE™ Low Quiescent Current Adjustable Switching Frequency, 2.5V ≤ VIN ≤ 36V, Burst Mode Operation
LTC1871-7
Flyback, Boost and SEPIC Controller
at Light Load
LTC3803/LTC3803-3/ 200kHz/300kHz Flyback DC/DC Controller
LTC3803-5
VIN and VOUT Limited Only by External Components, ThinSOT™ Package
3957f
28 Linear Technology Corporation
LT 0610 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
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