LT3973 - 42V, 750mA Step-Down Regulator with 2.5μA Quiescent Current and Integrated Diodes

LT3973/LT3973-3.3/LT3973-5
42V, 750mA Step-Down
Regulator with 2.5µA
Quiescent Current and
Integrated Diodes
FEATURES
DESCRIPTION
Ultralow Quiescent Current
nn 2.5µA I at 12V to 3.3V
Q
IN
OUT
nn Low Ripple Burst Mode® Operation
nn Output Ripple < 10mV
P-P
nn Wide Input Voltage Range: 4.2V to 42V Operating
nn Adjustable Switching Frequency: 200kHz to 2.2MHz
nn Integrated Boost and Catch Diodes
nn 750mA Output Current
nn Excellent Start-Up and Dropout Performance
nn Fixed Output Voltages: 3.3V, 5V
nn 1.9µA I at 12V
Q
IN
nn Accurate Programmable Undervoltage Lockout
nn Low Shutdown Current: I = 0.75µA
Q
nn Internal Catch Diode Current Limit
nn Power Good Flag
nn Thermal Shutdown
nn Small, Thermally Enhanced 10-Lead MSOP and
(3mm × 3mm) DFN Packages
The LT®3973 is an adjustable frequency monolithic buck
switching regulator that accepts a wide input voltage range
up to 42V, and consumes only 2.5µA of quiescent current.
A high efficiency switch is included on the die along with
the catch diode, boost diode, and the necessary oscillator,
control and logic circuitry. Low ripple Burst Mode operation
maintains high efficiency at low output currents while keeping the output ripple below 10mV in a typical application.
A minimum dropout voltage of 530mV is maintained when
the input voltage drops below the programmed output voltage, such as during automotive cold crank. Current mode
topology is used for fast transient response and good loop
stability. A catch diode current limit provides protection
against shorted outputs and overvoltage conditions, with
thermal shutdown providing additional fault protection. An
accurate programmable undervoltage lockout feature is
available, producing a low shutdown current of 0.75µA.
A power good flag signals when VOUT reaches 90% of
the programmed output voltage. The LT3973 is available
in small, thermally enhanced 10-lead MSOP and 3mm ×
3mm DFN packages.
nn
APPLICATIONS
Automotive Battery Regulation
Power for Portable Products
nn Industrial Supplies
nn Gate Drive Bias
nn
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
nn
TYPICAL APPLICATION
Efficiency
90
C3
0.47µF
VIN
BOOST
LT3973
OFF ON
C1
4.7µF
EN/UVLO
PG
RT
215k
f = 600kHz
GND
L1
15µH
VOUT
5V
750mA
SW
BD
OUT
15pF
1M
FB
C2
22µF
1000
VIN = 12V
80
100
70
10
60
1
50
0.1
POWER LOSS (mW)
VIN
5.6V TO 42V
EFFICIENCY (%)
5V Step-Down Converter
316k
3973 TA01a
40
0.01
0.1
1
100
10
LOAD CURRENT (mA)
3973 TA01b
For more information www.linear.com/LT3973
0.01
3973fb
1
LT3973/LT3973-3.3/LT3973-5
ABSOLUTE MAXIMUM RATINGS (Note 1)
VIN, EN/UVLO Voltage................................................42V
BOOST Pin Voltage....................................................55V
BOOST Pin Above SW Pin..........................................25V
FB/VOUT, RT, PG Voltage..............................................6V
BD Voltage.................................................................25V
OUT Voltage...............................................................14V
Operating Junction Temperature Range (Note 2)
LT3973E/LT3973E-X........................... –40°C to 125°C
LT3973I/LT3973I-X............................. –40°C to 125°C
LT3973H/LT3973H-X.......................... –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSE Only........................................................... 300°C
PIN CONFIGURATION
TOP VIEW
*FB/VOUT
1
OUT
2
EN/UVLO
3
VIN
4
GND
5
TOP VIEW
10 RT
11
GND
1
2
3
4
5
*FB/VOUT
OUT
EN/UVLO
VIN
GND
9 PG
8 BD
7 BOOST
6 SW
11
GND
10
9
8
7
6
RT
PG
BD
BOOST
SW
MSE PACKAGE
10-LEAD PLASTIC MSOP
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
θJA = 40°C/W, θJC = 5°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
* FB for LT3973, VOUT for LT3973-3.3, LT3973-5.
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3973EDD#PBF
LT3973EDD#TRPBF
LGCH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973IDD#PBF
LT3973IDD#TRPBF
LGCH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973HDD#PBF
LT3973HDD#TRPBF
LGCH
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 150°C
LT3973EMSE#PBF
LT3973EMSE#TRPBF
LTFYS
10-Lead Plastic MSOP
–40°C to 125°C
LT3973IMSE#PBF
LT3973IMSE#TRPBF
LTFYS
10-Lead Plastic MSOP
–40°C to 125°C
LT3973HMSE#PBF
LT3973HMSE#TRPBF
LTFYS
10-Lead Plastic MSOP
–40°C to 150°C
LT3973EMSE-3.3#PBF
LT3973EMSE-3.3#TRPBF LTGGB
10-Lead Plastic MSOP
–40°C to 125°C
LT3973IMSE-3.3#PBF
LT3973IMSE-3.3#TRPBF
10-Lead Plastic MSOP
–40°C to 125°C
LT3973HMSE-3.3#PBF
LT3973HMSE-3.3#TRPBF LTGGB
10-Lead Plastic MSOP
–40°C to 150°C
LT3973EDD-3.3#PBF
LT3973EDD-3.3#TRPBF
LGGC
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973IDD-3.3#PBF
LT3973IDD-3.3#TRPBF
LGGC
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973HDD-3.3#PBF
LT3973HDD-3.3#TRPBF
LGGC
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 150°C
LT3973EMSE-5#PBF
LT3973EMSE-5#TRPBF
LTGGD
10-Lead Plastic MSOP
–40°C to 125°C
LT3973IMSE-5#PBF
LT3973IMSE-5#TRPBF
LTGGD
10-Lead Plastic MSOP
–40°C to 125°C
LT3973HMSE-5#PBF
LT3973HMSE-5#TRPBF
LTGGD
10-Lead Plastic MSOP
–40°C to 150°C
2
LTGGB
3973fb
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LT3973/LT3973-3.3/LT3973-5
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3973EDD-5#PBF
LT3973EDD-5#TRPBF
LGGF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973IDD-5#PBF
LT3973IDD-5#TRPBF
LGGF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3973HDD-5#PBF
LT3973HDD-5#TRPBF
LGGF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping
container.Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBD = 3.3V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
Minimum Input Voltage (Note 3)
Quiescent Current from VIN
MIN
l
VEN/UVLO Low
VEN/UVLO High
VEN/UVLO High, –40°C to 125°C
VEN/UVLO High, –40°C to 150°C
4.2V < VIN < 40V
Switching Frequency
RT = 41.2k, VIN = 6V
RT = 158k, VIN = 6V
RT = 768k, VIN = 6V
Switch Current Limit
VIN = 5V, VFB = 0V
Catch Schottky Current Limit
VIN = 5V
Switch VCESAT
ISW = 500mA
V
0.75
1.8
1.3
2.8
6
12
µA
µA
µA
µA
1.21
1.21
1.225
1.235
V
V
l
3.26
3.234
3.3
3.3
3.34
3.366
V
V
l
4.94
4.9
5
5
5.06
5.1
V
V
0.1
20
nA
0.0002
0.01
%/V
1.72
632
156
2.15
790
195
2.58
948
234
MHz
kHz
kHz
l
1.237
1.65
1.98
A
l
0.92
1.15
1.44
A
l
250
Switch Leakage Current
0.05
ISCH = 200mA, VIN = VBD = NC
550
Catch Schottky Reverse Leakage
VSW = 12V
0.05
Boost Schottky Forward Voltage
ISCH = 50mA, VIN = NC, VBOOST = 0V
820
Catch Schottky Forward Voltage
Boost Schottky Reverse Leakage
VREVERSE = 12V
Minimum Boost Voltage (Note 4)
VIN = 5V
BOOST Pin Current
ISW = 500mA, VBOOST = 15V
Dropout Comparator Threshold
(VIN - OUT) Falling, VIN = 5V
Dropout Comparator Hysteresis
UNITS
1.195
1.185
LT3973-5 Output Voltage
FB/Output Voltage Line Regulation
4.2
l
LT3973-3.3 Output Voltage
VFB = 1.21V
MAX
3.8
l
l
LT3973 Feedback Voltage
LT3973 FB Pin Bias Current
TYP
l
l
400
mV
2
µA
mV
2
µA
mV
0.02
2
µA
1.4
1.8
V
10
13
mA
490
580
mV
40
mV
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LT3973/LT3973-3.3/LT3973-5
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBD = 3.3V unless otherwise noted. (Note 2)
EN/UVLO Pin Current
VEN/UVLO = 12V
EN/UVLO Voltage Threshold
EN/UVLO Falling, VIN ≥ 4.2V
l
EN/UVLO Voltage Threshold
EN/UVLO Rising, VIN ≥ 4.2V
l
1
30
nA
1.09
1.16
1.23
V
1.12
1.19
1.28
30
45
mV
10
13.5
%
1
µA
EN/UVLO Voltage Hysteresis
PG Threshold Offset from Feedback Voltage
VFB Rising
6.5
PG Hysteresis as % of Output Voltage
0.8
PG Leakage
VPG = 3V
PG Sink Current
VPG = 0.4V
0.01
l
Minimum Switch On-Time
Minimum Switch Off-Time (Note 5)
VIN = 10V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3973E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization, and correlation with statistical process controls. The
LT3973I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT3973H is guaranteed over the full –40°C to
150°C operating junction temperature range. High junction temperatures
degrade operating lifetimes. Operating lifetime is derated at junction
temperatures greater than 125°C. The junction temperature (TJ, in °C) is
calculated from the ambient temperature (TA in °C) and power dissipation
(PD, in Watts) according to the formula:
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance.
4
220
V
%
350
µA
70
ns
130
180
ns
Note 3: This is the minimum input voltage for operation with accurate FB
reference voltage.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
Note 5: The LT3973 contains circuitry that extends the maximum duty
cycle if there is sufficient voltage across the boost capacitor. See the
Application Information section for more details.
Note 6: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified maximum operating junction temperature may impair device
reliability or permanently damage the device.
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TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
Efficiency, VOUT = 3.3V
90
VIN = 12V
80
VIN = 12V
80
60
50
FRONT PAGE APPLICATION
VOUT = 3.3V
R1 = 1M
R2 = 576k
30
0
0.1
0.2 0.3 0.4 0.5
LOAD CURRENT (A)
0.6
VIN = 24V
70
VIN = 36V
60
50
0.7
VIN = 24V
FRONT PAGE APPLICATION
0
0.1
0.2 0.3 0.4 0.5
LOAD CURRENT (A)
0.6
20
0.01
0.7
VIN = 36V
50
FRONT PAGE APPLICATION
VOUT = 3.3V
R1 = 1M
R2 = 576k
30
3973 G01
0.1
1
10
100
LOAD CURRENT (mA)
3973 G02
Efficiency, VOUT = 5V
3973 G03
LT3973 Feedback Voltage
90
LT3973-3.3 Output Voltage
1.220
3.32
1.215
3.31
VIN = 24V
60
VIN = 36V
50
OUTPUT VOLTAGE (V)
70
FEEDBACK VOLTAGE (V)
VIN = 12V
80
1.210
1.205
1.200
40
30
0.01
FRONT PAGE APPLICATION
0.1
1
10
100
LOAD CURRENT (mA)
1.195
–50 –25
LT3973-5 Output Voltage
0
3.27
–50 –25
25 50
75 100 125 150
TEMPERATURE (°C)
5.02
SUPPLY CURRENT (µA)
4.98
4.96
3.0
2.5
2.0
25 50
75 100 125 150
TEMPERATURE (°C)
3973 G07
25 50
75 100 125 150
TEMPERATURE (°C)
No-Load Supply Current
FRONT PAGE APPLICATION
VOUT = 3.3V
R1 = 1M
R2 = 576k
LT3973-3.3
3.5
5.00
0
3973 G06
No-Load Supply Current
4.0
0
3.29
3973 G05
5.04
4.94
–50 –25
3.30
3.28
3973 G04
OUTPUT VOLTAGE (V)
60
40
40
30
VIN = 12V
70
EFFICIENCY (%)
VIN = 36V
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 24V
40
EFFICIENCY (%)
90
80
70
20
Efficiency, VOUT = 3.3V
Efficiency, VOUT = 5V
1.5
35
30
SUPPLY CURRENT (µA)
90
25
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 3.3V
R1 = 1M
R2 = 576k
20
15
10
5
5
10
30
15
25
20
INPUT VOLTAGE (V)
35
40
3973 G08
0
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
3973 G09
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LT3973/LT3973-3.3/LT3973-5
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
Maximum Load Current
Maximum Load Current
TYPICAL
1.2
LOAD CURRENT (A)
MINIMUM
1.0
0.8
0.6
0.4
LOAD CURRENT (A)
1.2
1.2
MINIMUM
0.9
0.6
0.3
FRONT PAGE APPLICATION
VOUT = 3.3V
0.2
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
40
0
45
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
40
3973 G10
Load Regulation
0.4
Switch Current Limit
SWITCH CURRENT LIMIT (A)
LOAD REGULATION (%)
0.6
0.10
0.05
0
–0.05
–0.10
LIMITED BY MAXIMUM
JUNCTION TEMPERATURE;
θJA = 45°C/W
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G12
Switch Current Limit
2.0
0.20
–0.15 FRONT PAGE APPLICATION
REFERENCED FROM VOUT AT 100mA LOAD
–0.20
100 200 300 400 500 600 700
0
2.0
1.8
SWITCH PEAK
CURRENT LIMIT
1.6
1.4
1.2
CATCH DIODE VALLEY CURRENT LIMIT
1.0
0.8
0
20
40
60
DUTY CYCLE (%)
LOAD CURRENT (mA)
3973 G13
SWITCH PEAK CURRENT LIMIT
1.6
1.4
1.2
CATCH DIODE VALLEY CURRENT LIMIT
1.0
0.8
–50 –25
100
80
1.8
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G14
Switching Frequency
3973 G15
Switch VCESAT (ISW = 500mA)
vs Temperature
Minimum Switch On-Time
150
2.4
2.2
350
LOAD CURRENT = 375mA
SWITCH ON-TIME (ns)
1.8
1.6
1.4
1.2
1.0
0.8
0.6
SWITCH VCESAT (mV)
125
2.0
FREQUENCY (MHz)
0.8
0
–50 –25
45
H-GRADE
3973 G11
0.25
0.15
LIMITED BY CURRENT LIMIT
1.0
0.2
FRONT PAGE APPLICATION
VOUT = 5V
SWITCH CURRENT LIMIT (A)
LOAD CURRENT (A)
1.4
TYPICAL
1.4
0
Maximum Load Current
1.5
1.6
100
75
50
300
250
25
0.4
0.2
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G16
6
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G17
200
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G18
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TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
BOOST PIN CURRENT (mA)
SWITCH VCESAT (mV)
500
400
300
200
200
400
600
800 1000
SWITCH CURRENT (mA)
5.0
20
4.5
15
10
5
100
0
25
INPUT VOLTAGE (V)
600
0
Minimum Input Voltage,
VOUT = 3.3V
BOOST Pin Current
Switch VCESAT
0
1200
Minimum Input Voltage,
VOUT = 5V
200
0
400
600
800 1000
SWITCH CURRENT (mA)
2.5
1200
0
100 200 300 400 500 600 700
LOAD CURRENT (mA)
3973 G21
Start-Up and Dropout
Performance
9
FRONT PAGE APPLICATION
VOUT = 5V
Minimum Input Voltage to Switch
4.0
FRONT PAGE APPLICATION
8
7
5.0
5
INPUT VOLTAGE (V)
TO START/TO RUN
3.5
VIN
6
VOLTAGE (V)
INPUT VOLTAGE (V)
TO START/TO RUN
3.5
3973 G20
6.0
5.5
4.0
3.0
3973 G19
6.5
FRONT PAGE APPLICATION
VOUT = 3.3V
VOUT
4
3
4.5
2
4.0
0
3.0
2.5
1
0
100 200 300 400 500 600 700
LOAD CURRENT (mA)
2.0
–50 –25
TIME
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G23
3973 G24
3973 G22
1.4
VFB Regulation Voltage
Boost Diode Forward Voltage
Catch Diode Forward Voltage
1.2
1.0
1.0
0.8
1.0
0.8
CATCH DIODE, VF (V)
BOOST DIODE VF (V)
VFB (V)
1.2
0.8
0.6
0.4
150°C
125°C
25°C
–50°C
0.2
0.6
2.0
2.5
3.0
3.5
4.0
INPUT VOLTAGE (V)
4.5
5.0
3973 G25
0
0
50
100
150
BOOST DIODE CURRENT (mA)
200
3973 G26
0.6
0.4
150°C
125°C
25°C
–50°C
0.2
0
0
600
200
400
800 1000
CATCH DIODE CURRENT (mA)
1200
3973 G27
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TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
Catch Diode Leakage
Power Good Threshold
200
150
100
1.245
91
1.220
90
89
50
0
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
88
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G28
1.195
1.170
1.145
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3973 G30
Transient Load Response; Load
Current is Stepped from 250mA
to 500mA
VOUT
100mV/DIV
VOUT
100mV/DIV
IL
200mA/DIV
IL
200mA/DIV
50µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
EN/UVLO RISING
3973 G29
Transient Load Response; Load
Current is Stepped from 50mA
(Burst Mode Operation) to 300mA
50µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
3973 G31
Switching Waveforms,
Burst Mode Operation
3973 G32
Switching Waveforms, Full
Frequency Continuous Operation
VSW
5V/DIV
VSW
5V/DIV
IL
200mA/DIV
IL
200mA/DIV
VOUT
10mV/DIV
VOUT
5mV/DIV
5µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
ILOAD = 15mA
f = 600kHz
8
EN/UVLO Threshold
92
THRESHOLD VOLTAGE (V)
VR = 12V
THRESHOLD (%)
CATCH DIODE LEAKAGE (µA)
250
3973 G33
1µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
ILOAD = 750mA
f = 600kHz
3973 G34
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PIN FUNCTIONS
FB (Pin 1, LT3973 Only): The LT3973 regulates the FB
pin to 1.21V. Connect the feedback resistor divider tap
to this pin.
VOUT (Pin 1, LT3973-3.3 and LT3973-5 Only): The
LT3973-3.3 and LT3973-5 regulate the VOUT pin to 3.3V
and 5V, respectively. This pin connects to the internal
feedback divider that programs the fixed output voltage.
OUT (Pin 2): The LT3973 regulates the VIN to VOUT voltage
for dropout conditions. It will also pull current from this
pin to charge the boost capacitor when needed. Connect
this pin to the output. If programmed output is greater
than 14V, tie this pin to GND.
EN/UVLO (Pin 3): The part is in shutdown when this pin is
low and active when this pin is high. The threshold voltage
is 1.19V going up with 30mV of hysteresis. Tie to VIN if
shutdown feature is not used. The EN/UVLO threshold is
accurate only when VIN is above 4.2V. If VIN is lower than
4.2V, ground EN/UVLO to place the part in shutdown.
GND (Pin 5, Exposed Pad Pin 11): Ground. The exposed
pad must be soldered to the PCB.
SW (Pin 6): The SW pin is the output of an internal power
switch. Connect this pin to the inductor.
BOOST (Pin 7): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
BD (Pin 8): This pin connects to the anode of the boost
diode. This pin also supplies current to the LT3973’s
internal regulator when BD is above 3.2V.
PG (Pin 9): The PG pin is the open-drain output of an
internal comparator. PG remains low until the FB pin is
within 10% of the final regulation voltage. PG is valid when
VIN is above 4.2V and EN/UVLO is high.
RT (Pin 10): A resistor is tied between RT and ground to
set the switching frequency.
VIN (Pin 4): The VIN pin supplies current to the LT3973’s
internal circuitry and to the internal power switch. This
pin must be locally bypassed.
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9
10
C1
VIN
RT
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–
+
GND
–
+
LT3973 ONLY
R2
1.09V
SHDN
FB
R1
–
ERROR
AMP
+
INTERNAL 1.21V REF
* LT3973-3.3: R1 = 12.72M, R2 = 7.39M
LT3973-5: R1 = 15.23M, R2 = 4.88M
PG
RT
EN/UVLO
1.21V
VIN
Burst Mode
DETECT
R1
LT3973-3.3 AND
LT3973-5 ONLY*
R2
VC
VOUT
OSCILLATOR
200kHz TO 2.2MHz
SLOPE COMP
+
–
S
R
Q
SWITCH LATCH
+
–
+
–
DCATCH
DBOOST
SW
BOOST
BD
OUT
C2
L1
C3
3973 BD
VOUT
LT3973/LT3973-3.3/LT3973-5
BLOCK DIAGRAM
3973fb
LT3973/LT3973-3.3/LT3973-5
OPERATION
The LT3973 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
sets an RS flip-flop, turning on the internal power switch.
An amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch-off when
this current reaches a level determined by the voltage at
VC (see Block Diagram). An error amplifier measures the
output voltage through an external resistor divider tied to
the FB pin and servos the VC node. If the error amplifier’s
output increases, more current is delivered to the output;
if it decreases, less current is delivered.
Another comparator monitors the current flowing through
the catch diode and reduces the operating frequency when
the current exceeds the 1.15A bottom current limit. This
foldback in frequency helps to control the output current
in fault conditions such as a shorted output with high
input voltage. Maximum deliverable current to the output
is therefore limited by both switch current limit and catch
diode current limit.
An internal regulator provides power to the control circuitry. The bias regulator normally draws power from
the VIN pin, but if the BD pin is connected to an external
voltage higher than 3.2V, bias power will be drawn from
the external source (typically the regulated output voltage).
This improves efficiency.
If the EN/UVLO pin is low, the LT3973 is shut down and draws
0.75µA from the input. When the EN/UVLO pin exceeds
1.19V, the switching regulator will become active. Undervoltage lockout is programmable via this pin.
The switch driver operates from either VIN or from the
BOOST pin. An external capacitor is used to generate a
voltage at the BOOST pin that is higher than the input
supply. This allows the driver to fully saturate the internal
bipolar NPN power switch for efficient operation.
To further optimize efficiency, the LT3973 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down reducing the input supply
current to 1.8µA.
If the input voltage decreases towards the programmed
output voltage, the LT3973 will start to skip switch-off times
and decrease the switching frequency to maintain output
regulation up to a maximum duty cycle of approximately
97.5%. When the OUT pin is tied to VOUT, the LT3973
regulates the output such that it stays more than 530mV
below VIN; this sets a minimum dropout voltage. This
enforced minimum dropout voltage limits the duty cycle
and keeps the boost capacitor charged during dropout
conditions. Since sufficient boost voltage is maintained,
the internal switch can fully saturate yielding good dropout
performance.
The LT3973 contains a power good comparator which
trips when the FB pin is at 90% of its regulated value. The
PG output is an open-drain transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3973 is
enabled and VIN is above 4.2V.
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FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
V

R1= R2  OUT – 1
 1.21 
Reference designators refer to the Block Diagram. Note
that choosing larger resistors will decrease the quiescent
current of the application circuit.
Setting the Switching Frequency
The LT3973 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2.2MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Table 1.
Table 1. Switching Frequency vs RT Value
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
0.3
0.4
0.5
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
2.2
732
475
340
267
215
150
115
90.9
73.2
61.9
51.1
43.2
36.5
Operating Frequency Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, and maximum input voltage.
The advantage of high frequency operation is that smaller
inductor and capacitor values may be used. The disadvantages are lower efficiency, and narrower input voltage
range at constant-frequency. The highest acceptable
switching frequency (fSW(MAX)) for a given application
can be calculated as follows:
fSW(MAX) =
12
VOUT + VD
tON(MIN) ( VIN – VSW + VD )
where VIN is the typical input voltage, VOUT is the output
voltage, VD is the integrated catch diode drop (~0.7V),
and VSW is the internal switch drop (~0.5V at max load).
This equation shows that slower switching frequency is
necessary to accommodate high VIN/VOUT ratio. This is
due to the limitation on the LT3973’s minimum on-time.
The minimum on-time is a strong function of temperature.
Use the minimum switch on-time curve (see Typical Performance Characteristics) to design for an application’s
maximum temperature, while adding about 30% for
part-to-part variation. The minimum duty cycle that can
be achieved taking this on-time into account is:
DCMIN = tON(MIN) • fSW
where fSW is the switching frequency, and the tON(MIN) is
the minimum switch on-time.
A good choice of switching frequency should allow adequate input voltage range (see next two sections) and
keep the inductor and capacitor values small.
Minimum Input Voltage Range
The minimum input voltage for regulation is determined
by either the LT3973’s minimum operating voltage of
4.2V, its maximum duty cycle, or the enforced minimum
dropout voltage. See the typical performance characteristics section for the minimum input voltage across load
for outputs of 3.3V and 5V.
The duty cycle is the fraction of time that the internal
switch is on during a clock cycle. Unlike many fixed frequency regulators, the LT3973 can extend its duty cycle
by remaining on for multiple clock cycles. The LT3973
will not switch off at the end of each clock cycle if there
is sufficient voltage across the boost capacitor (C3 in
the Block Diagram). Eventually, the voltage on the boost
capacitor falls and requires refreshing. When this occurs,
the switch will turn off, allowing the inductor current to
recharge the boost capacitor. This places a limitation on
the maximum duty cycle as follows:
DCMAX = 1/(1+1/ βSW)
where βSW is equal to the SW pin current divided by
the BOOST pin current (see the Typical Performance
Characteristics section), generally leading to a DCMAX of
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about 97.5%. This leads to a minimum input voltage of
approximately:
VIN(MIN1) =
VOUT + VD
– VD + VSW
DCMAX
Inductor Selection
where VOUT is the output voltage, VD is the catch diode
drop (~0.7V), VSW is the internal switch drop (~0.5V at
max load), and DCMAX is the maximum duty cycle.
The final factor affecting the minimum input voltage is the
minimum dropout voltage. When the OUT pin is tied to
VOUT, the LT3973 regulates the output such that it stays
more than 530mV below VIN. This enforced minimum
dropout voltage is due to reasons that are covered in a
later section. This places a limitation on the minimum
input voltage as follows:
VIN(MIN2) = VOUT + VDROPOUT(MIN)
where VOUT is the output voltage and VDROPOUT(MIN) is
the minimum dropout voltage (530mV).
Combining these factors leads to the overall minimum
input voltage:
VIN(MIN) = max(VIN(MIN1), VIN(MIN2), 4.2V)
Note that the LT3973 will begin switching at a lower input
voltage (typically 3V) but will regulate to a lower FB voltage
in this region of operation (see the Typical Performance
Characteristics section).
Maximum Input Voltage Range
The highest allowed VIN during normal operation (VIN(OPMAX)) is limited by minimum duty cycle and can be calculated by the following equation:
VIN(OP-MAX) =
be reduced below the programmed frequency to prevent
damage to the part. The output voltage ripple and inductor
current ripple may also be higher than in typical operation,
however the output will still be in regulation.
VOUT + VD
– VD + VSW
fSW • tON(MIN)
where tON(MIN) is the minimum switch on time.
However, the circuit will tolerate inputs up to the absolute
maximum ratings of the VIN and BOOST pins, regardless of
chosen switching frequency. During such transients where
VIN is higher than VIN(OP-MAX), the switching frequency will
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current increases with higher VIN or VOUT and
decreases with higher inductance and faster switching
frequency. A good starting point for selecting the inductor value is:
L = 1.5
VOUT + VD
fSW
where VD is the voltage drop of the catch diode (~0.7V),
L is in µH and fSW is in MHz. The inductor’s RMS current
rating must be greater than the maximum load current
and its saturation current should be about 30% higher.
For robust operation in fault conditions (start-up or short
circuit) and high input voltage (>30V), the saturation current should be above 1.5A. To keep the efficiency high,
the series resistance (DCR) should be less than 0.1Ω, and
the core material should be intended for high frequency
applications. Table 2 lists several inductor vendors.
Table 2. Inductor Vendors
VENDOR
URL
Coilcraft
www.coilcraft.com
Sumida
www.sumida.com
Toko
www.tokoam.com
Würth Elektronik
www.we-online.com
Coiltronics
www.cooperet.com
Murata
www.murata.com
This simple design guide will not always result in the
optimum inductor selection for a given application. As a
general rule, lower output voltages and higher switching
frequency will require smaller inductor values. If the application requires less than 750mA load current, then a
lesser inductor value may be acceptable. This allows use
of a physically smaller inductor, or one with a lower DCR
resulting in higher efficiency. There are several graphs in
the Typical Performance Characteristics section of this data
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Input Capacitor
Bypass the input of the LT3973 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 4.7µF ceramic capacitor is adequate to bypass the
LT3973 and will easily handle the ripple current. Note that
larger input capacitance is required when a lower switching
frequency is used (due to longer on-times). If the input
power source has high impedance, or there is significant
inductance due to long wires or cables, additional bulk
capacitance may be necessary. This can be provided with
a low performance electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3973 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7µF capacitor is capable of this task, but only if it is
placed close to the LT3973 (see the PCB Layout section).
A second precaution regarding the ceramic input capacitor
concerns the maximum input voltage rating of the LT3973.
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank
circuit. If the LT3973 circuit is plugged into a live supply,
the input voltage can ring to twice its nominal value, possibly exceeding the LT3973’s voltage rating. This situation
is easily avoided (see the Hot Plugging Safely section).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. It stores
energy in order to satisfy transient loads and stabilize the
LT3973’s control loop. Ceramic capacitors have very low
14
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
COUT =
50
VOUT • fSW
where fSW is in MHz and COUT is the recommended output
capacitance in μF. Use X5R or X7R types. This choice will
provide low output ripple and good transient response.
Transient performance can be improved with a higher value
capacitor if combined with a phase lead capacitor (typically
15pF) between the output and the feedback pin. A lower
value of output capacitor can be used to save space and
cost but transient performance will suffer.
The second function is that the output capacitor, along
with the inductor, filters the square wave generated by the
LT3973 to produce the DC output. In this role it determines
the output ripple, so low impedance (at the switching
frequency) is important. The output ripple decreases with
increasing output capacitance, down to approximately
1mV. See Figure 1. Note that a larger phase lead capacitor
should be used with a large output capacitor.
16
WORST-CASE OUTPUT RIPPLE (mV)
sheet that show the maximum load current as a function
of input voltage for several popular output voltages. Low
inductance may result in discontinuous mode operation,
which is acceptable but reduces maximum load current.
For details of maximum output current and discontinuous mode operation, see Application Note 44. Finally, for
duty cycles greater than 50% (VOUT/VIN > 0.5), there is
a minimum inductance required to avoid subharmonic
oscillations. See Application Note 19.
FRONT PAGE APPLICATION
14
12
10
8
6
VIN = 24V
VIN = 12V
4
2
0
0
20
60
40
COUT (µF)
80
100
3973 F01
Figure 1. Worst-Case Output Ripple Across Full Load Range
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor or one with a higher voltage
rating may be required. Table 3 lists several capacitor
vendors.
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MANUFACTURER
WEBSITE
AVX
www.avxcorp.com
Murata
www.murata.com
Taiyo Yuden
www.t-yuden.com
Vishay Siliconix
www.vishay.com
TDK
www.tdk.com
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3973 due to their piezoelectric nature.
When in Burst Mode operation, the LT3973’s switching
frequency depends on the load current, and at very light
loads the LT3973 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT3973
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If
this is unacceptable, use a high performance tantalum or
electrolytic capacitor at the output.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT3973. As previously mentioned, a ceramic input capacitor combined
with trace or cable inductance forms a high quality (under
damped) tank circuit. If the LT3973 circuit is plugged into a
live supply, the input voltage can ring to twice its nominal
value, possibly exceeding the LT3973’s rating. This situation is easily avoided (see the Hot Plugging Safely section).
Low Ripple Burst Mode Operation
To enhance efficiency at light loads, the LT3973 operates
in low ripple Burst Mode operation which keeps the output
capacitor charged to the proper voltage while minimizing
the input quiescent current. During Burst Mode operation, the LT3973 delivers single cycle bursts of current
to the output capacitor followed by sleep periods where
the output power is delivered to the load by the output
capacitor. Because the LT3973 delivers power to the output
with single, low current pulses, the output ripple is kept
below 10mV for a typical application. See Figure 2.
As the load current decreases towards a no load condition, the percentage of time that the LT3973 operates in
sleep mode increases and the average input current is
greatly reduced resulting in high efficiency even at very
low loads. Note that during Burst Mode operation, the
switching frequency will be lower than the programmed
switching frequency. See Figure 3.
At higher output loads (above 90mA for the front page
application) the LT3973 will be running at the frequency
programmed by the RT resistor, and will be operating in
standard PWM mode. The transition between PWM and
low ripple Burst Mode is seamless, and will not disturb
the output voltage.
VSW
5V/DIV
IL
200mA/DIV
VOUT
10mV/DIV
3973 F02
5µs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
ILOAD = 15mA
f = 600kHz
Figure 2. Burst Mode Operation
700
SWITCHING FREQUENCY (kHz)
Table 3. Recommended Ceramic Capacitor Vendors
FRONT PAGE APPLICATION
600
500
400
300
200
100
0
0
100
200 300 400 500 600
LOAD CURRENT (mA)
700
3973 F03
Figure 3. Switching Frequency in Burst Mode Operation
BOOST and BD Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see the
Block Diagram) are used to generate a boost voltage that
is higher than the input voltage. In most cases a 0.47µF
capacitor will work well. Figure 4 shows two ways to
arrange the boost circuit. The BOOST pin must be more
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5.0
FRONT PAGE APPLICATION
VOUT = 3.3V
4.5
INPUT VOLTAGE (V)
than 1.9V above the SW pin for best efficiency. For outputs of 2.2V and above, the standard circuit (Figure 4a) is
best. For outputs between 2.2V and 2.5V, use a 1µF boost
capacitor. For output voltages below 2.2V, the boost diode
can be tied to the input (Figure 4b), or to another external
supply greater than 2.2V. However, the circuit in Figure 4a
is more efficient because the BOOST pin current and BD
pin quiescent current come from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BD pins are not exceeded.
4.0
TO START/TO RUN
3.5
3.0
2.5
VOUT
0
100 200 300 400 500 600 700
LOAD CURRENT (mA)
BD
VIN
VIN
BOOST
6.5
C3
LT3973
FRONT PAGE APPLICATION
VOUT = 5V
SW
GND
INPUT VOLTAGE (V)
6.0
(4a) For VOUT ≥ 2.2V
BD
VIN
VIN
5.5
TO START/TO RUN
5.0
4.5
BOOST
C3
LT3973
SW
VOUT
GND
4.0
0
100 200 300 400 500 600 700
LOAD CURRENT (mA)
3973 F05
3973 F04
(4b) For VOUT < 2.2V; VIN < 25V
Figure 4. Two Circuits for Generating the Boost Voltage
Figure 5. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
Minimum Dropout Voltage
The LT3973 monitors the boost capacitor for sufficient
voltage such that the switch is allowed to fully saturate.
During start-up conditions when the boost capacitor may
not be fully charged, the switch will operate with about
1V of drop, and an internal current source will begin to
pull 70mA (typical) from the OUT pin which is typically
connected to VOUT. This current forces the LT3973 to
switch more often and with more inductor current, which
recharges the boost capacitor. When the boost capacitor
is sufficiently charged, the current source turns off, and
the part may enter Burst Mode. See Figure 5 for minimum
input voltage for outputs of 3.3V and 5V.
16
When the OUT pin is tied to VOUT, the LT3973 regulates
the output such that:
VIN – VOUT > VDROPOUT(MIN)
where VDROPOUT(MIN) is 530mV. This enforced minimum
dropout voltage keeps the boost capacitor charged regardless of load during dropout conditions. The LT3973
achieves this by limiting the duty cycle and forcing the
switch to turn off regularly to charge the boost capacitor. Since sufficient voltage across the boost capacitor is
maintained, the switch is allowed to fully saturate and the
internal switch drop stays low for good dropout performance. Figure 6 shows the overall VIN to VOUT performance
during start-up and dropout conditions.
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During dropout conditions when the output is below regulation, the output ripple may increase. At very high loads, this
ripple can increase to approximately 200mV for the front
page application. If lower output ripple is desired during
such conditions, a larger output capacitor can be used.
In order to not exceed the maximum voltage rating, tie
the OUT pin to GND for programmed outputs greater than
14V. Note that this will result in degraded start-up and
dropout performance.
9
VUVLO =
R3 + R4
• 1.19V
R4
where switching should not start until VIN is above VUVLO.
Note that due to the comparator’s hysteresis, switching
will not stop until the input falls slightly below VUVLO.
Undervoltage lockout is functional only when VUVLO is
greater than 5.5V.
VIN
FRONT PAGE APPLICATION
8
VOLTAGE (V)
7
6
5
4
LT3973
VIN
R3
1.19V
EN/UVLO
VIN
+
–
SHDN
R4
VOUT
3973 F07
Figure 7. Undervoltage Lockout
3
2
1
0
D4
TIME
3973 F06
VIN
Enable and Undervoltage Lockout
Figure 7 shows how to add undervoltage lockout (UVLO)
to the LT3973. Typically, UVLO is used in situations where
the input supply is current limited, or has a relatively high
source resistance. A switching regulator draws constant
power from the source, so source current increases as
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where the problems might occur. The UVLO threshold can
be adjusted by setting the values R3 and R4 such that they
satisfy the following equation:
BOOST
LT3973
Figure 6. VIN to VOUT Performance
The LT3973 is in shutdown when the EN/UVLO pin is low
and active when the pin is high. The rising threshold of the
EN/UVLO comparator is 1.19V, with a 30mV hysteresis.
This threshold is accurate when VIN is above 4.2V. If VIN
is lower than 4.2V, tie EN/UVLO pin to GND to place the
part in shutdown.
BD
VIN
EN/UVLO
SW
GND
FB
VOUT
+
BACKUP
3973 F08
Figure 8. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output. It Also Protects the Circuit
from a Reversed Input. The LT3973 Runs Only When the Input Is
Present
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively,
a LT3973 buck regulator will tolerate a shorted output.
There is another situation to consider in systems where the
output will be held high when the input to the LT3973 is
absent. This may occur in battery charging applications or
in battery backup systems where a battery or some other
supply is diode ORed with the LT3973’s output. If the VIN
pin is allowed to float and the EN/UVLO pin is held high
(either by a logic signal or because it is tied to VIN), then
the LT3973’s internal circuitry will pull its quiescent current
through its SW pin. This is fine if the system can tolerate
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a few µA in this state. If the EN/UVLO pin is grounded, the
SW pin current will drop to 0.75µA. However, if the VIN pin
is grounded while the output is held high, regardless of EN/
UVLO, parasitic diodes inside the LT3973 can pull current
from the output through the SW pin and the VIN pin. Figure
8 shows a circuit that will run only when the input voltage is
present and that protects against a shorted or reversed input.
GND
GND
1
10
2
9
EN/UVLO
3
8
VIN
4
7
5
6
PG
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 9 shows
the recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT3973’s VIN and SW pins, the internal catch
diode and the input capacitor. The loop formed by these
components should be as small as possible. These components, along with the inductor and output capacitor, should
be placed on the same side of the circuit board, and their
connections should be made on that layer. Place a local,
unbroken ground plane below these components. The SW
and BOOST nodes should be as small as possible. Finally,
keep the FB nodes small so that the ground traces will shield
them from the SW and BOOST nodes. The exposed pad on
the bottom must be soldered to ground so that the pad acts
as a heat sink. To keep thermal resistance low, extend the
ground plane as much as possible, and add thermal vias
under and near the LT3973 to additional ground planes
within the circuit board and on the bottom side.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3973 circuits. However, these capacitors can cause problems if the LT3973 is plugged into
a live supply. The low loss ceramic capacitor, combined
with stray inductance in series with the power source,
forms an under damped tank circuit, and the voltage at
the VIN pin of the LT3973 can ring to twice the nominal
input voltage, possibly exceeding the LT3973’s rating and
damaging the part. If the input supply is poorly controlled
or the user will be plugging the LT3973 into an energized
supply, the input network should be designed to prevent
this overshoot. See Application Note 88 for a complete
discussion.
18
VOUT
GND
VIAS TO LOCAL GROUND PLANE
VIAS TO VOUT
3973 F09
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
High Temperature Considerations
For higher ambient temperatures, care should be taken
in the layout of the PCB to ensure good heat sinking of
the LT3973. The exposed pad on the bottom must be
soldered to a ground plane. This ground should be tied to
large copper layers below with thermal vias; these layers
will spread the heat dissipated by the LT3973. Placing
additional vias can reduce thermal resistance further. The
maximum load current should be derated as the ambient
temperature approaches the maximum junction rating.
Power dissipation within the LT3973 can be estimated by
calculating the total power loss from an efficiency measurement and subtracting inductor loss. The die temperature
is calculated by multiplying the LT3973 power dissipation
by the thermal resistance from junction to ambient.
Finally, be aware that at high ambient temperatures the
internal Schottky diode will have significant leakage current (see the Typical Performance Characteristics section)
increasing the quiescent current of the LT3973 converter.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
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LT3973/LT3973-3.3/LT3973-5
TYPICAL APPLICATIONS
3.3V Step-Down Converter
VIN
4.2V TO 42V
C3
0.47µF
VIN
BOOST
LT3973
OFF ON
C1
4.7µF
5V Step-Down Converter
VIN
5.6V TO 42V
EN/UVLO
PG
RT
BD
OUT
15pF
1M
C2
22µF
FB
OFF ON
C1
4.7µF
RT
BOOST
LT3973-3.3
OFF ON
C1
4.7µF
EN/UVLO
PG
GND
215k
VOUT
3.3V
750mA
BD
OFF ON
BOOST
OFF ON
C1
4.7µF
EN/UVLO
PG
RT
215k
215k
R1
1M
22pF
R2
931k
C2
47µF
OFF ON
C1
4.7µF
215k
C1
4.7µF
EN/ULVO
PG
RT
215k
f = 600kHz
GND
FB
BOOST
GND
C3
0.47µF
L1
10µH
22pF
FB
VOUT
1.8V
750mA
R1
487k
R2
1M
C2
47µF
3973 TA07
5V, 2MHz Step-Down Converter
VIN
5.6V TO 28V
TRANSIENTS
TO 42V
C3
0.47µF
L1
22µH
15pF
R1
1M
R2
113k
C2
22µF
C3
0.47µF
VIN
BOOST
LT3973
VOUT
12V
750mA
SW
BD
OUT
3973 TA05
f = 600kHz
12V Step-Down Converter
OFF ON
C2
22µF
VOUT
EN/UVLO SW
BD
OUT
PG
RT
3973 TA06
BOOST
BD
LT3973
VOUT
2.5V
750mA
FB
LT3973
GND
VIN
L1
10µH
f = 600kHz
VIN
VOUT
5V
750mA
SW
VIN
4.2V TO 25V
BD
VIN
12.6V TO 42V
L1
15µH
1.8V Step-Down Converter
SW
GND
C3
0.47µF
f = 600kHz
C3
1µF
OUT
EN/UVLO
PG
RT
2.5V Step-Down Converter
LT3973
C2
22µF
R2
316k
OUT
C1
4.7µF
C2
22µF
VOUT
BOOST
LT3973-5
3973 TA04
VIN
FB
R1
1M
3973 TA03
VIN
L1
15µH
SW
VIN
4.2V TO 42V
15pF
5V Step-Down Converter
C3
0.47µF
f = 600kHz
BD
VIN
5.6V TO 42V
OUT
RT
VOUT
5V
750mA
f = 600kHz
3.3V Step-Down Converter
VIN
L1
15µH
SW
GND
215k
3973 TA02
VIN
4.2V TO 42V
EN/UVLO
PG
OUT
576k
f = 600kHz
BOOST
LT3973
VOUT
3.3V
750mA
SW
GND
215k
VIN
L1
15µH
C3
0.47µF
OFF ON
C1
2.2µF
3973 TA08
EN/UVLO
PG
RT
43.2k
f = 2MHz
GND
L1
10µH
VOUT
5V
750mA
SW
BD
OUT
FB
10pF
R1
1M
R2
316k
C2
10µF
3973 TA09
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19
LT3973/LT3973-3.3/LT3973-5
TYPICAL APPLICATIONS
5V Step-Down Converter with Undervoltage Lockout
VIN
6V TO 42V
kΩ
+
0.47µF
–
VIN
3.9M
BOOST
LT3973
976k
EN/UVLO
PG
4.7µF
RT
15µH
VOUT
5V
750mA
SW
BD
15pF
215k
FB
GND OUT
22µF
316k
3973 TA10a
f = 600kHz
Input Current During Start-Up
Start-Up from High Impedance Input Source
60
UVLO PROGRAMMED TO 6V
INPUT CURRENT (mA)
50
40
INPUT CURRENT
DROPOUT
CONDITIONS
30
FRONT PAGE
APPLICATION
10
0
20
VIN
5V/DIV
VOUT
2V/DIV
FRONT PAGE
APPLICATION
WITH UVLO
PROGRAMMED
TO 6V
20
–10
1M
0
2
6
8
4
INPUT VOLTAGE (V)
10
12
5ms/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
1k INPUT SOURCE RESISTANCE
2.5mA LOAD
3973 TA10c
3973 TA10b
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LT3973/LT3973-3.3/LT3973-5
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DD Package
DD Package
Plastic
DFN
(3mm × 3mm)
10-Lead10-Lead
Plastic DFN
(3mm
× 3mm)
(Reference
DWG # 05-08-1699
(Reference
LTC DWGLTC
# 05-08-1699
Rev C) Rev C)
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ±0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 ±0.10
(4 SIDES)
R = 0.125
TYP
6
0.40 ±0.10
10
1.65 ±0.10
(2 SIDES)
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
5
0.75 ±0.05
0.00 – 0.05
1
(DD) DFN REV C 0310
0.25 ±0.05
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
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21
LT3973/LT3973-3.3/LT3973-5
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev I)
BOTTOM VIEW OF
EXPOSED PAD OPTION
1.88 ±0.102
(.074 ±.004)
5.10
(.201)
MIN
1
0.889 ±0.127
(.035 ±.005)
1.68 ±0.102
(.066 ±.004)
0.05 REF
10
0.305 ± 0.038
(.0120 ±.0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
3.00 ±0.102
(.118 ±.004)
(NOTE 3)
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
10 9 8 7 6
DETAIL “A”
0° – 6° TYP
1 2 3 4 5
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
0.18
(.007)
SEATING
PLANE
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
NOTE:
BSC
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD
SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
22
0.497 ±0.076
(.0196 ±.003)
REF
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
0.254
(.010)
0.29
REF
1.68
(.066)
3.20 – 3.45
(.126 – .136)
0.50
(.0197)
BSC
1.88
(.074)
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE) 0213 REV I
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LT3973/LT3973-3.3/LT3973-5
REVISION HISTORY
REV
DATE
DESCRIPTION
A
4/12
Title and Features modified to include fixed output versions.
PAGE NUMBER
1
Absolute Maximum Ratings, Pin Configuration, and Order Information sections modified to include fixed
output versions.
2
Electrical Characteristics table modified to include fixed output versions.
3
Graphs modified to include fixed output versions.
Pin Functions and Block Diagram modified to include fixed output versions.
B
3/15
5
9, 10
Applications for fixed output versions added.
19
Clarified package designator from MS to MSE – All
2
Clarified Electrical Characteristics
3
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT3973
23
LT3973/LT3973-3.3/LT3973-5
TYPICAL APPLICATION
1.21V Step-Down Converter
VIN
4.2V TO 25V
VIN
BOOST
LT3973
OFF ON
C1
4.7µF
340k
C3
0.47µF
L1
10µH
EN/UVLO SW
BD
FB
PG
OUT
RT
GND
f = 400kHz
VOUT
1.21V
750mA
C2
47µF
3973 TA10
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3970/LT39703.3/LT3970-5
40V, 350mA, 2.2MHz High Efficiency Micropower Step-Down DC/DC
Converter with IQ = 2.5µA
VIN = 4.2V to 40V, VOUT(MIN) = 1.21V, IQ = 2.5µA, ISD < 1µA,
3mm × 2mm DFN-10, MSOP-10
LT3990
62V, 350mA, 2.2MHz High Efficiency Micropower Step-Down DC/DC
Converter with IQ = 2.5µA
VIN = 4.2V to 62V, VOUT(MIN) = 1.21V, IQ = 2.5µA, ISD < 1µA,
3mm × 3mm DFN-16, MSOP-16E
LT3971
38V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down DC/DC
Converter with IQ = 2.8µA
VIN = 4.3V to 38V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOPE-10
LT3991
55V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down DC/DC
Converter with IQ = 2.8µA
VIN = 4.3V to 55V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOPE-10
LT3682
36V, 60VMAX, 1A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter
VIN = 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 75µA, ISD < 1µA,
3mm × 3mm DFN-12
24 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT3973
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT3973
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LT 0315 REV B • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2011