SEMTECH SC427EVB

SC417/SC427
10A Integrated FET Regulator
with Programmable LDO
POWER MANAGEMENT
Features
Description
„
The SC417/SC427 is a stand-alone synchronous buck
power supply. It features integrated power MOSFETs, a
bootstrap switch, and a programmable LDO in a spacesaving MLPQ-5x5mm 32-pin package. The device is highly
efficient and uses minimal PCB area. It uses pseudo-fixed
frequency adaptive on-time operation to provide fast
transient response.
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Input voltage — 3V to 28V
Internal power MOSFETs — 10A
Integrated bootstrap switch
Smart power-save protection
Configurable 150mA LDO with bypass capability
TC compensated RDS(ON) sensed current limit
Pseudo-fixed frequency adaptive on-time control
Designed for use with ceramic capacitors
Programmable VIN UVLO threshold
Independent enable for switcher and LDO
Selectable ultra-sonic power-save (SC417)
Selectable power-save (SC427)
Internal soft-start and soft-shutdown at output
Internal reference — 1% tolerance
Over-voltage/under-voltage fault protection
Power good output
SmartDriveTM
Lead-free 5x5mm, 32 Pin MLPQ package
Fully WEEE and RoHS compliant
Applications
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Notebook, desktop, tablet, and server computers
Networking and telecommunication equipment
Printers, DSL, and STB applications
Embedded applications
Power supply modules
Point of load power supplies
The SC417/SC427 supports using standard capacitor types
such as electrolytic or special polymer, in addition to
ceramic, at switching frequencies up to 1MHz. The programmable frequency, synchronous operation, and selectable power-save provide high efficiency operation over a
wide load range.
The LDO output is programmable from 0.75V to 5.25V
using external resistors. The bias voltage for the device
can be supplied by the on-chip LDO when VIN > 4.5V, or by
an external 5V supply. When a separate source is used as
the bias supply, the LDO can be programmed to provide a
different voltage.
Additional features include cycle-by-cycle current limit,
soft-start, under and over-voltage protection, programmable over-current protection, soft shutdown, and selectable power-save. The device also provides separate enable
inputs for the PWM controller and LDO as well as a power
good output for the PWM controller.
The input voltage can range from 3V to 28V. The wide
input voltage range, programmable frequency, and programmable LDO make the device extremely flexible and
easy to use in a broad range of applications. Support is
provided for single cell or multi-cell battery systems in
addition to traditional DC power supply applications.
March 12, 2009
© 2009 Semtech Corporation
1
SC417/SC427
Typical Application Circuit
ENABLE/
PSAVE
ENABLE
LDO
PGOOD
RILIM
7.5KΩ
PAD 1
V5V is tied to VLDO
1
2
3
4
5
6
7
8
RLDO1
56.2KΩ
RLDO2
SC417/SC427
PAD 2
CBST
0.1μF
VIN
+12V
VIN
1μF
LX
LX
PGND
PGND
PGND
PGND
PGND
PGND
LX
PAD 3
RGND
CIN
22μF
0
1.05V @ 10A, 250kHz
L1
0.88μH
CFF
100pF
24
23
22
21
20
19
18
17
9
10
11
12
13
14
15
16
100nF
FB
FBL
V5V
AGND
VOUT
VIN
VLDO
BST
VIN
10KΩ
AGND
VIN
VIN
DH
LXBST
DL
PGND
PGND
Note:
ENL
TON
AGND
EN/PSV
LXS
ILIM
PGOOD
LX
32
31
30
29
28
27
26
25
RTON
154KΩ
COUT1
220μF
15mΩ
+
COUT2
220μF
15mΩ
+
VOUT
10nF
RFB1
11KΩ
RFB2
10KΩ
Key Components
Component
Value
Manufacturer
Part Number
CIN
22μF/25V
Murata
GRM32ER61E226KE15L
www.murata.com
COUT1, COUT2
220μF/15mΩ/6.3V
Panasonic
EEFUE0J221R
www.panasonic.com
L1
0.88μH/20A
Vishay
IHLP4040DZERR88M11
www.vishay.com
Web
All other small signal components (resistors and capacitors) are standard SMT devices.
2
SC417/SC427
VOUT
5
VIN
6
VLDO
7
BST
8
EN/PSV
ILIM
PGOOD
LX
AGND
LXS
TON
26
25
AGND
PAD 1
LX
PAD 3
VIN
PAD 2
VIN
9
10
11
12
13
14
15
16
PGND
4
27
PGND
AGND
28
DL
3
29
Top View
DH
V5V
30
LXBST
2
31
VIN
FBL
32
Ordering Information
VIN
FB
1
ENL
Pin Configuration
24
LX
23
LX
22
PGND
21
PGND
20
PGND
19
PGND
18
PGND
17
PGND
Device
Package
SC417MLTRT(1)(2)
MLPQ-32 5X5
SC427MLTRT(1)(2)
MLPQ-32 5X5
SC417EVB
Evaluation Board
SC427EVB
Evaluation Board
Notes:
1) Available in tape and reel only. A reel contains 3000 devices.
2) Lead-free packaging only. Device is WEEE and RoHS compliant.
MLPQ-32; 5x5, 32 LEAD
Marking Information
SC417
yyww
xxxxxx
xxxxxx
SC427
yyww
xxxxxx
xxxxxx
yyww = Date Code
xxxxxx = Semtech Lot Number
xxxxxx = Semtech Lot Number
yyww = Date Code
xxxxxx = Semtech Lot Number
xxxxxx = Semtech Lot Number
3
SC417/SC427
Absolute Maximum Ratings
Recommended Operating Conditions
LX to PGND (V). . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +30
Input Voltage (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.0 to 28
LX to PGND (V) (transient — 100ns max.) . . . . . . -2 to +30
V5V to PGND (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5 to 5.5
VIN to PGND (V). . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +30
VOUT to PGND (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.5 to 5.5
EN/PSV, PGOOD, ILIM, to GND (V) . . . . . . -0.3 to +(V5V + 0.3)
Thermal Information
VOUT, VLDO, FB, FBL, to GND (V) . . . . . . . -0.3 to +(V5V + 0.3)
V5V to PGND (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +6
Storage Temperature (°C) . . . . . . . . . . . . . . . . . . . . -60 to +150
TON to PGND (V) . . . . . . . . . . . . . . . . . . . . . -0.3 to +(V5V - 1.5)
Maximum Junction Temperature (°C) . . . . . . . . . . . . . . . 150
ENL (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to VIN
Operating Junction Temperature (°C) . . . . . . -40 to +125
BST to LX (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +6.0
Thermal resistance, junction to ambient (2) (°C/W)
BST to PGND (V) . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +35
High-side MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
AGND to PGND (V) . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 to +0.3
Low-side MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
ESD Protection Level(1) (kV) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
PWM controller and LDO thermal resistance . . . . . 50
Peak IR Reflow Temperature (°C) . . . . . . . . . . . . . . . . . . . . 260
Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters
specified in the Electrical Characteristics section is not recommended.
NOTES:
(1) Tested according to JEDEC standard JESD22-A114.
(2) Calculated from package in still air, mounted to 3 x 4.5 (in), 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards.
Electrical Characteristics
Unless specified: VIN =12V, TA = +25°C for Typ, -40 to +85 °C for Min and Max, TJ < 125°C, V5V = +5V, Typical Application Circuit
Parameter
Conditions
Min
Typ
Max
Units
3
28
V
4.5
5.5
V
Input Supplies
Input Supply Voltage
V5V Voltage
Sensed at ENL pin, rising edge
2.40
2.60
2.95
Sensed at ENL pin, falling edge
2.235
2.40
2.565
VIN UVLO Threshold(1)
VIN UVLO Hysteresis
V
EN/PSV = High
0.2
V
Measured at V5V pin, rising edge
3.7
3.9
4.1
Measured at V5V pin, falling edge
3.5
3.6
3.75
V5V UVLO Threshold
V
V5V UVLO Hysteresis
0.3
ENL , EN/PSV = 0V, VIN = 28V
8.5
Standby mode; ENL=V5V, EN/PSV = 0V
130
V
20
μA
VIN Supply Current
4
SC417/SC427
Electrical Characteristics (continued)
Parameter
Conditions
Min
Typ
Max
Units
ENL , EN/PSV = 0V
3
7
μA
SC417, EN/PSV = V5V, no load (fSW = 25kHz),
VFB > 500mV(2)
2
SC427, EN/PSV = V5V, no load, VFB > 500mV(2)
0.7
fSW = 250kHz, EN/PSV = floating , no load(2)
10
Input Supplies (continued)
V5V Supply Current
Static VIN and load, 0 to +85 °C
0.496
Static VIN and load, -40 to +85 °C
Continuous mode operation
0.500
mA
0.504
V
0.495
0.505
V
200
1000
FB On-Time Threshold
Frequency Range
kHz
Minimum fSW, (SC417 only), EN/PSV = V5V, no load
25
Bootstrap Switch Resistance
10
Ω
Timing
On-Time
Continuous mode operation,
VIN = 15V, VOUT = 5V, fSW= 300kHz, RTON = 133kΩ
999
1110
1220
ns
Minimum On-Time (2)
80
ns
Minimum Off-Time (2)
250
ns
850
μs
500
kΩ
Soft-Start
Soft-Start Ramp Time (2)
Analog Inputs/Outputs
VOUT Input Resistance
Current Sense
Zero-Crossing Detector Threshold
LX - PGND
-3
0
+3
mV
Power Good
Upper limit, VFB > internal 500mV reference
+20
%
Lower limit, VFB < internal 500mV reference
-10
%
Start-Up Delay Time
2
ms
Fault (noise immunity) Delay Time(2)
5
μs
Power Good Threshold
Leakage
Power Good On-Resistance
1
10
μA
Ω
5
SC417/SC427
Electrical Characteristics (continued)
Parameter
Conditions
Min
Typ
Max
Units
RILIM = 5.9k Ω
6
8
10
A
Fault Protection
Valley Current Limit
ILIM Source Current
ILIM Comparator Offset
10
With respect to AGND
-10
0
μA
+10
mV
Output Under-Voltage Fault
VFB with respect to internal 500mV reference,
8 consecutive clocks
-25
%
Smart Power-save Protection Threshold (2)
VFB with respect to internal 500mV reference
+10
%
Over-Voltage Protection Threshold
VFB with respect to internal 500mV reference
+20
%
5
μs
150
°C
Over-Voltage Fault Delay(2)
Over-Temperature Shutdown(2)
10°C hysteresis
Logic Inputs/Outputs
Logic Input High Voltage
ENL
Logic Input Low Voltage
ENL
2.0
V
0.4
V
2.2
5
V
EN/PSV Input for Forced Continuous Operation (2)
1
2
V
EN/PSV Input for Disabling Switcher (2)
0
0.8
V
-10
+10
μA
18
μA
+1
μA
EN/PSV Input for PSAVE Operation (2)
EN/PSV Input Bias Current
ENL Input Bias Current
FBL, FB Input Bias Current
V5V = 5V
EN/PSV= V5V or AGND
VIN = 28V
FBL, FB = V5V or AGND
11
-1
6
SC417/SC427
Electrical Characteristics (continued)
Parameter
Conditions
Min
Typ
Max
Units
VLDO load = 10mA
0.735
0.75
0.765
V
Linear Regulator (LDO)
FBL Accuracy
Start-up and foldback, VIN = 12V
85
mA
LDO Current Limit
Operating current limit, VIN = 12V
135
200
VLDO to VOUT Switch-over Threshold (3)
-140
+140
mV
VLDO to VOUT Non-switch-over Threshold (3)
-450
+450
mV
VLDO to VOUT Switch-over Resistance
LDO Drop Out Voltage (4)
VOUT = +5V
2
Ω
From VIN to VVLDO, VVLDO = +5V, IVLDO = 100mA
1.2
V
Notes:
(1) VIN UVLO is programmable using a resistor divider from VIN to ENL to AGND. The ENL voltage is compared to an internal reference.
(2) Guaranteed by design.
(3) The switch-over threshold is the maximum voltage differential between the VLDO and VOUT pins which ensures that VLDO will internally
switch-over to VOUT. The non-switch-over threshold is the minimum voltage differential between the VLDO and VOUT pins which ensures
that VLDO will not switch-over to VOUT.
(4) The LDO drop out voltage is the voltage at which the LDO output drops 2% below the nominal regulation point.
7
SC417/SC427
Typical Characteristics
Characteristics in this section are based on using the Typical Application Circuit on page 2 (SC417/SC427).
Efficiency vs. Load — Forced Continuous Mode
VOUT vs. Load — Forced Continuous Mode
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
100
1.100
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
90
85%
1.075
Efficiency (%)
+1%
VOUT (V)
80
70
1.050
-1%
1.025
60
50
0.0
1.0
2.0
3.0
4.0
5.0
6.0
IOUT (A)
7.0
8.0
9.0
10.0
1.000
0.0
Externally biased at VIN = 12V, V5V = 5V, VOUT = 1.050V
100
90
3.0
4.0
5.0
6.0
IOUT (A)
7.0
8.0
9.0
10.0
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
90
85%
85%
Efficiency (%)
Efficiency (%)
2.0
Efficiency vs. Load — Powersave Mode (SC417)
Efficiency vs. Load — Powersave Mode (SC417)
100
1.0
80
70
80
70
60
60
50
0.10
1.00
IOUT (A)
50
10.00
Efficiency vs. Load — Powersave Mode (SC427)
100
Externally biased at VIN = 12V, V5V = 5V, VOUT = 1.050V
0.10
1.00
IOUT (A)
10.00
Efficiency vs. Load — Powersave Mode (SC427)
100
90
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
90
Efficiency (%)
Efficiency (%)
85%
85%
80
70
60
50
0.1
80
70
60
50
1.0
IOUT (A)
10.0
0.1
1.0
IOUT (A)
10.0
8
SC417/SC427
Typical Characteristics (continued)
Characteristics in this section are based on using the Typical Application Circuit on page 2 (SC417/SC427).
VRIPPLE vs. Load — Forced Continuous Mode
Frequency vs. Load — Forced Continuous Mode
400
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
0.20
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
350
VRIPPLE (VP-P)
Freq (kHz)
0.15
+15%
300
250
-15%
200
0.10
50mV
0.05
150
VOUTP-P
100
0.0
1.0
2.0
3.0
4.0
5.0
6.0
IOUT (A)
7.0
8.0
9.0
0.00
0.0
10.0
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
1.075
2.0
3.0
4.0
5.0
6.0
IOUT (A)
7.0
8.0
9.0
10.0
VOUT vs. Load — Powersave Mode (SC427)
VOUT vs. Load — Powersave Mode (SC417)
1.100
1.0
1.100
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
1.050
+1%
VOUT (V)
VOUT (V)
+1%
1.050
-1%
1.000
-1%
1.025
0.950
1.000
0.10
10.00
1.00
IOUT (A)
0.900
0.1
1.0
IOUT (A)
10.0
VOUT vs. Line — Forced Continuous Mode
1.100
Internally biased at VLDO = 5V, VIN = 12V, VOUT = 1.050V
1.075
VOUT (V)
+1%
1.050
-1%
1.025
1.000
6.0
8.0
10.0
12.0
14.0
16.0
VIN (V)
18.0
20.0
22.0
24.0
9
SC417/SC427
Typical Characteristics (continued)
Characteristics in this section are based on using the Typical Application Circuit on page 2 (SC417/SC427).
Ultrasonic Powersave Mode — No Load (SC417)
Powersave Mode — No Load (SC427)
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = EN/PSV= ENL = 5V
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = EN/PSV= ENL = 5V
Δ V ~ 29mV
1.073V
(50mV/div)
(50mV/div)
1.044V
(10V/div)
f=26.22kHz
(10V/div)
(5V/div)
(5V/div)
Time (10ms/div)
Time (10μs/div)
Forced Continuous Mode — No Load
Self-Biased Start-Up — Power Good True
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = ENL = 5V, EN/PSV= float
VIN = 0V to 12V step, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = EN/PSV= ENL = 5V
1.078V
Δ V ~ 30mV
(50mV/div)
(10V/div)
1.048V
1.05V
f=227.4kHz
(10V/div)
ΔV/ΔT ~
(500mV/div)
(2V/div)
1.4 V/ms
0V
(10V/div)
(5V/div)
(5V/div)
Time (2μs/div)
Time (400μs/div)
Enabled Loaded Output — Full Scale
Enabled Loaded Output — Power Good True
VIN = 12V, VOUT = 1.05V, IOUT = 1A, VLDO = V5V = ENL = 5V. EN/PSV= 5V
VIN = 12V, VOUT = 1.05V, IOUT = 1A, VLDO = V5V = ENL = 5V. EN/PSV= 5V
1.05V
1.05V
ΔV/ΔT ~
(50mV/div)
0V
1.4V/ms
(500mV/div)
(10V/div)
(10V/div)
(5V/div)
(5V/div)
0V
~2ms
Time (100μs/div)
Time (400μs/div)
10
SC417/SC427
Typical Characteristics (continued)
Characteristics in this section are based on using the Typical Application Circuit on page 2 (SC417/SC427).
Transient Response — Load Rising (SC417)
VIN = 12V, VOUT = 1.05V, IOUT = 0A to 10A, VLDO = V5V = EN/PSV= ENL = 5V
Transient Response — Load Falling (SC417)
VIN = 12V, VOUT = 1.05V, IOUT = 10A to 0A, VLDO = V5V = EN/PSV= ENL = 5V
1.101V
1.063V
(50mV/div)
(50mV/div)
1.055V
1.025V
(10V/div)
(10V/div)
(10A/div)
(10A/div)
(5V/div)
(5V/div)
Time (10μs/div)
Time (10μs/div)
Transient Response — Load Rising (SC427)
VIN = 12V, VOUT = 1.05V, IOUT = 0A to 10A, VLDO = V5V = EN/PSV= ENL = 5V
Transient Response — Load Falling (SC427)
VIN = 12V, VOUT = 1.05V, IOUT = 10A to 0A, VLDO = V5V = EN/PSV= ENL = 5V
(50mV/div)
(50mV/div)
(10V/div)
(10V/div)
(5A/div)
(5A/div)
(5V/div)
(5V/div)
Time (10μs/div)
Time (10μs/div)
Output Under-voltage Response — Normal Operation
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = ENL = 5V, floating EN/PSV
Output Over-current Response — Normal Operation
VIN = 12V, VOUT = 1.05V, VLDO = V5V = ENL = 5V, EN/PSV= floating; IOUT ramped to trip point
(500mV/div)
1.05V
(10V/div)
0V
(500mV/div)
V2~710mV
(10A/div)
IOUT = 10.37A
(10V/div)
(5V/div)
(5V/div)
Time (100μs/div)
Time (100μs/div)
11
SC417/SC427
Typical Characteristics (continued)
Characteristics in this section are based on using the Typical Application Circuit on page 2 (SC417/SC427).
Shorted Output Response — Normal Operation
Shorted Output Response — Power-UP Operation
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = EN/PSV= ENL = 5V
VIN = 12V, VOUT = 1.05V, IOUT = 0A, VLDO = V5V = EN/PSV= ENL = 5V
(500mV/div)
1.05V
~1.7ms
(500mV/div)
0V
(10A/div)
(10A/div)
(10V/div)
(10V/div)
(5V/div)
(5V/div)
Time (40μs/div)
Time (400μs/div)
12
SC417/SC427
Pin Descriptions
Pin #
Pin Name
Pin Function
1
FB
Feedback input for switching regulator used to program the output voltage — connect to an external resistor divider from VOUT to AGND.
2
FBL
Feedback input for the LDO — connect to an external resistor divider from VLDO to AGND — used to program the LDO output.
3
V5V
5V power input for internal analog circuits and gate drives — connect to external 5V supply or configure
the LDO for 5V and connect to VLDO.
4, 30, PAD 1
AGND
Analog ground
5
VOUT
Switcher output voltage sense pin — also the input to the internal switch-over between VOUT and VLDO.
The voltage at this pin must be less than or equal to the voltage at the V5V pin.
6, 9-11,
PAD 2
VIN
7
VLDO
8
BST
Bootstrap pin — connect a capacitor of at least 100nF from BST to LX to develop the floating supply for the
high-side gate drive.
12
DH
High-side gate drive — do not connect this pin
13
LXBST
23-25, PAD 3
LX
Switching (phase) node
14
DL
Low-side gate drive — do not connect this pin
15-22
PGND
26
PGOOD
27
ILIM
Current limit sense pin — used to program the current limit by connecting a resistor from ILIM to LX.
28
LXS
LX sense — connects to RILIM.
Input supply voltage
LDO output — The voltage at this pin must be less than or equal to the voltage at the V5V pin.
LX Boost — connect to the BST capacitor.
Power ground
Open-drain power good indicator — high impedance indicates power is good. An external pull-up
resistor is required.
Enable/power-save input for the switching regulator — connect to AGND to disable the switching regulator.
Float to operate in forced continuous mode (power-save disabled). SC417 — connect to V5V to operate with
ultra-sonic power-save mode enabled. SC427 — connect to V5V to operate with power-save mode enabled
with no minimum frequency.
29
EN/PSV
31
TON
On-time programming input — set the on-time by connecting through a resistor to AGND
32
ENL
Enable input for the LDO — connect ENL to AGND to disable the LDO. Drive with logic to +3V for logic control, or program the VIN UVLO with a resistor divider between VIN, ENL, and AGND.
13
SC417/SC427
Block Diagram
V5V
3
VIN
EN/PSV
PGOOD
26
A
29
V5V
VIN
V5V
Bootstrap
Switch
AGND
D
Control & Status
Reference
8
BST
12
DH
B
LX
13
LXBST
28
LXS
C
PGND
27
ILIM
14
DL
DL
Hi-side
MOSFET
Soft Start
FB
Gate Drive
Control
On-- time
Generator
1
V5V
FB Comparator
TON
31
Zero Cross Detector
VOUT
5
Lo-side
MOSFET
Bypass Comparator
Valley Current Limit
A
VLDO
7
VIN
Y
B
LDO
VLDO Switchover MUX
FBL
2
32
ENL
A = connected to pins 6, 9-11, PAD 2
B = connected to pins 23-25, PAD 3
C = connected to pins 15-22
D = connect to pins 4, 30, PAD 1
14
SC417/SC427
Applications Information
Synchronous Buck Converter
The SC417/SC427 is a step down synchronous DC-DC buck
converter with integrated power MOSFETs and a programmable LDO. The device is capable of 10A operation at very
high efficiency. A space saving 5x5 (mm) 32-pin package
is used. The programmable operating frequency range of
200kHz to 1MHz enables optimizing the configuration for
PCB area and efficiency.
The buck controller uses a pseudo-fixed frequency adaptive on-time control. This control method allows fast transient response which permits the use of smaller output
capacitors.
The SC417/SC427 requires two input supplies for normal
operation: VIN and V5V. VIN operates over the wide range
from 3V to 28V. V5V requires a 5V supply input that can be
an external source or the internal LDO configured to
supply 5V.
Psuedo-fixed Frequency Adaptive On-time Control
The PWM control method used by the SC417/SC427 is
pseudo-fixed frequency, adaptive on-time, as shown in
Figure 1. The ripple voltage generated at the output
capacitor ESR is used as a PWM ramp signal. This ripple is
used to trigger the on-time of the controller.
TON
VLX
CIN
Q1
VFB
VLX
FB Threshold
VOUT
L
Q2
The advantages of adaptive on-time control are:
•
•
Input Voltage Requirements
VIN
The adaptive on-time is determined by an internal oneshot timer. When the one-shot is triggered by the output
ripple, the device sends a single on-time pulse to the highside MOSFET. The pulse period is determined by VOUT and
VIN; the period is proportional to output voltage and
inversely proportional to input voltage. With this adaptive
on-time arrangement, the device automatically anticipates the on-time needed to regulate VOUT for the present
VIN condition and at the selected frequency.
ESR
•
•
•
Predictable operating frequency compared to
other variable frequency methods.
Reduced component count by eliminating the
error amplifier and compensation components.
Reduced component count by removing the
need to sense and control inductor current.
Fast transient response — the response time is
controlled by a fast comparator instead of a typically slow error amplifier.
Reduced output capacitance due to fast transient response
One-Shot Timer and Operating Frequency
The one-shot timer operates as shown in Figure 2. The FB
Comparator output goes high when VFB is less than the
internal 500mV reference. This feeds into the gate drive
and turns on the high-side MOSFET, and also starts the
one-shot timer. The one-shot timer uses an internal comparator and a capacitor. One comparator input is connected to V OUT, the other input is connected to the
capacitor. When the on-time begins, the internal capacitor charges from zero volts through a current which is
proportional to VIN. When the capacitor voltage reaches
VOUT, the on-time is completed and the high-side MOSFET
turns off.
FB
+
COUT
Figure 1 — PWM Control Method, VOUT Ripple
15
SC417/SC427
Applications Information (continued)
FB Comparator
FB
500mV +
VOUT
VIN
RTON
Gate
Drives
One-Shot
Timer
VIN
DH
Q1
VLX
DL
Q2
VOUT
L
ESR
COUT
FB
+
Note that this control method regulates the valley of the
output ripple voltage, not the DC value. The DC output
voltage VOUT is offset by the output ripple according to the
following equation.
VOUT
When a large capacitor is placed in parallel with R1 (C TOP)
VOUT is shown by the following equation.
On-time = K x RTON x (VOUT/VIN)
Figure 2 — On-Time Generation
This method automatically produces an on-time that is
proportional to VOUT and inversely proportional to VIN.
Under steady-state conditions, the switching frequency
can be determined from the on-time by the following
equation.
VOUT
TON u VIN
fSW
The SC417/SC427 uses an external resistor to set the ontime which indirectly sets the frequency. The on-time can
be programmed to provide operating frequency from
200kHz to 1MHz using a resistor between the TON pin and
ground. The resistor value is selected by the following
equation.
RTON
(TON 10ns) uV IN
25pF uV OUT
The maximum RTON value allowed is shown by the following equation.
RTON _ MAX
VIN _ MIN
15PA
VOUT Voltage Selection
The switcher output voltage is regulated by comparing
VOUT as seen through a resistor divider at the FB pin to the
internal 500mV reference voltage, see Figure 3.
VOUT
§ R · §V
·
0.5 u ¨¨1 1 ¸¸ ¨ RIPPLE ¸
R
2
©
¹
2 ¹
©
To FB pin
R1
R2
Figure 3 — Output Voltage Selection
VOUT
§ R · §V
·
0.5 u ¨¨1 1 ¸¸ ¨ RIPPLE ¸ u
© R2 ¹ © 2 ¹
1 (R1ZCTOP )2
§ R u R1
·
1 ¨¨ 2
ZCTOP ¸¸
© R2 R1
¹
2
Enable and Power-save Inputs
The EN/PSV and ENL inputs are used to enable or disable
the switching regulator and the LDO. When EN/PSV is low
(grounded), the switching regulator is off and in its lowest
power state. When off, the output of the switching regulator soft-discharges the output into a 15Ω internal resistor
via the VOUT pin. When EN/PSV is allowed to float, the pin
voltage will float to 33% of the voltage at V5V. The switching regulator turns on with power-save disabled and all
switching is in forced continuous mode.
When EN/PSV is high (above 44% of the voltage at V5V)
for SC417, the switching regulator turns on with ultrasonic power-save enabled. The SC417 ultra-sonic powersave operation maintains a minimum switching frequency
of 25kHz, for applications with stringent audio
requirements.
When EN/PSV is high (above 44% of the voltage at V5V)
for SC427, the switching regulator turns on with powersave enabled. The SC427 power-save operation is designed
to maximize efficiency at light loads with no minimum
frequency limits. This makes the SC427 an excellent choice
for portable and battery-operated systems.
The ENL input is used to control the internal LDO. This
input serves a second function by acting as a VIN ULVO
sensor for the switching regulator. When ENL is low
(grounded), the LDO is off. When ENL is a logic high but
below the VIN UVLO threshold (2.6V typical), then the LDO
is on and the switcher is off. When ENL is above the VIN
16
SC417/SC427
Applications Information (continued)
UVLO threshold, the LDO is enabled and the switcher is
also enabled if the EN/PSV pin is not grounded.
The low-side MOSFET remains on until the inductor
current ramps down to zero, at which point the low-side
MOSFET is turned off.
Forced Continuous Mode Operation
The SC417/SC427 operates the switcher in Forced
Continuous Mode (FCM) by floating the EN/PSV pin (see
Figure 4). In this mode one of the power MOSFETs is
always on, with no intentional dead time other than to
avoid cross-conduction. This feature results in uniform
frequency across the full load range with the trade-off
being poor efficiency at light loads due to the high-frequency switching of the MOSFETs.
FB Ripple
Voltage (VFB)
FB threshold
(500mV)
Because the on-times are forced to occur at intervals no
greater than 40μs, the frequency will not fall below
~25kHz. Figure 5 shows ultra-sonic power-save
operation.
minimum fSW ~ 25kHz
FB Ripple
Voltage (VFB)
FB threshold
(500mV)
On-time
(TON)
DC Load Current
Inductor
Current
(0A)
Inductor
Current
DH On-time is triggered when
VFB reaches the FB Threshold
DH
40μs time-out
On-time
(TON)
DH on-time is triggered when
VFB reaches the FB Threshold.
DL
After the 40μsec time-out, DL drives high if VFB
has not reached the FB threshold.
DH
Figure 5 — Ultrasonic Power-save Operation
DL
Power-save Mode Operation (SC427)
DL drives high when on-time is completed.
DL remains high until VFB falls to the FB threshold.
Figure 4 — Forced Continuous Mode Operation
Ultra-sonic Power-save Operation (SC417)
The SC417 provides ultra-sonic power-save operation at
light loads, with the minimum operating frequency fixed
at 25kHz. This is accomplished using an internal timer that
monitors the time between consecutive high-side gate
pulses. If the time exceeds 40μs, DL drives high to turn the
low-side MOSFET on. This draws current from VOUT through
the inductor, forcing both VOUT and VFB to fall. When VFB
drops to the 500mV threshold, the next DH on-time is triggered. After the on-time is completed the high-side
MOSFET is turned off and the low-side MOSFET turns on.
The SC427 provides power-save operation at light loads
with no minimum operating frequency. With power-save
enabled, the internal zero crossing comparator monitors
the inductor current via the voltage across the low-side
MOSFET during the off-time. If the inductor current falls to
zero for 8 consecutive switching cycles, the controller
enters power-save operation. It will turn off the low-side
MOSFET on each subsequent cycle provided that the
current crosses zero. At this time both MOSFETs remain
off until VFB drops to the 500mV threshold. Because the
MOSFETs are off, the load is supplied by the output capacitor. If the inductor current does not reach zero on any
switching cycle, the controller immediately exits powersave and returns to forced continuous mode. Figure 6
shows power-save operation at light loads.
17
SC417/SC427
Applications Information (continued)
FB Ripple
Voltage
(VFB)
Dead time varies
according to load
FB threshold
(500mV)
VOUT drifts up to due to leakage
current flowing into COUT
Smart Power Save
Threshold (550mV)
VOUT discharges via inductor
and low-side MOSFET
Normal VOUT ripple
FB
threshold
DH and DL off
Inductor
Current
Zero (0A)
High-side
Drive (DH)
Single DH on-time pulse
after DL turn-off
On-time (TON)
DH On-time is triggered when
VFB reaches the FB Threshold.
DH
Low-side
Drive (DL)
DL turns on when Smart
PSAVE threshold is reached
Normal DL pulse after DH
on-time pulse
DL turns off when FB
threshold is reached
DL
Figure 7 — Smart Power-save
DL drives high when on-time is completed.
DL remains high until inductor current reaches zero.
Figure 6 — Power-save Operation
Smart Power-save Protection
Current Limit Protection
Active loads may leak current from a higher voltage into
the switcher output. Under light load conditions with
power-save enabled, this can force VOUT to slowly rise and
reach the over-voltage threshold, resulting in a hard shutdown. Smart power-save prevents this condition. When
the FB voltage exceeds 10% above nominal (exceeds
550mV), the device immediately disables power-save, and
DL drives high to turn on the low-side MOSFET. This draws
current from VOUT through the inductor and causes VOUT to
fall. When VFB drops back to the 500mV trip point, a normal
TON switching cycle begins. This method prevents a hard
OVP shutdown and also cycles energy from VOUT back to
VIN. It also minimizes operating power by avoiding forced
conduction mode operation. Figure 7 shows typical waveforms for the Smart Power-save feature.
The device features programmable current limiting, which
is accomplished by using the RDSON of the lower MOSFET
for current sensing. The current limit is set by RILIM resistor.
The RILIM resistor connects from the ILIM pin to the LX pin
which is also the drain of the low-side MOSFET. When the
low-side MOSFET is on, an internal ~10μA current flows
from the ILIM pin and through the RILIM resistor, creating a
voltage drop across the resistor. While the low-side
MOSFET is on, the inductor current flows through it and
creates a voltage across the RDSON. The voltage across the
MOSFET is negative with respect to ground. If this MOSFET
voltage drop exceeds the voltage across RILIM, the voltage
at the ILIM pin will be negative and current limit will activate. The current limit then keeps the low-side MOSFET on
and will not allow another high-side on-time, until the
current in the low-side MOSFET reduces enough to bring
the ILIM voltage back up to zero. This method regulates
the inductor valley current at the level shown by ILIM in
Figure 8.
SmartDriveTM
The DH drivers will turn on the high-side MOSFET at a
lower rate initially, allowing a softer, smooth turn-off of
the low-side diode. Once the diode is off, the SmartDrive
circuit automatically drives the high-side MOSFET on at a
rapid rate. This technique reduces switching while maintaining high efficiency and also avoids the need for snubbers or series resistors in the gate drive.
18
SC417/SC427
Inductor Current
Applications Information (continued)
IPEAK
ILOAD
ILIM
During soft-start the regulator turns off the low-side
MOSFET on any cycle if the inductor current falls to zero.
This prevents negative inductor current, allowing the
device to start into a pre-biased output.
Power Good Output
Time
Figure 8 — Valley Current Limit
Setting the valley current limit to 10A results in a peak
inductor current of 10A plus peak ripple current. In this
situation, the average (load) current through the inductor
is 10A plus one-half the peak-to-peak ripple current.
The internal 10μA current source is temperature compensated at 4100ppm in order to provide tracking with the
RDSON.
The RILIM value is calculated by the following equation.
RILIM = 735 x ILIM
When selecting a value for RILIM be sure not to exceed the
absolute maximum voltage value for the ILIM pin. Note
that because the low-side MOSFET with low RDSON is used
for current sensing, the PCB layout, solder connections,
and PCB connection to the LX node must be done carefully to obtain good results. RILIM should be connected
directly to LXS (pin 28).
Soft-Start of PWM Regulator
Soft-start is achieved in the PWM regulator by using an
internal voltage ramp as the reference for the FB
Comparator. The voltage ramp is generated using an
internal charge pump which drives the reference from
zero to 500mV in ~1.2mV increments, using an internal
~500kHz oscillator. When the ramp voltage reaches
500mV, the ramp is ignored and the FB comparator
switches over to a fixed 500mV threshold. During soft-start
the output voltage tracks the internal ramp, which limits
the start-up inrush current and provides a controlled softstart profile for a wide range of applications. Typical softstart ramp time is 850μs.
The power good (PGOOD) output is an open-drain output
which requires a pull-up resistor. When the output voltage
is 10% below the nominal voltage, PGOOD is pulled low. It
is held low until the output voltage returns above -8% of
nominal. PGOOD is held low during start-up and will not
be allowed to transition high until soft-start is completed
(when VFB reaches 500mV) and typically 2ms has passed.
PGOOD will transition low if the VFB pin exceeds +20% of
nominal, which is also the over-voltage shutdown threshold (600mV). PGOOD also pulls low if the EN/PSV pin is
low when V5V is present.
Output Over-Voltage Protection
Over-voltage protection becomes active as soon as the
device is enabled. The threshold is set at 500mV + 20%
(600mV). When VFB exceeds the OVP threshold, DL latches
high and the low-side MOSFET is turned on. DL remains
high and the controller remains off, until the EN/PSV input
is toggled or V5V is cycled. There is a 5μs delay built into
the OVP detector to prevent false transitions. PGOOD is
also low after an OVP event.
Output Under-Voltage Protection
When VFB falls 25% below its nominal voltage (falls to
375mV) for eight consecutive clock cycles, the switcher is
shut off and the DH and DL drives are pulled low to tristate the MOSFETs. The controller stays off until EN/PSV is
toggled or V5V is cycled.
V5V UVLO, and POR
Under-Voltage Lock-Out (UVLO) circuitry inhibits switching and tri-states the DH/DL drivers until V5V rises above
3.9V. An internal Power-On Reset (POR) occurs when V5V
exceeds 3.9V, which resets the fault latch and soft-start
counter to prepare for soft-start. The SC417/SC427 then
begins a soft-start cycle. The PWM will shut off if V5V falls
below 3.6V.
19
SC417/SC427
Applications Information (continued)
LDO Regulator
VVLDO Final
The device features an integrated LDO regulator with a
programmable output voltage from 0.75V to 5.25V using
external resistors. The feedback pin (FBL) for the LDO is
regulated to 750mV. There is also an enable pin (ENL) for
the LDO that provides independent control. The LDO
voltage can also be used to provide the bias voltage for
the switching regulator.
VLDO
Constant current startup
Figure 10 — LDO Start-Up
To FBL pin
RLDO1
LDO Switch-Over Operation
RLDO2
Figure 9 — LDO Start-Up
The LDO output voltage is set by the following equation.
VLDO
Voltage regulating with
~200mA current limit
90% of VVLDO Final
·
§
R
750mV u ¨¨1 LDO1 ¸¸
R
LDO 2 ¹
©
A minimum capacitance of 1μF referenced to AGND is
normally required at the output of the LDO for stability. If
the LDO is providing bias power to the device, then a
minimum 0.1μF capacitor referenced to AGND is required
along with a minimum 1.0μF capacitor referenced to
PGND to filter the gate drive pulses. Refer to the layout
guidelines section.
LDO Start-up
Before start-up, the LDO checks the status of the following
signals to ensure proper operation can be maintained.
1. ENL pin
2. VLDO output
3. VIN input voltage
When the ENL pin is high and VIN is above the UVLO point,
the LDO will begin start-up. During the initial phase, when
the LDO output voltage is near zero, the LDO initiates a
current-limited start-up (typically 85mA) to charge the
output capacitor. When VLDO has reached 90% of the final
value (as sensed at the FBL pin), the LDO current limit is
increased to ~200mA and the LDO output is quickly driven
to the nominal value by the internal LDO regulator.
The SC417/SC427 includes a switch-over function for the
LDO. The switch-over function is designed to increase
efficiency by using the more efficient DC-DC converter to
power the LDO output, avoiding the less efficient LDO
regulator when possible. The switch-over function connects the VLDO pin directly to the VOUT pin using an
internal switch. When the switch-over is complete the
LDO is turned off, which results in a power savings and
maximizes efficiency. If the LDO output is used to bias the
SC417/SC427, then after switch-over the device is selfpowered from the switching regulator with the LDO
turned off.
The switch-over logic waits for 32 switching cycles before
it starts the switch-over. There are two methods that
determine the switch-over of VLDO to VOUT.
In the first method, the LDO is already in regulation and
the DC-DC converter is later enabled. As soon as the
PGOOD output goes high, the 32 cycles are started. The
voltages at the VLDO and VOUT pins are then compared;
if the two voltages are within ±300mV of each other, the
VLDO pin connects to the VOUT pin using an internal
switch, and the LDO is turned off.
In the second method, the DC-DC converter is already
running and the LDO is enabled. In this case the 32 cycles
are started as soon as the LDO reaches 90% of its final
value. At this time, the VLDO and VOUT pins are compared,
and if within ±300mV the switch-over occurs and the LDO
is turned off.
20
SC417/SC427
Applications Information (continued)
Switch-over Limitations on VOUT and VLDO
ENL pin and VIN UVLO
Because the internal switch-over circuit always compares
the VOUT and VLDO pins at start-up, there are limitations
on permissible combinations of these pins. Consider the
case where VOUT is programmed to 3.0V and VLDO is programmed to 3.3V. After start-up, the device would connect
VOUT to VLDO and disable the LDO, since the two voltages are within the ±300mV switch-over window. To avoid
unwanted switch-over, the minimum difference between
the voltages for VOUT and VLDO should be ±500mV.
The ENL pin also acts as the switcher under-voltage
lockout for the VIN supply. The VIN UVLO voltage is programmable via a resistor divider at the VIN, ENL and AGND
pins.
It is not recommended to use the switch-over feature for
an output voltage less than 3V since this does not provide
sufficient voltage for the gate-source drive to the internal
p-channel switch-over MOSFET.
Switch-over MOSFET Parasitic Diodes
The switch-over MOSFET contains parasitic diodes that
are inherent to its construction, as shown in Figure 11.
Switchover
control
Switchover
MOSFET
VOUT
VLDO
ENL is the enable/disable signal for the LDO. In order to
implement the VIN UVLO there is also a timing requirement that needs to be satisfied.
If the ENL pin transitions low within 2 switching cycles and
is < 1V, then the LDO will turn off but the switcher remains
on. If ENL goes below the VIN UVLO threshold and stays
above 1V, then the switcher will turn off but the LDO
remains on.
The VIN UVLO function has a typical threshold of 2.6V on
the VIN rising edge. The falling edge threshold is 2.4V.
Note that it is possible to operate the switcher with the
LDO disabled, but the ENL pin must be below the logic
low threshold (0.4V maximum).
ENL Logic Control of PWM Operation
Parasitic diode
Parasitic diode
V5V
Figure 11— Switch-over MOSFET Parasitic Diodes
There are some important design rules that must be followed to prevent forward bias of these diodes. The following two conditions need to be satisfied in order for the
parasitic diodes to stay off.
•
•
V5V ≥ VLDO
V5V ≥ VOUT
If either VLDO or VOUT is higher than V5V, then the respective
diode will turn on and the SC417/SC427 operating current
will flow through this diode. This has the potential of
damaging the device.
When the ENL input is driven above 2.6V, it is impossible
to determine if the LDO output is going to be used to
power the device or not. In self-powered operation where
the LDO will power the device, it is necessary during the
LDO start-up to hold the PWM switching off until the LDO
has reached 90% of the final value. This prevents overloading the current-limited LDO output during the LDO
start-up. However, if the switcher was previously operating (with EN/PSV high but ENL at ground, and V5V supplied externally), then it is undesirable to shut down the
switcher.
To prevent this, when the ENL input is taken above 2.6V
(above the VIN UVLO threshold), the internal logic checks
the PGOOD signal. If PGOOD is high, then the switcher is
already running and the LDO will run through the start-up
cycle without affecting the switcher. If PGOOD is low, then
the LDO will not allow any PWM switching until the LDO
output has reached 90% of it’s final value.
21
SC417/SC427
Applications Information (continued)
Using the On-chip LDO to Bias the SC417/SC427
The following steps must be followed when using the onchip LDO to bias the device.
•
•
•
Connect V5V to VLDO before enabling the LDO.
The LDO has an initial current limit of 40mA at
start-up, therefore, do not connect any external
load to VLDO during start-up.
When VLDO reaches 90% of its final value, the
LDO current limit increases to 200mA. At this
time the LDO may be used to supply the required
bias current to the device.
Attempting to operate in self-powered mode in any other
configuration can cause unpredictable results and may
damage the device.
•
•
fSW = 250kHz
Load = 10A maximum
Frequency Selection
Selection of the switching frequency requires making a
trade-off between the size and cost of the external filter
components (inductor and output capacitor) and the
power conversion efficiency.
The desired switching frequency is 250kHz which results
from using components selected for optimum size and
cost .
A resistor (RTON) is used to program the on-time (indirectly
setting the frequency) using the following equation.
RTON
Design Procedure
When designing a switch mode supply the input voltage
range, load current, switching frequency, and inductor
ripple current must be specified.
The maximum input voltage (VINMAX) is the highest specified input voltage. The minimum input voltage ( VINMIN) is
determined by the lowest input voltage after evaluating
the voltage drops due to connectors, fuses, switches, and
PCB traces.
The following parameters define the design.
•
•
•
•
Nominal output voltage (VOUT )
Static or DC output tolerance
Transient response
Maximum load current (IOUT )
There are two values of load current to evaluate — continuous load current and peak load current. Continuous
load current relates to thermal stresses which drive the
selection of the inductor and input capacitors. Peak load
current determines instantaneous component stresses and
filtering requirements such as inductor saturation, output
capacitors, and design of the current limit circuit.
The following values are used in this design.
•
•
VIN = 12V + 10%
VOUT = 1.05V + 4%
(TON 10ns) u VIN
25pF u VOUT
To select RTON, use the maximum value for VIN, and for TON
use the value associated with maximum VIN.
T ON
V OUT
V INMAX u f SW
TON = 318 ns at 13.2VIN, 1.05VOUT, 250kHz
Substituting for RTON results in the following solution.
RTON = 154.9kΩ, use RTON = 154kΩ
Inductor Selection
In order to determine the inductance, the ripple current
must first be defined. Low inductor values result in smaller
size but create higher ripple current which can reduce
efficiency. Higher inductor values will reduce the ripple
current/voltage and for a given DC resistance are more
efficient. However, larger inductance translates directly
into larger packages and higher cost. Cost, size, output
ripple, and efficiency are all used in the selection process.
The ripple current will also set the boundary for powersave operation. The switching will typically enter powersave mode when the load current decreases to 1/2 of the
ripple current. For example, if ripple current is 4A then
Power-save operation will typically start for loads less than
2A. If ripple current is set at 40% of maximum load current,
then power-save will start for loads less than 20% of
maximum current.
22
SC417/SC427
Applications Information (continued)
The inductor value is typically selected to provide a ripple
current that is between 25% to 50% of the maximum load
current. This provides an optimal trade-off between cost,
efficiency, and transient performance.
During the DH on-time, voltage across the inductor is
(VIN - VOUT ). The equation for determining inductance is
shown next.
L
( VIN VOUT ) u TON
IRIPPLE
In this example, the inductor ripple current is set equal to
50% of the maximum load current. Therefore ripple
current will be 50% x 10A or 5A. To find the minimum
inductance needed, use the VIN and TON values that correspond to VINMAX.
(13.2 1.05) u 318ns
5A
0.77PH
A slightly larger value of 0.88μH is selected. This will
decrease the maximum IRIPPLE to 4.4A.
Note that the inductor must be rated for the maximum DC
load current plus 1/2 of the ripple current.
The ripple current under minimum VIN conditions is also
checked using the following equations.
TON _ VINMIN
IRIPPLE
The maximum ripple current of 4.4A creates a ripple
voltage across the ESR. The maximum ESR value allowed
is shown by the following equations.
ESRMAX
Example
L
The design goal is that the output voltage regulation be
±4% under static conditions. The internal 500mV reference tolerance is 1%. Allowing 1% tolerance from the FB
resistor divider, this allows 2% tolerance due to VOUT ripple.
Since this 2% error comes from 1/2 of the ripple voltage,
the allowable ripple is 4%, or 42mV for a 1.05V output.
25pF u RTON u VOUT
10ns
VINMIN
384ns
(10.8 1.05 ) u 384ns
0.88PH
4.25 A
Capacitor Selection
The output capacitors are chosen based on required ESR
and capacitance. The maximum ESR requirement is controlled by the output ripple requirement and the DC tolerance. The output voltage has a DC value that is equal to
the valley of the output ripple plus 1/2 of the peak-to-peak
ripple. Change in the output ripple voltage will lead to a
change in DC voltage at the output.
42mV
4 .4 A
IRIPPLEMAX
ESRMAX = 9.5 mΩ
The output capacitance is usually chosen to meet transient requirements. A worst-case load release, from
maximum load to no load at the exact moment when
inductor current is at the peak, determines the required
capacitance. If the load release is instantaneous (load
changes from maximum to zero in < 1μs), the output
capacitor must absorb all the inductor’s stored energy.
This will cause a peak voltage on the capacitor according
to the following equation.
COUTMIN
1
§
·2
L¨ IOUT u IRIPPLEMAX ¸
2
©
¹
2
VPEAK VOUT 2
Assuming a peak voltage VPEAK of 1.150 (100mV rise upon
load release), and a 10A load release, the required capacitance is shown by the next equation.
( VIN VOUT ) u TON
L
IRIPPLE _ VIN
VRIPPLE
COUTMIN
1
§
·2
0.88PH¨10 u 4.4 ¸
2
¹
©
2
1.15 1.05 2
COUTMIN = 595μF
If the load release is relatively slow, the output capacitance
can be reduced. At heavy loads during normal switching,
when the FB pin is above the 500mV reference, the DL
output is high and the low-side MOSFET is on. During this
time, the voltage across the inductor is approximately
-VOUT. This causes a down-slope or falling di/dt in the
23
SC417/SC427
Applications Information (continued)
inductor. If the load di/dt is not much faster than the
-di/dt in the inductor, then the inductor current will tend
to track the falling load current. This will reduce the excess
inductive energy that must be absorbed by the output
capacitor, therefore a smaller capacitance can be used.
The following can be used to calculate the needed capacitance for a given dILOAD/dt.
Peak inductor current is shown by the next equation.
ILPK = IMAX + 1/2 x IRIPPLEMAX
ILPK = 10 + 1/2 x 4.4 = 12.2A
Rate of change of Load Current
dlLOAD
dt
IMAX = maximum load release = 10A
COUT
ILPK u
I
I
Lu LPK MAX u dt
VOUT dlLOAD
2VPK VOUT Example
dlLOAD
dt
2 .5 A
Ps
Stability Considerations
Unstable operation is possible with adaptive on-time controllers, and usually takes the form of double-pulsing or
ESR loop instability.
Double-pulsing occurs due to switching noise seen at the
FB input or because the FB ripple voltage is too low. This
causes the FB comparator to trigger prematurely after the
250ns minimum off-time has expired. In extreme cases
the noise can cause three or more successive on-times.
Double-pulsing will result in higher ripple voltage at the
output, but in most applications it will not affect operation. This form of instability can usually be avoided by
providing the FB pin with a smooth, clean ripple signal
that is at least 10mVp-p, which may dictate the need to
increase the ESR of the output capacitors. It is also imperative to provide a proper PCB layout as discussed in the
Layout Guidelines section.
Another way to eliminate doubling-pulsing is to add a
small (~ 10pF) capacitor across the upper feedback resistor, as shown in Figure 13. This capacitor should be left
unpopulated until it can be confirmed that double-pulsing
exists. Adding the C TOP capacitor will couple more ripple
into FB to help eliminate the problem. An optional connection on the PCB should be available for this capacitor.
CTOP
This would cause the output current to move from 10A to
0A in 4μs, giving the minimum output capacitance
requirement shown in the following equation.
VOUT
COUT
12.2 u
12.2 10
u 1Ps
1.05 2.5
21.15 1.05 0.88PH u
COUT = 379 μF
Note that COUT is much smaller in this example, 379μF
compared to 595μF based on a worst-case load release. To
meet the two design criteria of minimum 379μF and
maximum 9mΩ ESR, select two capacitors rated at 220μF
and 15mΩ ESR.
It is recommended that an additional small capacitor be
placed in parallel with COUT in order to filter high frequency
switching noise.
To FB pin
R1
R2
Figure 13 — Capacitor Coupling to FB Pin
ESR loop instability is caused by insufficient ESR. The
details of this stability issue are discussed in the ESR
Requirements section. The best method for checking stability is to apply a zero-to-full load transient and observe
the output voltage ripple envelope for overshoot and
ringing. Ringing for more than one cycle after the initial
step is an indication that the ESR should be increased.
24
SC417/SC427
Applications Information (continued)
One simple way to solve this problem is to add trace resistance in the high current output path. A side effect of
adding trace resistance is a decrease in load regulation.
L
Highside
ESR Requirements
A minimum ESR is required for two reasons. One reason is
to generate enough output ripple voltage to provide
10mVp-p at the FB pin (after the resistor divider) to avoid
double-pulsing.
The second reason is to prevent instability due to insufficient ESR. The on-time control regulates the valley of the
output ripple voltage. This ripple voltage is the sum of the
two voltages. One is the ripple generated by the ESR, the
other is the ripple due to capacitive charging and discharging during the switching cycle. For most applications the minimum ESR ripple voltage is dominated by the
output capacitors, typically SP or POSCAP devices. For
stability the ESR zero of the output capacitor should be
lower than approximately one-third the switching frequency. The formula for minimum ESR is shown by the
following equation.
Lowside
3
2 u S u C OUT u f sw
Using Ceramic Output Capacitors
For applications using ceramic output capacitors, the ESR
is normally too small to meet the above ESR criteria. In
these applications it is necessary to add a small virtual ESR
network composed of two capacitors and one resistor, as
shown in Figure 14. This network creates a ramp voltage
across CL, analogous to the ramp voltage generated across
the ESR of a standard capacitor. This ramp is then capacitively coupled into the FB pin via capacitor CC.
CL
R1
CC
COUT
FB
pin
R2
Figure 14 — Virtual ESR Ramp Current
The component values used in this circuit are calculated
using the following procedure.
Select CL (100nF) and RL to provide a 25mV ripple across
CL (VCL).
RL
ESR MIN
RL
VIN VOUT
ICL
where
ICL
CL u 'VCL
TON
and
TON
VOUT
VIN u fSW
Next choose a value for CC so that
CC
TON
REQ
where
REQ
R1 u R 2
R1 R 2
The resistor values (R1 and R2) in the voltage divider circuit
set the VOUT for the switcher.
25
SC417/SC427
Applications Information (continued)
Dropout Performance
The output voltage adjust range for continuous-conduction operation is limited by the fixed 250ns (typical)
minimum off-time of the one-shot. When working with
low input voltages, the duty-factor limit must be calculated using worst-case values for on and off times.
The duty-factor limitation is shown by the next equation.
DUTY
TON(MIN)
TON(MIN) TOFF(MAX )
The inductor resistance and MOSFET on-state voltage
drops must be included when performing worst-case
dropout duty-factor calculations.
System DC Accuracy (VOUT Controller)
Three factors affect VOUT accuracy: the trip point of the FB
error comparator, the ripple voltage variation with line
and load, and the external resistor tolerance. The error
comparator offset is trimmed so that under static conditions it trips when the feedback pin is 500mV, 1%.
The on-time pulse from the SC417/SC427 in the design
example is calculated to give a pseudo-fixed frequency of
250kHz. Some frequency variation with line and load is
expected. This variation changes the output ripple
voltage. Because constant on-time converters regulate to
the valley of the output ripple, ½ of the output ripple
appears as a DC regulation error. For example, if the
output ripple is 50mV with VIN = 6 volts, then the measured
DC output will be 25mV above the comparator trip point.
If the ripple increases to 80mV with VIN = 25V, then the
measured DC output will be 40mV above the comparator
trip. The best way to minimize this effect is to minimize
the output ripple.
To compensate for valley regulation, it may be desirable to
use passive droop. Take the feedback directly from the
output side of the inductor and place a small amount of
trace resistance between the inductor and output capaci-
tor. This trace resistance should be optimized so that at
full load the output droops to near the lower regulation
limit. Passive droop minimizes the required output capacitance because the voltage excursions due to load steps
are reduced as seen at the load.
The use of 1% feedback resistors may result in up to 1%
error. If tighter DC accuracy is required, 0.1% resistors
should be used.
The output inductor value may change with current. This
will change the output ripple and therefore will have a
minor effect on the DC output voltage. The output ESR
also affects the output ripple and thus has a minor effect
on the DC output voltage.
Switching Frequency Variations
The switching frequency will vary depending on line and
load conditions. The line variations are a result of fixed
propagation delays in the on-time one-shot, as well as
unavoidable delays in the external MOSFET switching. As
VIN increases, these factors make the actual DH on-time
slightly longer than the ideal on-time. The net effect is
that frequency tends to falls slightly with increasing input
voltage.
The switching frequency also varies with load current as a
result of the power losses in the MOSFETs and the inductor. For a conventional PWM constant-frequency converter, as load increases the duty cycle also increases
slightly to compensate for IR and switching losses in the
MOSFETs and inductor. A constant on-time converter
must also compensate for the same losses by increasing
the effective duty cycle (more time is spent drawing
energy from VIN as losses increase). The on-time is essentially constant for a given VOUT/VIN combination, to offset
the losses the off-time will tend to reduce slightly as load
increases. The net effect is that switching frequency
increases slightly with increasing load.
26
SC417/SC427
Applications Information (continued)
•
PCB Layout Guidelines
The optimum layout for the SC417/SC427 is shown in
Figure 15. This layout shows an integrated FET buck regulator with a maximum current of 10A. The total PCB area is
approximately 20 x 25 mm.
Critical Layout Guidelines
The following critical layout guidelines must be followed
to ensure proper performance of the device.
•
•
•
•
•
•
IC Decoupling capacitors
PGND plane
AGND island
FB, VOUT, and other analog control signals
BST, ILIM, and LX
CIN and COUT placement and Current Loops
IC Decoupling Capacitors
A 0.1 μF capacitor must be located as close as
possible to the IC and directly connected to pins
3 (V5V) and 4 (AGND).
All other decoupling capacitors must be located
as close as possible to the IC.
PGND Plane
PGND requires its own copper plane with no
other signal traces routed on it.
Copper planes, multiple vias and wide traces are
needed to connect PGND to input capacitors,
output capacitors, and the PGND pins on the IC.
The PGND copper area between the input
capacitors, output capacitors and PGND pins
must be as tight and compact as possible to
reduce the area of the PCB that is exposed to
noise due to current flow on this node.
Connect PGND to AGND with a short trace or
0Ω resistor. This connection should be as close
to the IC as possible.
•
•
•
•
•
V5V Decoupling Capacitor
RGND — AGND connects to
PGND close to SC417/SC427
AGND plane on
inner layer
RILIM
RLDO2
RLDO1
CLDO
RFB1
RFB2
CFF
CBST
CIN
PGND on
Top Layer
SC417/SC427
with vias for LX,
AGND, VIN
CV5V
All components
shown Top Side
Pin 1 marking
VIN plane on inner
or bottom layer
COUT
VOUT Plane
on Top layer
L
PGND on inner
or bottom layer
LX plane on inner
or bottom layer
PGND
Figure 15 — PCB Layout
27
SC417/SC427
Applications Information (continued)
AGND Island
AGND should have its own island of copper with
no other signal traces routed on this layer that
connects the AGND pins and pad of the IC to the
analog control components.
All of the components for the analog control circuitry should be located so that the connections
to AGND are done by wide copper traces or vias
down to AGND.
Connect PGND to AGND with a short trace or 0Ω
resistor. This connection should be as close to
the IC as possible.
•
•
•
FB, VOUT, and Other Analog Control Signals
The connection from the V OUT power to the
analog control circuitry must be routed from the
output capacitors and located on a quiet layer.
The traces between Vout and the analog control
circuitry (VOUT, and FB pins) must be short and
routed away from noise sources, such as BST, LX,
VIN, and PGND between the input capacitors,
output capacitors, and the IC.
ILIM and TON nodes must be as short as possible
to ensure the best accuracy in current limit and
on time.
RILIM should be close to the IC and connected to
LX with a Kelvin trace to pin 28 on the IC. All of
the LX pins are connected to the LX PAD on the
IC, which should be a sufficient connection and
will prevent the need to connect the resistor
further into the LX plane.
The feedback components for the switcher and
the LDO need to be as close to the FB and FBL
pins of the IC as possible to reduce the possibility of noise corrupting these analog signals.
•
•
•
•
BST, ILIM and LX
LX and BST are very noisy nodes and must be
routed to minimized the PCB area that is exposed
to these signals.
The connections for the boost capacitor
between the IC and LX must be short and directly
connected to the LXBST (pin 13).
The connections for the current limit resistor
between the ILIM pin and LX must be as short as
possible and directly connected to pin 28 (LXS).
The LX node between the IC and the inductor
should be wide enough to handle the inductor
current and short enough to eliminate the possibility of LX noise corrupting other signals.
Multiple vias should be used to provide a good
connection to LX between the IC and the
inductor.
•
•
•
•
•
Capacitors and Current Loops
The current loops between the input capacitors,
the IC, the inductor, and the output capacitors
must be as close as possible to each other to
reduce IR drop across the copper.
All bypass and output capacitors must be connected as close as possible to the pin on the IC.
•
•
•
28
SC417/SC427
Outline Drawing — MLPQ-5x5-32
A
DIMENSIONS
MILLIMETERS
INCHES
DIM
MIN NOM MAX MIN NOM MAX
1.00
.031
.039 0.80
A
0.05
.000
.002 0.00
A1
(.008)
(0.20)
A2
b
.007 .010 .012 0.18 0.25 0.30
D
.193 .197 .201 4.90 5.00 5.10
D1 .076 .078 .080 1.92 1.97 2.02
E
.193 .197 .201 4.90 5.00 5.10
E1 .135 .137 .139 3.43 3.48 3.53
e
.020 BSC
0.50 BSC
L
.012 .016 .020 0.30 0.40 0.50
32
32
N
aaa
0.08
.003
0.10
bbb
.004
B
D
PIN 1
INDICATOR
(LASER MARK)
E
A2
A
aaa
SEATING
PLANE
C
C
A1
3.48
D1
0.76
1.05
LxN
1.49
E1
3.61
1.66
2
1
0.76
N
R0.20
PIN 1
IDENTIFICATION
bxN
e
bbb
C A
B
NOTES:
1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS THE TERMINALS.
29
SC417/SC427
Land Pattern — MLPQ-5x5-32
3.48
K1
K
DIMENSIONS
1.74
H2
1.74
(C)
H
3.61
G
Z
H1
Y
X
P
DIM
INCHES
MILLIMETERS
C
(.195)
(4.95)
G
.165
4.20
H
.137
3.48
H1
.059
1.49
H2
.065
1.66
K
.078
1.97
K1
.041
1.05
P
.020
0.50
X
.012
0.30
Y
.030
0.75
Z
.224
5.70
NOTES:
1.
CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
3. THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED PAD
SHALL BE CONNECTED TO A SYSTEM GROUND PLANE.
FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR
FUNCTIONAL PERFORMANCE OF THE DEVICE.
4. SQUARE PACKAGE-DIMENSIONS APPLY IN BOTH X AND Y DIRECTIONS.
Contact Information
Semtech Corporation
Power Mangement Products Division
200 Flynn Road, Camarillo, CA 93012
Phone: (805) 498-2111 Fax: (805) 498-3804
www.semtech.com
30