G FEATURINperature Range July 2000 Tem ommercial Extended C 70˚C to C 0˚ -2 ment dheld Equip an H e bl ta or for P ML4950* Adjustable Output, Low Current Single Cell Boost Regulator with Detect GENERAL DESCRIPTION FEATURES The ML4950 is a low power boost regulator designed for low voltage DC to DC conversion in single cell battery powered systems. The maximum switching frequency can exceed 100kHz, allowing the use of small, low cost inductors. ■ Guaranteed full load start-up and operation at 1V input ■ Pulse Frequency Modulation (PFM) and internal synchronous rectification for high efficiency ■ Minimum external components ■ Low ON resistance internal switching FETs ■ Micropower operation ■ Adjustable output voltage (2V to 3V) ■ Low battery detect The combination of integrated synchronous rectification, variable frequency operation, and low supply current make the ML4950 ideal for single cell applications. The ML4950 is capable of start-up with input voltages as low as 1V, and the output voltage can be set anywhere between 2V and 3V. An integrated synchronous rectifier eliminates the need for an external Schottky diode and provides a lower forward voltage drop, resulting in higher conversion efficiency. In addition, low quiescent battery current and variable frequency operation result in high efficiency even at light loads. The ML4950 requires a minimum number of external components and is capable of achieving conversion efficiencies in excess of 90%. The circuit also contains a RESET output which goes low when the IC can no longer function due to low input voltage, or when the DETECT input drops below 200mV. *Some Packages Are Obsolete BLOCK DIAGRAM 7 RESET DETECT 4 6 VL + COMP VREF 1 – VIN START-UP + VOUT 5 – S Q R Q 5µs ONE SHOT PWR GND 8 SENSE 3 – + REFERENCE VREF VREF GND 2 1 ML4950 PIN CONFIGURATION ML4950 8-Pin SOIC (S08) VIN 1 8 PWR GND GND 2 7 RESET SENSE 3 6 VL DETECT 4 5 VOUT TOP VIEW PIN DESCRIPTION PIN NAME FUNCTION PIN NAME FUNCTION 1 VIN Battery input voltage 5 VOUT Boost regulator output 2 GND Analog signal ground 6 VL Boost inductor connection 3 SENSE Programming pin for setting the output voltage 7 RESET Output goes low when regulation cannot be achieved, or when DETECT goes below 200mV 4 DETECT Pulling this pin below 200mV causes the RESET pin to go low 8 PWR GND Return for the NMOS output transistor 2 ML4950 ABSOLUTE MAXIMUM RATINGS OPERATING CONDITIONS Absolute maximum ratings are those values beyond which the device could be permanently damaged. Absolute maximum ratings are stress ratings only and functional device operation is not implied. Temperature Range ML4950CS-X .............................................. 0ºC to 70ºC ML4950ES-X ........................................... -20ºC to 70ºC VIN Operating Range ML4950CS-X ................................. 1.0V to VOUT - 0.2V ML4950ES-X ................................. 1.1V to VOUT - 0.2V VOUT Operating Range ....................................... 2V to 3V VOUT ............................................................................................... 7V Voltage on Any Other Pin ..... GND - 0.3V to VOUT + 0.3V Peak Switch Current (IPEAK) .......................................... 1A Average Switch Current (IAVG) .............................. 250mA Junction Temperature .............................................. 150ºC Storage Temperature Range...................... –65ºC to 150ºC Lead Temperature (Soldering, 10 sec) ...................... 150ºC Thermal Resistance (qJA) .................................... 160ºC/W ELECTRICAL CHARACTERISTICS Unless otherwise specified, VIN = Operating Voltage Range, TA = Operating Temperature Range (Note 1) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 50 60 µA 8 10 µA 1 µA SUPPLY IIN VIN Current IOUT(Q) IL(Q) VIN = VOUT - 0.2V VOUT Quiescent Current VL Quiescent Current PFM REGULATOR tON VSENSE Pulse Width 4.5 5 5.5 µs SENSE Compator Threshold Voltage 196 201 208 mV 2.425 2.5 2.575 V 0.85 0.95 V 200 206 mV 100 nA Load Regulation See Figure 1 VIN = 1.2V, IOUT £ 25mA Undervoltage Lockout Threshold RESET COMPARATOR Note 1: DETECT Threshold Voltage 194 DETECT Bias Current -100 RESET Output High Voltage IOH = -100µA RESET Output Low Voltage IOL = 100µA VOUT-0.2 V 0.2 V Limits are guaranteed by 100% testing, sampling, or correlation with worst-case test conditions. 3 ML4950 27µH (Sumida CD75) ML4950 VIN VIN PWR GND RESET GND 100µF SENSE VL DETECT IOUT VOUT VOUT 464kΩ 0.1% 100µF 40.2kΩ 0.1% Figure 1. Application Test Circuit. L1 VIN 6 VL START-UP Q2 VOUT + A2 – + 5 C1 Q1 S Q R Q 5µs ONE SHOT – R1 SENSE 3 – A1 + VREF R2 Figure 2. PFM Regulator Block Diagram. INDUCTOR CURRENT Q(ONE SHOT) Q1 ON Q2 ON Q1 ON Q2 ON Q1 & Q2 OFF Figure 3. PFM Inductor Current Waveforms and Timing. 4 VOUT ML4950 FUNCTIONAL DESCRIPTION DESIGN CONSIDERATIONS The ML4950 combines Pulse Frequency Modulation (PFM) and synchronous rectification to create a boost converter that is both highly efficient and simple to use. A PFM regulator charges a single inductor for a fixed period of time and then completely discharges before another cycle begins, simplifying the design by eliminating the need for conventional current limiting circuitry. Synchronous rectification is accomplished by replacing an external Schottky diode with an on-chip PMOS device, reducing switching losses and external component count. INDUCTOR REGULATOR OPERATION A block diagram of the boost converter is shown in Figure 2. The circuit remains idle when VOUT is at or above the desired output voltage, drawing 50µA from VIN, and 8µA from VOUT through the feedback resistors R1 and R2. When VOUT drops below the desired output level, the output of amplifier A1 goes high, signaling the regulator to deliver charge to the output. Since the output of amplifier A2 is normally high, the flip-flop captures the A1 set signal and creates a pulse at the gate of the NMOS transistor Q1. The NMOS transistor will charge the inductor L1 for 5µs, resulting in a peak current given by: IL(PEAK) t VIN 5ms VIN = ON = L1 L1 (1) For reliable operation, L1 should be chosen so that IL(PEAK) does not exceed 1A. When the one-shot times out, the NMOS transistor releases the VL pin, allowing the inductor to fly-back and momentarily charge the output through the body diode of PMOS transistor Q2. But as the voltage across the PMOS transistor changes polarity, its gate will be driven low by the current sense amplifier A2, causing Q2 to short out its body diode. The inductor then discharges into the load through Q2. The output of A2 also serves to reset the flipflop and one-shot in preparation for the next charging cycle. A2 releases the gate of Q2 when its current falls to zero. If VOUT is still low, the flip-flop will immediately initiate another pulse. The output capacitor (C1) filters the inductor current, limiting output voltage ripple. Inductor current and one-shot waveforms are shown in Figure 3. RESET COMPARATOR An additional comparator is provided to detect low VIN or any other error condition that is important to the user. The inverting input of the comparator is internally connected to VREF, while the non-inverting input is provided externally at the DETECT pin. The output of the comparator is the RESET pin, which swings from VOUT to GND when an error is detected. Selecting the proper inductor for a specific application usually involves a trade-off between efficiency and maximum output current. Choosing too high a value will keep the regulator from delivering the required output current under worst case conditions. Choosing too low a value causes efficiency to suffer. It is necessary to know the maximum required output current and the input voltage range to select the proper inductor value. The maximum inductor value can be estimated using the following formula: LMAX = VIN( MIN) 2 t ON( MIN) h 2 VOUT IOUT( MAX) (2) where h is the efficiency, typically between 0.8 and 0.9. Note that this is the value of inductance that just barely delivers the required output current under worst case conditions. A lower value may be required to cover inductor tolerance, the effect of lower peak inductor currents caused by resistive losses, and minimum dead time between pulses. Another method of determining the appropriate inductor value is to make an estimate based on the typical performance curves given in Figures 4 and 5. Figure 4 shows maximum output current as a function of input voltage for several inductor values. These are typical performance curves and leave no margin for inductance and ON-time variations. To accommodate worst case conditions, it is necessary to derate these curves by at least 10% in addition to inductor tolerance. For example, a single cell to 2.5 V application requires 20mA of output current while using an inductor with 15% tolerance. The output current should be derated by 25% to 25mA to cover the combined inductor and ON-time tolerances. Assuming that 1V is the end of life voltage of a single cell input, Figure 4 shows that with the ML4950 delivers 25mA at 2.5V with a 27µH inductor. Figure 5 shows efficiency under the conditions used to create Figure 4. It can be seen that efficiency is mostly independent of input voltage and is closely related to inductor value. This illustrates the need to keep the inductor value as high as possible to attain peak system efficiency. As the inductor value goes down to 18µH, the efficiency drops to between 75% and 80%. With 68µH, the efficiency exceeds 90% and there is little room for improvement. At values greater than 100µH, the operation of the synchronous rectifier becomes unreliable because the inductor current is so small that it is difficult for the control circuitry to detect. The data used to generate Figures 4 and 5 is provided in Table 1. 5 ML4950 DESIGN CONSIDERATIONS (Continued) After the appropriate inductor value is chosen, it is necessary to find the minimum inductor current rating required. Peak inductor current is determined from the following formula: IL(PEAK) = t ON( MAX) VIN( MAX) (3) LMIN In the single cell application previously described, a maximum input voltage of 1.6V would give a peak current of 383mA. When comparing various inductors, it is important to keep in mind that suppliers use different criteria to determine their ratings. Many use a conservative current level, where inductance has dropped to 90% of its normal level. In any case, it is a good idea to try inductors of various current ratings with the ML4950 to determine which inductor is the best choice. Check efficiency and maximum output current, and if a current probe is available, look at the inductor current to see if it looks like the waveform shown in Figure 3. For additional information, see Application Note 29. Suitable inductors can be purchased from the following suppliers: Coilcraft (847) 639-6400 Coiltronics (561) 241-7876 Dale (605) 665-9301 Sumida (847) 956-0666 Capacitor Equivalent Series Resistance (ESR) and Equivalent Series Inductance (ESL), also contribute to the output ripple due to the inductor discharge current waveform. Just after the NMOS transistor turns off, the output current ramps quickly to match the peak inductor current. This fast change in current through the output capacitor’s ESL causes a high frequency (5ns) spike that can be over 1V in magnitude. After the ESL spike settles, the output voltage still has a ripple component equal to the inductor discharge current times the ESR. This component will have a sawtooth shape and a peak value equal to the peak inductor current times the ESR. ESR also has a negative effect on efficiency by contributing I2R losses during the discharge cycle. An output capacitor with a capacitance of 100µF, an ESR of less than 0.1W, and an ESL of less than 5nH is a good general purpose choice. Tantalum capacitors which meet these requirements can be obtained from the following suppliers: AVX (207) 282-5111 Sprague (207) 324-4140 If ESL spikes are causing output noise problems, an EMI filter can be added in series with the output. INPUT CAPACITOR XFMRS, Inc. (317) 834-1066 OUTPUT CAPACITOR The choice of output capacitor is also important, as it controls the output ripple and optimizes the efficiency of the circuit. Output ripple is influenced by three parameters: capacitance, ESR, and ESL. The contribution due to capacitance can be determined by looking at the change in capacitor voltage required to store the energy delivered by the inductor in a single charge-discharge cycle, as determined by the following formula: DVOUT = t ON2 VIN2 2 L C VOUT - VIN 1 6 (4) For a 1.2V input, a 2.5V output, a 27µH inductor, and a 47µF capacitor, the expected output ripple due to capacitor value is 11mV. Unless the input source is a very low impedance battery, it will be necessary to decouple the input with a capacitor with a value of between 47µF and 100µF. This prevents input ripple from affecting the ML4950 control circuitry, and it also improves efficiency by reducing I2R losses during the charge and discharge cycles of the inductor. Again, a low ESR capacitor (such as tantalum) is recommended. SETTING THE OUTPUT VOLTAGE The adjustable output can be set to any voltage between 2V and 3V by connecting a resistor divider to the SENSE pin as shown in the block diagram. The resistor values R1 and R2 can be calculated using the following equation: VOUT = 0.2 1 2 R2 (5) The value of R2 should be 40kW or less to minimize bias current errors. R1 is then found by rearranging the equation: R1 = R 2 6 1R + R 6 V - 1 0.2 OUT (6) ML4950 140 95 L = 18µH VOUT = 2V VOUT = 2V 120 L = 68µH 90 IOUT (mA) L = 33µH 80 L = 47µH 60 EFFICIENCY (%) 100 L = 47µH L = 33µH 85 L = 68µH 40 80 L = 18µH 20 0 1.0 1.2 1.4 75 1.0 1.8 1.6 1.2 VIN (V) 1.4 1.6 1.8 VIN (V) 140 95 L = 18µH VOUT = 2.5V VOUT = 2.5V 120 L = 68µH 90 100 IOUT (mA) 80 L = 47µH 60 EFFICIENCY (%) L = 47µH L = 33µH L = 33µH 85 L = 18µH 40 80 L = 68µH 20 0 1.0 1.2 1.4 75 1.0 1.8 1.6 1.2 VIN (V) 1.8 95 VOUT = 3V L = 18µH VOUT = 3V 70 L = 68µH L = 33µH 50 40 L = 47µH EFFICIENCY (%) 90 60 IOUT (mA) 1.6 VIN (V) 90 80 1.4 L = 47µH L = 33µH 85 L = 18µH 30 L = 68µH 20 80 10 0 1.0 1.2 1.4 1.6 VIN (V) Figure 4. Output Current vs Input Voltage 1.8 75 1.0 1.2 1.4 1.6 1.8 VIN (V) Figure 5. Typical Efficiency as a Function of VIN 7 ML4950 DESIGN CONSIDERATIONS (Continued) It is important to note that the accuracy of these resistors directly affects the accuracy of the output voltage. The SENSE pin threshold variation is ±3%, and the tolerance of R1 and R2 will add to this to determine the total output variation. LAYOUT Under some circumstances, input ripple cannot be reduced effectively. This occurs primarily in applications where inductor currents are high, causing excess output ripple due to “pulse grouping”, where the chargedischarge pulses are not evenly spaced in time. In such cases it may be necessary to add a small 20pF to 100pF ceramic feedforward capacitor (CFF) from the VIN pin to the SENSE pin. This is particularly true if the ripple voltage at VIN is greater than 100mV. ■ Use adequate ground and power traces or planes ■ Keep components as close as possible to the ML4950 ■ Use short trace lengths from the inductor to the VL pin and from the output capacitor to the VOUT pin ■ Use a single point ground for the ML4950 PWR GND pin and the input and output capacitors, and connect GND to PWR GND with a separate trace SETTING THE RESET THRESHOLD To use the RESET comparator as an input voltage monitor, it is necessary to use an external resistor divider tied to the DETECT pin as shown in Figure 7. The resistor values RA and RB can be calculated using the following equation: VIN( MIN) = 0.2 1R A + RB 6 V 0.2 RB IN( MIN) A typical PC board layout is shown in Figure 8. 7 VIN RESET 1 (7) The value of RB should be 100kW or less to minimize bias current errors. RA is then found by rearranging the equation: R A = RB Good PC board layout practices will ensure the proper operation of the ML4950. Important layout considerations include: -1 RA DETECT 4 + COMP RB VREF – (8) FROM START-UP CIRCUITRY Figure 7. Battery Monitoring Circuit 80 60 IIN (µA) 2V OUTPUT 3V OUTPUT 40 20 0 1.0 1.5 2.0 2.5 3.0 VIN (V) Figure 6. No Load Input Current vs. VIN For 2V, R1 = 365kW, R2 = 40.2kW For 3V, R1 = 562kW, R2 = 40.2kW L = Sumida CD43, 22µH 8 Figure 8. Typical PC Board Layout ML4950 VOUT = 2V VIN L = 18µH 1.0 1.2 1.4 1.6 1.8 L = 33µH 1.0 1.2 1.4 1.6 1.8 L = 47µH 1.0 1.2 1.4 1.6 1.8 L = 68µH 1.0 1.2 1.4 1.6 1.8 VOUT = 2.5V IOUT (mA) EFFICIENCY (%) 45.8 65.8 89.4 111.7 132.9 77.5 78.0 78.3 78.9 79.8 27.9 41.9 57.8 73.7 89.8 83.6 84.5 85.0 85.6 86.4 20.1 31.6 43.7 57.5 70.1 85.1 86.6 87.4 87.5 88.9 14.9 23.3 32.6 43.5 54.6 86.5 88.5 89.1 89.6 90.9 VIN IOUT (mA) EFFICIENCY (%) 38.3 54.5 74.6 98.5 124 80.0 80.2 80.5 80.8 81.0 22.7 33.0 47.0 61.8 79.3 84.2 85.2 86.0 86.6 87.0 16.4 24.1 33.4 46.1 58.6 86.2 87.1 88.1 89.0 89.5 12.6 17.4 25.2 33.9 43.8 86.4 87.9 89.2 90.1 90.9 L = 18µH 1.0 1.2 1.4 1.6 1.8 L = 33µH 1.0 1.2 1.4 1.6 1.8 L = 47µH 1.0 1.2 1.4 1.6 1.8 L = 68µH 1.0 1.2 1.4 1.6 1.8 VOUT = 3V VIN L = 18µH 1.0 1.2 1.4 1.6 1.8 L = 33µH 1.0 1.2 1.4 1.6 1.8 L = 47µH 1.0 1.2 1.4 1.6 1.8 L = 68µH 1.0 1.2 1.4 1.6 1.8 IOUT (mA) EFFICIENCY (%) 30.9 43.8 55.4 65.6 84.4 81.6 81.9 82.1 82.2 82.3 18.7 27.9 38.1 50.6 65.4 85.1 85.6 86.4 86.9 87.5 13.3 19.9 27.9 36.7 47.4 84.7 86.5 87.7 88.5 89.1 9.2 14.0 20.5 27.6 35.7 84.7 86.3 88.2 89.1 90.2 Table 1. Typical IOUT and Efficiency vs. VIN 9 ML4950 PHYSICAL DIMENSIONS inches (millimeters) Package: S08 8-Pin SOIC 0.189 - 0.199 (4.80 - 5.06) 8 PIN 1 ID 0.148 - 0.158 0.228 - 0.244 (3.76 - 4.01) (5.79 - 6.20) 1 0.017 - 0.027 (0.43 - 0.69) (4 PLACES) 0.050 BSC (1.27 BSC) 0.059 - 0.069 (1.49 - 1.75) 0º - 8º 0.055 - 0.061 (1.40 - 1.55) 0.012 - 0.020 (0.30 - 0.51) 0.004 - 0.010 (0.10 - 0.26) 0.015 - 0.035 (0.38 - 0.89) 0.006 - 0.010 (0.15 - 0.26) SEATING PLANE ORDERING INFORMATION © Micro Linear 1998. PART NUMBER TEMPERATURE RANGE PACKAGE ML4950CS ML4950ES (Obsolete) 0°C to 70°C –20°C to 70°C 8-Pin SOIC (S08) 8-Pin SOIC (S08) is a registered trademark of Micro Linear Corporation. All other trademarks are the property of their respective owners. Products described herein may be covered by one or more of the following U.S. patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940; 5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; 5,652,479; 5,661,427; 5,663,874; 5,672,959; 5,689,167; 5,714,897; 5,717,798; 5,742,151; 5,754,012; 5,757,174. Japan: 2,598,946; 2,619,299; 2,704,176. Other patents are pending. Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design. Micro Linear does not assume any liability arising out of the application or use of any product described herein, neither does it convey any license under its patent right nor the rights of others. The circuits contained in this data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel before deciding on a particular application. 10 DS4950-01 2092 Concourse Drive San Jose, CA 95131 Tel: 408/433-5200 Fax: 408/432-0295 www.microlinear.com 7/18/98 Printed in U.S.A.