TI TPS61175

TPS61175
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SLVS892A – DECEMBER 2008 – REVISED OCTOBER 2010
3-A High Voltage Boost Converter with Soft-start and Programmable Switching Frequency
Check for Samples: TPS61175
FEATURES
1
•
•
•
•
2
•
•
•
•
DESCRIPTION
2.9-V to 18-V Input Voltage Range
3A, 40V Internal Switch
High Efficiency Power Conversion: Up to 93%
Frequency Set by External Resistor: 200-kHz
to 2.2-MHz
Synchronous External Switching Frequency
User Defined Soft Start into Full Load
Skip-Switching Cycle for Output Regulation at
Light Load
14-pin HTSSOP Package with PowerPad™
The TPS61175 is a monolithic switching regulator
with integrated 3-A, 40-V power switch. It can be
configured in several standard switching-regulator
topologies, including boost, SEPIC and flyback. The
device has a wide input voltage range to support
application with input voltage from multi-cell batteries
or regulated 5-V, 12-V power rails.
The TPS61175 regulates the output voltage with
current mode PWM (pulse width modulation) control.
The switching frequency of PWM is either set by an
external resistor or an external clock signal. The user
can program the switching frequency from 200-kHz to
2.2-MHz.
APPLICATIONS
•
•
•
•
5V to 12V, 24V power conversion
Supports SEPIC, Flyback topology
ADSL Modems
TV Tuner
The device features a programmable soft-start
function to limit inrush current during start-up, and
has built-in other protection features, such as
pulse-by-pulse over current limit and thermal
shutdown. The TPS61175 is available in 14-pin
HTSSOP package with Powerpad.
TYPICAL APPLICATION FOR BOOST CONVERTER
VIN
D1
L1
VOUT
C2
C1
TPS61175
VIN
SW
EN
SW
FREQ
SS
FB
PGND
COMP PGND
R4
C3
Syn
R3
AGND
R1
R2
PGND
NC
C4
Figure 1. Typical Application
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPad is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2010, Texas Instruments Incorporated
TPS61175
SLVS892A – DECEMBER 2008 – REVISED OCTOBER 2010
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION (1)
(1)
(2)
TA
PART NUMBER
PACKAGE (2)
–40°C to 85°C
TPS61175PWP
HTSSOP-14
For the most current package and ordering information, see the TI Web site at www.ti.com.
The PWP package is available in tape and reel. Add R suffix (TPS61175PWPR) to order quantities of 2000 parts per reel. Without suffix,
the TPS61175PWP is shipped in tubes with 90 parts per tube.
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
Supply Voltages on pin VIN
Voltages on pins EN
(2)
(2)
VALUE
UNIT
–0.3 to 20
V
–0.3 to 20
V
Voltage on pin FB, FREQ and COMP (2)
–0.3 to 3
V
Voltage on pin SYNC, SS (2)
–0.3 to 7
V
Voltage on pin SW (2)
–0.3 to 40
V
Continuous Power Dissipation
See Dissipation Rating Table
Operating Junction Temperature Range
–40 to +150
°C
Storage Temperature Range
–65 to +150
°C
(1)
(2)
Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltage values are with respect to network ground terminal.
DISSIPATION RATINGS
(1)
PACKAGE
THERMAL RESISTANCE
RqJA
TA ≤ 25°C
POWER RATING
TA = 85°C
POWER RATING
14 pin PWP (1)
44.5°C/W
2.25 W
0.9 W
Rating based on JEDEC high thermal conductivity (High K) board with 2x2 array of thermal vias. See Texas Instruments application
report (SLMA002) regarding thermal characteristics of the PowerPAD package
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
UNIT
VIN
Input voltage range
2.9
18
VO
Output voltage range
VIN
38
V
L
Inductor (1)
4.7
47
mH
fSW
Switching frequency
200
2200
kHz
CI
Input Capacitor
4.7
CO
Output Capacitor
4.7
VSYN
External Switching Frequency Logic
TA
Operating ambient temperature
TJ
Operating junction temperature
(1)
2
V
mF
mF
5
V
–40
85
°C
–40
125
°C
The inductance value depends on the switching frequency and end application. While larger values may be used, values between
4.7-mH and 47-mH have been successfully tested in various applications. Refer to the Inductor Selection for detail.
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ELECTRICAL CHARACTERISTICS
FSW = 1.2 MHz (Rfreq=80 kΩ), Vin=3.6V, TA = –40°C to +85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY CURRENT
VIN
Input voltage range
IQ
Operating quiescent current into Vin
Device PWM switching without load
2.9
ISD
Shutdown current
EN=GND
VUVLO
Under-voltage lockout threshold
2.5
Vhys
Under-voltage lockout hysteresis
130
18
V
3.5
mA
1.5
mA
2.7
V
mV
ENABLE AND REFERENCE CONTROL
Venh
EN logic high voltage
Vin = 2.9 V to 18 V
Venl
EN logic low voltage
Vin = 2.9 V to 18 V
VSYNh
SYN logic high voltage
VSYNl
SYN logic low voltage
Ren
EN pull down resistor
Toff
Shutdown delay, SS discharge
1.2
V
0.4
V
0.4
V
1.2
400
EN high to low
800
1600
kΩ
10
ms
VOLTAGE AND CURRENT CONTROL
VREF
Voltage feedback regulation voltage
1.204
1.229
1.254
V
IFB
Voltage feedback input bias current
Isink
Comp pin sink current
VFB = VREF + 200 mV, VCOMP = 1 V
50
mA
Isource
Comp pin source current
VFB = VREF –200 mV, VCOMP = 1 V
130
mA
VCCLP
Comp pin Clamp Voltage
High Clamp, VFB = 1 V
Low Clamp, VFB = 1.5 V
3
0.75
V
VCTH
Comp pin threshold
Duty cycle = 0%
Gea
Error amplifier transconductance
Rea
Error amplifier output resistance
fea
Error amplifier crossover frequency
200
nA
0.95
240
340
V
440
mmho
10
MΩ
500
KHz
FREQUENCY
fS
Oscillator frequency
Dmax
Maximum duty cycle
VFREQ
FREQ pin voltage
Tmin_on
Minimum on pulse width
Rfreq = 480 kΩ
0.16
0.21
Rfreq = 80 kΩ
1.0
1.2
1.4
Rfreq = 40 kΩ
1.76
2.2
2.64
VFB = 1.0 V, Rfreq = 80 kΩ
89%
Rfreq = 80 kΩ
0.26
MHz
93%
1.229
V
60
ns
POWER SWITCH
RDS(ON)
N-channel MOSFET on-resistance
VIN = VGS = 3.6 V
VIN = VGS = 3.0 V
ILN_NFET
N-channel leakage current
VDS = 40 V, TA = 25°C
0.13
0.13
0.25
0.3
Ω
1
mA
OC, OVP AND SS
ILIM
N-Channel MOSFET current limit
D = Dmax
ISS
Soft start bias current
Vss = 0 V
3
3.8
5
A
6
mA
160
°C
15
°C
THERMAL SHUTDOWN
Tshutdown
Thermal shutdown threshold
T hysteresis
Thermal shutdown threshold hysteresis
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DEVICE INFORMATION
PIN ASSIGNMENTS
TSSOP 14-pin
(TOP VIEW)
SW
SW
VIN
EN
SS
SYNC
AGND
1
2
3
4
5
6
7
PGND
PGND
PGND
NC
FREQ
FB
COMP
14
13
12
11
10
9
8
PIN FUNCTIONS
PIN
NAME
NO.
DESCRIPTION
I/O
VIN
3
I
The input supply pin for the IC. Connect VIN to a supply voltage between 2.9V and 18V. It is acceptable for
the voltage on the pin to be different from the boost power stage input for applications requiring voltage
beyond VIN range.
SW
1,2
I
This is the switching node of the IC. Connect SW to the switched side of the inductor.
FB
9
I
Feedback pin for positive voltage regulation. Connect to the center tap of a resistor divider to program the
output voltage.
EN
4
I
Enable pin. When the voltage of this pin falls below the enable threshold for more than 10ms, the IC turns
off.
COMP
8
O
Output of the internal transconductance error amplifier. An external RC network is connected to this pin to
compensate the regulator.
SS
5
O
Soft start programming pin. A capacitor between the SS pin and GND pin programs soft start timing. See
application section for information on how to size the SS capacitor.
FREQ
10
O
Switch frequency program pin. An external resistor is connected to this pin to set switch frequency. See
application section for information on how to size the FREQ resistor.
AGND
7
I
Signal ground of the IC
PGND
12,13,14
I
Power ground of the IC. It is connected to the source of the PWM switch.
SYNC
6
I
Switch frequency synchronous pin. Customers can use an external signal to set the IC switch frequency
between 200-kHz and 2.2-MHz. If not used, this pin should be tied to AGND as short as possbile to avoid
noise coupling.
NC
11
I
Reserved pin. Must connect this pin to ground.
Thermal Pad
4
The thermal pad should be soldered to the analog ground. If possible, use thermal via to connect to top and
internal ground plane layers for ideal power dissipation.
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SLVS892A – DECEMBER 2008 – REVISED OCTOBER 2010
FUNCTIONAL BLOCK DIAGRAM
L1
D1
C1
C2
R1
FB
SW
VIN
R2
FB
EA
EN
Gate
Dirver
1.229 V
Reference
COMP
PWM Control
R3
C4
Ramp
Generator
Current
Sensor
+
Oscillator
SS
C3
FREQ
SYNC
AGND
PGND
R4
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
Circuit of Figure 1; L1 = D104C2-10mH; D1 = SS3P6L-E3/86A, R4 = 80kΩ, R3 = 10kΩ, C4 = 22nF,
C2 = 10mF;Vin = 5V, Vout = 24V, Iout = 200mA; unless otherwise noted
FIGURE
Efficiency
VIN = 5V, Vout = 12V, 24V, 35V
2
Efficiency
VIN = 5V, 12V; Vout = 24V
3
Error amplifier transconductance
vs Temperature
4
Switch current limit
vs Temperature
5
Switch current limit
vs Duty cycle
6
FB accuracy
vs Temperature
7
Line transient response
Vin = 4.5 V to 5 V
8
Load transient response
Iout = 100 mA to 300 mA; refer to 'compensating the control loop' for optimization
PWM Operation
9
10
Pulse skipping
No load
11
Start-up
C3 = 47 nF
12
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EFFICIENCY
vs
OUTPUT CURRENT
EFFICIENCY
vs
OUTPUT CURRENT
100
100
VI = 5 V
VO = 12 V
90
VI = 12 V
90
VI = 5 V
Efficiency - %
Efficiency - %
VO = 24 V
80
VO = 35 V
70
80
70
60
60
VO = 24 V
50
50
0.2
0.4
0.6
0.8
IO - Output Current - A
1
0
1.2
ERROR AMPLIFIER TRANSCONDUCTANCE
vs
FREE-AIR TEMPERATURE
OVERCURRENT LIMIT
vs
DUTY CYCLE
5
380
4.5
360
1
1.2
4
3.5
340
-20
0
20
40
60
80
TA - Free-Air Temperature - °C
100
120
3
0.2
Figure 4.
6
0.4
0.6
0.8
IO - Output Current - A
Figure 3.
400
320
-40
0.2
Figure 2.
Overcurrent Limit - A
EA Transconductance - mhos
0
0.4
0.6
Duty Cycle - %
0.8
1
Figure 5.
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OVERCURRENT LIMIT
vs
FREE-AIR TEMPERATURE
FB VOLTAGE
vs
FREE-AIR TEMPERATURE
4
1240
3.9
FB Voltage - mV
Overcurrent Limit - A
1235
3.8
3.7
1230
1225
3.6
3.5
-40
-20
0
80
60
20
40
TA - Free-Air Temperature - °C
100
120
1220
-40
-20
0
20
40
60
80
TA - Free-Air Temperature - °C
Figure 6.
Figure 7.
Line Transient Response
Load Transient Response
100
120
VOUT
500 mV/div
AC
VIN
1 V/div
AC
VOUT
100 mV/div
AC
ILOAD
200 mA/div
t - 100 ms/div
t - 200 ms/div
Figure 8.
Figure 9.
PWM Operation
Pulse Skipping
VOUT
1 V/div
20 V offset
SW
20 V/div
VOUT
20 mV/div
AC
IL
500 mA/div
IL
100 mA/div
VOUT
100 mV/div
AC
t - 400 ms/div
t - 400 ns/div
Figure 10.
Figure 11.
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Soft Startup
EN
2 V/div
VOUT
5 V/div
IL
500 mA/div
t - 1 ms/div
Figure 12.
DETAILED DESCRIPTION
OPERATION
The TPS61175 integrates a 40-V low side switch FET for up to 38-V output. The device regulates the output with
current mode PWM (pulse width modulation) control. The PWM control circuitry turns on the switch at the
beginning of each switching cycle. The input voltage is applied across the inductor and stores the energy as
inductor current ramps up. During this portion of the switching cycle, the load current is provided by the output
capacitor. When the inductor current rises to the threshold set by the error amplifier output, the power switch
turns off and the external Schottky diode is forward biased. The inductor transfers stored energy to replenish the
output capacitor and supply the load current. This operation repeats each every switching cycle. As shown in the
block diagram, the duty cycle of the converter is determined by the PWM control comparator which compares the
error amplifier output and the current signal. The switching frequency is programmed by the external resistor or
synchronized to an external clock signal.
A ramp signal from the oscillator is added to the current ramp to provide slope compensation. Slope
compensation is necessary to avoid subharmonic oscillation that is intrinsic to the current mode control at duty
cycle higher than 50%. If the inductor value is lower than 4.7mH, the slope compensation may not be adequate.
The feedback loop regulates the FB pin to a reference voltage through a transconductance error amplifier. The
output of the error amplifier is connected to the COMP pin. An external RC compensation network is connected
to the COMP pin to optimize the feedback loop for stability and transient response.
SWITCHING FREQUENCY
The switch frequency is set by a resistor (R4) connected to the FREQ pin of the TPS61175. Do not leave this pin
open. A resistor must always be connected for proper operation. See Table 1 and Figure 13 for resistor values
and corresponding frequencies.
Table 1. Switching Frequency vs External Resistor
8
R4 (kΩ)
fSW (kHz)
443
240
256
400
176
600
80
1200
51
2000
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3500
3000
f - Frequency - kHz
2500
2000
1500
1000
500
0
10
100
External Resistor - kW
1000
Figure 13. Switching Frequency vs External Resistor
Alternatively, the TPS61175 switching frequency will synchronize to an external clock signal that is applied to the
SYNC pin. The logic level of the external clock is shown in the specification table. The duty cycle of the clock is
recommended in the range of 10% to 90%. The resistor also must be connected to the FREQ pin when IC is
switching by the external clock. The external clock frequency must be within ±20% of the corresponding
frequency set by the resistor. For example, if the corresponding frequency as set by a resistor on the FREQ pin
is 1.2-MHz, the external clock signal should be in the range of 0.96-MHz to 1.44-MHz.
If the external clock signal is higher than the frequency per the resistor on the FREQ pin, the maximum duty
cycle specification (DMAX) should be lowered by 2%. For instance, if the resistor set value is 2.5MHz, and the
external clock is 3MHz, DMAX is 87% instead of 89%.
SOFT START
The TPS61175 has a built-in soft start circuit which significantly reduces the start-up current spike and output
voltage overshoot. When the IC is enabled, an internal bias current (6-mA typically) charges a capacitor (C3) on
the SS pin. The voltage at the capacitor clamps the output of the internal error amplifier that determines the duty
cycle of PWM control, thereby the input inrush current is eliminated. Once the capacitor reaches 1.8-V, the soft
start cycle is completed and the soft start voltage no longer clamps the error amplifier output. Refer to Figure 13
for the soft start waveform. See Table 2 for C3 and corresponding soft start time. A 47-nF capacitor eliminates
the output overshoot and reduces the peak inductor current for most applications.
Table 2. Soft Start Time vs C3
Vin (V)
5
12
Vout (V)
24
35
Load (A)
0.4
0.6
Cout (mF)
10
10
fSW (MHz)
1.2
2
C3 (nF)
tSS(ms)
Overshot (mV)
47
4
none
10
0.8
210
100
6.5
none
10
0.4
300
When the EN is pulled low for 10-ms, the IC enters shutdown and the SS capacitor discharges through a 5kΩ
resistor for the next soft start.
OVERCURRENT PROTECTION
The TPS61175 has a cycle-by-cycle overcurrent limit protection that turns off the power switch once the inductor
current reaches the overcurrent limit threshold. The PWM circuitry resets itself at the beginning of the next switch
cycle. During an overcurrent event, the output voltage begins to droop as a function of the load on the output.
When the FB voltage drops lower than 0.9-V, the switching frequency is automatically reduced to 1/4 of the set
value. The switching frequency does not reset until the overcurrent condition is removed. This feature is disabled
during soft start.
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ENABLE AND THERMAL SHUTDOWN
The TPS61175 enters shutdown when the EN voltage is less than 0.4-V for more than 10-ms. In shutdown, the
input supply current for the device is less than 1.5-mA (max). The EN pin has an internal 800-kΩ pull down
resistor to disable the device when it is floating.
An internal thermal shutdown turns off the device when the typical junction temperature of 160°C is exceeded.
The IC restarts when the junction temperature drops by 15°C.
UNDER VOLTAGE LOCKOUT (UVLO)
An under voltage lockout circuit prevents mis-operation of the device at input voltages below 2.5-V (typical).
When the input voltage is below the under voltage threshold, the device remains off and the internal switch FET
is turned off. The under voltage lockout threshold is set below minimum operating voltage of 2.9V to avoid any
transient VIN dip triggering the UVLO and causing the device to reset. For the input voltages between UVLO
threshold and 2.9V, the device attempts to operate, but the specifications are not ensured.
MINIMUM ON TIME and PULSE SKIPPING
Once the PWM switch is turned on, the TPS61175 has minimum ON pulse width of 60-ns. This sets the limit of
the minimum duty cycle of the PWM switch, and it is independent of the set switching frequency. When operating
conditions result in the TPS61175 having a minimum ON pulse width less than 60-ns, the IC enters
pulse-skipping mode. In this mode, the device keeps the power switch off for several switching cycles to keep the
output voltage from rising above the regulated voltage. This operation typically occurs in light load condition
when the PWM operates in discontinuous mode. Pulse skipping increases the output voltage ripple, see the
Figure 14.
10
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APPLICATION INFORMATION
The following section provides a step-by-step design approach for configuring the TPS61175 as a voltage
regulating boost converter, as shown in Figure 14. When configured as SEPIC or flyback converter, a different
design approach is required.
VIN
D1
L1
VOUT
C2
C1
TPS61175
VIN
SW
EN
SW
FREQ
SS
FB
PGND
COMP PGND
R4
C3
R3
Syn
AGND
R1
R2
PGND
NC
C4
Figure 14. Boost Converter Configuration
DETERMINING THE DUTY CYCLE
The TPS61175 has a maximum worst case duty cycle of 89% and a minimum on time of 60 ns. These two
constraints place limitations on the operating frequency that can be used for a given input to output conversion
ratio. The duty cycle at which the converter operates is dependent on the mode in which the converter is running.
If the converter is running in discontinuous conduction mode (DCM), where the inductor current ramps to zero at
the end of each cycle, the duty cycle varies with changes to the load much more than it does when running in
continuous conduction mode (CCM). In continuous conduction mode, where the inductor maintains a dc current,
the duty cycle is related primarily to the input and output voltages as computed below:
V
+ VD - VIN
D = OUT
VOUT + VD
(1)
In discontinuous mode the duty cycle is a function of the load, input and output voltages, inductance and
switching frequency as computed below:
2 ´ (VOUT + VD ) ´ IOUT ´ L ´ ¦ SW
D=
VIN 2
(2)
All converters using a diode as the freewheeling or catch component have a load current level at which they
transition from discontinuous conduction to continuous conduction. This is the point where the inductor current
just falls to zero. At higher load currents, the inductor current does not fall to zero but remains flowing in a
positive direction and assumes a trapezoidal wave shape as opposed to a triangular wave shape. This load
boundary between discontinuous conduction and continuous conduction can be found for a set of converter
parameters as follows.
IOUT(crit) =
(VOUT + VD - VIN ) ´ VIN2
2 ´ (VOUT + VD ) 2 ´ ¦ SW ´ L
(3)
For loads higher than the result of the equation above, the duty cycle is given by Equation 1 and for loads less
that the results of Equation 2, the duty cycle is given Equation 3. For Equation 1 through Equation 3, the variable
definitions are as follows.
• VOUT is the output voltage of the converter in V
• VD is the forward conduction voltage drop across the rectifier or catch diode in V
• VIN is the input voltage to the converter in V
• IOUT is the output current of the converter in A
• L is the inductor value in H
• fSW is the switching frequency in Hz
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Unless otherwise stated, the design equations that follow assume that the converter is running in continuous
mode.
SELECTING THE INDUCTOR
The selection of the inductor affects steady state operation as well as transient behavior and loop stability. These
factors make it the most important component in power regulator design. There are three important inductor
specifications, inductor value, DC resistance and saturation current. Considering inductor value alone is not
enough.
Inductor values can have ±20% tolerance with no current bias. When the inductor current approaches saturation
level, its inductance can fall to some percentage of its 0-A value depending on how the inductor vendor defines
saturation current. For CCM operation, the rule of thumb is to choose the inductor so that its inductor ripple
current (ΔIL) is no more than a certain percentage (RPL% = 20–40%) of its average DC value (IIN(AVG) = IL(AVG))
V ´ D
(VOUT + VD - VIN ) ´ (1 - D)
1
=
=
D IL = IN
L ´ ¦SW
L ´ ¦S W
é
æ
1
1 öù
+
êL ´ ¦ SW ´ ç
÷ú
êë
è VO UT + VD - VIN VIN ø úû
PO UT
£ RPL% ´
VIN ´ ηes t
(4)
Rearranging and solving for L gives
ηest ´ VIN
L £
é
æ
1
1 öù
+
ê ¦S W ç
÷ ú ´ RPL% POUT
êë
è VOUT + VD - VIN VIN ø úû
(5)
Choosing the inductor ripple current to closer to 20% of the average inductor current results in a larger
inductance value, maximizes the converter’s potential output current and minimizes EMI. Choosing the inductor
ripple current closer to 40% of IL(AVG) results in a smaller inductance value, and a physically smaller inductor,
improves transient response but results in potentially higher EMI and lower efficiency if the DCR of the smaller
packaged inductor is significantly higher. Using an inductor with a smaller inductance value than computed
above may result in the converter operating in DCM. This reduces the boost converter’s maximum output current,
causes larger input voltage and output ripple and typically reduces efficiency. Table 3 lists the recommended
inductor for the TPS61175.
Table 3. Recommended Inductors for TPS61175
PART NUMBER
L
(mH)
DCR MAX
(mΩ)
SATURATION CURRENT
(A)
SIZE
(L × W × H mm)
VENDOR
D104C2
10
44
3.6
10.4x10.4x4.8
TOKO
VLF10040
15
42
3.1
10.0x9.7x4.0
TDK
CDRH105RNP
22
61
2.9
10.5x10.3x5.1
Sumida
MSS1038
15
50
3.8
10.0x10.2x3.8
Coilcraft
The device has built-in slope compensation to avoid subharmonic oscillation associated with current mode
control. If the inductor value is lower than 4.7mH, the slope compensation may not be adequate, and the loop can
be unstable. Applications requiring inductors above 47mH have not been evaluated. Therefore, the user is
responsible for verifying operation if they select an inductor that is outside the 4.7mH–47mH recommended range.
COMPUTING THE MAXIMUM OUTPUT CURRENT
The over-current limit for the integrated power FET limits the maximum input current and thus the maximum input
power for a given input voltage. Maximum output power is less than maximum input power due to power
conversion losses. Therefore, the current limit setting, input voltage, output voltage and efficiency can all change
the maximum current output (IOUT(MAX)). The current limit clamps the peak inductor current, therefore the ripple
has to be subtracted to derive maximum DC current.
12
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IOUT(max) =
VIN(min) ´ ηest
VOUT
=
VIN(min) ´ ILIM ´ (1 - %RPL/2) ´ ηest
VOUT
(6)
where
ILIM = over current limit
hest= efficiency estimate based on similar applications or computed above
For instance, when VIN = 12-V is boosted to VOUT = 24-V, the inductor is 10-uH, the Schottky forward voltage is
0.4-V and the switching frequency is 1.2-MHz; then the maximum output current is 1.2-A in typical condition,
assuming 90% efficiency and a %RPL = 20%.
SETTING OUTPUT VOLTAGE
To set the output voltage in either DCM or CCM, select the values of R1 and R2 according to the following
equation.
æ R1
ö
Vout = 1.229 V ´ ç
+ 1÷
è R2
ø
æ Vout
ö
- 1÷
R1 = R2 ´ ç
1.229V
è
ø
(7)
Considering the leakage current through the resistor divider and noise decoupling into FB pin, an optimum value
for R2 is around 10k. The output voltage tolerance depends on the VFB accuracy and the tolerance of R1 and
R2.
SETTING THE SWITCHING FREQUENCY
Choose the appropriate resistor from the resistance versus frequency table Table 1 or graph Figure 13. A resistor
must be placed from the FREQ pin to ground, even if an external oscillation is applied for synchronization.
Increasing switching frequency reduces the value of external capacitors and inductors, but also reduces the
power conversion efficiency. The user should set the frequency for the minimum tolerable efficiency.
SETTING THE SOFT START TIME
Choose the appropriate capacitor from the soft start table Table 2. Increasing the soft start time reduces the
overshoot during start-up.
SELECTING THE SCHOTTKY DIODE
The high switching frequency of the TPS61175 demands a high-speed rectification for optimum efficiency.
Ensure that the diode’s average and peak current rating exceed the average output current and peak inductor
current. In addition, the diode’s reverse breakdown voltage must exceed the switch FET rating voltage of 40V.
So, the VISHAY SS3P6L-E3/86A is recommended for TPS61175. The power dissipation of the diode's package
must be larger than IOUT(max) x VD
SELECTING THE INPUT AND OUTPUT CAPACITORS
The output capacitor is mainly selected to meet the requirements for the output ripple and load transient. Then
the loop is compensated for the output capacitor selected. The output ripple voltage is related to the capacitor’s
capacitance and its equivalent series resistance (ESR). Assuming a capacitor with zero ESR, the minimum
capacitance needed for a given ripple can be calculated by
Cout =
(VOUT
- VIN )Iout
VOUT ´ Fs ´ Vripple
(8)
where, Vripple= peak to peak output ripple. The additional output ripple component caused by ESR is calculated
using:
Vripple_ESR = I × RESR
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Due to its low ESR, Vripple_ESR can be neglected for ceramic capacitors, but must be considered if tantalum or
electrolytic capacitors are used.
The minimum ceramic output capacitance needed to meet a load transient requirement can be estimated by the
equation below:
ΔITRAN
COUT =
2 ´ p ´ fLOOP-BW ´ ΔVTRAN
(9)
Where
ΔITRAN is the transient load current step
ΔVTRAN is the allowed voltage dip for the load current step
fLOOP-BW is the control loop bandwidth (i.e., the frequency where the control loop gain crosses zero).
Care must be taken when evaluating a ceramic capacitor’s derating under dc bias, aging and AC signal. For
example, larger form factor capacitors (in 1206 size) have their self resonant frequencies in the range of the
switching frequency. So the effective capacitance is significantly lower. The DC bias can also significantly reduce
capacitance. Ceramic capacitors can loss as much as 50% of its capacitance at its rated voltage. Therefore, one
must add margin on the voltage rating to ensure adequate capacitance at the required output voltage.
For a typical boost converter implementation, at least 4.7mF of ceramic input and output capacitance is
recommended. Additional input and output capacitance may be required to meet ripple and/or transient
requirements.
The popular vendors for high value ceramic capacitors are:
TDK (http://www.component.tdk.com/components.php)
Murata (http://www.murata.com/cap/index.html)
COMPENSATING THE SMALL SIGNAL CONTROL LOOP
All continuous mode boost converters have a right half plane zero (ƒRHPZ) due to the inductor being removed
from the output during charging. In a traditional voltage mode controlled boost converter, the inductor and output
capacitor form a small signal double pole. For a negative feedback system to be stable, the fed back signal must
have a gain less than 1 before having 180 degrees of phase shift. With its double pole and RHPZ all providing
phase shift, voltage mode boost converters are a challenge to compensate. In a converter with current mode
control, there are essentially two loops, an inner current feedback loop created by the inductor current
information sensed across RSENSE (40mΩ) and the output voltage feedback loop. The inner current loop allows
the switch, inductor and modulator to be lumped together into a small signal variable current source controlled by
the error amplifier, as shown in Figure 15.
(1-D)
RSENSE
R1
+
_
C2
Vref
C4
RO
2
C5
(optional)
R3
R2
RESR
Figure 15. Small Signal Model of a Current Mode Boost in CCM
The new power stage, including the slope compensation, small signal model becomes:
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æ
öæ
s
s
ç1 +
÷ ç1 2 ´ p ´ ¦ESR ø è
2 ´ p ´ ¦RHPZ
ROUT ´ (1 - D) è
´
GPS (s) =
s
2 ´ RSENSE
1+
2 ´ p ´ ¦P
ö
÷
ø ´ He(s)
(10)
Where
¦P =
2
2 p ´ RO ´ C2
¦ESR »
¦RHPZ =
(11)
1
2p ´ RESR ´ C2
(12)
2
æ V
ö
´ ç IN ÷
2p ´ L
è VOUT ø
RO
(13)
And
He(s) =
1
éæ
ù
Se ö
s ´ êç1 +
÷ ´ (1 - D) - 0.5ú
Sn ø
s2
ëè
û
+
1+
2
¦S W
( p ´ ¦ SW )
(14)
He(s) models the inductor current sampling effect as well as the slope compensation effect on the small signal
response. Note that if Sn > Se, e.g., when L is smaller than recommended, the converter operates as a voltage
mode converter and the above model no longer holds.
The slope compensation in TPS61175 is shown as follow
V
+ VD - VIN
Sn = OUT
´ RSENSE
L
0.32 V / R4
0.5 mA
Se =
+
16 ´ (1 - D )´ 6pF
6 pF
Where R4 is the frequency setting resistor
(15)
(16)
Figure 16 shows a bode plot of a typical CCM boost converter power stage
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180
120
Gain
0
Phase – °
Gain − dB
60
–60
Phase
–120
–180
fP
f − Frequency − kHz
Figure 16. Bode Plot of Power Stage Gain and Phase
The TPS61175 COMP pin is the output of the internal trans-conductance amplifier. Equation 17 shows the
equation for feedback resistor network and the error amplifier.
s
1+
2 ´ p ´ ¦Z
R2
HEA = GEA ´ REA ´
´
R2 + R1 æ
ö æ
ö
s
s
ç1 +
÷ ´ ç1 +
÷
2
2
´
p
´
¦
´
p
´
¦
P1 ø è
P2 ø
è
(17)
where GEA and REA are the amplifier’s trans-conductance and output resistance located in the ELECTRICAL
CHARACTERISTICS table.
¦ P1 =
¦P2
1
2 p ´ R EA ´ C4
(18)
1
=
(optional)
2p ´ R3 ´ C5
C5 is optional and can be modeled as 10 pF stray capacitance.
(19)
and
¦Z =
1
2p ´ R3 ´ C4
(20)
Figure 17 shows a typical bode plot for transfer function H(s).
16
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180
90
Phase – °
Gain − dB
Phase
0
Kcomp
–90
Gain
–180
<–fp1
fC
fZ
f − Frequency − kHz
fp2
Figure 17. Bode Plot of Feedback Resistors and Compensated Amplifier Gain and Phase
The next step is to choose the loop crossover frequency, fC. The higher in frequency that the loop gain stays
above zero before crossing over, the faster the loop response will be and therefore the lower the output voltage
will droop during a step load. It is generally accepted that the loop gain cross over no higher than the lower of
either 1/5 of the switching frequency, fSW, or 1/3 of the RHPZ frequency, fRHPZ. To approximate a single pole
roll-off up to fP2, select R3 so that the compensation gain, KCOMP, at fC on Figure 17 is the reciprocal of the gain,
KPW, read at frequency fC from the Figure 16 bode plot or more simply
KCOMP(fC) = 20 × log(GEA × R3 × R2/(R2+R1)) = 1/KPW(fC)
This makes the total loop gain, T(s) = GPS(s) × HEA(s), zero at the fC. Then, select C4 so that fZ ≅ fC/10 and
optional fP2> fC *10. Following this method should lead to a loop with a phase margin near 45 degrees. Lowering
R3 while keeping fZ ≅ fC/10 increases the phase margin and therefore increases the time it takes for the output
voltage to settle following a step load.
In the TPS61175, if the FB pin voltage changes suddenly due to a load step on the output voltage, the error
amplifier increases its transconductance for 8-ms in an effort to speed up the IC’s transient response and reduce
output voltage droop due to the load step. For example, if the FB voltage decreases 10-mV due to load change,
the error amplifier increases its source current through COMP by 5 times; if FB voltage increases 11-mV, the
sink current through COMP is increased to 3.5 times normal value. This feature often results in saw tooth ringing
on the output voltage, shown as Figure 9. Designing the loop for greater than 45 degrees of phase margin and
greater than 10db gain margin minimizes the amplitude of this ringing. This feature is disabled during soft start.
LAYOUT CONSIDERATIONS
As for all switching power supplies, especially those running at high switching frequency and high currents,
layout is an important design step. If layout is not carefully done, the regulator could suffer from instability as well
as noise problems. To maximize efficiency, switch rise and fall times are very fast. To prevent radiation of high
frequency noise (eg. EMI), proper layout of the high frequency switching path is essential. Minimize the length
and area of all traces connected to the SW pin and always use a ground plane under the switching regulator to
minimize interplane coupling. The high current path including the switch, Schottky diode, and output capacitor,
contains nanosecond rise and fall times and should be kept as short as possible. The input capacitor needs not
only to be close to the VIN pin, but also to the GND pin in order to reduce the Iinput supply ripple.
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VIN
INPUT
CAPACITOR
VOUT
INDUCTOR
SCHOTTKEY
OUTPUT
CAPACITOR
SW
Minimize the area
of SW trace
SW
PGND
SW
PGND
VIN
PGND
PGND
SS
Thermal Pad
EN
NC
FREQ
SYNC
FB
AGND
COMP
Place enough
VIAs around
thermal pad to
enhance thermal
performance
AGND
FEEDBACK
COMPESNATION
NETWORK
Figure 18. TPS61175 Layout
THERMAL CONSIDERATIONS
The maximum IC junction temperature should be restricted to 125°C under normal operating conditions. This
restriction limits the power dissipation of the TPS61175. Calculate the maximum allowable dissipation, PD(max),
and keep the actual dissipation less than or equal to PD(max). The maximum-power-dissipation limit is
determined using the following equation:
125 °C - TA
PD(max) =
R qJA
(21)
where, TA is the maximum ambient temperature for the application. RqJA is the thermal resistance
junction-to-ambient given in Power Dissipation Table.
The TPS61175 comes in a thermally enhanced TSSOP package. This package includes a thermal pad that
improves the thermal capabilities of the package. The RqJA of the TSSOP package greatly depends on the PCB
layout and thermal pad connection. The thermal pad must be soldered to the analog ground on the PCB. Using
thermal vias underneath the thermal pad.
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REVISION HISTORY
Changes from Original (December 2008) to Revision A
•
Page
Changed the Ordering Information table - Part Number From: TPS61175 To: TPS61175PWP; Removed the
Package Marking column ..................................................................................................................................................... 2
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PACKAGE OPTION ADDENDUM
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13-Oct-2010
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package
Drawing
Pins
Package Qty
Eco Plan
(2)
Lead/
Ball Finish
MSL Peak Temp
(3)
Samples
(Requires Login)
TPS61175PWP
ACTIVE
HTSSOP
PWP
14
90
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Purchase Samples
TPS61175PWPR
ACTIVE
HTSSOP
PWP
14
2000
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Request Free Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
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continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
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