Dual 8-,10-,12-Bit High Bandwidth Multiplying DACs with Serial Interface AD5429/AD5439/AD5449 FEATURES FUNCTIONAL BLOCK DIAGRAM VREFA AD5429/AD5439/AD5449 RFB R VDD RFBA SYNC SCLK SHIFT REGISTER INPUT REGISTER DAC REGISTER IOUT1A 8-/10-/12-BIT R-2R DAC A IOUT2A SDIN SDO LDAC CLR POWER-ON RESET INPUT REGISTER DAC REGISTER IOUT1B 8-/10-/12-BIT R-2R DAC B IOUT2B RFB R LDAC RFBB VREFB 04464-0-001 10 MHz multiplying bandwidth 50 MHz serial interface 2.5 V to 5.5 V supply operation ±10 V reference input Pin compatible 8-, 10-, and 12-bit DACs Extended temperature range: −40°C to +125°C 16-lead TSSOP package Guaranteed monotonic Power-on reset Daisy-chain mode Readback function 0.5 µA typical current consumption Figure 1. APPLICATIONS Portable battery-powered applications Waveform generators Analog processing Instrumentation applications Programmable amplifiers and attenuators Digitally controlled calibration Programmable filters and oscillators Composite video Ultrasound Gain, offset, and voltage trimming GENERAL DESCRIPTION The AD5429/AD5439/AD54491 are CMOS 8-, 10-, and 12-bit dual-channel current output digital-to-analog converters, respectively. These devices operate from a 2.5 V to 5.5 V power supply, making them suited to battery-powered and other applications. The applied external reference input voltage (VREF) determines the full-scale output current. An integrated feedback resistor (RFB) provides temperature tracking and full-scale voltage output when combined with an external current-to-voltage precision amplifier. These DACs utilize a double-buffered, 3-wire serial interface that is compatible with SPI®, QSPI™, MICROWIRE™, and most DSP interface standards. In addition, a serial data out pin (SDO) allows daisy-chaining when multiple packages are used. Data readback allows the user to read the contents of the DAC register via the SDO pin. On power-up, the internal shift register and latches are filled with zeros and the DAC outputs are at zero scale. As a result of manufacture on a CMOS submicron process, these parts offer excellent 4-quadrant multiplication characteristics, with large signal multiplying bandwidths of 10 MHz. The AD5429/AD5439/AD5449 DAC are available in 16-lead TSSOP packages. 1 US Patent Number 5,689,257. Rev. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 © 2004 Analog Devices, Inc. All rights reserved. AD5429/AD5439/AD5449 TABLE OF CONTENTS Specifications..................................................................................... 3 Adding Gain................................................................................ 18 Timing Characteristics..................................................................... 5 Divider or Programmable Gain Element ................................ 18 Absolute Maximum Ratings............................................................ 7 Reference Selection .................................................................... 19 ESD Caution.................................................................................. 7 Amplifier Selection .................................................................... 19 Pin Configuration and Function Descriptions............................. 8 Serial Interface ................................................................................ 20 Terminology ...................................................................................... 9 Microprocessor Interfacing....................................................... 22 Typical Performance Characteristics ........................................... 10 PCB Layout and Power Supply Decoupling................................ 24 General Description ....................................................................... 15 Power Supplies for the Evaluation Board................................ 24 Unipolar Mode............................................................................ 15 Evaluation Board for the DACs................................................ 24 Bipolar Operation....................................................................... 16 Overview of AD54xx Devices....................................................... 28 Stability ........................................................................................ 16 Outline Dimensions ....................................................................... 29 Single-Supply Applications........................................................ 17 Ordering Guide .......................................................................... 29 Positive Output Voltage ............................................................. 17 REVISION HISTORY 7/04—Revision 0: Initial Version Rev. 0 | Page 2 of 32 AD5429/AD5439/AD5449 SPECIFICATIONS VDD = 2.5 V to 5.5 V, VREF = 10 V, IOUT2A, IOUT2B = 0 V. All specifications TMIN to TMAX, unless otherwise noted. DC performance measured with OP1177, ac performance with AD9631, unless otherwise noted. Temperature range for Y version is −40°C to +125°C. Table 1. Parameter STATIC PERFORMANCE AD5429 Resolution Relative Accuracy Differential Nonlinearity AD5439 Resolution Relative Accuracy Differential Nonlinearity AD5449 Resolution Relative Accuracy Differential Nonlinearity Gain Error Gain Error Temp Coefficient1 Output Leakage Current REFERENCE INPUT1 Reference Input Range VREFA,VREFB Input Resistance VREFA/B Input Resistance Mismatch DIGITAL INPUTS/OUTPUT1 Input High Voltage, VIH Input Low Voltage, VIL Input Leakage Current, IIL Input Capacitance VDD = 4.5 V to 5.5 V Output Low Voltage, VOL Output High Voltage, VOH VDD = 2.5 V to 3.6 V Output Low Voltage, VOL Output High Voltage, VOH DYNAMIC PERFORMANCE1 Reference Multiplying BW Output Voltage Settling Time AD5429 AD5439 AD5449 Digital Delay Digital-to-Analog Glitch Impulse Multiplying Feedthrough Error Min Typ Max Unit Conditions 8 ±0.5 ±1 Bits LSB LSB Guaranteed monotonic 10 ±0.5 ±1 Bits LSB LSB Guaranteed monotonic 12 ±1 −1/+2 ±10 ±5 ±10 Bits LSB LSB mV ppm FSR/°C nA nA ±10 10 1.6 12 2.5 V kΩ % 0.8 0.7 1 10 V V V µA pF VDD = 2.5 V to 5.5 V VDD = 2.7 V to 5.5 V VDD = 2.5 V to 2.7 V 0.4 V V ISINK = 200 µA ISOURCE = 200 µA 0.4 V V ISINK = 200 µA ISOURCE = 200 µA 10 MHz 50 55 100 110 ns ns 90 20 3 160 40 ns ns nV-s ±5 8 1.7 VDD − 1 VDD − 0.5 −75 dB Rev. 0 | Page 3 of 32 Guaranteed monotonic Data = 0000H, TA = 25°C, IOUT1 Data = 0000H, IOUT1 Typical resistor TC = −50 ppm/°C DAC input resistance Typ = 25°C, max = 125°C VREF = 5 V p-p, DAC loaded all 1s Measured to ±4 mV of FS, RLOAD = 100 Ω, CLOAD = 0s DAC latch alternately loaded with 0s and 1s RLOAD = 100 Ω, CLOAD = 15 pF 1 LSB change around major carry, VREF = 0 V DAC latch loaded with all 0s, reference = 10 kHz AD5429/AD5439/AD5449 Parameter Output Capacitance Digital Feedthrough 5 Unit pF pF nV-s Total Harmonic Distortion −75 −75 dB dB 25 nV/√Hz 55 63 65 dB dB dB 50 60 62 dB dB dB Output Noise Spectral Density SFDR PERFORMANCE (Wideband) Clock = 10 MHz 500 kHz fout 100 kHz fout 50 kHz fout Clock = 25 MHz 500 kHz fout 100 kHz fout 50 kHz fout SFDR PERFORMANCE (Narrow Band) Clock = 10 MHz 500 kHz fout 100 kHz fout 50 kHz fout Clock = 25 MHz 500 kHz fout 100 kHz fout 50 kHz fout INTERMODULATION DISTORTION Clock = 10 MHz f1 = 400 kHz, f2 = 500 kHz f1 = 40 kHz, f2 = 50 kHz Clock = 25 MHz f1 = 400 kHz, f2 = 500 kHz f1 = 40 kHz, f2 = 50 kHz POWER REQUIREMENTS Power Supply Range IDD Power Supply Sensitivity1 1 Min Typ Max 2 4 Conditions DAC latches loaded with all 0s DAC latches loaded with all 1s Feedthrough to DAC output with CS high and alternate loading of all 0s and all 1s VREF = 5 V p-p, all 1s loaded, f = 1 kHz VREF = 5 V, sine wave generated from digital code @ 1 kHz AD5449, 65 k codes, VREF = 3.5 V AD5449, 65 k codes, VREF = 3.5 V 73 80 87 dB dB dB 70 75 80 dB dB dB AD5449, 65 k codes, VREF = 3.5 V 65 72 dB dB 51 65 dB dB 2.5 5.5 10 0.001 V µA %/% Guaranteed by design and characterization, not subject to production test. Rev. 0 | Page 4 of 32 Logic inputs = 0 V or VDD ∆VDD = ±5% AD5429/AD5439/AD5449 TIMING CHARACTERISTICS VDD = 2.5 V to 5.5 V, VREF = 5 V, IOUT2 = 0 V. All specifications TMIN to TMAX, unless otherwise noted. See Figure 2 and Figure 3. Temperature range for Y version is −40°C to +125°C. Guaranteed by design and characterization, not subject to production test. All input signals are specified with tr = tf = ns (10% to 90% of VDD) and timed from a voltage level of (VIL + VIH)/2. Table 2. Parameter fSCLK t1 t2 t3 t4 t5 t6 t7 t8 t9 t10 t11 t122 1 2 Limit at TMIN, TMAX 50 20 8 8 13 5 4 5 30 0 12 10 25 60 Unit MHz max ns min ns min ns min ns min ns min ns min ns min ns min ns min ns min ns min ns min ns min Conditions/Comments1 Max clock frequency SCLK cycle time SCLK high time SCLK low time SYNC falling edge to SCLK falling edge setup time Data setup time Data hold time SYNC rising edge to SCLK falling edge Minimum SYNC high time SCLK falling edge to LDAC falling edge LDAC pulse width SCLK falling edge to LDAC rising edge SCLK active edge to SDO valid, strong SDO driver SCLK active edge to SDO valid, weak SDO driver Falling or rising edge as determined by the control bits of the serial word. Strong or weak SDO driver selected via the control register. Daisy-chain and readback modes cannot operate at maximum clock frequency. SDO timing specifications are measured with a load circuit, as shown in Figure 4. Rev. 0 | Page 5 of 32 AD5429/AD5439/AD5449 t1 SCLK t2 t4 t8 t3 t7 SYNC t6 t5 DB0 DB15 DIN t9 t10 LDAC1 t11 LDAC2 04464-0-002 NOTES 1ASYNCHRONOUS LDAC UPDATE MODE 2SYNCHRONOUS LDAC UPDATE MODE ALTERNATIVELY, DATA CAN BE CLOCKED INTO INPUT SHIFT REGISTER ON RISING EDGE OF SCLK AS DETERMINED BY CONTROL BITS. TIMING AS ABOVE, WITH SCLK INVERTED. Figure 2. Standalone Mode Timing Diagram t1 SCLK t2 t4 t3 t7 SYNC t6 t8 t5 SDIN DB0 (N) DB15 (N) DB15 (N+1) DB0 (N+1) DB15 (N) DB0 (N) SDO ALTERNATIVELY, DATA CAN BE CLOCKED INTO INPUT SHIFT REGISTER ON RISING EDGE OF SCLK AS DETERMINED BY CONTROL BITS. IN THIS CASE, DATA WOULD BE CLOCKED OUT OF SDO ON FALLING EDGE OF SCLK. TIMING AS ABOVE, WITH SCLK INVERTED. Figure 3. Daisy-Chain and Readback Modes Timing Diagram 200µA VOH (MIN) + VOL (MAX) 2 CL 50pF 200µA IOH Figure 4. Load Circuit for SDO Timing Specifications Rev. 0 | Page 6 of 32 04464-0-004 TO OUTPUT PIN IOL 04464-0-003 t12 AD5429/AD5439/AD5449 ABSOLUTE MAXIMUM RATINGS Table 3. Parameter VDD to GND VREF, RFB to GND IOUT1, IOUT2 to GND Input Current to Any Pin except Supplies Logic Inputs and Output1 Operating Temperature Range Extended (Y Version) Storage Temperature Range Junction Temperature 16-Lead TSSOP θJA Thermal Impedance Lead Temperature, Soldering (10 s) IR Reflow, Peak Temperature (< 20 s) 1 Rating −0.3 V to +7 V −12 V to +12 V −0.3 V to +7 V ±10 mA −0.3 V to VDD + 0.3 V −40°C to +125°C −65°C to +150°C 150°C 150°C/W 300°C 235°C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Only one absolute maximum rating may be applied at any one time. Transient currents of up to 100 mA do not cause SCR latch-up. TA = 25°C unless otherwise noted. Overvoltages at SCLK, SYNC, and DIN are clamped by internal diodes. Current should be limited to the maximum ratings given. ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. 0 | Page 7 of 32 AD5429/AD5439/AD5449 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS IOUT1A 1 16 IOUT2A 2 15 IOUT2B RFBA 3 14 RFBB VREFA 4 GND 5 LDAC 6 SCLK 7 10 SDIN 8 9 TOP VIEW (Not to Scale) 13 VREFB 12 VDD 11 CLR SYNC SDO NC = NO CONNECT 04464-0-005 AD5429/ AD5439/ AD5449 IOUT1B Figure 5. Pin Configuration Table 4. Pin Function Descriptions Pin No. 1 2 Mnemonic IOUT1A IOUT2A 3 4 5 6 RFBA VREFA GND LDAC 7 SCLK 8 SDIN 9 SDO 10 SYNC 11 CLR 12 13 14 15 VDD VREFB RFBB IOUT2B 16 IOUT1B Function DAC A Current Output. DAC A Analog Ground. This pin should typically be tied to the analog ground of the system, but can be biased to achieve single-supply operation. DAC Feedback Resistor Pin. Establish voltage output for the DAC by connecting to an external amplifier output. DAC A Reference Voltage Input Pin. Ground Pin. Load DAC Input. Allows asynchronous or synchronous updates to the DAC output. The DAC is asynchronously updated when this signal goes low. Alternatively, if this line is held permanently low, an automatic or synchronous update mode is selected whereby the DAC is updated on the 16th clock falling edge when the device is in standalone mode, or on the rising edge of SYNC when in daisy-chain mode. Serial Clock Input. By default, data is clocked into the input shift register on the falling edge of the serial clock input. Alternatively, by means of the serial control bits, the device can be configured such that data is clocked into the shift register on the rising edge of SCLK. Serial Data Input. Data is clocked into the 16-bit input register on the active edge of the serial clock input. By default, on power-up, data is clocked into the shift register on the falling edge of SCLK. The control bits allow the user to change the active edge to rising edge. Serial Data Output. This allows a number of parts to be daisy-chained. By default, data is clocked into the shift register on the falling edge and out via SDO on the rising edge of SCLK. Data is always clocked out on the alternate edge to loading data to the shift register. Writing the readback control word to the shift register makes the DAC register contents available for readback on the SDO pin, clocked out on the next 16 opposite clock edges to the active clock edge. Active Low Control Input. This is the frame synchronization signal for the input data. When SYNC goes low, it powers on the SCLK and DIN buffers, and the input shift register is enabled. Data is loaded to the shift register on the active edge of the following clocks. In standalone mode, the serial interface counts clocks, and data is latched to the shift register on the16th active clock edge. Active Low Control Input. This pin clears the DAC output, input, and DAC registers. Configuration mode allows the user to enable the hardware CLR pin as a clear to zero scale or midscale as required. Positive Power Supply Input. These parts can be operated from a supply of 2.5 V to 5.5 V. DAC B Reference Voltage Input Pin. DAC B Feedback Resistor Pin. Establish voltage output for the DAC by connecting to an external amplifier output. DAC B Analog Ground. This pin typically should be tied to the analog ground of the system, but can be biased to achieve single-supply operation. DAC B Current Output. Rev. 0 | Page 8 of 32 AD5429/AD5439/AD5449 TERMINOLOGY Relative Accuracy Relative accuracy or endpoint nonlinearity is a measure of the maximum deviation from a straight line passing through the endpoints of the DAC transfer function. It is measured after adjusting for zero and full scale and is typically expressed in LSBs or as a percentage of full-scale reading. Differential Nonlinearity Differential nonlinearity is the difference between the measured change and the ideal 1 LSB change between any two adjacent codes. A specified differential nonlinearity of ±1 LSB maximum over the operating temperature range ensures monotonicity. Gain Error Gain error or full-scale error is a measure of the output error between an ideal DAC and the actual device output. For these DACs, ideal maximum output is VREF − 1 LSB. Gain error of the DACs is adjustable to zero with external resistance. Output Leakage Current Output leakage current is current that flows in the DAC ladder switches when these are turned off. For the IOUT1 terminal, it can be measured by loading all 0s to the DAC and measuring the IOUT1 current. Minimum current flows in the IOUT2 line when the DAC is loaded with all 1s. Output Capacitance Capacitance from IOUT1 or IOUT2 to AGND. Digital Crosstalk The glitch impulse transferred to the outputs of one DAC in response to a full-scale code change (all 0s to all 1s and vice versa) in the input register of the other DAC. It is expressed in nV-s. Analog Crosstalk The glitch impulse transferred to the output of one DAC due to a change in the output of another DAC. It is measured by loading one of the input registers with a full-scale code change (all 0s to all 1s and vice versa), while keeping LDAC high. Then pulse LDAC low and monitor the output of the DAC whose digital code was not changed. The area of the glitch is expressed in nV-s. Channel-to-Channel Isolation The proportion of input signal from the reference input of one DAC that appears at the output of the other DAC. It is expressed in dB. Total Harmonic Distortion (THD) The DAC is driven by an ac reference. The ratio of the rms sum of the harmonics of the DAC output to the fundamental value is the THD. Usually only the lower-order harmonics are included, such as second to fifth. THD = 20 log Output Current Settling Time The amount of time needed for the output to settle to a specified level for a full-scale input change. For these devices, it is specified with a 100 Ω resistor to ground. Digital-to-Analog Glitch lmpulse The amount of charge injected from the digital inputs to the analog output when the inputs change state. This is normally specified as the area of the glitch in either pA-s or nV-s, depending upon whether the glitch is measured as a current or voltage signal. V1 Intermodulation Distortion The DAC is driven by two combined sine wave references of frequencies fa and fb. Distortion products are produced at sum and difference frequencies of mfa ± nfb, where m, n = 0, 1, 2, 3… Intermodulation terms are those for which m or n is not equal to zero. The second-order terms include (fa + fb) and (fa − fb) and the third-order terms are (2fa + fb), (2fa − fb), (f + 2fa + 2fb) and (fa − 2fb). IMD is defined as IMD = 20 log Digital Feedthrough When the device is not selected, high frequency logic activity on the device digital inputs is capacitively coupled through the device to show up as noise on the IOUT pins and subsequently into the following circuitry. This noise is digital feedthrough. (V22 +V32 + V42 + V52 ) (rms sum of the sum and diff distortion products ) rms amplitude of the fundamental Compliance Voltage Range The maximum range of (output) terminal voltage for which the device provides the specified characteristics. Multiplying Feedthrough Error The error due to capacitive feedthrough from the DAC reference input to the DAC IOUT1 terminal, when all 0s are loaded to the DAC. Rev. 0 | Page 9 of 32 AD5429/AD5439/AD5449 TYPICAL PERFORMANCE CHARACTERISTICS 0.20 0.20 0.10 0.05 0.05 DNL (LSB) 0.10 0 0 –0.05 –0.05 –0.10 –0.10 –0.15 –0.15 –0.20 0 50 TA = 25°C VREF = 10V VDD = 5V 0.15 100 150 200 250 CODE –0.20 04462-0-007 0 50 200 250 Figure 9. DNL vs. Code (8-Bit DAC) 0.5 TA = 25°C VREF = 10V VDD = 5V 0.4 0.3 0.3 0.2 0.1 0.1 DNL (LSB) 0.2 0 –0.1 0 –0.1 –0.2 –0.2 –0.3 –0.3 –0.4 –0.4 200 400 600 800 1000 CODE –0.5 04462-0-008 –0.5 0 TA = 25°C VREF = 10V VDD = 5V 0.4 0 200 400 600 800 1000 CODE Figure 7. INL vs. Code (10-Bit DAC) 04462-0-011 0.5 INL (LSB) 150 CODE Figure 6. INL vs. Code (8-Bit DAC) Figure 10. DNL vs. Code (10-Bit DAC) 1.0 1.0 TA = 25°C VREF = 10V VDD = 5V 0.8 0.6 0.6 0.4 0.2 0.2 DNL (LSB) 0.4 0 –0.2 0 –0.2 –0.4 –0.4 –0.6 –0.6 –0.8 –0.8 –1.0 0 500 1000 TA = 25°C VREF = 10V VDD = 5V 0.8 1500 2000 2500 3000 CODE 3500 4000 04462-0-009 INL (LSB) 100 Figure 8. INL vs. Code (12-Bit DAC) –1.0 0 500 1000 1500 2000 2500 3000 CODE Figure 11. DNL vs. Code (12-Bit DAC) Rev. 0 | Page 10 of 32 3500 4000 04462-0-012 INL (LSB) 0.15 04462-0-010 TA = 25°C VREF = 10V VDD = 5V AD5429/AD5439/AD5449 0.6 8 0.5 7 TA = 25°C 0.4 6 MAX INL CURRENT (mA) INL (LSB) 0.3 0.2 TA = 25°C VREF = 10V VDD = 5V 0.1 5 VDD = 5V 4 3 0 MIN INL 2 –0.1 –0.2 1 –0.3 0 VDD = 3V 3 4 5 6 7 8 9 10 REFERENCE VOLTAGE 0 Figure 12. INL vs. Reference Voltage 0.5 1.5 1.0 2.0 2.5 3.0 3.5 INPUT VOLTAGE (V) 4.0 4.5 5.0 04462-0-022 2 04462-0-013 VDD = 2.5V Figure 15. Supply Current vs. Logic Input Voltage –0.40 1.6 TA = 25°C VREF = 10V VDD = 5V –0.45 1.4 1.2 IOUT1 VDD 5V IOUT LEAKAGE (nA) DNL (LSB) –0.50 –0.55 –0.60 MIN DNL 1.0 0.8 IOUT1 VDD 3V 0.6 0.4 –0.65 3 4 5 6 7 8 9 10 REFERENCE VOLTAGE 0 –40 0 20 60 80 100 120 Figure 16. IOUT1 Leakage Current vs. Temperature 5 0.50 4 0.45 TA = 25°C VDD = 5V 3 40 TEMPERATURE (°C) Figure 13. DNL vs. Reference Voltage VDD = 5V 0.40 2 CURRENT (µA) 0.35 1 0 VDD = 2.5V –1 ALL 0s 0.30 0.15 0.10 VREF = 10V –5 –60 –40 –20 VDD = 2.5V 0.20 –3 –4 ALL 1s 0.25 –2 ALL 1s ALL 0s 0.05 0 20 40 60 80 100 TEMPERATURE (°C) 120 140 04462-0-015 ERROR (mV) –20 Figure 14. Gain Error vs. Temperature 0 –60 –40 –20 0 20 40 60 80 100 TEMPERATURE (°C) Figure 17. Supply Current vs. Temperature Rev. 0 | Page 11 of 32 120 140 04462-0-024 2 04462-0-014 –0.70 04462-0-023 0.2 AD5429/AD5439/AD5449 14 3 VDD = 5V 0 GAIN (dB) 8 6 VDD = 3V –3 4 VREF = ±2V, AD8038 CC 1.47pF VREF = ±2V, AD8038 CC 1pF VREF = ±0.15V, AD8038 CC 1pF VREF = ±0.15V, AD8038 CC 1.47pF VREF = ±3.51V, AD8038 CC 1.8pF –6 VDD = 2.5V 2 1 10 100 1k 10k 100k 1M 10M –9 10k 04462-0-025 0 100M FREQUENCY (Hz) 0.045 ALL ON DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DB3 DB2 DB1 DB0 10 100 1M 10M 100M OUTPUT VOLTAGE (V) TA = 25°C VDD = 5V VREF = ±3.5V INPUT CCOMP = 1.8pF AD8038 AMPLIFIER 1k 10k 100k FREQUENCY (Hz) 10M 100M TA = 25°C VREF = 0V AD8038 AMPLIFIER CCOMP = 1.8pF VDD = 5V 0.035 ALL OFF 1 7FF TO 800H 0.040 0.030 0.025 VDD = 3V 0.020 0.015 800 TO 7FFH 0.010 VDD = 3V 0.005 0 –0.005 04462-0-026 VDD = 5V –0.010 0 20 40 60 80 100 120 140 160 180 200 TIME (ns) Figure 19. Reference Multiplying Bandwidth vs. Frequency and Code Figure 22. Midscale Transition, VREF = 0 V 0.2 –1.68 TA = 25°C VREF = 3.5V AD8038 AMPLIFIER CCOMP = 1.8pF 7FF TO 800H –1.69 VDD = 5V OUTPUT VOLTAGE (V) –1.70 –0.2 –0.4 TA = 25°C VDD = 5V VREF = ±3.5V CCOMP = 1.8pF AD8038 AMPLIFIER –0.6 10 100 –1.72 VDD = 3V –1.73 VDD = 5V –1.74 VDD = 3V –1.76 800 TO 7FFH –0.8 1 –1.71 –1.75 1k 10k 100k 1M 10M 100M FREQUENCY (Hz) 04462-0-027 GAIN (dB) 0 –1.77 0 20 40 60 80 100 120 140 160 TIME (ns) Figure 23. Midscale Transition, VREF = 3.5 V Figure 20. Reference Multiplying Bandwidth–All 1s Loaded Rev. 0 | Page 12 of 32 180 200 04462-0-042 GAIN (dB) TA = 25°C LOADING ZS TO FS 1M FREQUENCY (Hz) Figure 21. Reference Multiplying Bandwidth vs. Frequency and Compensation Capacitor Figure 18. Supply Current vs. Update Rate 6 0 –6 –12 –18 –24 –30 –36 –42 –48 –54 –60 –66 –72 –78 –84 –90 –96 –102 100k 04462-0-028 10 04462-0-041 12 IDD (mA) TA = 25°C VDD = 5V TA = 25°C LOADING ZS TO FS AD5429/AD5439/AD5449 20 90 TA = 25°C VDD = 3V AMP = AD8038 0 80 MCLK = 5MHz 70 MCLK = 10MHz –20 SFDR (dB) PSRR (dB) 60 –40 FULL SCALE –60 ZERO SCALE 50 MCLK = 25MHz 40 30 –80 20 –100 TA = 25°C VREF = 3.5V AD8038 AMPLIFIER 1 100 10 1k 10k 100k 1M 10M FREQUENCY (Hz) 0 0 100 200 300 400 500 600 700 900 1000 fOUT (kHz) Figure 24. Power Supply Rejection vs. Frequency Figure 27. Wideband SFDR vs. fOUT Frequency –60 0 TA = 25°C VDD = 3V VREF = 3.5V p-p –65 800 04462-0-046 –120 04462-0-043 10 TA = 25°C VDD = 5V AMP = AD8038 65k CODES –10 –20 –30 SFDR (dB) THD + N (dB) –70 –75 –40 –50 –60 –80 –70 –85 1 10 100 1k 10k 100k 1M FREQUENCY (Hz) –90 04462-0-044 –90 0 2 4 6 8 FREQUENCY (MHz) 10 12 04462-0-047 –80 Figure 28. Wideband SFDR, fOUT = 100 kHz, Clock = 25 MHz Figure 25. THD + Noise vs. Frequency 100 0 MCLK = 1MHz TA= 25°C VDD = 5V AMP = AD8038 65k CODES –10 80 –20 SFDR (dB) MCLK = 200kHz 60 MCLK = 0.5MHz 40 –40 –50 –60 –70 –80 20 TA = 25°C VREF = 3.5V AD8038 AMPLIFIER 0 20 40 60 80 100 120 140 160 180 fOUT (kHz) 200 –90 –100 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 FREQUENCY (MHz) 4.0 4.5 5.0 Figure 29. Wideband SFDR, fOUT = 500 kHz, Clock = 10 MHz Figure 26. Wideband SFDR vs. fOUT Frequency Rev. 0 | Page 13 of 32 044620-048 0 04462-0-045 SFDR (dB) –30 AD5429/AD5439/AD5449 0 0 TA = 25°C VDD = 5V AMP = AD8038 65k CODES –10 –20 –20 –30 –40 –40 (dB) –50 –50 –60 –60 –70 –70 –80 –80 0.5 1.0 1.5 2.0 2.5 3.0 3.5 FREQUENCY (MHz) 4.0 4.5 5.0 Figure 30. Wideband SFDR, fOUT = 50 kHz, Clock = 10 MHz 0 –100 70 –20 85 95 90 100 105 FREQUENCY (MHz) 0 110 115 120 TA= 25°C VDD = 5V AMP = AD8038 65k CODES –10 –20 –30 –40 –40 (dB) –30 –50 –50 –60 –60 –70 –70 –80 –80 –90 –90 300 350 400 450 500 550 600 FREQUENCY (MHz) 650 700 750 04462-0-050 –100 250 80 Figure 33. Narrow-Band IMD, fOUT = 90 kHz, 100 kHz, Clock = 10 MHz TA= 25°C VDD = 3V AMP = AD8038 65k CODES –10 75 04462-0-052 0 04462-0-049 –90 –90 –100 0 Figure 31. Narrow-Band Spectral Response, fOUT = 500 kHz, Clock = 25 MHz 20 100 150 200 250 FREQUENCY (kHz) 300 350 400 Figure 34. Wideband IMD, fOUT = 90 kHz, 100 kHz, Clock = 25 MHz 300 TA= 25°C VDD = 3V AMP = AD8038 65k CODES 0 50 04462-0-053 SFDR (dB) –30 SFDR (dB) TA= 25°C VDD = 3V AMP = AD8038 65k CODES –10 TA = 25°C AMP = AD8038 ZERO SCALE LOADED TO DAC 250 MIDSCALE LOADED TO DAC OUTPUT NOISE (nV/ Hz) FULL SCALE LOADED TO DAC –40 –60 –80 150 100 50 60 70 80 90 100 110 120 FREQUENCY (MHz) 130 140 150 04462-0-051 –100 –120 50 200 0 100 1k 10k FREQUENCY (Hz) Figure 32. Narrow-Band SFDR, fOUT = 100 kHz, Clock = 25 MHz Figure 35. Output Noise Spectral Density Rev. 0 | Page 14 of 32 100k 04462-0-054 SFDR (dB) –20 AD5429/AD5439/AD5449 GENERAL DESCRIPTION The AD5429/AD5439/AD5449 are 8-, 10-, and 12-bit dualchannel current output DACs consisting of a standard inverting R−2R ladder configuration. A simplified diagram of one DAC channel for the AD5449 is shown in Figure 36. The feedback resistor RFB has a value of R. The value of R is typically 10 kΩ (minimum 8 kΩ and maximum 12 kΩ). If IOUT1 and IOUT2 are kept at the same potential, a constant current flows in each ladder leg, regardless of digital input code. Therefore, the input resistance presented at VREF is always constant. R R When an output amplifier is connected in unipolar mode, the output voltage is given by VOUT = − VREF × D / 2n where D is the fractional representation of the digital word loaded to the DAC, and n is the number of bits. D = 0 to 255 (AD5429) = 0 to 1023 (AD5439) = 0 to 4095 (AD5449) R VREFA 2R 2R 2R 2R S1 S2 S3 S12 With a fixed 10 V reference, the circuit shown in Figure 37 gives a unipolar 0 V to −10 V output voltage swing. When VIN is an ac signal, the circuit performs 2-quadrant multiplication. 2R 2R RFBA IOUT1A IOUT2A Table 5 shows the relationship between digital code and the expected output voltage for unipolar operation for the AD5429. 04464-0-006 DAC DATA LATCHES AND DRIVERS Figure 36. Simplified Ladder Table 5. Unipolar Code Table Access is provided to the VREF, RFB, IOUT1, and IOUT2 terminals of the DACs, making the devices extremely versatile and allowing them to be configured in several operating modes, such as unipolar mode, bipolar output mode, or single-supply mode. Digital Input 1111 1111 1000 0000 0000 0001 0000 0000 UNIPOLAR MODE Analog Output (V) −VREF (4095/4096) −VREF (2048/4096) = −VREF/2 −VREF (1/4096) −VREF (0/4096) = 0 Using a single op amp, these devices can easily be configured to provide 2-quadrant multiplying operation or a unipolar output voltage swing, as shown in Figure 37. VDD R2 VDD VREF VREF R1 AD5429/ AD5439/ AD5449 SYNC SCLK SDIN C1 RFBA IOUT1A A1 IOUT2A GND VOUT = 0V TO –VREF µCONTROLLER Figure 37. Unipolar Operation Rev. 0 | Page 15 of 32 04464-0-007 AGND NOTES: 1. R1 AND R2 USED ONLY IF GAIN ADJUSTMENT IS REQUIRED. 2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED IF A1 IS A HIGH SPEED AMPLIFIER. 3. DAC B OMITTED FOR CLARITY. AD5429/AD5439/AD5449 BIPOLAR OPERATION STABILITY In some applications, it might be necessary to generate full 4-quadrant multiplying operation or a bipolar output swing. This can be easily accomplished by using another external amplifier and three external resistors, as shown in Figure 38. In the I-to-V configuration, the IOUT of the DAC and the inverting node of the op amp must be connected as closely as possible, and proper PCB layout techniques must be employed. Because every code change corresponds to a step function, gain peaking can occur, if the op amp has limited GBP and there is excessive parasitic capacitance at the inverting node. This parasitic capacitance introduces a pole into the open loop response, which can cause ringing or instability in the closedloop applications circuit. When VIN is an ac signal, the circuit performs 4-quadrant multiplication. When connected in bipolar mode, the output voltage is ( ) VOUT = VREF × D / 2n −1 − VREF As shown in Figure 37 and Figure 38, an optional compensation capacitor, C1, can be added in parallel with RFB for stability. Too small a value of C1 can produce ringing at the output, while too large a value can adversely affect the settling time. C1 should be found empirically, but 1 pF to 2 pF is generally adequate for the compensation. where D is the fractional representation of the digital word loaded to the DAC, and n is the number of bits. D = 0 to 255 (AD5429) = 0 to 1023 (AD5439) = 0 to 4095 (AD5449) Table 6 shows the relationship between digital code and the expected output voltage for bipolar operation with the AD5429. Table 6. Bipolar Code Table Analog Output (V) +VREF (2047/2048) 0 −VREF (2047/2048) −VREF (2048/2048) R3 20kΩ VDD VDD R1 VREF ±10V VREF AD5429/ AD5439/ AD5449 SYNC SCLK SDIN µCONTROLLER R2 RFBA IOUT1A R5 20kΩ C1 A1 R4 10kΩ A2 IOUT2A VOUT = –VREF TO +VREF GND AGND NOTES: 1. R1 AND R2 USED ONLY IF GAIN ADJUSTMENT IS REQUIRED. ADJUST R1 FOR VOUT = 0V WITH CODE 10000000 LOADED TO DAC. 2. MATCHING AND TRACKING IS ESSENTIAL FOR RESISTOR PAIRS R3 AND R4. 3. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED, IF A1/A2 IS A HIGH SPEED AMPLIFIER. 4. DAC B AND ADDITIONAL PINS OMITTED FOR CLARITY. Figure 38. Bipolar Operation Rev. 0 | Page 16 of 32 04464-0-008 Digital Input 1111 1111 1000 0000 0000 0001 0000 0000 AD5429/AD5439/AD5449 SINGLE-SUPPLY APPLICATIONS POSITIVE OUTPUT VOLTAGE Voltage-Switching Mode The output voltage polarity is opposite to the VREF polarity for dc reference voltages. To achieve a positive voltage output, an applied negative reference to the input of the DAC is preferred over the output inversion through an inverting amplifier because of the resistor’s tolerance errors. To generate a negative reference, the reference can be level-shifted by an op amp such that the VOUT and GND pins of the reference become the virtual ground and −2.5 V, respectively, as shown in Figure 40. Figure 39 shows the DACs operating in voltage-switching mode. The reference voltage, VIN, is applied to the IOUT1 pin, IOUT2 is connected to AGND, and the output voltage is available at the VREF terminal. In this configuration, a positive reference voltage results in a positive output voltage, making single-supply operation possible. The output from the DAC is voltage at a constant impedance (the DAC ladder resistance). Therefore, an op amp is necessary to buffer the output voltage. The reference input no longer sees a constant input impedance, but one that varies with code. So, the voltage input should be driven from a low impedance source. Note that VIN is limited to low voltages, because the switches in the DAC ladder no longer have the same source-drain drive voltage. As a result, their on resistance differs and this degrades the integral linearity of the DAC. Also, VIN must not go negative by more than 0.3 V or an internal diode turns on, exceeding the maximum ratings of the device. In this type of application, the DAC’s full range of multiplying capability is lost. VDD R1 RFB IOUT1 VIN R2 VDD VOUT VREF IOUT2 NOTES: 1. ADDITIONAL PINS OMITTED FOR CLARITY. 2. C1 PHASE COMPENSATION (1pF–2pF) MAY BE REQUIRED IF A1 IS A HIGH SPEED AMPLIFIER. 04464-0-009 GND Figure 39. Single-Supply Voltage-Switching Mode VDD = +5V ADR03 VOUT VIN GND +5V –2.5V 1/2 AD8552 VREF 8-/10-/12-BIT DAC RFB C1 IOUT1 IOUT2 GND –5V VOUT = 0V TO +2.5V 1/2 AD8552 NOTES 1. ADDITIONAL PINS OMITTED FOR CLARITY. 2. C1 PHASE COMPENSATION (1pF–2pF) MAY BE REQUIRED IF A1 IS A HIGH SPEED AMPLIFIER. Figure 40. Positive Voltage Output with Minimum Components Rev. 0 | Page 17 of 32 04464-0-010 VDD AD5429/AD5439/AD5449 ADDING GAIN In applications in which the output voltage is required to be greater than VIN, gain can be added with an additional external amplifier, or it can be achieved in a single stage. Be sure to take into consideration the effect of temperature coefficients of the thin film resistors of the DAC. Simply placing a resistor in series with the RFB resistor causes mismatches in the temperature coefficients, resulting in larger gain temperature coefficient errors. Instead, the circuit of Figure 41 is a recommended method of increasing the gain of the circuit. R1, R2, and R3 should all have similar temperature coefficients, but they need not match the temperature coefficients of the DAC. This approach is recommended in circuits in which gains of > 1 are required. As D is reduced, the output voltage increases. For small values of the digital fraction D, it is important to ensure that the amplifier does not saturate and also that the required accuracy is met. For example, an 8-bit DAC driven with the binary code 0 × 10 (00010000)—that is, 16 decimal—in the circuit of Figure 42 should cause the output voltage to be 16 × VIN. However, if the DAC has a linearity specification of ±0.5 LSB, then D can, in fact, have a weight in the range 15.5/256 to 16.5/256, so that the possible output voltage is in the range 15.5 VIN to 16.5 VIN with an error of +3%, even though the DAC itself has a maximum error of 0.2%. DAC leakage current is also a potential error source in divider circuits. The leakage current must be counterbalanced by an opposite current supplied from the op amp through the DAC. Because only a fraction D of the current into the VREF terminal is routed to the IOUT1 terminal, the output voltage has to change as follows: DIVIDER OR PROGRAMMABLE GAIN ELEMENT Current-steering DACs are very flexible and lend themselves to many different applications. If this type of DAC is connected as the feedback element of an op amp, and RFB is used as the input resistor, as shown in Figure 42, then the output voltage is inversely proportional to the digital input fraction D. For D = 1 − 2n the output voltage is where R is the DAC resistance at the VREF terminal. For a DAC leakage current of 10 nA, R = 10 kΩ and a gain (that is, 1/D) of 16, the error voltage is 1.6 mV. ) VDD VIN R2 VREF C1 RFB 8-/10-/12-BIT DAC IOUT1 VOUT IOUT2 R3 GND GAIN = R2 R2 + R3 R2 R2R3 R1 = R2 + R3 NOTES: 1. ADDITIONAL PINS OMITTED FOR CLARITY. 2. C1 PHASE COMPENSATION (1pF TO 2pF) MAY BE REQUIRED IF A1 IS A HIGH SPEED AMPLIFIER. Figure 41. Increasing Gain of Current Output DAC VDD VIN RFB VDD IOUT1 VREF IOUT2 GND VOUT NOTE: 1. ADDITIONAL PINS OMITTED FOR CLARITY. Figure 42. Current-Steering DAC Used as a Divider or Programmable Gain Element Rev. 0 | Page 18 of 32 04464-0-011 VDD 04464-0-012 VOUT = − VIN / D = − VIN / (1 − 2 −n Output Error Voltage Due to DAC Leakage = (Leakage × R)/D AD5429/AD5439/AD5449 REFERENCE SELECTION When selecting a reference for use with the AD5429/AD5439/ AD5449 family of current output DACs, pay attention to the reference’s output voltage temperature coefficient specification. This parameter affects not only the full-scale error, but also the linearity (INL and DNL) performance. The reference temperature coefficient should be consistent with the system accuracy specifications. For example, an 8-bit system required to hold its overall specification to within 1 LSB over the temperature range 0°C to 50°C dictates that the maximum system drift with temperature should be less than 78 ppm/°C. A 12-bit system with the same temperature range to overall specification within 2 LSBs requires a maximum drift of 10 ppm/°C. By choosing a precision reference with low output temperature coefficient, this error source can be minimized. Table 7 lists some of the references available from Analog Devices that are suitable for use with this range of current output DACs. AMPLIFIER SELECTION The primary requirement for the current-steering mode is an amplifier with low input bias currents and low input offset voltage. The input offset voltage of an op amp is multiplied by the variable gain (due to the code-dependent output resistance of the DAC) of the circuit. A change in this noise gain between two adjacent digital fractions produces a step change in the output voltage due to the amplifier’s input offset voltage. This output voltage change is superimposed upon the desired change in output between the two codes and gives rise to a differential linearity error, which, if large enough, could cause the DAC to be nonmonotonic. The input bias current of an op amp also generates an offset at the voltage output as a result of the bias current flowing in the feedback resistor RFB. Most op amps have input bias currents low enough to prevent any significant errors in 12-bit applications. Common-mode rejection of the op amp is important in voltage-switching circuits, because it produces a codedependent error at the voltage output of the circuit. Most op amps have adequate common-mode rejection for use at 8-, 10-, and 12-bit resolution. Provided that the DAC switches are driven from true wideband low impedance sources (VIN and AGND), they settle quickly. Consequently, the slew rate and settling time of a voltageswitching DAC circuit is determined largely by the output op amp. To obtain minimum settling time in this configuration, it is important to minimize capacitance at the VREF node (voltage output node in this application) of the DAC. This is done by using low input capacitance buffer amplifiers and careful board design. Most single-supply circuits include ground as part of the analog signal range, which in turns requires an amplifier that can handle rail-to-rail signals. Analog Devices supplies a large range of single-supply amplifiers. Table 7. Suitable ADI Precision References Recommended for Use with AD5429/AD5439/AD5449 DACs Reference ADR01 ADR02 ADR03 ADR425 Output Voltage 10 V 5V 2.5 V 5V Initial Tolerance 0.1% 0.1% 0.2% 0.04% Temperature Drift 3 ppm/°C 3 ppm/°C 3 ppm/°C 3 ppm/°C 0.1 Hz to 10 Hz Noise 20 µV p-p 10 µV p-p 10 µV p-p 3.4 µV p-p Package SC70, TSOT, SOIC SC70, TSOT, SOIC SC70, TSOT, SOIC MSOP, SOIC Table 8. Precision ADI Op Amps Suitable for Use with AD5429/AD5439/AD5449 DACs Part No. OP97 OP1177 AD8551 Max Supply Voltage (V) ±20 ±18 ±6 VOS (max) µV 25 60 5 IB (max) nA 0.1 2 0.05 GBP MHz 0.9 1.3 1.5 Slew Rate V/µs 0.2 0.7 0.4 VOS (max) µV 1500 1000 3000 IB max (nA) 0.01 1000 0.75 Table 9. High Speed ADI Op Amps Suitable for Use with AD5429/AD5439/AD5449 DACs Part No. AD8065 AD8021 AD8038 Max Supply Voltage (V) ±12 ±12 ±5 BW @ ACL (MHz) 145 200 350 Slew Rate (V/µs) 180 100 425 Rev. 0 | Page 19 of 32 AD5429/AD5439/AD5449 SERIAL INTERFACE SDO Control (SDO1 and SDO2) The AD5429/AD5439/AD5449 have an easy to use, 3-wire interface that is compatible with SPI, QSPI, MICROWIRE, and DSP interface standards. Data is written to the device in 16-bit words. This 16-bit word consists of 4 control bits and either 8, 10, or 12 data bits, as shown in Figure 43, Figure 44, and Figure 45. The SDO bits enable the user to control the SDO output driver strength, disable the SDO output, or configure it as an opendrain driver. The strength of the SDO driver affects the timing of t12, and, when stronger, allows a faster clock cycle. Table 10. SDO Control Bits Low Power Serial Interface SDO2 0 0 1 1 To minimize the power consumption of the device, the interface powers up fully only when the device is being written to, that is, on the falling edge of SYNC. The SCLK and DIN input buffers are powered down on the rising edge of SYNC. SDO1 0 1 0 1 Function Implemented Full SDO driver SDO configured as open-drain Weak SDO driver Disable SDO output DAC Control Bits C3–C0 Control bits C3 to C0 allow control of various functions of the DAC, as shown in Table 11. Default setting of the DAC at power-on are as follows. Daisy-Chain Control (DSY) DSY allows the enabling or disabling of daisy-chain mode. A 1 enables daisy-chain mode, and 0 disables daisy-chain mode. When disabled, a readback request is accepted, SDO is automatically enabled, the DAC register contents of the relevant DAC are clocked out on SDO, and, when complete, SDO is disabled again. Data is clocked into the shift register on falling clock edges; daisy-chain mode is enabled. The device powers on with zeroscale load to the DAC register and IOUT lines. The DAC control bits allow the user to adjust certain features at power-on; for example, daisy-chaining can be disabled if not in use, active clock edge can be changed to rising edge, and DAC output can be cleared to either zero scale or midscale. The user can also initiate a readback of the DAC register contents for verification. Hardware CLR Bit (HCLR) The default setting for the hardware CLR bit is to clear the registers and DAC output to zero code. A 1 in the HCLR bit allows the CLR pin to clear the DAC outputs to midscale and a 0 clears to zero scale. Control Register (Control Bits = 1101) While maintaining software compatibility with the singlechannel current output DACs (AD5426/AD5432/AD5443), these DACs also feature some additional interface functionality. Set the control bits to 1101 to enter control register mode. Figure 46 shows the contents of the control register. The following sections describe the functions of the control register. Active Clock Edge (SCLK) The default active clock edge is falling edge. Write a 1 to this bit to clock data in on the rising edge, or a 0 for falling edge. C3 C2 DB0 (LSB) C1 C0 DB7 DB6 DB5 DB4 DB3 CONTROL BITS DB2 DB1 DB0 0 0 0 0 DATA BITS 04464-0-013 DB15 (MSB) Figure 43. AD5429 8-Bit Input Shift Register Contents C3 C2 DB0 (LSB) C1 C0 DB9 DB8 DB7 DB6 DB5 CONTROL BITS DB4 DB3 DB2 DB1 DB0 0 0 DATA BITS 04464-0-014 DB15 (MSB) Figure 44. AD5439 10-Bit Input Shift Register Contents C3 C2 DB0 (LSB) C1 CONTROL BITS C0 DB11 DB10 DB9 DB8 DB7 DB6 DB5 DB4 DATA BITS Figure 45. AD5449 12-Bit Input Shift Register Contents Rev. 0 | Page 20 of 32 DB3 DB2 DB1 DB0 04464-0-015 DB15 (MSB) AD5429/AD5439/AD5449 SYNC Function SYNC is an edge-triggered input that acts as a frame synchronization signal and chip enable. Data can be transferred into the device only while SYNC is low. To start the serial data transfer, SYNC should be taken low, observing the minimum SYNC falling to SCLK falling edge setup time, t4. Daisy-Chain Mode Daisy-chain mode is the default power-on mode. To disable the daisy-chain function, write 1001 to the control word. In daisychain mode, the internal gating on SCLK is disabled. The SCLK is continuously applied to the input shift register when SYNC is low. If more than 16 clock pulses are applied, the data ripples out of the shift register and appears on the SDO line. This data is clocked out on the rising edge of SCLK (this is the default, use the control word to change the active edge) and is valid for the next device on the falling edge (default). By connecting this line to the SDIN input on the next device in the chain, a multidevice interface is constructed. For each device in the system, 16 clock pulses are required. Therefore, the total number of clock cycles must equal 16, where N is the total number of devices in the chain. See Figure 3. When the serial transfer to all devices is complete, SYNC should be taken high. This prevents additional data from being clocked into the input shift register. A burst clock containing the exact number of clock cycles can be used and SYNC taken high some time later. After the rising edge of SYNC, data is automatically transferred from each device’s input shift register to the addressed DAC. When control bits = 0000, the device is in no operation mode. This might be useful in daisy-chain applications, in which the user does not wish to change the settings of a particular DAC in the chain. Write 0000 to the control bits for that DAC, and the following data bits are ignored. Standalone Mode After power-on, write 1001 to the control word to disable daisychain mode. The first falling edge of SYNC resets a counter that counts the number of serial clocks to ensure that the correct number of bits are shifted in and out of the serial shift registers. A SYNC edge during the 16-bit write cycle causes the device to abort the current write cycle. After the falling edge of the 16th SCLK pulse, data is automatically transferred from the input shift register to the DAC. In order for another serial transfer to take place, the counter must be reset by the falling edge of SYNC. LDAC Function The LDAC function allows asynchronous or synchronous updates to the DAC output. The DAC is asynchronously updated when this signal goes low. Alternatively, if this line is held permanently low, an automatic or synchronous update mode is selected, whereby the DAC is updated on the 16th clock falling edge when the device is in standalone mode, or on the rising edge of SYNC when in daisy-chain mode. Table 11. DAC Control Bits C2 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 C1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 C0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 DAC A and B A A A B B B A and B A and B - Function Implemented No operation (power-on default) Load and update Initiate readback Load input register Load and update Initiate readback Load input register Update DAC outputs Load input registers Daisy chain disable Clock data to shift register on rising edge Clear DAC output to zero scale Clear DAC output to midscale Control word Reserved No operation DB15 (MSB) 1 1 DB0 (LSB) 0 1 SDO2 SDO1 DSY HCLR SCLK X X CONTROL BITS Figure 46. Control Register Loading Sequence Rev. 0 | Page 21 of 32 X X X X X 04464-0-016 C3 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 AD5429/AD5439/AD5449 Software LDAC Function Load and update mode can also function as a software update function, irrespective of the voltage level on the LDAC pin. Communication between two devices at a given clock speed is possible when the following specifications are compatible: frame sync delay and frame sync setup-and-hold, data delay and data setup-and-hold, and SCLK width. The DAC interface expects a t4 SYNC falling edge to SCLK falling edge setup time) of 13 ns minimum. See the ADSP-21xx User Manual for details on clock and frame sync frequencies for the SPORT register. MICROPROCESSOR INTERFACING Microprocessor interfacing to this family of DACs is via a serial bus that uses standard protocol compatible with microcontrollers and DSP processors. The communications channel is a 3-wire interface consisting of a clock signal, a data signal, and a synchronization signal. The AD5429/AD5439/AD5449 require a 16-bit word with the default being data valid on the falling edge of SCLK, but this is changeable via the control bits in the data-word. Table 12 shows how the SPORT control register must be set up. Table 12. Name TFSW INVTFS DTYPE ISCLK TFSR ITFS SLEN ADSP-21xx to AD5429/AD5439/AD5449 Interface The ADSP-21xx family of DSPs is easily interfaced to this family of DACs without the need for extra glue logic. Figure 47 is an example of an SPI interface between the DAC and the ADSP-2191M. SCK of the DSP drives the serial data line, DIN. SYNC is driven from one of the port lines, in this case SPIxSEL. AD5429/AD5439/ AD5449* SPIxSEL 80C51/80L51 to AD5429/AD5439/AD5449 Interface SYNC SDIN SCK SCLK 04464-0-027 MOSI *ADDITIONAL PINS OMITTED FOR CLARITY Figure 47. ADSP-2191 SPI to AD5429/AD5439/AD5449 Interface A serial interface between the DAC and DSP SPORT is shown in Figure 48. In this interface example, SPORT0 is used to transfer data to the DAC shift register. Transmission is initiated by writing a word to the Tx register after the SPORT has been enabled. In a write sequence, data is clocked out on each rising edge of the DSP’s serial clock and clocked into the DAC input shift register on the falling edge of its SCLK. The update of the DAC output takes place on the rising edge of the SYNC signal. AD5429/AD5439/ AD5449* TFS SYNC DT SDIN SCLK A serial interface between the DAC and the 80C51/80L51 is shown in Figure 49. TxD of the 80C51/80L51drives SCLK of the DAC serial interface, while RxD drives the serial data line, DIN. P1.1 is a bit-programmable pin on the serial port and is used to drive SYNC. When data is to be transmitted to the switch, P1.1 is taken low. The 80C51/80L51 transmit data only in 8-bit bytes; thus, only eight falling clock edges occur in the transmit cycle. To load data correctly to the DAC, P1.1 is left low after the first eight bits are transmitted, and a second write cycle is initiated to transmit the second byte of data. Data on RXD is clocked out of the microcontroller on the rising edge of TXD and is valid on the falling edge. As a result, no glue logic is required between the DAC and microcontroller interface. P1.1 is taken high following the completion of this cycle. The 80C51/80L51 provide the LSB of the SBUF register as the first bit in the data stream. The DAC input register requires its data with the MSB as the first bit received. The transmit routine should take this into account. SCLK *ADDITIONAL PINS OMITTED FOR CLARITY Figure 48. ADSP-2101/ADSP-2103/ADSP-2191 SPORT to AD5429/AD5439/AD5449 Interface AD5429/AD5439/ AD5449* 80C51* 04464-0-028 ADSP-2101/ ADSP-2103/ ADSP-2191* Description Alternate framing Active low frame signal Right-justify data Internal serial clock Frame every word Internal framing signal 16-bit data-word TxD SCLK RxD SDIN P1.1 SYNC *ADDITIONAL PINS OMITTED FOR CLARITY Figure 49. 80C51/80L51 to AD5429/AD5439/AD5449 Interface Rev. 0 | Page 22 of 32 04464-0-029 ADSP-2191* Setting 1 1 00 1 1 1 1111 AD5429/AD5439/AD5449 MC68HC11 to AD5429/AD5439/AD5449 Interface MICROWIRE to AD5429/AD5439/AD5449 Interface Figure 50 is an example of a serial interface between the DAC and the MC68HC11 microcontroller. The serial peripheral interface (SPI) on the MC68HC11 is configured for master mode (MSTR) = 1, clock polarity bit (CPOL) = 0, and the clock phase bit (CPHA) = 1. The SPI is configured by writing to the SPI control register (SPCR)—see the 68HC11 User Manual. The SCK of the 68HC11 drives the SCLK of the DAC interface; the MOSI output drives the serial data line (DIN) of the AD5429/AD5439/AD5449. Figure 51 shows an interface between the DAC and any MICROWIRE compatible device. Serial data is shifted out on the falling edge of the serial clock, SK, and is clocked into the DAC input shift register on the rising edge of SK, which corresponds to the falling edge of the DAC’s SCLK. MICROWIRE* PIC16C6x/7x to AD5429/AD5439/AD5449 The PIC16C6x/7x synchronous serial port (SSP) is configured as an SPI master with the clock polarity bit (CKP) = 0. This is done by writing to the synchronous serial port control register (SSPCON). See the PIC16/17 Microcontroller User Manual. In this example, the I/O port RA1 is used to provide a SYNC signal and enable the serial port of the DAC. This microcontroller transfers only eight bits of data during each serial transfer operation; therefore, two consecutive write operations are required. Figure 52 shows the connection diagram. SYNC SCLK MOSI SDIN *ADDITIONAL PINS OMITTED FOR CLARITY PIC16C6x/7x* Figure 50. MCH68HC11/68L11 to AD5429/AD5439/AD5449 Interface If the user wants to verify the data previously written to the input shift register, the SDO line can be connected to MISO of the MC68HC11, and, with SYNC low, the shift register clocks data out on the rising edges of SCLK. SYNC Figure 51. MICROWIRE to AD5429/AD5439/AD5449 Interface 04464-0-030 PC7 SDIN CS *ADDITIONAL PINS OMITTED FOR CLARITY AD5429/AD5439/ AD5449* SCK SCLK AD5429/AD5439/ AD5449* SCK/RC3 SCLK SDI/RC4 SDIN RA1 SYNC *ADDITIONAL PINS OMITTED FOR CLARITY Figure 52. PIC16C6x/7x to AD5429/AD5439/AD5449 Interface Rev. 0 | Page 23 of 32 04464-0-032 MC68HC11* SK SO 04464-0-031 The SYNC signal is derived from a port line (PC7). When data is being transmitted to the AD5429/AD5439/AD5449, the SYNC line is taken low (PC7). Data appearing on the MOSI output is valid on the falling edge of SCK. Serial data from the 68HC11 is transmitted in 8-bit bytes with only 8 falling clock edges occurring in the transmit cycle. Data is transmitted MSB first. To load data to the DAC, PC7 is left low after the first eight bits are transferred, and a second serial write operation is performed to the DAC. PC7 is taken high at the end of this procedure. AD5429/AD5439/ AD5449* AD5429/AD5439/AD5449 PCB LAYOUT AND POWER SUPPLY DECOUPLING In any circuit where accuracy is important, careful consideration of the power supply and ground return layout helps to ensure the rated performance. The printed circuit board on which the DAC is mounted should be designed so that the analog and digital sections are separated and confined to certain areas of the board. If the DAC is in a system in which multiple devices require an AGND to DGND connection, the connection should be made at one point only. The star ground point should be established as close as possible to the device. It is good practice to employ compact, minimum lead-length PCB layout design. Leads to the input should be as short as possible to minimize IR drops and stray inductance. These DACs should have ample supply bypassing of 10 µF in parallel with 0.1 µF on the supply located as close to the package as possible, ideally right up against the device. The 0.1 µF capacitor should have low effective series resistance (ESR) and effective series inductance (ESI), like the common ceramic types that provide a low impedance path to ground at high frequencies, to handle transient currents due to internal logic switching. Low ESR 1 µF to 10 µF tantalum or electrolytic capacitors should also be applied at the supplies to minimize transient disturbance and filter out low frequency ripple. The board requires ±12 V and +5 V supplies. The 12 V VDD and VSS are used to power the output amplifier, while the 5 V is used to power the DAC (VDD1) and transceivers (VCC). Fast switching signals such as clocks should be shielded with digital ground to avoid radiating noise to other parts of the board, and should never be run near the reference inputs. Avoid crossover of digital and analog signals. Traces on opposite sides of the board should run at right angles to each other. This reduces the effects of feedthrough on the board. A microstrip technique is by far the best, but not always possible with a double-sided board. In this technique, the component side of the board is dedicated to the ground plane, while signal traces are placed on the soldered side. The PCB metal traces between VREF and RFB should also be matched to minimize gain error. To maximize high frequency performance, the I-to-V amplifier should be located as close to the device as possible. POWER SUPPLIES FOR THE EVALUATION BOARD Both supplies are decoupled to their respective ground plane with 10 µF tantalum and 0.1 µF ceramic capacitors. EVALUATION BOARD FOR THE DACS The evaluation board includes a DAC from the AD5429/ AD5439/AD5449 family and a current-to-voltage amplifier, AD8065. On the evaluation board is a 10 V reference, ADR01. An external reference can also be applied via an SMB input. The evaluation kit consists of a CD-ROM with self-installing PC software to control the DAC. The software allows the user to write a code to the device. Rev. 0 | Page 24 of 32 Figure 53. Schematic of the Evaluation Board Rev. 0 | Page 25 of 32 04464-0-023 P1–19 P1–20 P1–21 P1–22 P1–23 P1–24 P1–25 P1–26 P1–27 P1–28 P1–29 P1–30 P1–6 P1–13 P1–5 P1–4 P1–2 P1–3 P2–4 P2–1 P2–2 P2–3 LDAC SDIN SCLK C15 0.1µF C13 0.1µF C11 0.1µF + + + A B C16 10µF C14 10µF C12 10µF LK2 R1 10kΩ VDD R2 10kΩ VDD R3 10kΩ VDD SDO VSS VDD1 AGND VDD J6 J5 J4 J3 CLR C3 10µF J7 LDAC SYNC SDIN SCLK CLR SDO + VDD C4 0.1µF RFBB VREFA IOUT2A IOUT1A RFBB IOUT2B IOUT1B 5 TRIM 3 +V IN 4 GND U2 J8 1 LK1 B C5 0.1µF VREFB J9 VREFA VOUT 4 VREFB VREFA 5 13 4 2 1 3 15 16 14 C1 0.1µF + C2 10µF VREFB GND LDAC SYNC SDIN SCLK 12 VDD AD5429/AD5439/ AD5449 11 9 6 10 8 7 U1 VDD1 A C6 1.8pF C17 1.8pF 3 VDD 7 V– V+ U3 4 7 V– V+ 4 2 VSS VDD U4 3 2 VSS C10 0.1µF + C9 10µF AD8065AR 6 C8 0.1µF + C7 10µF C21 0.1µF + C20 10µF AD8065AR 6 C19 0.1µF + C18 10µF TP1 TP2 J1 J2 VOUT A VOUT B AD5429/AD5439/AD5449 04464-0-024 AD5429/AD5439/AD5449 04464-0-025 Figure 54. Component-Side Artwork Figure 55. Silkscreen—Component-Side View (Top) Rev. 0 | Page 26 of 32 04464-0-026 AD5429/AD5439/AD5449 Figure 56. Solder-Side Artwork Rev. 0 | Page 27 of 32 AD5429/AD5439/AD5449 OVERVIEW OF AD54xx DEVICES Table 13. Part No. AD5424 AD5426 AD5428 AD5429 AD5450 AD5432 AD5433 AD5439 AD5440 AD5451 AD5443 AD5444 AD5415 AD5445 AD5447 AD5449 AD5452 AD5446 AD5453 AD5553 AD5556 AD5555 AD5557 AD5543 AD5546 AD5545 AD5547 Resolution 8 8 8 8 8 10 10 10 10 10 12 12 12 12 12 12 12 14 14 14 14 14 14 16 16 16 16 No. DACs 1 1 2 2 1 1 1 2 2 1 1 1 2 2 2 2 1 1 1 1 1 2 2 1 1 2 2 INL (LSB) ±0.25 ±0.25 ±0.25 ±0.25 ±0.25 ±0.5 ±0.5 ±0.5 ±0.5 ±0.25 ±1 ±0.5 ±1 ±1 ±1 ±1 ±0.5 ±1 ±2 ±1 ±1 ±1 ±1 ±2 ±2 ±2 ±2 Interface Parallel Serial Parallel Serial Serial Serial Parallel Serial Parallel Serial Serial Serial Serial Parallel Parallel Serial Serial Serial Serial Serial Parallel Serial Parallel Serial Parallel Serial Parallel Package RU-16, CP-20 RM-10 RU-20 RU-10 RJ-8 RM-10 RU-20, CP-20 RU-16 RU-24 RJ-8 RM-10 RM-8 RU-24 RU-20, CP-20 RU-24 RU-16 RJ-8, RM-8 RM-8 UJ-8, RM-8 RM-8 RU-28 RM-8 RU-38 RM-8 RU-28 RU-16 RU-38 Rev. 0 | Page 28 of 32 Features 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 50 MHz Serial 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 50 MHz Serial 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 58 MHz Serial 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 17 ns CS Pulse Width 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 10 MHz BW, 50 MHz Serial 4 MHz BW, 50 MHz Serial Clock 4 MHz BW, 20 ns WR Pulse Width 4 MHz BW, 50 MHz Serial Clock 4 MHz BW, 20 ns WR Pulse Width 4 MHz BW, 50 MHz Serial Clock 4 MHz BW, 20 ns WR Pulse Width 4 MHz BW, 50 MHz Serial Clock 4 MHz BW, 20 ns WR Pulse Width AD5429/AD5439/AD5449 OUTLINE DIMENSIONS 5.10 5.00 4.90 16 9 4.50 4.40 4.30 6.40 BSC 1 8 PIN 1 1.20 MAX 0.15 0.05 0.20 0.09 0.65 BSC 0.30 0.19 COPLANARITY 0.10 SEATING PLANE 0.75 0.60 0.45 8° 0° COMPLIANT TO JEDEC STANDARDS MO-153AB Figure 57. 16-Lead Thin Shrink Small Outline Package [TSSOP] (RU-16) Dimensions shown in millimeters ORDERING GUIDE Model AD5429YRU AD5429YRU-REEL AD5429YRU-REEL7 AD5439YRU AD5439YRU-REEL AD5439YRU-REEL7 AD5449YRU AD5449YRU-REEL AD5449YRU-REEL7 EVAL-AD5429EB EVAL-AD5439EB EVAL-AD5449EB Resolution 8 8 8 10 10 10 12 12 12 INL (LSBs) ±0.5 ±0.5 ±0.5 ±0.5 ±0.5 ±0.5 ±1 ±1 ±1 Temperature Range −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C Rev. 0 | Page 29 of 32 Package Description TSSOP TSSOP TSSOP TSSOP TSSOP TSSOP TSSOP TSSOP TSSOP Evaluation Board Evaluation Board Evaluation Board Package Option RU-16 RU-16 RU-16 RU-16 RU-16 RU-16 RU-16 RU-16 RU-16 AD5429/AD5439/AD5449 NOTES Rev. 0 | Page 30 of 32 AD5429/AD5439/AD5449 NOTES Rev. 0 | Page 31 of 32 AD5429/AD5439/AD5449 NOTES © 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D04464–0–7/04(0) Rev. 0 | Page 32 of 32